CN114142760B - Discrete control method and device for three-phase full-bridge inverter - Google Patents

Discrete control method and device for three-phase full-bridge inverter Download PDF

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CN114142760B
CN114142760B CN202111536472.5A CN202111536472A CN114142760B CN 114142760 B CN114142760 B CN 114142760B CN 202111536472 A CN202111536472 A CN 202111536472A CN 114142760 B CN114142760 B CN 114142760B
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duty ratio
load voltage
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bridge inverter
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CN114142760A (en
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沙金
王一帆
许建平
魏红昊
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Southwest Jiaotong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/20—Active power filtering [APF]

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Abstract

The invention belongs to the technical field of three-phase alternating current power supplies, and particularly relates to a discrete control method and a discrete control device for a three-phase full-bridge inverter. The method of the invention is as follows, according to the control requirement of the three-phase load voltage, in one switching periodAt the beginning time, the controller samples the input bus capacitor voltage, the three-phase load voltage and the three-phase load current and calculates to obtain three-phase discrete P H And P L A duty cycle; at the same time, three-phase load voltages are sampled and compared with preset three-phase load voltage set respectively, and the three-phase load voltages are obtained in three phases P H And P L The proper duty ratio is selected as the effective duty ratio of the phase control pulse in the next switching period, so that the control of the three-phase load voltage is realized. The scheme of the invention is realized under a static coordinate system, a load current control loop is not needed, a proportional-integral amplifier and a corresponding compensation link are not needed, the control is simple, and the control cost is low.

Description

Discrete control method and device for three-phase full-bridge inverter
Technical Field
The invention belongs to the technical field of three-phase alternating current power supplies, and particularly relates to a discrete control method and a discrete control device for a three-phase full-bridge inverter.
Background
Based on the advantages of low voltage and current stress and high power density of the three-phase full-bridge inverter, the three-phase full-bridge inverter is widely applied to high-power industrial occasions such as three-phase uninterruptible power supplies, alternating current asynchronous motors, permanent magnet synchronous motor alternating current-direct current-traffic frequency converters, power system static reactive power compensation devices, three-phase active power filter devices and the like which are powered by independent single-phase and three-phase loads. These applications require that the three-phase full-bridge inverter power supply have good steady-state performance, rapid input and load dynamic performance, and have frequency modulation and voltage regulation functions.
The control strategy of the three-phase full-bridge inverter can be divided into the following two types according to the coordinate system of the implementation control mode:
the first type is a static coordinate system control mode, namely, a control target is realized under an A-B-C three-phase static coordinate system or an alpha-beta two-phase static coordinate system. The advantages are that: the control system does not need a software phase-locked loop (and a rotating coordinate transformation link, is simple to realize, has lower software cost, can realize broadband control performance by adopting a traditional proportional-integral (PI) controller, can realize no static difference control performance under an alternating current control system by adopting a traditional proportional-resonance (PR) controller or a quasi-proportional-resonance controller (Q-PR), has the following defects that because of the PI controllerThe influence of the middle integral compensation link is that the control static difference exists in a three-phase alternating current control system by adopting the traditional PI control; (see document (1): L.Huber, M.Kumar and M.M).
Figure GDA0003461846090000011
"Implementation and performance comparison of five dsp-based control methods for three-phase six-switch boost pfc rectifier", 2015-IEEE Applied Power Electronics Conference and Exposition (APEC), pp.101-108,2015.) due to the resonant frequency parameter ω in PR controller r Is based on the influence of the angular frequency omega of the controlled object of the alternating current system and the resonant frequency parameter omega of the PR controller r When frequency offset occurs, the gain and control performance of the controller are reduced, so that the multi-frequency control can only be realized through a plurality of PR controllers, and the realization complexity of the system is increased; (see, document (2) I.Etxeberria-Otadui, A.L. de Herodia, H.Gaztaaga, S.Bacha and M.R. Reyero, "A Single Synchronous Frame Hybrid (SSFH) multifrequency controller for power active filters", IEEE Trans.Ind. Electron., vol.53, no.5, pp.1640-1648, oct.2006; document (3): M.Liserre, R.Teodorescu, and F.Blaabjerg, "Multiple harmonics control for three-phase grid converter systems with the use of PI-RES current controller in a rotating frame", IEEE Trans.Power Electron., vol.21, pp.836-841, may 2006.) Q-PR controller introduces a control parameter cut-off frequency ω on the basis of PR controller c To improve the control performance at the time of frequency offset, but the transfer function and parameter setting process of the controller are relatively complex compared with the traditional PR controller, and the frequency bandwidth of the controller is subject to the cutoff frequency parameter omega c Is limited by the number of (a). ( See document (4): chen and F.Wu, "Research and Implementation of Single-phase AC Electronic Load Based on Quasi-PR Control",2018 International Conference on Advanced Mechatronic Systems (ICAMechs), pp.157-161,2018. )
The second type is a rotating coordinate system control mode, namely, a control target is realized under a d-q two-phase rotating coordinate system. The advantages are that: under the rotating coordinate system, the control target is converted from alternating current to direct current, so that no-static-difference control can be realized by using a traditional PI controller. The defects are as follows: the control system needs a PLL module and a rotary coordinate transformation module, so that the control operand and the software cost are increased; the control system depends on the PLL module, so that the control system can only realize fixed frequency control and cannot be used in three-phase inversion variable-frequency power supply application occasions requiring frequency adjustment. (see, document (5): Z.Shuai, Y.Li, W.Wu, C.Tu, A.Luo and Z.J.Screen, "digital DQ small-signal model: a new perspective for the stability analysis of three-phase grid-fed indexes", IEEE Trans.Ind. Electron., vol.66, no.8, pp.6493-6504, sep.2019).
Disclosure of Invention
In order to solve the above problems, the present invention provides a discrete control method and device suitable for a three-phase full-bridge inverter, so as to optimize performance of the conventional control strategy described in the background description.
The technical scheme of the invention is as follows:
a discrete control method of a three-phase full-bridge inverter comprises the following steps:
s1, setting given values of three-phase load voltages of a kth switching period to be u respectively Ar * (k)、u Br * (k)、u Cr * (k)；
S2, at the starting time t of the kth switching period, obtaining the input bus capacitor voltage u of the three-phase full-bridge inverter through sampling dc (t), three-phase load voltage u Ar (t)、u Br (t)、u Cr (t) three-phase load current i Ar (t)、i Br (t)、i Cr (t) signal, and the signal is converted by A/D conversion module to obtain correspondent digital signal u dc (k)、u Ar (k)、u Br (k)、u Cr (k)、i Ar (k)、i Br (k)、i Cr (k)；
S3, according to the data of the steps S1 and S2, calculating the three-phase P of the next switching period H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1), specifically:
Figure GDA0003461846090000021
Figure GDA0003461846090000022
wherein the method comprises the steps of
Figure GDA0003461846090000031
Figure GDA0003461846090000032
Omega is the angular frequency of a three-phase fundamental wave of the system, L and C are LC filtering parameters of a three-phase full-bridge inverter, and load current i αr (k) And i βr (k) By three-phase load current i Ar (k)、i Br (k)、i Cr (k) Obtained through 3s-2s coordinate transformation:
Figure GDA0003461846090000033
s4, u is Ar (k)、u Br (k)、u Cr (k) And u is equal to Ar * (k)、u Br * (k)、u Cr * (k) Comparing, according to the control rule, the three phases P generated in step S3 H Three-phase P L Three-phase duty ratio D of the next switching cycle is selected and generated from the duty ratios A (k+1)、D B (k+1)、D C (k+1); the control rule is as follows: when the A-phase load voltage feedback value u Ar (k) Less than the phase A load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AH (k+1); when the A-phase load voltage feedback value u Ar (k) Greater than phase a load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AL (k+1), B and C phases stationary coordinate systemFinal duty cycle D in (2) B (k+1) and D C (k+1) generating rules as same as the A-phase stationary coordinate system;
s5, according to the obtained three-phase duty ratio D A (k+1)、D B (k+1)、D C (k+1) generating a corresponding three-phase control signal S A (k+1)、S B (k+1)、S C (k+1) realizing control of three-phase load voltage by controlling the three-phase full-bridge inverter power device.
The discrete control device of the three-phase full-bridge inverter comprises the three-phase full-bridge inverter and a discrete controller, wherein the discrete controller comprises a voltage and current sampling and A/D conversion module, a duty ratio calculation module, a duty ratio selection module, a DPWM and a driving module;
the voltage and current sampling and A/D conversion module is used for sampling the input bus capacitor voltage u of the three-phase full-bridge inverter at the starting time of the kth switching period dc (t) three-phase load voltage u Ar (t)、u Br (t)、u Cr (t) three-phase load current i Ar (t)、i Br (t)、i Cr (t) signal, and the signal is converted by A/D conversion module to obtain correspondent digital signal u dc (k)、u Ar (k)、u Br (k)、u Cr (k)、i Ar (k)、i Br (k)、i Cr (k) Outputting the obtained digital signal to a duty ratio calculation module;
the duty ratio calculation module is used for calculating the three-phase P of the next switching period according to the received data H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1), specifically:
Figure GDA0003461846090000041
Figure GDA0003461846090000042
wherein the method comprises the steps of
Figure GDA0003461846090000043
Figure GDA0003461846090000044
u Ar * (k)、u Br * (k)、u Cr * (k) The third switching period is the set value of the three-phase load voltage of the kth switching period, the omega is the angular frequency of the three-phase fundamental wave of the system, L and C are LC filtering parameters of the three-phase full-bridge inverter, and the load current i αr (k) And i βr (k) By three-phase load current i Ar (k)、i Br (k)、i Cr (k) Obtained through 3s-2s coordinate transformation:
Figure GDA0003461846090000051
the duty ratio calculation module obtains three phases P H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1) output to the duty cycle selection module;
the duty ratio selection module selects the duty ratio according to the set value u Ar * (k)、u Br * (k)、u Cr * (k) And feedback value u Ar (k)、u Br (k)、u Cr (k) The relationship between the three phases P H Duty cycle and three-phase P L Of the duty ratios, three-phase duty ratio D of the next switching cycle is selected A (k+1)、D B (k+1)、D C (k+1), specifically: when the A-phase load voltage feedback value u Ar (k) Less than the phase A load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AH (k+1); when the A-phase load voltage feedback value u Ar (k) Greater than phase a load voltage set point u Ar * (k) In the time-course of which the first and second contact surfaces,the A phase duty cycle of the next switching period is D A (k+1)＝D AL (k+1), final duty cycle D in stationary coordinate systems of B and C phases B (k+1) and D C (k+1) generating rules as same as the A-phase stationary coordinate system; the duty ratio selection module outputs the obtained three-phase duty ratio to the DPWM and the driving module;
the DPWM and the driving module generate a three-phase control signal S of the three-phase full-bridge inverter according to the received three-phase duty ratio A (k+1)、S B (k+1)、S C And (k+1) is used for driving the power devices of the three-phase full-bridge inverter so as to realize the control of the three-phase load voltage of the converter.
The beneficial effects of the invention are as follows:
1. compared with the traditional two-phase rotating coordinate system-three-phase full-bridge inverter control method, the control system is realized under the static coordinate system, the rotating coordinate transformation and the PLL link are not needed, the control is realized simply, and the control cost is low;
2. compared with the traditional voltage-current double closed-loop control method of the three-phase full-bridge inverter, the control system only has a load voltage control loop, and good control performance on alternating current load voltage can be realized without an integral compensation link in the voltage control loop, so that complicated control parameter setting work is not required; in addition, a current loop and a corresponding controller design are not needed, and the complexity of the controller is low;
3. compared with the traditional control method of the single-closed-loop three-phase full-bridge inverter, the invention introduces load current information into a load voltage control loop of a control system, and can realize the following steps: (1) under different load conditions, the system has good steady-state control performance; (2) fast load dynamic control performance;
4. compared with the traditional three-phase full-bridge inversion PR control system, the control system can have higher fault tolerance on the setting of fundamental wave angular frequency parameter omega by utilizing control performance, realize broadband control within a certain frequency range, and can be used for three-phase inversion power supply application occasions needing transformation/frequency conversion; in addition, the discrete control method has higher fault tolerance to L, C filtering parameters in the three-phase full-bridge inverter system, the robustness of the control system is strong, and a parameter identification link is not needed in the control system.
Drawings
Fig. 1 is a schematic diagram of a discrete control method of a three-phase inverter according to the present invention.
Fig. 2 is a schematic diagram of the structure of the duty ratio calculation module.
Fig. 3 shows an embodiment of the three-phase full-bridge inverter under full load (three-phase output active power P o =1200W) steady-state time domain simulation waveform diagram of three-phase load voltage, current, and three-phase active power.
Fig. 4 is a three-phase load voltage simulation THD data histogram of the three-phase full-bridge inverter according to the embodiment under different load conditions (three-phase output active power per unit value 0.1p.u-1.0 p.u).
Fig. 5 is a time domain simulation waveform diagram of the three-phase load voltage, current and dc input voltage of the control three-phase full-bridge inverter according to the present invention when the input voltage suddenly changes (the dc input voltage suddenly rises from 300V to 400V at t=0.04 s).
Fig. 6 is a time-domain simulation waveform diagram and a dynamic detail waveform diagram of three-phase load voltage, current and three-phase active power of the control three-phase full-bridge inverter according to the present invention when a sudden change occurs in load (the three-phase output power suddenly rises from 600W to 1200W at t=0.04 s).
Fig. 7 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =1200W), the load voltage amplitude when the filter parameter set value is relatively erroneous controls the error data line graph.
Fig. 8 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =1200W), a load voltage phase control error data plot when a relative error occurs in the filter parameter set point.
Fig. 9 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =600W), the load voltage amplitude when the filter parameter set value is relatively error-controlling error data line graph.
FIG. 10 shows a three-phase full-bridge inversion according to an embodiment of the inventionWhen the transformer is in half-load working condition (three-phase output active power P o =600W), a load voltage phase control error data plot when a relative error occurs in the filter parameter set point.
Fig. 11 shows a three-phase full-bridge inverter according to an embodiment (three-phase output active power P o =1200w), three-phase load voltage instantaneous value, amplitude and phase time domain simulation waveform diagram when the relative error of the filter parameter set value is 50%.
Fig. 12 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =1200w), a load voltage amplitude error data line graph when a relative error occurs between the fundamental angular frequency parameter actual value and the set value (the embodiment of the present invention is set to ω=100pi rad/s, the corresponding frequency parameter set value is 50 Hz).
Fig. 13 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =1200w), a load voltage phase control error data line graph when a relative error occurs between the fundamental angular frequency parameter actual value and the set value (the embodiment of the present invention is set to ω=100pi rad/s, the corresponding frequency parameter set value is 50 Hz).
Fig. 14 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =600w), a load voltage amplitude control error data line graph when a relative error occurs between an actual value and a set value of the fundamental angular frequency parameter (set to ω=100pi rad/s for the embodiment of the present invention, and 50Hz for the frequency parameter set).
Fig. 15 shows a three-phase full-bridge inverter according to an embodiment of the invention (three-phase output active power P o =600w), a load voltage phase control error data line graph when a relative error occurs between an actual value and a set value of the fundamental angular frequency parameter (set to ω=100pi rad/s for the embodiment of the present invention, and 50Hz for the frequency parameter set).
Fig. 16 shows the three-phase full-bridge inverter according to the embodiment (three-phase output active power P o =1200W), the fundamental wave frequency parameter setting value is 50Hz, and the actual value is 10HzVoltage instantaneous value, amplitude and phase time domain simulation waveform diagram.
Fig. 17 shows the three-phase full-bridge inverter according to the embodiment (three-phase output active power P o =1200w), the fundamental frequency parameter setting value is 50Hz, and the three-phase load voltage instantaneous value, amplitude and phase time domain simulation waveform diagram when the actual value is 100 Hz.
Fig. 18 is a three-phase load voltage time domain simulation waveform diagram of the three-phase full-bridge inverter according to the embodiment, wherein the load voltage is given with amplitude of 0-110V and frequency of 10-200 Hz when the fundamental frequency parameter setting value is 50 Hz.
Detailed Description
The invention is described in detail below with reference to the drawings and examples.
As shown in fig. 1, the discrete control method and apparatus for a three-phase full-bridge inverter provided by the present invention include a three-phase full-bridge inverter and a discrete controller, where the discrete controller includes: the device comprises a voltage and current sampling and A/D conversion module, a duty ratio calculation module, a duty ratio selection module, a DPWM and a driving module.
As shown in fig. 2, the duty ratio calculation module in the controller includes: 3s-2s coordinate transformation module and duty ratio D AI (k+1)-D BI (k+1)-D CI (k+1) calculation module, duty cycle D AII (k+1)-D BII (k+1)-D CII (k+1) calculation module, three-phase P H Duty ratio calculation module and three-phase P L And a duty cycle calculation module.
The discrete control method suitable for the three-phase full-bridge inverter is realized by the following steps:
step one, setting the given values of the three-phase load voltage of the kth switching period to be u respectively according to the control requirement of the three-phase load voltage Ar * (k)、u Br * (k)、u Cr * (k)；
Step two, at the starting time of the switching period, the sampling module samples the input bus capacitor voltage u of the three-phase full-bridge inverter dc (t) three-phase load voltage u Ar (t)、u Br (t)、u Cr (t)Three-phase load current i Ar (t)、i Br (t)、i Cr (t) signal, and the signal is converted by A/D conversion module to obtain correspondent digital signal u dc (k)、u Ar (k)、u Br (k)、u Cr (k)、i Ar (k)、i Br (k)、i Cr (k)；
Step three, the duty ratio calculation module calculates three-phase P of the next switching period H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1)；
Three-phase P H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL Each of (k+1) contains a two-part duty cycle: duty cycle D AI (k+1)、D BI (k+1)、D CI (k+1) and duty cycle D AII (k+1)、D BII (k+1)、D CII (k+1) wherein three phases P H Duty ratios are respectively D AH (k+1)＝D AI (k+1)+|D AII (k+1)|、D BH (k+1)＝D BI (k+1)+|D BII (k+1) | and D CH (k+1)＝D CI (k+1)+|D CII (k+1) |, three-phase P L Duty ratios are respectively D AH (k+1)＝D AI (k+1)-|D AII (k+1)|、D BH (k+1)＝D BI (k+1)-|D BII (k+1) | and D CH (k+1)＝D CI (k+1)-|D CII (k+1)|；
Duty cycle D AI (k+1)、D BI (k+1)、D CI (k+1) the input bus capacitor voltage u dc (k) The given value of the three-phase load voltage is u Ar * (k)、u Br * (k)、u Cr * (k) The calculation formula is as follows:
Figure GDA0003461846090000081
l and C are LC filtering parameters of the three-phase full-bridge inverter, and omega is the angular frequency of a three-phase fundamental wave of the system.
Duty cycle D AII (k+1)、D BII (k+1)、D CII (k+1) the input bus capacitor voltage u dc (k) Three-phase load voltage u Ar (k)、u Br (k)、u Cr (k) Load current i in alpha-beta two-phase stationary coordinate system αr (k)、i βr (k) Three-phase load current i Ar (k)、i Br (k)、i Cr (k) The calculation formula is as follows:
Figure GDA0003461846090000082
wherein i is αr (k) And i βr (k) By three-phase load current i Ar (k)、i Br (k)、i Cr (k) Obtained through 3s-2s coordinate transformation, and the calculation formula is as follows:
Figure GDA0003461846090000083
to sum up, three phases P H The duty cycle may be obtained by:
Figure GDA0003461846090000084
to sum up, three phases P L The duty cycle may be obtained by:
Figure GDA0003461846090000091
the output end of the duty ratio calculation module is connected with the input end of the duty ratio selection module.
Step four, according to the given value u of the three-phase load voltage Ar * (k)、u Br * (k)、u Cr * (k) And feedback value u Ar (k)、u Br (k)、u Cr (k) Between themThe duty ratio selection module is in three phases P H Duty cycle and three-phase P L Of the duty ratios, three-phase duty ratio D of the next switching cycle is selected A (k+1)、D B (k+1)、D C (k+1)；
Three-phase duty ratio D of next switching period in step four A (k+1)、D B (k+1)、D C The generation rule of (k+1) is as follows:
taking phase A as an example, when phase A load voltage feedback u Ar (k) Less than the phase A load voltage given u Ar * (k) At the time, the A phase duty ratio D of the next switching period A (k+1)＝D AH (k+1); when A phase load voltage feedback u Ar (k) Greater than phase A load voltage given u Ar * (k) At the time, the A phase duty ratio D of the next switching period A (k+1)＝D AL (k+1)。
Duty ratio D of next switching cycle in B and C phase stationary coordinate system B (k+1) and D C The (k+1) generation rule is the same as the stationary coordinate system of phase A.
Step five, the selected three-phase duty ratio generates a three-phase control signal S of the three-phase full-bridge inverter through the DPWM and the driving module A (k+1)、S B (k+1)、S C (k+1)) for driving the power devices of the three-phase full-bridge inverter to realize control of three-phase load voltage.
The simulation results of the embodiment adopting the scheme of the invention are shown in fig. 3-18. Fig. 3 is a simulation result of the three-phase full-bridge inverter in the full-load steady-state condition according to the embodiment of the invention. Fig. 4 is a three-phase load voltage THD simulation result of the three-phase full-bridge inverter according to the embodiment of the present invention under different load steady-state conditions. Fig. 5 is a simulation result of the control of the three-phase full-bridge inverter according to the present invention when the input voltage is changed (the dc input voltage is suddenly increased from 300V to 400V at t=0.04 s). Fig. 6 is a simulation result of the present invention controlling the three-phase full-bridge inverter when the load changes (three-phase load power suddenly rises from 600W to 1200W at t=0.04 s). Fig. 7-10 are simulation results of load voltage control performance when the filter parameters are relatively error in the half-load and full-load steady-state conditions according to an embodiment of the present invention. FIG. 11 is a simulation result of the instantaneous value, amplitude and phase of the load voltage with a relative error of 50% for the filter parameters under full load conditions according to an embodiment of the present invention. Fig. 12-15 are simulation results of load voltage control performance when the fundamental angular frequency parameter actual value and the set value (ω=100pi rad/s, corresponding to the frequency parameter set value of 50 Hz) have relative errors under the half-load and full-load steady-state conditions according to the embodiment of the present invention. Fig. 16 is a simulation result of the load voltage instantaneous value, amplitude and phase of the three-phase full-bridge inverter according to the embodiment of the present invention when the fundamental frequency parameter setting value is 50Hz and the actual fundamental frequency is 10 Hz. Fig. 17 is a simulation result of the load voltage instantaneous value, amplitude and phase of the three-phase full-bridge inverter according to the embodiment of the present invention when the fundamental frequency parameter setting value is 50Hz and the actual fundamental frequency is 100 Hz. Fig. 18 is a simulation result of the transformation frequency conversion control of the three-phase full-bridge inverter according to the embodiment of the invention, wherein the given amplitude of the load voltage is 0-110V and the given frequency is 10-200 Hz when the fundamental frequency parameter setting value is 50 Hz.
The three-phase full-bridge inverter parameters in fig. 1 are as follows: the switching frequency is set at 20KHz, the input direct current voltage is set at 300V, the rated load voltage amplitude is set at 110V, the three-phase rated output power is 1200W, the three-phase output filter inductance L=2.5 mH, the three-phase output filter capacitance C=5.7 uF, and the power electronic simulation software PLECS is adopted to perform time domain simulation on the control method of the embodiment, and the result is as follows: FIG. 3 shows the output of active power P in a full load steady state condition (three phases o =1200W), wherein THD of the three-phase load voltage is respectively: 1.46%, 1.77%, 1.83% and better waveform quality.
FIG. 4 shows the values of the three-phase output active power per unit (P) at different load steady-state conditions (0.1. P.u-1.0.P.u o Three-phase load voltage simulation THD data of =600w-1200W), the THD of the three-phase load voltage is limited below 2.5%, which indicates that the discrete control method of the present invention can obtain three-phase output voltage with higher waveform quality under wide load conditions (10% full load-100% full load).
Fig. 5 shows three-phase load voltage and dc input voltage waveforms when the input dc voltage jumps (the dc input voltage rises from 300V to 400V at t=0.04 s), and the three-phase load voltage has no obvious dynamic process at t=0.04 s, which indicates that the discrete control method of the present invention has good input dynamic performance.
Fig. 6 shows a three-phase load voltage and a three-phase output active power waveform in which the load power jumps (the three-phase load output power rises from 600W to 1200W at t=0.04 s) when the dc input voltage is 300V, and the instantaneous drop value of the load voltage is 25V (which is 23% of the load voltage amplitude), which illustrates that the discrete control method of the present invention has good load dynamic performance.
FIGS. 7-8 show the power at full load steady state conditions (three phase output active power P o =1200W), the load voltage control relative error when the set values of the filter parameters L and C and the actual values have relative errors, wherein: FIG. 7 is a graph showing the relative amplitude error, wherein the absolute amplitude error is 0.006% at maximum; fig. 8 shows a phase relative error, where the absolute value of the phase relative error is 2.1 degrees, and illustrates that the discrete control method of the present invention has better fault tolerance, i.e. robustness, for setting the filtering parameters.
FIGS. 9-10 show the output of active power P in half-load steady-state conditions (three phases o =600W), the load voltage control relative error when the set values of the filter parameters L and C and the actual values have relative errors, wherein: FIG. 9 is a graph of amplitude relative error, with an absolute amplitude error maximum of 0.0089%; fig. 10 shows a phase relative error, where the absolute value of the phase relative error is 1.06 degrees, and illustrates that the discrete control method of the present invention has better fault tolerance, i.e. robustness, for setting the filtering parameters.
The simulation results of fig. 7-10 are combined to illustrate that the discrete control method of the present invention does not need to perform parameter identification on the filtering parameters L and C.
FIG. 11 shows the output of active power P in a full load steady state condition (three phases o =1200W), simulation results of the relative error of the set values of the filter parameters L and C to the actual values being 50%: the three-phase load voltage waveform has good quality, and THD values are respectively: 2.12%, 2.25% and 2.05%; load voltage amplitude baseThe present stability is at a given value of 110V.
FIGS. 12-13 show the power at full load steady state conditions (three phase output active power P o =1200w), a load voltage control error when a relative error occurs between an actual value and a set value (set to ω=100pi rad/s in the embodiment of the present invention), wherein: FIG. 12 is a graph of amplitude relative error, with an absolute amplitude error maximum of 0.0092%; fig. 13 shows a phase relative error, wherein the maximum value of the absolute value of the phase relative error is 3 degrees, which illustrates that the discrete control method of the present invention has good fault tolerance, i.e. robustness, for setting the fundamental wave angular frequency parameter.
FIGS. 14-15 show the output of active power P in half-load steady-state conditions (three phases o Actual value and set value of fundamental angular frequency parameter (the embodiment of the invention is set as P o =1200W) a load voltage control error when a relative error occurs, wherein: FIG. 14 is a graph showing the relative amplitude error, wherein the absolute amplitude error is 0.009% at the maximum; fig. 15 shows the relative phase error, the absolute value of the relative phase error is 1.51 degrees, which illustrates that the discrete control method of the present invention has good fault tolerance, i.e. robustness, for setting the fundamental wave angular frequency parameter.
The simulation results of fig. 12-15 are combined to show that the discrete control method of the invention does not need to perform parameter identification work on fundamental wave angular frequency parameter omega, and can realize broadband control, namely variable frequency control, within a certain frequency range.
FIG. 16 shows the output of active power P in a full load steady state condition (three phases o =1200W), the fundamental frequency parameter set value is 50Hz, the actual value is the simulation result at 10Hz, wherein: the waveform quality of the three-phase load voltage is good; the load voltage amplitude is substantially stable at a given value of 110V.
FIG. 17 shows the output of active power P in a full load steady state condition (three phases o =600W), the fundamental frequency parameter set value is 50Hz, the actual value is simulation result at 100Hz, wherein: the waveform quality of the three-phase load voltage is good; the load voltage amplitude is substantially stable at a given value of 110V.
FIG. 18 shows simulation results of variable-voltage variable-frequency control with a load voltage amplitude of 0-110V ramp and a frequency of 10-220Hz ramp given at a fundamental frequency parameter set value of 50 Hz. The waveform quality of the three-phase load voltage is good; according to the amplitude waveform of the load voltage, the device can realize the dynamic voltage regulation process within the range of 0-110V; according to the phase waveform of the load voltage, the device can realize the dynamic frequency modulation process within the range of 10-220 Hz. The discrete control method can realize variable-voltage variable-frequency control, and can be used for three-phase inversion alternating-current power supply occasions requiring variable-voltage variable-frequency control.

Claims (2)

1. The discrete control method of the three-phase full-bridge inverter is characterized by comprising the following steps of:
s1, setting the setting values of the three-phase load voltages of the kth switching period to be u respectively Ar * (k)、u Br * (k)、u Cr * (k)；
S2, at the starting time t of the kth switching period, obtaining the input bus capacitor voltage u of the three-phase full-bridge inverter through sampling dc (t), three-phase load voltage u Ar (t)、u Br (t)、u Cr (t) three-phase load current i Ar (t)、i Br (t)、i Cr (t) signal, and the signal is converted by A/D conversion module to obtain correspondent digital signal u dc (k)、u Ar (k)、u Br (k)、u Cr (k)、i Ar (k)、i Br (k)、i Cr (k)；
S3, according to the data of the steps S1 and S2, calculating the three-phase P of the next switching period H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1), specifically:
Figure FDA0004123696260000011
Figure FDA0004123696260000012
wherein the method comprises the steps of
Figure FDA0004123696260000013
Figure FDA0004123696260000014
Omega is the angular frequency of a three-phase fundamental wave of the system, L and C are LC filtering parameters of a three-phase full-bridge inverter, and load current i αr (k) And i βr (k) By three-phase load current i Ar (k)、i Br (k)、i Cr (k) Obtained through 3s-2s coordinate transformation:
Figure FDA0004123696260000021
s4, u is Ar (k)、u Br (k)、u Cr (k) And u is equal to Ar * (k)、u Br * (k)、u Cr * (k) Comparing, according to the control rule, the three phases P generated in step S3 H Three-phase P L Three-phase duty ratio D of the next switching cycle is selected and generated from the duty ratios A (k+1)、D B (k+1)、D C (k+1); the control rule is as follows: when the A-phase load voltage feedback value u Ar (k) Less than the phase A load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AH (k+1); when the A-phase load voltage feedback value u Ar (k) Greater than phase a load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AL (k+1), final duty cycle D in stationary coordinate systems of B and C phases B (k+1) and D C (k+1) generating rules as same as the A-phase stationary coordinate system;
s5, according to the obtained three-phase duty ratio D A (k+1)、D B (k+1)、D C (k+1) rawTo form corresponding three-phase control signal S A (k+1)、S B (k+1)、S C (k+1) realizing control of three-phase load voltage by controlling the three-phase full-bridge inverter power device.
2. The discrete control device of the three-phase full-bridge inverter comprises the three-phase full-bridge inverter and is characterized by further comprising a discrete controller, wherein the discrete controller comprises a voltage and current sampling and A/D conversion module, a duty ratio calculation module, a duty ratio selection module, a DPWM and a driving module;
the voltage and current sampling and A/D conversion module is used for sampling the input bus capacitor voltage u of the three-phase full-bridge inverter at the starting time of the kth switching period dc (t) three-phase load voltage u Ar (t)、u Br (t)、u Cr (t) three-phase load current i Ar (t)、i Br (t)、i Cr (t) signal, and the signal is converted by A/D conversion module to obtain correspondent digital signal u dc (k)、u Ar (k)、u Br (k)、u Cr (k)、i Ar (k)、i Br (k)、i Cr (k) Outputting the obtained digital signal to a duty ratio calculation module;
the duty ratio calculation module is used for calculating the three-phase P of the next switching period according to the received data H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1), specifically:
Figure FDA0004123696260000022
Figure FDA0004123696260000031
wherein the method comprises the steps of
Figure FDA0004123696260000032
Figure FDA0004123696260000033
u Ar * (k)、u Br * (k)、u Cr * (k) The third switching period is the set value of the three-phase load voltage of the kth switching period, the omega is the angular frequency of the three-phase fundamental wave of the system, L and C are LC filtering parameters of the three-phase full-bridge inverter, and the load current i αr (k) And i βr (k) By three-phase load current i Ar (k)、i Br (k)、i Cr (k) Obtained through 3s-2s coordinate transformation:
Figure FDA0004123696260000034
the duty ratio calculation module obtains three phases P H Duty cycle D AH (k+1)、D BH (k+1)、D CH (k+1) and three phases P L Duty cycle D AL (k+1)、D BL (k+1)、D CL (k+1) output to the duty cycle selection module;
the duty ratio selection module selects the duty ratio according to the set value u Ar * (k)、u Br * (k)、u Cr * (k) And feedback value u Ar (k)、u Br (k)、u Cr (k) The relationship between the three phases P H Duty cycle and three-phase P L Of the duty ratios, three-phase duty ratio D of the next switching cycle is selected A (k+1)、D B (k+1)、D C (k+1), specifically: when the A-phase load voltage feedback value u Ar (k) Less than the phase A load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AH (k+1); when the A-phase load voltage feedback value u Ar (k) Greater than phase a load voltage set point u Ar * (k) At the next switching period, the A phase duty ratio is D A (k+1)＝D AL (k+1), final duty cycle D in stationary coordinate systems of B and C phases B (k+1) and D C (k+1) generating rules as same as the A-phase stationary coordinate system; the duty ratio selection module outputs the obtained three-phase duty ratio to the DPWM and the driving module;
the DPWM and the driving module generate a three-phase control signal S of the three-phase full-bridge inverter according to the received three-phase duty ratio A (k+1)、S B (k+1)、S C And (k+1) is used for driving the power devices of the three-phase full-bridge inverter so as to realize the control of the three-phase load voltage of the converter.
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