CN113630009A

CN113630009A - High-performance non-isolated bidirectional direct current converter and control method thereof

Info

Publication number: CN113630009A
Application number: CN202111088954.9A
Authority: CN
Inventors: 秦岭; 饶家齐; 张雷; 刘宇涵; 周磊; 王亚芳
Original assignee: Nantong University
Current assignee: Nantong University
Priority date: 2021-09-16
Filing date: 2021-09-16
Publication date: 2021-11-09
Anticipated expiration: 2041-09-16
Also published as: CN113630009B

Abstract

The invention discloses a high-performance non-isolated bidirectional direct current converter and a control method thereof.A positive electrode of a low-voltage direct current power supply is connected with one end of a first inductor and a positive electrode of a fourth capacitor; the other end of the first inductor is connected with the drain electrode of the first switch tube, the source electrode of the second switch tube and the cathode of the second capacitor; the source electrode of the first switching tube is connected with the negative electrode of the low-voltage direct-current power supply, the negative electrode of the fourth capacitor, the negative electrode of the first capacitor, the negative electrode of the third capacitor and the negative electrode of the high-voltage direct-current power supply; the drain electrode of the second switch tube is connected with one end of the second inductor and the anode of the first capacitor; the positive electrode of the second capacitor is connected with the other end of the second inductor and the source electrode of the third switching tube; the drain electrode of the third switching tube is connected with the anode of the third capacitor and the anode of the high-voltage direct-current power supply; the control method is that the driving signal of the second switching tube is the same as the driving signal of the third switching tube and is complementary with the driving signal of the first switching tube.

Description

High-performance non-isolated bidirectional direct current converter and control method thereof

Technical Field

The invention belongs to the technical field of bidirectional DC/DC converters, and particularly relates to a high-performance non-isolated bidirectional DC converter and a control method thereof.

Background

The bidirectional DC/DC converter can serve as an interface between a DC bus and an energy storage device, control energy flow, and provide a desired voltage level, and is widely used in various fields such as fuel cells, photovoltaic power generation systems, uninterruptible power supplies, and electric vehicles in recent years. Bidirectional DC/DC converters are mainly classified into two main categories: isolated and non-isolated. In the application without electrical isolation, the non-isolated DC/DC converter has the advantages of simple design and structure, low loss and small volume.

The bidirectional buck/boost converter has fewer passive and active elements and is a non-isolated bidirectional DC/DC converter with the simplest structure. However, it has limited boosting capability in boost mode. Therefore, in recent years, many researchers have proposed various bidirectional buck/boost converters with high boost capability, which mostly have multiple inductors and capacitors, and thus have large volume, low power density and high cost. Increasing the switching frequency improves power density and dynamic response, however, switching losses increase and conversion efficiency decreases. These problems can be overcome by introducing soft switching techniques. There are various soft switching high gain non-isolated bidirectional DC/DC converters. These topologies have the following problems in common: (1) the number of devices is large, and the structure is complex; (2) part of power tubes still work in a hard switching state, so that the efficiency is difficult to further improve; (3) the power tube has higher voltage stress, and a high-voltage-resistant semiconductor device is required, so that the on-state loss is larger, and the cost is higher; (4) the input and output are not in common, resulting in a complex voltage sampling circuit and also causing EMI problems.

Disclosure of Invention

In view of the above, the present invention aims to provide a high-performance non-isolated bidirectional dc converter and a control method thereof, wherein the high-performance non-isolated bidirectional dc converter has a simple structure, a small number of devices, a low cost, a high step-up/step-down capability, all switching tubes capable of realizing soft switching, a high conversion efficiency, a low voltage stress, and is suitable for occasions such as an uninterruptible power supply system, a photovoltaic power generation system, a fuel cell, an electric vehicle, and the like.

In order to achieve the purpose, the technical scheme provided by the invention is as follows:

a high-performance non-isolated bidirectional DC converter comprises a low-voltage side DC power supply U_LHigh-voltage side DC power supply U_HA first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄A first inductor L₁A second inductor L₂A first switch tube S₁A second switch tube S₂A third switch tube S₃；

The low-voltage side DC power supply U_LAnd the first inductor L₁One terminal of, the fourth capacitance C₄The positive electrode of (1) is connected;

the first inductor L₁And the other end of the first switch tube S₁Drain electrode of (1), the second switching tube S₂Source electrode of, the second capacitor C₂The negative electrode of (1) is connected;

the first switch tube S₁Source electrode of and the low-voltage side direct current power supply U_LNegative pole of (1), the fourth capacitor C₄Negative pole of (1), the first capacitor C₁Negative pole of (2), the third capacitor C₃Negative pole of the power supply, the high-voltage side direct current power supply U_HThe negative electrode of (1) is connected;

the second switch tube S₂And the second inductor L₂One terminal of, the first capacitor C₁The positive electrode of (1) is connected;

the second capacitor C₂And the second inductor L₂The other end of the third switching tube S₃Is connected to the source of (a);

the third switch tube S₃And the third capacitor C₃The positive electrode and the high-voltage side direct-current power supply U_HThe positive electrode of (1) is connected;

wherein the first switch tube S₁The second switch tube S₂And the third switching tube S₃All the metal oxide semiconductor field effect transistors are provided with reverse parallel diodes;

the first inductor L₁The inductance value of (a) satisfies:

in the above formula, L₁Is a first inductance L₁Inductance value of, U_LIs a low-side DC supply voltage, U_HIs a high-side DC supply voltage, I_L,maxIs the maximum average value of the current on the low-voltage side, f_sDelta% is the first inductance L for the switching frequency₁Allowable maximum current ripple and first inductor L₁A percentage of maximum average current;

the second inductor L₂The inductance value of (a) satisfies:

in the above formula, L₂Is a second inductance L₂Inductance value of (1)_H,maxIs the maximum average value of the high side current;

the invention also provides a control method of the high-performance non-isolated bidirectional direct current converter, which is used for performing double closed-loop control on the voltage of the high-voltage side and the current of the low-voltage side, and specifically comprises the following steps:

for high-voltage side DC power supply U_HThe terminal voltage is sampled to obtain a high-voltage side voltage sampling value u_H,fSampling value u of the voltage on the high voltage side_H,fAnd a high-side voltage reference value u_H,refComparing to obtain a first error signal;

the first error signal passes through a voltage controller G in sequence_uH(s) and an amplitude limiting link Lim are processed to obtain a low-voltage side current reference value i_L,ref；

For the first inductance L₁The current is sampled to obtain a low-voltage side current sampling value i_L,fSampling the current of the low voltage side i_L,fAnd a low-voltage side current reference value i_L,refComparing to obtain a second error signal, and passing the second error signal through a current controller G_iL(s) processing with a unipolar triangular carrier u_cCrossing to generate a first switch tube S₁The PWM driving signal of (1);

opening the first openingThe PWM driving signal of the switch tube S1 is inverted to obtain a second switch tube S₂And a third switching tube S₃The drive signal of (1).

Preferably, the ideal voltage gain in the boost mode of the high-performance non-isolated bidirectional direct current converter is (1+ D)₁)/(1-D₁) The ideal voltage gain in buck mode is D₂/(2-D₂) Wherein D is₁Is a first switch tube S₁Duty ratio of the drive signal of D₂Is a second switch tube S₂And a third switching tube S₃The duty cycle of the drive signal.

Preferably, the voltage stress of the first switching tube, the voltage stress of the second switching tube and the voltage stress of the third switching tube are all (U)_H+U_L)/2。

Preferably, the first inductor L₁And a second inductance L₂Are operated in current continuous mode.

Compared with the prior art, the high-performance non-isolated bidirectional direct current converter provided by the invention has strong voltage rising/reducing capacity; has less device number and lower cost, and the second inductor L₂The current is changed in a bidirectional linear mode, the required inductance value is small, the size of the system is further reduced, and the power density of the system is improved; the voltage stress of the first switch tube, the second switch tube and the third switch tube is (U)_H+U_L) A low-voltage-resistant device can be adopted, so that the system cost and the loss are reduced; ZVS soft switching is realized for all switching tubes, and the conversion efficiency is high; the input and the output are connected to the ground, and the sampling circuit has a simple structure.

Drawings

Fig. 1 is a schematic circuit diagram of a high-performance non-isolated bidirectional dc converter according to the present invention;

FIG. 2 is a control block diagram of a high performance non-isolated bi-directional DC converter provided by the present invention;

fig. 3(a) -3 (f) are equivalent diagrams of 6 operating modes of the converter shown in fig. 1 in a boost mode during a switching period;

4(a) -4 (f) are equivalent diagrams of 6 operating modes of the converter shown in FIG. 1 in a buck mode within one switching period;

FIG. 5(a) shows the converter of FIG. 1 operating in boost mode for a switching period T_sThe main working waveform diagram in the inner part;

FIG. 5(b) shows the converter of FIG. 1 operating in buck mode for a switching period T_sThe main working waveform diagram in the inner part;

FIG. 6(a) is an equivalent circuit schematic of the average current of the converter of FIG. 1 when operating in boost mode;

FIG. 6(b) is an equivalent circuit schematic diagram of the average current of the converter shown in FIG. 1 when operating in buck mode;

fig. 7(a) -7 (e) are simulated waveforms of the converter shown in fig. 1 in the boost mode. Fig. 7(a) is a simulated waveform diagram of driving signals of the first switching tube, the second switching tube, and the third switching tube, low-voltage side current, and first inductor and second inductor current; FIG. 7(b) is a simulated waveform diagram of the low-side power supply voltage, the high-side power supply voltage, and the voltages at the two ends of the first capacitor and the second capacitor; fig. 7(c) - (e) are simulation waveforms of soft switching implementation of the first switching tube, the second switching tube, and the third switching tube, respectively.

Fig. 8(a) -8 (e) are simulation waveforms of the converter shown in fig. 1 in buck mode. The specific distribution is the same as the simulation oscillogram in the boost mode.

Fig. 9 is a schematic diagram of a dynamic adjustment process when the converter shown in fig. 1 is switched from the boost mode to the buck mode.

Detailed Description

The technical solutions in the embodiments of the present application will be described clearly and completely with reference to the drawings in the embodiments of the present application, and it is obvious that the described embodiments are only a part of the embodiments of the present application, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.

The invention provides a high-performance non-isolated bidirectional direct current converter and a control method thereof, and the circuit structure is shown in figure 1. It comprises a low-voltage side direct current power supply U_LHigh voltage side straightCurrent source U_HA first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄A first inductor L₁A second inductor L₂A first switch tube S₁A second switch tube S₂A third switch tube S₃；

Low-voltage direct-current power supply U_LPositive pole and first inductance L₁One terminal of (1), a fourth capacitor C₄The positive electrode of (1) is connected;

first inductance L₁And the other end of the first switch tube S₁Drain electrode of the first switching tube S₂Source electrode of the first capacitor C₂The negative electrode of (1) is connected;

first switch tube S₁Source and low-voltage side DC power supply U_LNegative electrode of (1), fourth capacitor C₄Negative electrode of (1), first capacitor C₁Negative electrode of (1), third capacitor C₃Negative electrode, high voltage side DC power supply U_HThe negative electrode of (1) is connected;

a second switch tube S₂Drain electrode of and second inductor L₂One terminal of (1), a first capacitor C₁The positive electrode of (1) is connected;

second capacitor C₂Positive pole and second inductance L₂The other end of the third switch tube S₃Is connected to the source of (a);

third switch tube S₃And the third capacitor C₃The positive electrode and the high-voltage side direct-current power supply U_HThe positive electrode of (1) is connected;

in this embodiment, the first switch tube S₁The second switch tube S₂And the third switching tube S₃Are metal-oxide-semiconductor field effect transistors (MOSFETs) with antiparallel diodes.

First inductance L₁The inductance value of (a) satisfies:

in the above formula, L₁Is a first inductance L₁Inductance value of, U_LIs a low-side DC supply voltage, U_HIs a high-voltage side direct currentSupply voltage, I_L,maxIs the maximum average value of the current on the low-voltage side, f_sDelta% is the first inductance L for the switching frequency₁Allowable maximum current ripple and first inductor L₁A percentage of maximum average current;

the second inductor L₂The inductance value of (a) satisfies:

in this embodiment, the control method of the high-performance non-isolated bidirectional dc converter includes: the voltage on the high-voltage side and the current on the low-voltage side are subjected to double closed-loop control, and the control block diagram is shown in fig. 2.

Sampling value u of high-voltage side voltage_H,fWith reference value u of the high-side voltage_H,refComparing, the error signal is sent to the voltage controller G_uH(s) and a voltage controller G_uHAfter the output signal of(s) passes through an amplitude limiting link Lim, a reference value i of the low-voltage side current is obtained_L,ref(ii) a Sampling value i of low-voltage side current_L,fWith reference value i of the low-side current_L,refComparing, and sending the error signal to the current controller G_iL(s)；

Current controller G_iL(s) output signal and unipolar triangular carrier u_cIntercept to generate a first switch tube S₁The PWM driving signal of (1); the driving signal is inverted to control the second switch tube S₂And a third switching tube S₃。

The operation of the high performance non-isolated bi-directional dc converter shown in fig. 1 is described below.

To simplify the analysis, the following assumptions were made: first switch tube S₁A second switch tube S₂A third switch tube S₃A first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄A first inductor L₁A second inductor L₂Are all ideal devices; a first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄Large enough that voltage ripple is negligible; low-voltage side DC power supply U_LThe negative end is a zero potential reference point; power tube S₁、S₂、S₃Respectively of body diodes D_S1、D_S2、D_S3(ii) a Power tube S₁、S₂、S₃Respectively is C_S1、C_S2、C_S3。

Based on the above assumptions, after entering the steady state, the working process of the high-performance non-isolated bidirectional dc converter of the present invention in a switching period in the boost mode can be divided into 6 modes. The equivalent circuits of the modes are shown in fig. 3(a) to 3 (f). The main waveforms in one switching cycle are shown in fig. 5 (a).

The following are distinguished:

t₀before the moment, the potential of the point a is 0, D_S1And conducting. L is₁And L₂Subject to a forward voltage (U respectively)_LAnd U_C1-U_C2) First inductor current i_L1Linearly increasing in the forward direction, second inductor current i_L2The inverse linearity decreases.

(1) Mode 1, t₀～t₁Stage (2): t is t₀At the moment, zero voltage turns on S₁The equivalent circuit is shown in fig. 3 (a); i.e. i_L1And i_L2Maintain the original slope to continue changing i_L2The reverse linearity is reduced to 0 and then the forward linearity is increased, and the expression is as follows:

in the formula of U_LIs a low-side DC supply voltage, U_C1And U_C2Are respectively a first capacitor C₁And a second capacitor C₂The terminal voltage of (c).

(2) Mode 2, t₁～t₂Stage (2): t is t₁At time, turn off S₁The equivalent circuit is shown in fig. 3 (b). i.e. i_L1And i_L2All of the partial currents of (1) flow into a node a, which is C_S1Charging and pumping away C_S2A charge on; i.e. i_L2Flows into node b, and draws off C_S3The charge on the substrate. The potential at the point a is continuously increased from 0 and the potential at the point b is increased from U_C2And is rising continuously. The modal duration is short, approximately considering the first inductor current i_L1And a second inductor current i_L2Remain unchanged.

(3) Mode 3, t₂～t₃Stage (2): t is t₂Time of day, C_S1、C_S2、C_S3After the completion of charging and discharging, the potential at the point a rises to U_C1B point potential rises to U_C1+U_C2，D_S2And D_S3Are all conducted in the forward direction, S₂、S₃The terminal voltage drops to 0, and the equivalent circuit is shown in fig. 3 (c). First inductance L₁A second inductor L₂Respectively bear reverse voltage U_C1-U_L、U_C2，i_L1、i_L2The forward linearity decreases, and the expression is:

(4) mode 4, t₃～t₄：t₃At the moment, zero voltage turns on S₂、S₃The equivalent circuit is shown in fig. 3 (d). i.e. i_L1、i_L2All keep the original linear change of slope i_L2The forward linearity decreases to 0 before the reverse linearity increases.

(5) Mode 5, t₄～t₅：t₄At time, turn off S₂The equivalent circuit is shown in fig. 3 (e). i.e. i_L1And i_L2All the partial currents flow out of the node a, and C is pumped away_S1Of a charge of C_S2Charging; i.e. i_L2The other part of the current flowing out of the node b is C_S3And (6) charging. The potentials of the point a and the point b are gradually reduced. The modal duration is short, approximately considered as the first inductor current i_L1And a second inductor current i_L2Remain unchanged.

(6) Mode 6, t₅～t₆：t₅Time of day, C_S1、C_S2、C_S3After the charging and discharging are completed, the potential drop at the point a is 0, D_S1Conduction, S₁The terminal voltage drops to 0, and the equivalent circuit is shown in fig. 3 (f). L is₁、L₂Respectively bear forward voltage U_LAnd U_C1-U_C2First inductor current i_L1Linearly increasing in the forward direction, second inductor current i_L2The inverse linearity decreases. t is t₆At the moment, zero voltage turns on S₁Mode 6 ends and the next switching cycle is entered.

Based on the above operating principle, the steady-state characteristics of the converter of the present invention in boost mode are analyzed below.

From the volt-second balance of the first inductance and the second inductance, we can obtain:

in addition, as shown in fig. 3(d) (mode 4 equivalent circuit diagram):

U_C1+U_C2＝U_H (4)

according to the equations (3) and (4), the ideal voltage gain G of the converter in boost mode can be obtained as follows:

the voltage stress of the first capacitor and the second capacitor is as follows:

the voltage stress of the power tube is as follows:

after entering a steady state, the first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄The average current of (a) is zero, so that an equivalent circuit diagram of the average current in the boost mode can be obtained, which can be obtained from fig. 6 (a):

in the above formula, I_L1Is a first inductance L₁Average current value of (1)_L2Is a second inductance L₂Average current value of (1)_S1Is a first switch tube S₁Average current value of (1)_S2Is a second switch tube S₂Average current value of (1)_LIs the average value of the current on the low-voltage side, I_HIs the average value of the high side current.

The working process of the high-performance non-isolated bidirectional direct-current converter in a buck mode in one switching period can be divided into 6 modes. The equivalent circuits of the modes are shown in fig. 4(a) to 4 (f). The main waveform in one switching cycle is shown in fig. 5 (b).

The following are distinguished:

t₀before the moment, the potential of the point a is 0, D_S1And conducting. L is₁And L₂Subject to a forward voltage (U respectively)_LAnd U_C1-U_C2) First inductor current i_L1A first inductor current i_L2The inverse linearity decreases.

(1) Mode 1, t₀～t₁Stage (2): t is t₀At the moment, zero voltage turns on S₁The equivalent circuit is shown in fig. 4 (a); i.e. i_L1And i_L2Maintain the original slope to continue changing i_L2The reverse linearity is reduced to 0 and then the forward linearity is increased, and the expression is as follows:

in the formula of U_LIs low voltage side straightVoltage of a current source, U_C1And U_C2Are respectively a first capacitor C₁And a second C₂The terminal voltage of (c).

(2) Mode 2, t₁～t₂Stage (2): t is t₁At time, turn off S₁The equivalent circuit is shown in FIG. 4 (b). I is_L1And i_L2All of the partial currents of (1) flow into a node a, which is C_S1Charging and pumping away C_S2A charge on; i.e. i_L2Flows into node b, and draws off C_S3The charge on the substrate. The potential at the point a is continuously increased from 0 and the potential at the point b is increased from U_C2And is rising continuously. The modal duration is short, approximately considering the first inductor current i_L1And a second inductor current i_L2Remain unchanged.

(3) Mode 3, t₂～t₃Stage (2): t is t₂Time of day, C_S1、C_S2、C_S3After the completion of charging and discharging, the potential at the point a rises to U_C1B point potential rises to U_C1+U_C2，D_S2And D_S3Are all conducted in the forward direction, S₂、S₃The terminal voltage drops to 0, and the equivalent circuit is shown in fig. 4 (c). First inductance L₁A second inductor L₂Respectively bear reverse voltage U_C1-U_L、U_C2，i_L1Inverse linear increase, i_L2The forward linearity decreases, and the expression is:

(4) mode 4, t₃～t₄Stage (2): t is t₃At the moment, zero voltage turns on S₂、S₃The equivalent circuit is shown in fig. 4 (d). i.e. i_L1、i_L2All keep the original linear change of slope i_L2The forward linearity decreases to 0 before the reverse linearity increases.

(5) Mode 5, t₄～t₅Stage (2): t is t₄At time, turn off S₂The equivalent circuit is shown in fig. 4 (e). i.e. i_L1And i_L2All the partial currents flow out of the node a, and C is pumped away_S1Of a charge of C_S2Charging; i.e. i_L2The other part of the current flowing out of the node b is C_S3And (6) charging. The potentials of the point a and the point b are gradually reduced. The modal duration is short, approximately considered as the first inductor current i_L1And a second inductor current i_L2Remain unchanged.

(6) Mode 6, t₅～t₆Stage (2): t is t₅Time of day, C_S1、C_S2、C_S3After the charging and discharging are completed, the potential drop at the point a is 0, D_S1Conduction, S₁The terminal voltage drops to 0, and the equivalent circuit is shown in fig. 4 (f). L is₁、L₂Respectively bear forward voltage U_LAnd U_C1-U_C2First inductor current i_L1A second inductor current i_L2The inverse linearity decreases. t is t₆At the moment, zero voltage turns on S₁Mode 6 ends and the next switching cycle is entered.

Based on the above working principle, the steady-state characteristics of the converter of the present invention when operating in buck mode are analyzed below.

According to the volt-second balance of the currents of the first inductor and the second inductor, the following results are obtained:

in addition, as shown in fig. 4(d) (mode 4 equivalent circuit diagram):

U_C1+U_C2＝U_H (12)

according to the equations (11) and (12), the ideal voltage gain G of the converter in buck mode can be obtained as follows:

the voltage stress of the power tube is as follows:

after entering a steady state, the first capacitor C₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄The average current of (a) is zero, so that an equivalent circuit diagram of the average current in buck mode can be obtained, as shown in fig. 6 (b). It can be seen that in buck mode, the first inductor L₁A second inductor L₂First switch tube S₁A second switch tube S₂The average current of (2) satisfies the formula (8).

The voltage stress of the first capacitor and the voltage stress of the second capacitor in both the boost mode and the buck mode of the converter can meet the following conditions:

the voltage stress of the power tube satisfies the following conditions:

first inductance L₁A second inductor L₂First switch tube S₁A second switch tube S₂The average current of (2) satisfies the formula (8).

The ZVS soft switching condition of the present invention is discussed below.

When the converter works in boost mode, the first switch tube S₁The key to achieving ZVS switching on is to ensure that during mode 5, there are:

i_L1(t)+i_L2(t)＜0，t∈(t₄,t₅) (18)

only then can the current be supplied to the first switching tube S respectively₁A second switch tube S₂A third switch tube S₃Parasitic capacitance C of_S1、C_S2And C_S3Charging and discharging are carried out so that the switching tube S₁The voltage across the terminals gradually decreases to 0.

In the same way, the second switch tube S₂A third switch tube S₃The key to realizing ZVS switching-on is to ensure that in the mode 2 process:

i_L1(t)+i_L2(t)＞0，t∈(t₁,t₂) (19)

obviously, this condition is always satisfied.

Similarly, when the converter operates in buck mode, the second switch tube S₂A third switch tube S₃The key to realizing ZVS switching-on is to ensure that in the mode 2 process:

i_L1(t)+i_L2(t)＞0，t∈(t₁,t₂) (20)

first switch tube S₁The key to achieving ZVS switching on is to ensure that during mode 5, there are:

i_L1(t)+i_L2(t)＜0，t∈(t₄,t₅) (21)

because of T_ch2<<T_s，T_ch5<<T_s(T_ch2And T_ch5Duration of mode 2 and mode 5, respectively, T_sOne switching cycle duration) for simplicity of analysis, the first inductor current i during charging and discharging may be assumed_L1And a second inductor current i_L2Is kept constant, in boost mode there are:

similarly, in buck mode there are:

namely, the ZVS soft switching conditions are:

wherein, I_LIs the average value of the current on the low-voltage side, I_HIs the average value of the high side current, Δ I_L1Is a first inductance L₁Current pulsation amount of, Δ I_L2Is a second inductance L₂The amount of current ripple.

The high gain converter of the present invention is designed with the following parameters.

The design criteria of the converter are: switching frequency f_s100kHz, low side DC supply voltage U_L48V, high-side DC power supply voltage U_H400V, maximum output power P_o,max＝250W。

The duty ratio D of the converter can be obtained from the formula (5) and the formula (13) according to the index₁、D₂Comprises the following steps:

if the first inductance L₁Current pulsation amount Δ I of_L1Not exceeding its maximum average current I_L1,max30% of (I), i.e. Δ I_L1≤0.3I_L1,maxThen, there are:

because of Δ I_L1≤0.3I_L1,maxFrom equation (26), sufficient conditions for ZVS soft switching can be obtained:

namely:

wherein, I_H,maxIs the maximum average value of the high side current, I_L,maxThe maximum average value of the low side current.

Based on the above modal analysis, operating condition analysis and parameter design of the transducer of the present invention, it was verified by simulation using Saber simulation software as follows:

the steady-state characteristics of the converter provided by the invention are verified through open-loop simulation, and the specific technical indexes and circuit parameters are as follows: u shape_L＝48V，U_H＝400V，f_s＝100kHz，P_o,max250W; a first capacitor C₁47 muF, second capacitance C₂47 muF, third capacitance C₃4.7 muF, fourth capacitance C₄22 μ F; first inductance L₁250uH, second inductance L₂25 uH; theoretical voltage gain G ═ 8.33.

Fig. 7(a) -7 (e) are simulated waveforms of the converter of fig. 1 operating in boost mode.

As can be seen from FIG. 7(a), i_L1Successive, i_L2Positive and negative alternation, illustrating the first inductance L₁And a second inductance L₂In CCM and BCM, respectively. As can be seen from FIGS. 7(c) - (e), in the case of the driving signal u_gs1、u_gs2And u_gs3Before the positive pressure comes, the first switch tube S₁A second switch tube S₂And a third switching tube S₃Terminal voltage u of_ds1、u_ds2And u_ds3Respectively reduced to zero, which shows that the three realize zero voltage switching-on; in addition, as can be seen from fig. 7(a) and 7(b), when the voltage gain G is 8.33, the duty ratio D is actually measured₁Is 0.796 and the theoretical duty cycle

Almost identical; as can be seen from FIGS. 7(b) to 7(e), the power tube S₁、S₂、S₃The voltage stress is 223.9V, 224.7V and 224.7V respectively, and the first capacitor C₁A second capacitor C₂The voltage stress was 223.8V and 175.9V, respectively, which were substantially consistent with the theoretical values.

Fig. 8(a) -8 (e) are simulation waveforms of the converter shown in fig. 1 when the converter operates in buck mode, and it can be seen that the simulation results in buck mode are also basically consistent with theory, thereby verifying the correctness of theoretical analysis.

And then, performing closed-loop simulation on the voltage at the high-voltage side and the current at the low-voltage side to verify the feasibility of the control scheme provided by the invention, wherein the specific technical indexes and circuit parameters are as follows: u shape_L＝48V，U_H＝400V，f_s＝100kHz，P_o,max250W; a first capacitor C₁47 muF, second capacitance C₂47 muF, third capacitance C₃4.7 muF, fourth capacitance C₄22 μ F; first inductance L₁250uH, second inductance L₂25 uH; theoretical voltage gain G ═ 8.33; the high-voltage side is connected with a resistive load in series by adopting a variable power supply to realize the switching of two modes. The resistive load R is designed to be 32 Ω, and the variable supply voltage is switched from 380V to 420V instantaneously when the simulation proceeds to 100 ms.

FIG. 9 shows the dynamic regulation process of the converter shown in FIG. 1 when switching from boost mode to buck mode, and it can be seen that at 100ms, the first inductor L₁The current changes from positive to negative, the energy flow direction of the converter changes, and the working mode is switched from the boost mode to the buck mode; before and after mode switching, the voltage of the high-voltage side of the converter is stabilized at 400V, which is consistent with the theory, so that the feasibility of the control scheme provided by the invention is verified.

The high-performance non-isolated bidirectional direct current converter provided by the invention has the following advantages: (1) the voltage gain in boost mode is (1+ D)₁)/(1-D₁) Voltage gain in buck mode is D₂/(2-D₂) The pressure rising/reducing capability is strong; (2) the number of power devices is small, and the structure is simple; (3) all power tubes realize ZVS soft switching; (4) the voltage stress of the first switch tube, the voltage stress of the second switch tube and the voltage stress of the third switch tube are equal and are (U)_H+U_L) The voltage stress is lower, and devices with low rated voltage and low cost can be selected; (5) the input and the output are connected to the ground, and the sampling circuit has a simple structure.

It is noted that, herein, relational terms such as first and second, and the like may be used solely to distinguish one entity or action from another entity or action without necessarily requiring or implying any actual such relationship or order between such entities or actions. Also, the terms "comprises," "comprising," or any other variation thereof, are intended to cover a non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements does not include only those elements but may include other elements not expressly listed or inherent to such process, method, article, or apparatus. Without further limitation, an element defined by the phrase "comprising an … …" does not exclude the presence of other identical elements in a process, method, article, or apparatus that comprises the element.

The above description of the embodiments is only intended to facilitate the understanding of the method of the invention and its core idea, and not to limit it. It should be noted that, for those skilled in the art, without departing from the principle of the present invention, several improvements and modifications can be made to the present invention, and these improvements and modifications also fall into the protection scope of the present invention.

Claims

1. A high-performance non-isolated bidirectional DC converter is characterized by comprising a low-voltage side DC power supply U_LHigh-voltage side DC power supply U_HA first capacitorC₁A second capacitor C₂A third capacitor C₃A fourth capacitor C₄A first inductor L₁A second inductor L₂A first switch tube S₁A second switch tube S₂A third switch tube S₃；

the first inductor L₁The inductance value of (a) satisfies:

in the above formula, the first and second carbon atoms are,L₁is a first inductance L₁Inductance value of, U_LIs a low-side DC supply voltage, U_HIs a high-side DC supply voltage, I_L,maxIs the maximum average value of the current on the low-voltage side, f_sDelta% is the first inductance L for the switching frequency₁Allowable maximum current ripple and first inductor L₁A percentage of maximum average current;

the second inductor L₂The inductance value of (a) satisfies:

in the above formula, L₂Is a second inductance L₂Inductance value of (1)_H,maxThe maximum average value of the high side current.

2. A method of controlling a high performance non-isolated bi-directional dc converter according to claim 1, comprising the steps of:

inverting the PWM driving signal of the first switching tube S1 to obtain a second switching tube S₂And a third switching tube S₃The drive signal of (1).

3. The high performance non-isolated bi-directional DC converter according to claim 1, wherein the ideal voltage gain of the converter in boost mode is (1+ D)₁)/(1-D₁) The ideal voltage gain in buck mode is D₂/(2-D₂) Wherein D is₁Is a first switch tube S₁Duty ratio of the drive signal of D₂Is a second switch tube S₂And a third switching tube S₃The duty cycle of the drive signal.

4. The high performance non-isolated bi-directional DC converter according to claim 1, wherein the voltage stress of each of the first, second and third switching tubes is (U)_H+U_L)/2。