CN113594684A - Dual-frequency antenna based on dielectric integrated waveguide - Google Patents

Dual-frequency antenna based on dielectric integrated waveguide Download PDF

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Publication number
CN113594684A
CN113594684A CN202110991238.5A CN202110991238A CN113594684A CN 113594684 A CN113594684 A CN 113594684A CN 202110991238 A CN202110991238 A CN 202110991238A CN 113594684 A CN113594684 A CN 113594684A
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dual
cavity
qmsiw
stub
frequency
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蒋溱
陈国胜
李瑞兵
夏琦
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Shengweilun Shenzhen Communication Technology Co ltd
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Shengweilun Shenzhen Communication Technology Co ltd
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/50—Structural association of antennas with earthing switches, lead-in devices or lightning protectors
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/10—Resonant antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/20—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements characterised by the operating wavebands
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way
    • H01Q5/314—Individual or coupled radiating elements, each element being fed in an unspecified way using frequency dependent circuits or components, e.g. trap circuits or capacitors
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way
    • H01Q5/314—Individual or coupled radiating elements, each element being fed in an unspecified way using frequency dependent circuits or components, e.g. trap circuits or capacitors
    • H01Q5/335—Individual or coupled radiating elements, each element being fed in an unspecified way using frequency dependent circuits or components, e.g. trap circuits or capacitors at the feed, e.g. for impedance matching

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Abstract

The invention provides a dual-frequency antenna based on a dielectric integrated waveguide, which consists of two semicircular HMSIW cavity resonators, wherein each semicircular cavity resonator is realized by adopting a metal through hole, the radius of the through hole and a gap between the through holes are used for reducing the cavity and leakage from the cavity, the top of the antenna is provided with an open metal groove by an etching process to divide the semicircular HMSIW cavity into two different QMSIW cavities, each QMSIW cavity comprises a stub line with a rectangular open circuit, the stub line is connected to a strong field region of the QMSIW cavity resonator, and the dual-frequency antenna also comprises a varactor diode and a bias circuit, the varactor diode is connected with the stub line and is fed by one bias circuit, and the varactor diode is used as a frequency tuning element and is connected with the stub line. By adopting the dual-frequency antenna based on the dielectric integrated waveguide, the design cost of the dual-frequency antenna is low, the dual-frequency antenna has the advantages of low profile, excellent parameters and small volume, and the product competitiveness is strong.

Description

Dual-frequency antenna based on dielectric integrated waveguide
Technical Field
The invention belongs to the technical field of communication, and particularly relates to a double-frequency adjustable antenna structure at Sub-6 GHz.
Background
In the technological path of the rapid development of the new generation of 5G mobile and wireless communication systems, the dual-band or multi-band communication mode has become a technical means for increasing the network communication speed and increasing the network capacity, for example, the new generation of WIFI systems all use two frequency bands of 2.4GHz and 5GHz, and the 5G frequency band of the chinese mobile is also fixed at two frequency bands of 2.6GHz (2.5-2.7 GHz) and 4.9GHz (4.8-5.0 GHz), so the dual-band or multi-band wireless and mobile communication system will be a major trend of the future communication technology.
Chinese patent application No. 2021100625718 proposes a small-size dual-band antenna and communication equipment, this dual-band antenna includes dielectric substrate and sets up the first radiating element on dielectric substrate, the second radiating element, balun structure and feeder line, wherein, the first radiating element radiates the radiation signal of first frequency channel and second frequency channel respectively with the second radiating element, realize dual-band radiation, need not to set up two sets of antennas alone, and simultaneously, the second radiating element is buckled and is set up, reduce the size of dual-band antenna, first radiating element sets up and is the ladder impedance transform structure in the both sides of balun structure, the second radiating element sets up in the both sides of balun structure and both sides are buckled towards the balun structure.
The dual-frequency antenna of the technical scheme has a complex design structure and a narrow frequency range, and still cannot meet the diversified demands of the market.
Therefore, it is necessary to redesign the technical solution of the dual-band antenna to satisfy a wider antenna resonant frequency range.
Disclosure of Invention
The present patent of invention aims to provide a solution for a low-cost, low-profile and small-volume dual-frequency antenna, which is implemented based on a dielectric integrated waveguide and has the function of frequency selection and tuning by direct current.
Aiming at the problems in the prior art, the invention provides a low-cost, low-profile and small-size dual-frequency antenna based on a Quarter-dielectric Integrated Waveguide (QMSIW) cavity technology, and the technical scheme of the invention is as follows:
the invention provides a dual-frequency antenna based on a dielectric integrated waveguide, which consists of two semicircular HMSIW cavity resonators, wherein each semicircular cavity resonator is realized by adopting a metal through hole, the radius of the through hole and a gap between the through holes are used for reducing the cavity and leakage from the cavity, the top of the antenna is provided with an open metal groove by an etching process to divide the semicircular HMSIW cavity into two different QMSIW cavities, each QMSIW cavity comprises a rectangular open stub line which is connected to a strong field region of the QMSIW cavity resonator, and the dual-frequency antenna also comprises a varactor diode and a bias circuit which are connected with the stub line, each varactor diode is fed by one bias circuit, and the varactor diode is used as a frequency tuning element and is connected with the stub line.
Further, each QMSIW cavity is excited by a 50 Ω microstrip line.
Further, the bias circuit includes at least an inductive element and a resistive element.
Further, the QMSIW cavity resonator is equivalent to an RLC circuit, and the input impedance of the equivalent circuit model is calculated by using formula (1) and public expression (2):
Zin =jωLeq (1−ω2LstubCv )/1−ω2Cv (Leq+Lstub ) (1)
ω0=1/(Cv (Leq + Lstub ))1/2 (2)
wherein Z isinIs the input impedance, j is an imaginary number, ω is the angular frequency, ω0Is the resonant frequency, LeqIs the inductance, L, of a QMSIW cavity resonatorstubIs an inductance of the load stub, CvIs the capacitance of the varactor.
Further, the resonant frequency of the QMSIW cavity resonator varies inversely with the length of the stub in three modes of modes 1 to 3, and varies proportionally with the width of the stub.
Further, the resonant frequency of the QMSIW cavity resonator varies inversely proportional to the capacitance value.
Further, the resonant frequency of the QMSIW cavity resonator is inversely proportional to the cavity radius in the modes 1 and 3, and the resonant frequency is directly proportional to the cavity radius in the mode 2, and the specific variation relationship is shown in the formula (3):
(fres) 01=2.404c/2πRcav (
Figure 880154DEST_PATH_IMAGE001
)1/2 (3)
wherein f isresIs the resonant frequency of mode 1, 2.404 is the Bessel function value of mode 1, RcavIs the cavity radius of a circular SIW cavity,
Figure 266136DEST_PATH_IMAGE001
is the relative dielectric constant of the substrate.
Further, the QMSIW cavity resonator has a quality factor that varies inversely with variations in the insertion feed length and width.
Further, the QMSIW cavity resonator has a quality factor that varies in proportion to the stub length and the capacitance of the varactor, the quality factor of the QMSIW cavity resonator being calculated using equation (4):
Qex =f0/f±90（4）
wherein Q isexIs an external quality factor, f0Is the center frequency, f, obtained from the peak of the group delay plot±90Is the center frequency f in the S11 phase0Of ± 90.
Further, the varactor is of the type SMV 1405.
Furthermore, the tuning frequency of the dual-frequency antenna is respectively between two frequency bands of 3.77 GHz-4.59 GHz and 4.96 GHz-6.1 GHz, and the isolation between two resonators in the two frequency bands is less than 21 dB.
By adopting the dual-frequency antenna based on the dielectric integrated waveguide, the design cost of the dual-frequency antenna is low, the dual-frequency antenna has the advantages of low profile, excellent parameters and small volume, and the product competitiveness is strong.
Drawings
FIG. 1: a single unit QMSIW cavity resonator of the present invention is schematically illustrated.
FIG. 2: the invention discloses an equivalent circuit model schematic diagram of a QMSIW cavity resonator.
FIG. 3: the length L of the short section of the QMSIW cavity resonator of the inventionsWaveform diagram for resonant frequency.
FIG. 4: the invention QMSIW cavity resonator has a stub width WsWaveform diagram for resonant frequency.
FIG. 5: the invention QMSIW cavity resonator varactor capacitance CvWaveform diagram for resonant frequency.
FIG. 6: radius R of QMSIW cavity resonator of the inventioncavWaveform diagram for resonant frequency.
FIG. 7: the external quality factor schematic diagram of the dual-frequency antenna of the dielectric integrated waveguide is under different feed parameters.
FIG. 8: the external quality factor of the dual-frequency antenna of the dielectric integrated waveguide is schematically shown under different parameters of the stub and the capacitance value.
FIG. 9: the cross section of the dual-frequency antenna of the dielectric integrated waveguide is schematic.
FIG. 10: the invention discloses a schematic diagram of a dual-frequency antenna packaging structure of a dielectric integrated waveguide.
FIG. 11: the dual-frequency antenna of the dielectric integrated waveguide of the invention has S11 and S22 waveform diagrams at Ls 1.
FIG. 12: the dual-frequency antenna of the dielectric integrated waveguide of the invention has S11 and S22 waveform diagrams at Ls 2.
FIG. 13: the dual-frequency antenna of the dielectric integrated waveguide of the invention has S21 and S22 waveform diagrams at Ls 3.
FIG. 14: the dual-frequency antenna of the dielectric integrated waveguide of the invention has S21 and S22 waveform diagrams at Ls 4.
FIG. 15: the invention discloses a schematic diagram of S parameter waveforms of simulation waveforms and measurement waveforms of a first embodiment of a dual-frequency antenna of a dielectric integrated waveguide.
FIG. 16: the invention discloses a schematic diagram of S parameter waveforms of a simulation waveform and a measurement waveform of a second embodiment of a dual-frequency antenna of a dielectric integrated waveguide.
FIG. 17: the dual-frequency antenna of the dielectric integrated waveguide of the present invention has the simulated and measured directional schematic diagrams (E-plane and H-plane) of the antenna at 0.63pF (30V), 1.34pF (3V) and 2.67pF (0V).
FIG. 18: the invention discloses a physical schematic diagram of a dual-frequency antenna of a dielectric integrated waveguide.
Description of the drawing reference numbers: a short stub 101, a strong field region 102, a varactor 103, and a bias circuit 104.
Detailed Description
The dual-band antenna design method based on dielectric integrated waveguides proposed by the present invention is illustrated below by way of specific examples, examples of which are shown in the accompanying drawings, wherein like reference numerals refer to the like meanings throughout, and the embodiments described below by reference to the drawings are exemplary only and are not to be construed as limiting the invention.
Referring to fig. 1, a single-unit stub QMSIW cavity resonator of the present invention is schematically illustrated, which comprises a rectangular open-circuited stub 101 connected to a strong field region 102 of the stub QMSIW cavity resonator, a varactor 103 connected to the stub 101 along with a bias circuit 104 comprising at least an inductive element and a resistive element.
It is known that a quarter-mode cavity still maintains the same resonance frequency at three-quarters of the reduced resonator size, and that a 75% miniaturized circuit is realized compared to a conventional SIW (dielectric Integrated Waveguide), and that further miniaturization of a QMSIW cavity resonator is realized by connecting the stub 101 to the strong field region 102 of a QMSIW cavity resonator, which is conventionally equivalent to an RLC (Radio Link Control) circuit, but such a resonator operating below the cutoff frequency can be modeled as an equivalent inductor.
FIG. 2 is a schematic diagram of an equivalent circuit model of the QMSIW cavity resonator of the present invention, in which Z is set to be ZinIs the input impedance, j is an imaginary number, ω is the angular frequency, the coupling between the QMSIW cavity resonator and the source is transferred as an impedance by an impedance transformer, LeqIs the inductance, L, of a QMSIW cavity resonatorstubIs an inductance of the load stub, CvThe input impedance of the equivalent circuit model is calculated by adopting a formula (1) and a public expression (2) to obtain the capacitance of the variable capacitance diode:
Zin =jωLeq (1−ω2LstubCv )/1−ω2Cv (Leq+Lstub ) (1)
ω0=1/(Cv (Leq + Lstub ))1/2 (2)
as can be seen from equation (2), the resonant frequency ω0And CvAnd LstubIs related to the value of CvAnd LstubValue of (d) and resonant frequency ω0There is an inverse relationship, and the size of the QMSIW cavity resonator can be further reduced by increasing the length of the stub 101 and the capacitance value of the varactor 103.
Further referring to fig. 3, length L of short section of QMSIW cavity resonator of the present inventionsSchematic diagram of waveform of resonance frequency, fig. 4 short section width W of QMSIW cavity resonator of the inventionsSchematic diagram of wave form of resonant frequency, and figure 5 QMSIW cavity resonator varactor capacitance C of the inventionvWaveform diagram of resonance frequency, cavity radius R of QMSIW cavity resonator of FIG. 6cavIn the QMSIW cavity resonator of the present invention, experiments were conducted on the influence of the parameters of the stub, the cavity size, and the capacitance of the varactor diode 103 on the resonant frequencies of the three modes, i.e., mode 1(TM01), mode 2(TM02), and mode 3(TM20), and it can be seen that the resonant frequencies of the three modes of mode 1 to mode 3 follow LsIs changed in inverse proportion toAs shown in FIG. 3 and equation (2), the length L of the stub 101 is increasedsAn additional inductive effect is caused and the resonance frequency is shifted to a lower frequency. Width W of stub 101sThe influence on the mode is shown in FIG. 4, the width W of the stub 101sThe influence on the modes 1-3 is small, and the three resonant modes are all equal to WsAnd (4) positively correlating.
As shown in fig. 5, as the capacitance of the varactor diode 103 increases, the resonant frequencies of all three modes 1 to 3 shift to low frequencies, and the capacitance CvIn the denominator of equation (2), the capacitance value C is thus increasedvThe resonant frequency of the antenna is lowered and, in addition, the capacitance value C is apparent from the formula (2)vInversely proportional to the resonance frequency and can therefore be verified by parametric analysis. As shown in FIG. 6, the pattern analysis is for different lumen radii RcavBy making a radius RcavThe specific variation relationship is shown in formula (3) and is in inverse proportion to mode 1 and mode 3, but in proportion to mode 2:
(fres) 01=2.404c/2πRcav (
Figure 514715DEST_PATH_IMAGE001
)1/2 (3)
wherein f isresIs the resonant frequency of mode 1(TM01), 2.404 is the Bessel function value of mode 1(TM01), RcavIs the cavity radius of a circular SIW cavity, and
Figure 582028DEST_PATH_IMAGE001
is the relative dielectric constant of the substrate.
As can be seen from equation (3), the parameter analysis verifies RcavIn relation to the resonant frequency of mode 1(TM01), experimental results show that: the length L of the stub 101sAnd the capacitance C of the varactor 103vHas a significant influence on the resonance frequency of mode 1(TM01), and the width W of the stub 101sThe influence on the resonant frequency is small, and the QMSIW cavity resonator has small size advantage, so that larger manufacturing error can be allowed。
Referring to fig. 7, the external quality factor of the dual-band antenna of the dielectric integrated waveguide of the present invention under different feeding parameters is schematically illustrated, the present invention employs a 50 Ω microstrip transmission line and an embedded feeding technique for exciting the QMSIW cavity resonator, as shown in fig. 7, an embedded feeding length p and an embedded feeding width g for controlling the external quality factor of the QMSIW cavity resonator, and when the inserted feeding length p and width g are increased, the quality factor is decreased;
please refer to fig. 8, which is a schematic diagram of the external quality factor of the dual-band antenna of the dielectric integrated waveguide of the present invention under different parameters of the stub and the capacitance value, because the stub 101 is also coupled to the QMSIW cavity resonator, the length L of the stub 101sAnd the capacitance C of the varactor 103vAlso has an effect on the external quality factor, which varies with the length L of the stub 101sAnd the capacitance C of the varactorvIncreased, the external quality factor of the QMSIW cavity resonator is calculated using equation (4):
Qex =f0/f±90（4）
wherein Q isexIs an external quality factor, f0Is the center frequency, f, obtained from the peak of the group delay plot±90Is the center frequency f in the S11 phase0Of ± 90.
Please refer to fig. 9, which is a schematic cross-sectional view of a dual-band antenna of the dielectric integrated waveguide of the present invention, and fig. 18, which is a schematic physical view of the dual-band antenna of the dielectric integrated waveguide of the present invention.
The self-duplexed tuned antenna consists of two QMSIW cavity resonators of unequal size, each obtained from a HMSIW (Half-Mode Substrate Integrated Waveguide) cavity resonator.
The self-duplex tuning antenna consists of two semicircular HMSIW cavity resonators, each SIW cavity resonator is realized by adopting a metal through hole, the radius of the through hole and the clearance between the through holes are used for reducing the cavity and the leakage from the cavity, the top of the antenna is provided with an open metal groove by an etching process and is used for separating the semicircular HMSIW cavities into two different QMSIW cavities, and each QMSIW cavity is respectively provided with one QMSIW cavityA 50 omega microstrip line excitation, a stub 101 connected to the QMSIW cavity resonator for reducing the cavity size, varactors 103 connected to the stub as frequency tuning elements, each varactor 103 fed by a bias circuit 104, and a self-duplexed tuned antenna having a length LsubWidth of Wsub，Rc1And Rc2Radius of two QMSIW cavity resonators, Ls1And Ls3Length of two QMSIW cavity resonators, Ls2And Ls4Respectively, the lengths from the short stub to the varactor of the two QMSIW cavity resonators, p is the feed length, g is the embedded feed width, and W is the lengthgFor the distance between two different sized QMSIW cavity resonators, V1 and V2 are the voltages applied by the two bias circuits, LtlIs the length of a 50 omega fed microstrip transmission line.
Referring to fig. 10, the dual-band antenna package structure of the dielectric integrated waveguide of the present invention is a schematic diagram of a receiver or a transmitter using the self-duplex tuned antenna of the present invention, and the PCB board inside the package structure is provided with surface-mounted electronic circuit devices, such as any one of a resistor, an inductor, a capacitor, an integrated circuit, a sensor and a battery, and a combination thereof. In simulation experiments, different package structures were placed on a PCB board and assigned various material properties, e.g., FR-4 (R) ((R))
Figure 865241DEST_PATH_IMAGE001
1mm thickness of = 4.4 and tan δ = 0.02) and alumina: (a)
Figure 422125DEST_PATH_IMAGE001
A thickness of 0.25mm of = 9.8 and tan δ = 0.006) is used for the PCB and the device housing, respectively, and the size of the package structure in fig. 10 is, as an example: length 59mm, width 38mm, height 11.6 mm.
In practical application, the self-duplex tuning antenna and other electronic or radio frequency components are positioned in a device, and the self-duplex tuning antenna has two resonances at 4.59GHz (| S11|) and 6.1GHz (| S22|), has better isolation, and is better than 24dB at a low frequency band and better than 29dB at a high frequency band.
With further reference to fig. 11, the dual-band antenna of the dielectric integrated waveguide of the present invention is at Ls1The waveform diagrams of S11 and S22 show that, as the length of the stub 101 increases, the resonant frequencies of the three modes, mode 1(TM01), mode 2(TM02) and mode 3(TM20), move to a lower frequency, and in order to further verify the concept of miniaturization, the invention uses the stub load to simulate the effect of the key parameters on the resonant frequency of the antennas1The effect on | S11|, | S21|, and | S22| performance as shown in FIG. 11, we observe that with Ls1The resonance frequency | S11| is reduced without affecting | S22 |; by mixing Ls1Changing from 1mm to 5mm, the resonant frequency of | S11| is shifted from 4.94GHz to 4.17GHz, as shown in FIG. 11, for any Ls1Is changed, | S22| remains stable; when L iss1When changing from 1mm to 5mm, the isolation between port 1 and port 2 changes from 22dB to 24dB, and therefore, it can be concluded that the resonance frequency | S11| can be easily controlled without affecting the resonance frequencies of the other ports.
Referring to fig. 12, the dual-band antenna of the dielectric integrated waveguide of the present invention is at Ls2The S11 and S22 waveforms at time Ls2The effect on S11, S21, and S22 is shown in FIG. 12, with Ls2The resonance frequency of S11 is shifted to a lower frequency, obviously, equal to Ls1In contrast, because Ls1Directly connected to QMSIW cavity resonators, Ls2Is cut off to the varactor diode, so Ls2The resonant frequency of S11 is affected little.
Referring further to fig. 13, the dual-band antenna of the dielectric integrated waveguide of the present invention is at Ls3The waveforms of S21 and S22 are shown, and it can be seen that the resonant frequency of S22 follows Ls3Is increased and decreased, while S11 is unchanged, when L iss3Varying from 1.1mm to 4.1mm, the resonant frequency of S22 was reduced from 6.1GHz to 5.01GHz without affecting the resonant frequency of S11, so it can be concluded that the isolation between the two ports varies with Ls3Is increased because of the QMSIW cavityIs connected to the main stub through a varactor diode, the parameter Ls4The influence on S22 is small.
Referring to fig. 14, the dual-band antenna of the dielectric integrated waveguide of the present invention is at Ls4Waveform diagrams of S21 and S22, with Ls4The S22 is gradually decreased without affecting the S11, and the analysis of the results of the simulation experiment is consistent with the results of formula (2), and formula (2) shows that the resonant frequency and the inductance of the stub have an inverse relationship.
In practical applications, the QMSIW cavity back self-duplex tuned antenna of the present invention is equipped with necessary circuit elements and rf components of PCB and is located inside the device (not shown in the figure), considering real-time applications, the device is used as a receiver or a transmitter, the QMSIW cavity back self-duplex tuned antenna of the present invention is simulated and tested inside the device, and we use HFSS (High Frequency Structure Simulator) to simulate, as shown in fig. 7, in order to verify the simulation results, the QMSIW cavity back self-duplex tuned antenna of the present invention is manufactured in Rogers RT/Duroid 5880 (High Frequency Structure simulation) with a thickness of 1.575mm
Figure 158000DEST_PATH_IMAGE001
=2.2) material, PCB for mounting circuits and radio frequency components is made of commercial FR-4 material, the PCB has a thickness of 1mm, electronic components are surface soldered, a 0.25mm thick aluminum material is used to design a device package, the QMSIW cavity back self-duplex tuned antenna and PCB are placed in a rectangular device (e.g. a quasi-transceiver) and tested experimentally, the lid of the device is fixed to the device container using epoxy, varactor diodes from SKYWORK model SMV1405 are surface mounted as frequency tuning elements, each varactor diode is powered by a current limiting resistor with a resistance of 10k Ω and a radio frequency choke of inductor L47 nH.
With further reference to fig. 15, S parameter waveform diagrams of simulation waveforms and measured waveforms of the first embodiment of the dual-band antenna of the dielectric integrated waveguide of the present invention are shown, for simplicity and clarity of description, the simulation and measurement results of the antenna of two frequency bands are only shown as waveforms of 0V, 3V and 30V, as shown in fig. 15, the low frequency band employs Cv1Frequency of operationTuning, holding Cv2Voltage 30V is constant, Cv1The resonant frequency of the low frequency band can be tuned from 3.77GHz to 4.56GHz from 30V to 0V at Cv1At =30V (0.63pF), the resonant frequency of the lower band is 4.59GHz with a measurement bandwidth of 2.8%; reducing the voltage Cv1=3V (1.34pF), the resonance frequency dropped to 4.24GHz, and the measured partial bandwidth was 2.1%; further decrease of Cv1To 0V (2.67 pF), the resonant frequency was decreased to 3.77GHz, at which time the measurement bandwidth was 1.3%, and it can be seen that C isv1Does not affect the resonant frequency of port 2 (S22). When C is presentv1When the voltage drops from 30V to 0V, the isolation measurement value of the port 1 and the port 2 drops from 24dB to 21 dB; if it is kept Cv1Is 30V when Cv2When the voltage is changed from 30V to 0V, the resonance frequency is changed from 4.96GHz to 6.1 GHz; at Cv2The resonance frequency of the second frequency band is 6.1GHz, and the measurement bandwidth is 3.2%; when C is presentv2When the voltage is reduced to 3V, the resonance frequency of S22 is 5.6 GHz, and the measurement bandwidth is 2.6%; with Cv2Further down to 0V (2.67 pF) and the resonance frequency down to 4.96GHz (measurement bandwidth = 2.2%).
Referring to fig. 16, S parameter waveforms of the simulation waveform and the measured waveform of the second embodiment of the dual-band antenna of the dielectric integrated waveguide according to the present invention, the simulation result shows that Cv2Does not affect the resonant frequency of port 1 (| S11|), when Cv1The measured isolation between port 1 and port 2 was from 29dB to 25dB when changing from 30V to 0V. It can therefore be concluded that: the resonant frequencies of the two ports (| S11| and | S22|) can be independently tuned, so that the self-duplex tuning antenna designed by the invention is proved to be suitable for a tunable radio frequency front end, and the simulation experiment results are consistent with the conclusion of the formula (2), namely the resonant frequency and the capacitance of the varactor are in an inverse proportion relation.
With further reference to fig. 17, simulation and measurement directional diagrams (E-plane and H-plane) of the self-duplex tuned antenna of the present invention at 0.63pF (30V), 1.34pF (3V) and 2.67pF (0V) for the tuning points of the self-duplex tuned antenna corresponding to the resonant frequencies are 0.67pF (30V), 1.34pF (3V), 2.67pF (0V), respectively, and from the simulation and measurement results, the self-duplex tuned antenna designed by the present invention exhibits unidirectional directivity patterns over all tuning ranges of the two bands, with the measured gain varying from 4.85dBi to 5.87dBi when port 1 is excited; when port 2 was energized, the measured gain varied from 4.87dBi to 6.5 dBi.
By adopting the dual-frequency antenna based on the dielectric integrated waveguide, the design cost of the dual-frequency antenna is low, the dual-frequency antenna has the advantages of low profile and small volume, and the product competitiveness is strong.
The above description is only a preferred embodiment of the present invention, and is not intended to limit the scope of the present invention, and all equivalent designs made within the spirit and scope of the present invention are within the scope of the present invention.

Claims (11)

1. A dual-band antenna based on dielectric integrated waveguide, characterized in that the dual-band antenna is composed of two semicircular HMSIW cavity resonators, each semicircular cavity resonator is realized by a metal through hole, the radius of the through hole and the gap between the through holes are used for reducing the cavity and the leakage from the cavity, the top of the antenna is provided with an open metal slot by etching process to divide the semicircular HMSIW cavity into two different QMSIW cavities, each QMSIW cavity comprises a rectangular open stub connected to the strong field region of the QMSIW cavity resonator, a varactor connected to the stub and a bias circuit, each varactor is fed by a bias circuit, and the varactor is connected with the stub as a frequency tuning element.
2. The dual-band dielectric integrated waveguide-based antenna according to claim 1, wherein each QMSIW cavity is excited by a 50 Ω microstrip line.
3. The dual-band dielectric integrated waveguide-based antenna of claim 1, wherein the bias circuit comprises at least one inductive element and one resistive element.
4. The dual-band dielectric integrated waveguide-based antenna according to claim 1, wherein the QMSIW cavity resonator is equivalent to an RLC circuit, and the input impedance of the equivalent circuit model is calculated by using formula (1) and public notation (2):
Zin =jωLeq (1−ω2LstubCv )/1−ω2Cv (Leq+Lstub ) (1)
ω0=1/(Cv (Leq + Lstub ))1/2 (2)
wherein Z isinIs the input impedance, j is an imaginary number, ω is the angular frequency, ω0Is the resonant frequency, LeqIs the inductance, L, of a QMSIW cavity resonatorstubIs an inductance of the load stub, CvIs the capacitance of the varactor.
5. The dielectric integrated waveguide-based dual-band antenna of claim 1, wherein the resonant frequency of the QMSIW cavity resonator varies inversely with the length of the stub and proportionally with the width of the stub in three modes, mode 1-mode 3.
6. The dual-band dielectric integrated waveguide-based antenna of claim 1, wherein the resonant frequency of the QMSIW cavity resonator varies inversely proportional to the capacitance value.
7. The dual-band dielectric integrated waveguide-based antenna according to claim 1, wherein the resonant frequency of the QMSIW cavity resonator is inversely related to the cavity radius in mode 1 and mode 3, and the resonant frequency is directly related to the cavity radius in mode 2, and the specific variation relationship is as shown in formula (3):
(fres) 01=2.404c/2πRcav(
Figure 420129DEST_PATH_IMAGE001
)1/2 (3)
wherein f isresIs the resonant frequency of mode 1, 2.404 is the Bessel function value of mode 1, RcavIs the cavity radius of a circular SIW cavity,
Figure 451539DEST_PATH_IMAGE001
is the relative dielectric constant of the substrate.
8. The dual-band dielectric integrated waveguide-based antenna of claim 1, wherein the QMSIW cavity resonator has a quality factor that varies inversely with variations in inserted feed length and width.
9. The dual-band dielectric integrated waveguide-based antenna of claim 1, wherein the QMSIW cavity resonator has a quality factor that varies in direct proportion to a stub length and a capacitance of a varactor, the quality factor of the QMSIW cavity resonator being calculated using equation (4):
Qex =f0/f ±90（4）
wherein Q isexIs an external quality factor, f0Is the center frequency, f, obtained from the peak of the group delay plot±90Is the center frequency f in the S11 phase0Of ± 90.
10. The dual-band dielectric integrated waveguide-based antenna of claim 1, wherein the varactor is of the type SMV 1405.
11. The dual-band dielectric integrated waveguide-based antenna according to claim 1, wherein the tuning frequency of the dual-band antenna is between two bands of 3.77 GHz-4.59 GHz and 4.96 GHz-6.1 GHz, respectively, and the isolation between the two resonators in the two bands is less than 21 dB.
CN202110991238.5A 2021-08-26 2021-08-26 Dual-frequency antenna based on dielectric integrated waveguide Pending CN113594684A (en)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114725686A (en) * 2022-05-17 2022-07-08 安徽大学 Log-periodic antenna based on half-mode rectangular metal waveguide excitation
CN115275583A (en) * 2022-09-23 2022-11-01 盛纬伦(深圳)通信技术有限公司 Broadband multi-beam antenna array element and array applied to decimeter wave frequency band vehicle-mounted communication

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114725686A (en) * 2022-05-17 2022-07-08 安徽大学 Log-periodic antenna based on half-mode rectangular metal waveguide excitation
CN114725686B (en) * 2022-05-17 2024-03-12 安徽大学 Logarithmic periodic antenna based on half-module rectangular metal waveguide excitation
CN115275583A (en) * 2022-09-23 2022-11-01 盛纬伦(深圳)通信技术有限公司 Broadband multi-beam antenna array element and array applied to decimeter wave frequency band vehicle-mounted communication

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