CN113556049B

CN113556049B - Non-isolated three-phase buck-boost rectifier converter and control method

Info

Publication number: CN113556049B
Application number: CN202110712489.5A
Authority: CN
Inventors: 李伦全; 陈乾宏; 刘斌
Original assignee: Shenzhen Gaoyi Intelligent Electrical Co ltd; Nanjing University of Aeronautics and Astronautics
Current assignee: Shenzhen Gaoyi Intelligent Electrical Co ltd; Nanjing University of Aeronautics and Astronautics
Priority date: 2021-06-25
Filing date: 2021-06-25
Publication date: 2023-11-03
Anticipated expiration: 2041-06-25
Also published as: CN113556049A

Abstract

The application relates to a non-isolated three-phase buck-boost rectifier converter and a control method thereof, wherein the non-isolated three-phase buck-boost rectifier converter comprises an input rectifier bridge group, a buck switch unit and an energy storage freewheel unit; the input rectifier bridge group comprises first to third rectifier bridges, the step-down switch unit comprises first to sixth switch tubes, and the energy storage freewheel unit comprises nineteenth to twentieth diodes, a seventh switch tube, first to second freewheel inductors and first to fourth filter capacitors. The application realizes the maximum multiplexing of the input rectifying bridge group and the step-down switch unit in the three-phase rectifying converter by applying the PWM driving signal of the medium mode and the PWM driving signal of the high mode to the switch tube, is suitable for the wide output voltage range of medium and small power or the output voltage betweenMultiple times toMultiple input phase voltage ranges and where high efficiency and high power density are required.

Description

Non-isolated three-phase buck-boost rectifier converter and control method

Technical Field

The application relates to the technical field of power electronics, in particular to a non-isolated three-phase buck-boost rectifier converter and a control method.

Background

The current electric equipment is more and more in power, and the electric equipment adopting a three-phase power supply mode is more and more, if the electric equipment does not have a Power Factor Correction (PFC) function, the electric energy quality of a power grid is greatly damaged, and even the power grid is paralyzed when serious. In order to meet the power grid quality requirement, harmonic pollution to the power grid is reduced or unnecessary conveying burden of the power distribution network is caused, so that the three-phase electric equipment must have PFC function or be added with a filter device to meet the related regulation requirement.

Generally, if a PFC (power factor correction) function is required for a three-phase ac input rectifying/converting circuit, the circuit is usually mainly of a two-level or three-level boosting type, but after boosting, the output voltage is high, and the use of a converter or a load connected to the back end is limited, for example, the output is generally set to about 720V or even up to 800V when a nominal three-phase three-wire 380VAC is input, and when the back end output voltage is still required to be regulated by the converter, a conventional power tube with better performance is below 650V, and in recent years, a novel switching device such as SiC with a slightly high voltage and a 1200V with better high-frequency switching performance is provided, but the cost is high; in order to solve the limitation of the power device of the dc converter at the back end of the rectifier converter and to consider the efficiency and other factors, the buck-type two-level rectifier converter has become a hot spot of research in recent years, such as the buck PFC of fig. 1, which needs to be subjected to a reduction process when the initial voltage is low, and the buck PFC of fig. 2, which is shown in fig. 2, can theoretically stably output a rated voltage of up to 1.5 times the peak voltage of the phase voltage if the buck PFC is adopted, and if the output required voltage exceeds the voltage range but does not reach The back end of the phase voltage peak value is required to be additionally provided with a one-stage non-isolated DC/DC direct current conversion circuit (such as a boost scheme) to convert the phase voltage peak value into the required output voltage, and the phase voltage peak value is realized by adopting a boost or buck scheme and then performing one-stage DC/DC voltage stabilizing conversion, so that the cost of the two-stage scheme is higher, and meanwhile, the efficiency is reduced due to the two-stage conversion.

Disclosure of Invention

The invention aims to provide a non-isolated three-phase buck-boost rectifier converter and a control method thereof, which solve the technical problems that in the prior art, a two-stage converter needs multiple conversion, flow guide passage devices are more, and the flow guide capacity of a buck switching device cannot be fully utilized, so that the loss is large, and the non-isolated three-phase buck-boost rectifier converter is not suitable for being applied to places with relatively high cost requirements.

The first technical scheme adopted by the invention is as follows: a non-isolated three-phase buck-boost rectifier converter comprises an input rectifier bridge group, a buck switch unit and an energy storage freewheel unit; the input rectifier bridge group comprises first to third rectifier bridges, each of the first to third rectifier bridges comprises four diodes, the four diodes are respectively connected in parallel in pairs to form two bridge arm groups with the same function, the two bridge arm groups are connected in parallel to form two alternating current input ports, namely the midpoints of the two diodes in the bridge arm groups connected in series, one rectifier output positive end, namely the cathode of the bridge arm group, and one rectifier output negative end, namely the anode of the bridge arm group; the step-down switching unit comprises thirteenth to eighteenth diodes and first to sixth switching tubes, and the energy storage freewheeling unit comprises first to second freewheeling inductors, a seventh switching tube, nineteenth to twentieth diodes and first to fourth filter capacitors;

The first rectifier bridge comprises first to fourth diodes, wherein one alternating current input port of the first rectifier bridge, namely the anode of the first diode, is connected with A of a three-phase three-wire power supply, and the other alternating current input port of the first rectifier bridge, namely the anode of the second diode, is connected with B of the three-phase three-wire power supply; the positive end of the rectification output of the first rectification bridge, namely the cathodes of the first diode and the second diode are respectively connected with the anode of the thirteenth diode and one end of the first filter capacitor, and the negative end of the rectification output of the first rectification bridge, namely the anodes of the third diode and the fourth diode are respectively connected with the cathode of the fourteenth diode and the other end of the first filter capacitor; the cathode of the thirteenth diode is connected with the drain electrode of the first switching tube, and the anode of the fourteenth diode is connected with the source electrode of the second switching tube; the second rectifier bridge comprises fifth to eighth diodes, wherein one alternating current input port of the second rectifier bridge, namely the anode of the fifth diode, is connected with the B phase of the three-phase three-wire power supply, and the other alternating current input port of the second rectifier bridge, namely the anode of the sixth diode, is connected with the C phase of the three-phase three-wire power supply; the positive end of the rectification output of the second rectification bridge, namely the cathodes of the fifth diode and the sixth diode are respectively connected with the anode of the fifteenth diode and one end of the second filter capacitor, and the negative end of the rectification output of the second rectification bridge, namely the anodes of the seventh diode and the eighth diode are respectively connected with the cathode of the sixteenth diode and the other end of the second filter capacitor; the cathode of the fifteenth diode is connected with the drain electrode of the third switching tube, and the anode of the sixteenth diode is connected with the source electrode of the fourth switching tube; the third rectifier bridge comprises ninth to twelfth diodes, one alternating current input port of the third rectifier bridge, namely the anode of the ninth diode, is connected with the A phase of the three-phase three-wire power supply, and the other alternating current input port of the third rectifier bridge, namely the anode of the twelfth diode, is connected with the C phase of the three-phase three-wire power supply; the positive end of the rectification output of the third rectification bridge, namely the cathodes of the ninth diode and the twelfth diode are respectively connected with the anode of the seventeenth diode and one end of the third filter capacitor, and the negative end of the rectification output of the third rectification bridge, namely the anodes of the eleventh diode and the twelfth diode are respectively connected with the cathode of the eighteenth diode and the other end of the third filter capacitor; the cathode of the seventeenth diode is connected with the drain electrode of the fifth switching tube, and the anode of the eighteenth diode is connected with the source electrode of the sixth switching tube; one end of the first follow current inductor is respectively connected with a source electrode of the first switching tube, a source electrode of the third switching tube, a source electrode of the fifth switching tube and a cathode of the nineteenth diode, the other end of the first follow current inductor is respectively connected with an anode of the twentieth diode and a drain electrode of the seventh switching tube, and one end of the fourth filter capacitor is connected with the cathode of the twentieth diode to form a positive output end of the rectifier converter; one end of the second follow current inductor is respectively connected with the drain electrode of the second switching tube, the drain electrode of the fourth switching tube, the drain electrode of the sixth switching tube and the anode of the nineteenth diode, and the other end of the second follow current inductor is respectively connected with the source electrode of the seventh switching tube and the other end of the fourth filter capacitor to form a negative output end of the rectifier converter.

Further, the thirteenth to eighteenth diodes are high-frequency diodes, the first to seventh switching transistors are semiconductor devices for controlling on and off by high-frequency driving signals, and the first to seventh switching transistors are provided with anti-parallel diodes, and the anti-parallel diodes are integrated diodes, parasitic diodes or externally-added diodes; the first to third filter capacitors are nonpolar capacitors, and the fourth filter capacitor is a nonpolar capacitor or a polar capacitor; the positive electrode of the capacitor with polarity is connected with the cathode of the twentieth diode, and the negative electrode of the capacitor with polarity is connected with the source electrode of the seventh switching tube.

Further, the three-phase three-wire power supply also comprises an input filter, wherein the input filter is connected between the three-phase three-wire power supply and the input rectifier bridge group.

The second technical scheme adopted by the invention is as follows: a three-phase rectifier transformer, comprising at least two non-isolated three-phase buck-boost rectifier transformers according to the first aspect, each of the non-isolated three-phase buck-boost rectifier transformers being connected in parallel, and the operating phases of the first to sixth switching tubes of each of the non-isolated three-phase buck-boost rectifier transformers being staggered by 1/N high frequency switching cycles, wherein N is the total number of non-isolated three-phase buck-boost rectifier transformers connected in parallel.

The third technical scheme adopted by the invention is as follows: a control method of a non-isolated three-phase buck-boost rectifier converter is used for controlling the non-isolated three-phase buck-boost rectifier converter according to the first technical scheme, and comprises the following steps:

s100: according to the phase lock of the input three-phase three-wire power supply voltage signal, analyzing the phase and interval section of each phase power supply at the current moment;

s200: analyzing the instantaneous value of the voltage of each phase power supply in each interval section according to the phase;

s300: applying a driving signal to a buck switch unit under the current interval section to perform PWM driving control so that the two-phase current with a higher instantaneous value is conducted first; then the step-down switch circuit on the conducting instantaneous value secondary high-phase current loop is turned off, so that the current of the phase with the highest instantaneous value and the current of the phase with the lowest instantaneous value are continuously conducted; the specific method comprises the following steps: simultaneously applying a high-mode PWM driving signal with the same duty ratio to a corresponding switching tube in a two-phase alternating current loop with the highest instantaneous value and the lowest instantaneous value, simultaneously applying a medium-mode PWM driving signal to a corresponding switching tube in a current loop with the second highest amplitude value, and turning off the switching tube applying the high-mode PWM driving signal in each interval section and turning off the switching tube applying the medium-mode PWM driving signal;

S400: when the two-phase current with higher instantaneous value is conducted, judging whether the maximum value of the inter-phase instantaneous value voltage difference of the conducted two phases is larger than or equal to the set value of the output voltage, and performing PWM driving control on the seventh switching tube; if the voltage is larger than or equal to the set value of the output voltage, the seventh switching tube is not required to be turned on, and if the voltage is smaller than the set value of the output voltage, the seventh switching tube is required to be turned on; when the seventh switching tube is in an on state, the PWM switching frequency of the seventh switching tube is consistent with the PWM switching frequencies of the first to sixth switching tubes;

s500: all driving signals of the buck switch unit are turned off, and the energy storage freewheel unit is used for freewheel, so that each phase of current can be conducted in each switch period.

Further, in steps S300 to S500, a driving signal identical to the "high" mode PWM driving signal or a driving signal turned off simultaneously with the "high" mode PWM driving signal is applied to the switching tube which does not form the current path.

Further, the time of each phase of conducting current is in direct proportion to the instantaneous value of the phase voltage, and the current conducting time of the phase with the largest instantaneous value is equal to the sum of the current conducting time of other two phases.

The fourth technical scheme adopted by the invention is as follows: a control method of three-phase rectification converter, is used for controlling a three-phase rectification converter stated in the second technical scheme, use the following step to control each non-isolated three-phase step-up and step-down rectification converter in the stated three-phase rectification converter separately, the concrete step is:

The beneficial technical effects of the invention are as follows:

(1) The high-voltage power converter overcomes the defect of high voltage at the rear end of the traditional boost three-phase rectification conversion circuit in terms of structure and performance, avoids the complexity of multi-stage circuit conversion, reduces the limitation of a power device of a direct-current converter at the rear end, and has larger optional scope;

(2) The realization mode of the traditional step-up or step-down three-phase rectification conversion circuit is changed, the output voltage has smaller limitation compared with the alternating current input, can be step-up, step-down and even can be the voltage in the phase difference amplitude, namely the output voltage is between Multiple to->In the input phase voltage range, the advantages are obvious in replacing the traditional passive PFC, and especially the traditional three-phase passive PFC below 30kW is replaced;

(3) The invention changes the realization channel form of the traditional buck three-phase rectification conversion circuit, has lower loop conduction impedance, and is particularly suitable for occasions with high efficiency and high power density requirements because the loop conduction impedance is only half of that of the existing known scheme under the condition of using the same switching tube in a buck mode;

(4) Because of the structural simplification, only the conduction of the output positive end or the output negative end of the buck switch unit is controlled, so that the control difficulty is reduced, and from the aspect of the switching operation of the PFC function, the control method is simplified by applying regular or logical combined PWM driving signals to the switching tube of each phase; simultaneously, the positive and negative current loop impedance between the parallel circuits is changed by adjusting the on time of each phase of loop, so that cross loop current is avoided;

(5) The input rectifier bridge group can be replaced by a common power frequency rectifier, and the electrodeless capacitor connected in parallel behind the rectifier bridge can absorb part of high-frequency current, so that current oscillation of an external filter is reduced, and the EMI is improved.

Drawings

In order to more clearly illustrate the technical solutions of the embodiments of the present application, the drawings that are needed in the embodiments will be briefly described below, and it is obvious that the drawings in the following description are only some embodiments of the present application, and other drawings may be obtained according to these drawings without inventive effort for a person skilled in the art.

Fig. 1 is a schematic diagram of a conventional classical boost PFC circuit;

FIG. 2 is a schematic diagram of a prior art buck rectifier circuit;

FIG. 3 is a schematic diagram of a prior art DC output block;

FIG. 4 is a schematic diagram of a non-isolated three-phase buck-boost rectifier converter according to embodiment 1 of the present application;

FIG. 5 is a schematic diagram showing waveforms of three-phase voltages and definition of junction in embodiment 1 of the present application;

FIG. 6 is a schematic diagram of an AC-BC interval AB phase conduction loop of embodiment 1 of the present application, which is in a buck inductor energy storage mode;

FIG. 7 is a schematic diagram of the AC-O interval BC phase-sequence loop of embodiment 1 of the present application;

FIG. 8 is a schematic diagram of an O-BC interval AC phase-sequence loop in accordance with embodiment 1 of the present application;

FIG. 9 is a schematic diagram of an AC-BC interval inductor current freewheel loop in accordance with embodiment 1 of the present application;

FIG. 10 is a schematic diagram of an AC-BC interval AB phase conduction loop 2 of embodiment 1 of the present application, which is in a boost inductor energy storage mode;

FIG. 11 is a schematic diagram of the AB phase conduction loop in the AC-BC segment in embodiment 1 of the present invention, which is a boost inductor energy release mode;

FIG. 12 is an equivalent transformation schematic 1 of embodiment 1 of the present invention;

FIG. 13 is an equivalent transformation schematic diagram 2 of embodiment 1 of the present invention;

FIG. 14 is an equivalent transformation schematic 3 of embodiment 1 of the present invention;

FIG. 15 is a schematic diagram showing the relationship between driving waveforms of each switch group in a three-phase AC period according to embodiment 1 of the present invention;

fig. 16 is a schematic structural view of embodiment 2 of the present invention.

Reference numerals explain: FB1, FB2, FB3, third rectifier bridge, D1, first diode, D2, second diode, D3, third diode, D4, fourth diode, D5, fifth diode, D6, sixth diode, D7, seventh diode, D8, eighth diode d9.ninth diode, D10.tenth diode, D11.eleventh diode, D12.twelfth diode, D13.thirteenth diode, D14.fourteenth diode, D15.fifteenth diode, D16.sixteenth diode, D17.seventeenth diode the circuit comprises a diode D18, a diode D19, a diode nineteenth, a diode D20, a diode twenty, a first switching tube Q1, a second switching tube Q2, a third switching tube Q3, a fourth switching tube Q4, a fifth switching tube Q5, a sixth switching tube Q6, a seventh switching tube Q7, a first freewheeling inductor L1, a second freewheeling inductor L2, a first filter capacitor C1, a second filter capacitor C2, a third filter capacitor C3, a fourth filter capacitor C4, a Phase A and Phase B input and a Phase C and Phase C input.

Detailed Description

In order that the above-recited objects, features and advantages of the present application will be more clearly understood, a more particular description of the application will be rendered by reference to the appended drawings and appended detailed description. In the following description, numerous specific details are set forth in order to provide a thorough understanding of the present application, however, the present application may be practiced in other ways than those described herein, and therefore the present application is not limited to the specific embodiments disclosed below.

Unless defined otherwise, technical or scientific terms used herein should be given the ordinary meaning as understood by one of ordinary skill in the art to which this application pertains. The terms "first," "second," and the like in the description and in the claims, are not used for any order, quantity, or importance, but are used for distinguishing between different elements. Likewise, the terms "a" or "an" and the like do not denote a limitation of quantity, but rather denote the presence of at least one. The terms "connected" or "connected," and the like, are not limited to physical or mechanical connections, but may include electrical connections, whether direct or indirect. "upper", "lower", "left", "right", etc. are used merely to indicate a relative positional relationship, which changes accordingly when the absolute position of the object to be described changes.

Example 1:

as shown in fig. 4, a non-isolated three-phase buck-boost rectifier converter includes an input rectifier bridge group, a buck switch unit and an energy storage freewheel unit; the input rectifier bridge group comprises first to third rectifier bridges FB1 to FB3, wherein the first to third rectifier bridges FB1 to FB3 comprise four diodes, the four diodes are respectively connected in parallel in pairs to form two bridge arm groups with the same function, the two bridge arm groups are connected in parallel to form two alternating current input ports, namely the midpoints of the series connection of the two diodes in the bridge arm groups, one rectifier output positive end, namely the cathode of the bridge arm group, and one rectifier output negative end, namely the anode of the bridge arm group; the step-down switching unit comprises thirteenth to eighteenth diodes D13 to D18 and first to sixth switching tubes Q1 to Q6, and the energy storage freewheeling unit comprises first to second freewheeling inductors L1 to L2, a seventh switching tube Q7, nineteenth to twentieth diodes D19 to D20 and first to fourth filter capacitors C1 to C4;

the first rectifier bridge FB1 comprises first to fourth diodes D1 to D4, and one ac input port of the first rectifier bridge FB1, that is, an anode of the first diode D1 is connected with an a of the three-phase three-wire power supply, and the other ac input port, that is, an anode of the second diode D2 is connected with a B of the three-phase three-wire power supply; the positive rectification output end of the first rectification bridge FB1, namely the cathodes of the first diode D1 and the second diode D2 are respectively connected with the anode of the thirteenth diode D13 and one end of the first filter capacitor C1, and the negative rectification output end of the first rectification bridge FB1, namely the anodes of the third diode D3 and the fourth diode D4 are respectively connected with the cathode of the fourteenth diode D14 and the other end of the first filter capacitor C1; the cathode of the thirteenth diode D13 is connected with the drain electrode of the first switching tube Q1, and the anode of the fourteenth diode D14 is connected with the source electrode of the second switching tube Q2; the second rectifier bridge FB2 includes fifth to eighth diodes D5 to D8, and one ac input port of the second rectifier bridge FB2, that is, an anode of the fifth diode D5 is connected to B of the three-phase three-wire power supply, and another ac input port, that is, an anode of the sixth diode D6 is connected to C of the three-phase three-wire power supply; the positive rectification output end of the second rectification bridge FB2, namely, the cathodes of the fifth diode D5 and the sixth diode D6 are respectively connected with the anode of the fifteenth diode D15 and one end of the second filter capacitor C2, and the negative rectification output end of the second rectification bridge FB2, namely, the anodes of the seventh diode D7 and the eighth diode D8 are respectively connected with the cathode of the sixteenth diode D16 and the other end of the second filter capacitor C2; the cathode of the fifteenth diode D15 is connected with the drain electrode of the third switching tube Q3, and the anode of the sixteenth diode D16 is connected with the source electrode of the fourth switching tube Q4; the third rectifier bridge FB3 includes ninth to twelfth diodes D9 to D12, and one ac input port of the third rectifier bridge FB3, that is, an anode of the ninth diode D9 is connected to the a of the three-phase three-wire power supply, and another ac input port, that is, an anode of the twelfth diode D12 is connected to the C of the three-phase three-wire power supply; the positive rectification output end of the third rectification bridge FB3, namely, the cathodes of the ninth diode D9 and the tenth diode D12 are respectively connected with the anode of the seventeenth diode D17 and one end of the third filter capacitor C3, and the negative rectification output end of the third rectification bridge FB3, namely, the anodes of the eleventh diode D11 and the twelfth diode D12 are respectively connected with the cathode of the eighteenth diode D18 and the other end of the third filter capacitor C3; a cathode of the seventeenth diode D17 is connected to the drain of the fifth switching tube Q5, and an anode of the eighteenth diode D18 is connected to the source of the sixth switching tube Q6; one end of the first follow current inductor L1 is respectively connected with the source electrode of the first switching tube Q1, the source electrode of the third switching tube Q3, the source electrode of the fifth switching tube Q5 and the cathode of the nineteenth diode D19, the other end is respectively connected with the anode of the twentieth diode D20 and the drain electrode of the seventh switching tube Q7, and one end of the fourth filter capacitor C4 is connected with the cathode of the twentieth diode D20 to form a positive output end of the rectifier converter; one end of the second follow current inductor L2 is respectively connected with the drain electrode of the second switching tube Q2, the drain electrode of the fourth switching tube Q4, the drain electrode of the sixth switching tube Q6 and the anode of the nineteenth diode D19, and the other end of the second follow current inductor L2 is respectively connected with the source electrode of the seventh switching tube Q7 and the other end of the fourth filter capacitor C4 to form a negative output end of the rectifier converter.

In embodiment 1, the device further comprises an input filter, wherein the input filter is connected between the three-phase three-wire power supply and the input rectifier bridge group, and is used for filtering the input power supply and filtering and attenuating internal clutter reflection to the input end. The thirteenth to eighteenth diodes D13 to D18 are high-frequency diodes, the first to seventh switching transistors Q1 to Q7 are semiconductor devices for controlling on and off by a high-frequency driving signal, and the first to seventh switching transistors Q1 to Q7 are provided with anti-parallel diodes, which are integrated diodes, parasitic diodes, or external diodes; the first to seventh switching transistors Q1 to Q7 may be MOS transistors or IGBT transistors, and those skilled in the art should understand that the present invention is not limited to the above two semiconductor power switches, but may be other power devices capable of performing high frequency switching operations. An independent driving power supply is used between the first switching tube Q1 to the seventh switching tube Q7; the first switching tube Q1, the third switching tube Q3, and the fifth switching tube Q5 may share one driving power source. The first to third filter capacitors C1 to C3 are nonpolar capacitors, and the fourth filter capacitor C4 is a nonpolar capacitor or a polar capacitor; if the fourth filter capacitor C4 is a capacitor with a polarity, the positive electrode of the capacitor is connected to the cathode of the twentieth diode D20, and the negative electrode is connected to the source of the seventh switching tube Q7. The thirteenth to eighteenth diodes D13 to D18 are respectively connected in series with the first to sixth switching tubes Q1 to Q6 to form unidirectional current channels, so that the reverse parallel diodes of the first to sixth switching tubes Q1 to Q6 can be prevented from reversely inputting the voltages of other rectification loops to form clamping. Therefore, the input rectifier bridge group is only ordinary power frequency rectification, an ordinary power frequency rectifier can be used for replacing the input rectifier bridge group, meanwhile, the first filtering capacitor C1 to the third filtering capacitor C3 which are connected in parallel behind the first rectifier bridge FB1 to the third rectifier bridge FB3 can absorb part of high-frequency current, current oscillation of an external filter and high-frequency voltage peak caused by a switching tube when the switching tube is opened at high frequency are reduced, and therefore EMI is improved.

The control method adopted in the embodiment 1 is as follows: a control method of a non-isolated three-phase buck-boost rectifier converter comprises the following steps:

In steps S300 to S500, a driving signal identical to the "high" mode PWM driving signal or a driving signal turned off simultaneously with the "high" mode PWM driving signal is applied to the switching tube which does not form a current path. The time of each phase of conducting current is in direct proportion to the instantaneous value of the phase voltage, and the current conducting time of the phase with the maximum instantaneous value is equal to the sum of the current conducting time of other two phases. The method for judging the magnitude of the instantaneous value is to compare the absolute value magnitude of the instantaneous value of each phase.

As shown in fig. 5, the three-Phase ac power input in embodiment 1 includes an a-Phase input Phase a, a B-Phase input Phase B, and a C-Phase input Phase C, and for convenience of description, three-Phase voltages are set to be 120 ° different and sinusoidal, one cycle at each 360 °; in view of visual convenience of expression, the points of 30 degrees to 390 degrees, namely 30 degrees of the next period are taken as a complete period, and each intersection point is respectively defined as AC (30 degrees), BC (90 degrees), BA (150 degrees), CA (210 degrees), CB (270 degrees), AB (330 degrees), AC (30 degrees or 390 degrees); zero crossings are marked as "0" points.

As shown in fig. 4, a load or a circuit equivalent to a load may be connected between the positive output terminal and the negative output terminal. According to the basic principle of circuit voltage reduction, the output voltage should be lower than the input voltage to form voltage reduction. Therefore, in embodiment 1, the two phases with the largest instantaneous value form the opposite output end to form a voltage difference, and the 0 ° or origin of the a phase in fig. 5 is taken as a reference, and the lowest point of the instantaneous difference of the voltage difference should be the 30 °, 90 °, 150 °, 210 °, 270 °, 330 ° points of the a phase, or similar periodic phase difference relation point, where the lowest value is 1+1/2 times the highest amplitude of the phase voltage; the highest point of the instantaneous difference of the voltage difference is 60 DEG, 120 DEG, 180 DEG, 240 DEG, 300 DEG, 360 DEG of the A phase, or the similar periodic phase difference relation point, and the highest value is The highest amplitude of the phase voltage is multiplied.

When the output voltage is set to be smaller thanWherein->The effective phase voltage value is smaller than the minimum voltage difference between the two phases of the three-phase voltage at any time, and the output working state of the embodiment 1 is in a full-step-down mode. When the output voltage is set to be greater than +>The output operation state of example 1 is in the boost mode, which is higher than the maximum voltage difference between the three-phase voltages and the two phases at any time. When the output voltage is between->Is->In the working mode of example 1, both the voltage is increased and decreased.

(1) Determining a buck mode based on output voltage demand

As shown in fig. 6, in the AC-BC interval from the AC point to the BC point, the absolute value of the instantaneous value of the a-phase and B-phase voltages is higher than that of the C-phase, so that the internal positive-side diodes (i.e., the first diode D1 and the ninth diode D9), the thirteenth diode D13 and the seventeenth diode D17, which are connected to the a-phase, of the first rectifier bridge FB1 and the third rectifier bridge FB3 are turned on by the forward bias voltage, and the voltage is denoted as Va; the first rectifier bridge FB1 connected with B and the negative end diodes (namely the fourth diode D4 and the seventh diode D7), the fourteenth diode D14 and the sixteenth diode D16 in the second rectifier bridge FB2 are turned on by the positive bias voltage, and the voltage is denoted as Vb; the internal diodes of the second rectifier bridge FB2 and the third rectifier bridge FB3 connected to C, that is, the sixth diode D6, the eighth diode D8, the tenth diode D10 and the twelfth diode D12 are reversely biased by the voltages Va and Vb to be turned on, and neither the third switching tube Q3 nor the sixth switching tube Q6 passes current, that is, the voltage of the third switching tube Q3 is clamped to Va, and the voltage of the sixth switching tube Q6 is clamped to Vb. When the PWM driving on signals are simultaneously applied to the first to sixth switching tubes Q1 to Q6, the corresponding switching tubes of the first to sixth switching tubes Q1 to Q6 are turned on. The current of the phase a can flow through the first freewheeling inductor L1 and the twentieth diode D20, through the fourth filter capacitor C4 or the load, through the second freewheeling inductor L2, through the second switch tube Q2, the fourteenth diode D14 and the branch of the fourth diode D4 Hou Cheng, or through the fourth switch tube Q4, the sixteenth diode D16 and the seventh diode D7 back to the B-phase current source via the branch of the first diode D1, the thirteenth diode D13 and the first switch tube or the branch of the ninth diode D9, the seventeenth diode D17 and the fifth switch tube Q5. At this time, the first freewheeling inductor L1 and the second freewheeling inductor L2 are in a step-down energy storage state, and the current passes through two parallel paths of the input rectifier bridge group and the step-down switch unit.

As shown in fig. 7, in the AC-0 interval, after the driving of the first switching tube Q1, the fifth switching tube Q5, and the sixth switching tube Q6 or the driving of the first switching tube Q1 and the fifth switching tube Q5 is turned off; at this time, the current cannot be immediately reversed due to the existence of the first freewheeling inductor L1 and the second freewheeling inductor L2 in the loop, the electromotive force of the inductor can be reversed, and the inductance releases energy and freewheels; meanwhile, the sixth diode D6 and the fifteenth diode D15 connected to the C phase are turned on by the forward bias voltage, and the third switching tube Q3 is turned on all the time because the third switching tube Q3 is turned on by the on driving signal, the current flows from the C phase through the sixth diode D6, the fifteenth diode D15 and the third switching tube Q3, through the first freewheeling inductor L1, the twentieth diode D20, the fourth filter capacitor C4 and the second freewheeling inductor L2, and returns to the B phase-current source through the branches of the second switching tube Q2, the fourteenth diode D14 and the fourth diode D4 Hou Cheng or the fourth switching tube Q4, the sixteenth diode D16 and the seventh diode D7.

As shown in fig. 8, in the 0-BC interval, after the driving of the second switching tube Q2, the third switching tube Q3 and the fourth switching tube Q4 or the driving of the second switching tube Q2 and the fourth switching tube Q4 is turned off, the current cannot be immediately reversed due to the existence of the first freewheeling inductor L1 and the second freewheeling inductor L2 in the loop, the electromotive force of the inductor is reversed, and the energy released by the inductor freewheels; the twelfth diode D12 and the eighteenth diode D18 which are connected with the C are always conducted because the sixth switching tube Q6 is driven by the on driving signal, and the current flows through the first freewheeling inductor L1, the twentieth diode D20, the fourth filter capacitor C4 and the second freewheeling inductor L2 from the A phase through the branch formed by the first diode D1, the thirteenth diode D13 and the first switching tube or the branch formed by the ninth diode D9, the seventeenth diode D17 and the fifth switching tube Q5 and returns to the C intersecting current source through the sixth switching tube Q6, the eighteenth diode D18 and the twelfth diode D12.

From the above, it can be seen that the key to realize that each phase can conduct current in each switching period, so as to realize high PF value and low THDI is that two phases with higher instantaneous value and opposite polarity are conducted first, and energy is stored in the inductance of the loop, and then the switching tube in the conducting loop of the next highest phase with absolute value of the instantaneous value is turned off, so that the freewheeling current passes through the lowest phase of the instantaneous value. Therefore, in each switching period, the current loop with the instantaneous value of the next highest phase is turned off first, the PWM driving signal mode of the switching tube turned off first is marked as 'middle', and the PWM driving signal mode of the switching tube turned off later is marked as 'high'. Although the switching tube driving of the lowest instantaneous value phase can also apply a PWM driving signal in a high mode, the switching tube needs to be turned on after the PWM driving signal in a medium mode is turned off, and the PWM driving signal mode is named as a low mode. Therefore, in the actual control of embodiment 1, although there may be three duty ratios at which the switching tube is turned on, the PWM driving for each period normally has two values to satisfy the control.

As shown in fig. 9, when all PWM on voltages applied to the switching tube are turned off, all current loops input after the switching tube is turned off are cut off, and the first freewheeling inductor L1 and the second freewheeling inductor L2 inevitably maintain freewheeling due to the fact that the current of the inductor cannot be transient, so that the nineteenth diode D19 is turned on by forward bias. The current is returned to the positive end of the fourth filter capacitor C4 or the equivalent load positive end of the circuit output end through the second freewheeling inductor L2, the nineteenth diode D19, the first freewheeling inductor L1 and the twentieth diode D20 to form a current freewheeling loop.

According to the above operation principle of embodiment 1, in the step-down switching unit, in the same switching period, the PWM drive signal pattern in which the PWM drive signal of the switching transistor on the turn-on loop is turned off first is denoted as "medium", and the PWM drive signal pattern in which the PWM drive signal is turned off later is denoted as "high". In each switching period, the control method firstly conducts two phases with relatively higher instantaneous values and opposite polarities, the inductance of the conducting loop generates voltage drop and energy storage, and then switches off the switching tube in the path of the next highest phase of the instantaneous values, so that the follow current of the switching tube passes through the lowest phase of the absolute value of the instantaneous values. Therefore, in each switching period, three phases have current flowing, if the duty ratio of the PWM driving signal is modulated according to real-time control, the current waveform and the voltage waveform can be consistent, and therefore a higher PF value can be obtained, namely, the PFC correcting function is realized.

In addition, in the case where the same effect is only required regardless of the control complexity, another control mode may be adopted: and applying driving signals to the switching tubes of all the buck loops at different time, firstly applying signals to two phases with higher instantaneous values and opposite polarities to conduct the switching tubes, then switching off the switching tube of the instantaneous value next-higher phase current path in the two phases being conducted, and applying driving signals to the switching tube on the lowest instantaneous value phase-crossing loop to conduct the switching tube, so that the follow current passes through the phase with the lowest instantaneous value, and then controlling to turn off the switching tube in the buck switching unit. Therefore, in each switching period, the current loop of the higher phase with the same amplitude is turned off first, the PWM driving signal mode turned off first is marked as "middle", the PWM driving signal mode turned on later is marked as "low", and the PWM driving signal mode turned on first and turned off last is marked as "high". This approach does not depart from our previous "high" and "medium" control strategy and will not be described in detail below.

(2) Determining boost mode based on output voltage demand

The first to sixth switching transistors Q1 to Q6 need to be driven correspondingly in accordance with the PWM driving control method shown in fig. 15, and PWM driving signals need to be applied to the seventh switching transistor Q7. As shown in fig. 10, when the seventh switching tube Q7 is turned on, the current will be directly shorted by the seventh switching tube Q7 to form a reflux loop, and the voltage of the ac source is completely applied to the first freewheeling inductor L1 and the second freewheeling inductor L2, so that the inductance stores energy. As shown in fig. 11, when the drive of the seventh switching tube Q7 is turned off, the current cannot be reversed due to the existence of the first freewheeling inductor L1 and the second freewheeling inductor L2, and the current continues to remain in the original direction, so that the inductance electromotive force is reversed, and is connected in series with the input voltage, and the energy is released to the output or load side together with the input power source.

According to the analysis of the foregoing working principle, the equivalent transformation of the circuit of embodiment 1 in each working mode can be performed: in embodiment 1, when the switching tubes of the A, B two phases are turned on, the equivalent circuit of fig. 4 is shown in fig. 12, and the circuit has one more parallel diversion path formed by the seventh switching tube Q7 than the circuit shown in fig. 2. Fig. 13 is obtained by simplifying fig. 12 in terms of symmetry and switching functionality.

Under transient conditions, the alternating current source can be equivalently a direct current source after being rectified by a diode, or the alternating current source and the diode can be regarded as a direct current source in an instantaneous circuit; meanwhile, the combined switch tube in the alternating current loop can be simplified and equivalent to a switch, so that the circuit shown in fig. 13 can be further equivalent to the circuit shown in fig. 14. The circuit of embodiment 1 can be regarded as a step-up/down circuit in practice after the above equivalent, and thus has typical step-down and step-up functions. Considering the loss of device conduction in the circuit, dead zone, drive delay and other duty ratio conduction angle loss and necessary power factor correction functions, compared with the traditional two-stage conversion mode, the amplitude range of the output voltage of embodiment 1 is optimally the effective value of the input three-phase voltageThe ratio may slightly exceed this range.

For other intervals, the control method is similar to the AC-BC interval. For the BC-BA interval, in the BC-0 interval, the switching tube driving signal of the A, B two-phase current path is a high-mode PWM driving signal, and the switching tube driving signal of the C-phase current path is a medium-mode PWM driving signal, namely, the C-phase loop is firstly turned off; in the interval of 0-BA, the driving signal of the switching tube of the C, B two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the A-phase current path is a 'medium' mode PWM driving signal, namely the A-phase loop is turned off first.

For the BA-CA interval, in the BA-0 interval, the driving signal of the switching tube of the A, C two-phase current path is a high-mode PWM driving signal, and the driving signal of the switching tube of the B-phase current path is a medium-mode PWM driving signal, namely, the B-phase loop is firstly turned off; in the interval of 0-CA, the driving signal of the switching tube of the A, B two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the C-phase current path is a 'medium' mode PWM driving signal, namely, the C-phase loop is firstly turned off.

For the CA-CB interval, in the CA-0 interval, the driving signal of the switching tube of the B, C two-phase current path is a high-mode PWM driving signal, and the driving signal of the switching tube of the A-phase current path is a medium-mode PWM driving signal, namely, the A-phase loop is firstly turned off; in the 0-CB interval, the driving signal of the switching tube of the A, C two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the B-phase current path is a 'medium' mode PWM driving signal, namely, the B-phase loop is firstly turned off.

For the CB-AB section, in the CB-0 section, the driving signal of the switching tube of the B, A two-phase current path is a high-mode PWM driving signal, and the driving signal of the switching tube of the C-phase current path is a medium-mode PWM driving signal, namely, the C-phase loop is firstly turned off; in the 0-AB interval, the driving signal of the switching tube of the B, C two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the A-phase current path is a 'medium' mode PWM driving signal, namely the A-phase loop is turned off first.

For the AB-AC interval, in the AB-0 interval, the driving signal of the switching tube of the C, A two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the B-phase current path is a 'medium' mode PWM driving signal, namely, the B-phase loop is firstly turned off; in the 0-AC interval, the driving signal of the switching tube of the B, A two-phase current path is a 'high' mode PWM driving signal, and the driving signal of the switching tube of the C-phase current path is a 'medium' mode PWM driving signal, namely the C-phase loop is turned off first.

Since the three-phase voltage is not necessarily completely ideal in reality, and there are changes in phase, amplitude and direction, the driving waveform of each section can only be determined according to the actual phase lock, and therefore the driving waveform of each section should be determined according to the characteristics of the instantaneous waveform of each ac voltage of the section, rather than the ideal angle, and the driving waveform of each section can be divided into twelve sections according to the characteristics of the three-phase power signal, and the twelve sections are shown in table 1 according to the above principle.

Table 1 switching tube driving state logic table

The "low" mode means that the same drive signal as the switching tube of the maximum instantaneous value can be applied according to the control method described above, or the drive signal constituting the freewheel can be applied to the switching tube of the other phase in the same direction as the instantaneous value before the drive signal of the switching tube of the maximum instantaneous value is turned off at the latest, and the duty ratio is recorded as "high-medium". The "low 1" mode means that the driving signal does not need to be applied or a signal of an arbitrary duty ratio can be applied during the on period of the switching tube of the maximum instantaneous value phase. Therefore, in view of simplification and normalization of control, both the "low" and "low 1" modes can apply driving signals consistent with the "high" mode without affecting the function implementation. At this time, table 1 can be simplified into a switching tube driving state logic table as shown in table 2:

Table 2 simplified switching tube drive state logic table

According to the switching tube driving state logic table shown in table 2, one control period is divided into 12 sections in total, and the following control method is performed:

detecting input alternating voltage, judging whether various indexes of the input voltage meet working conditions, and continuing waiting if the indexes of the input voltage do not meet the working conditions; if the condition is met, starting to work, and analyzing the phase and the interval section of each phase of power supply at the current moment according to the phase locking judgment of the input three-phase three-wire power supply voltage signal; analyzing the absolute value of the instantaneous value of the voltage of each phase of power supply; and judges whether embodiment 1 operates in the step-up mode or the step-down mode based on the magnitude of the input interphase voltage instantaneous value and the output voltage set value. If the voltage boosting mode is adopted, PWM driving is required to be applied according to the operation result to turn on the seventh switching tube Q7, otherwise, the voltage reducing mode is adopted, and the seventh switching tube Q7 is not required to be turned on. Meanwhile, a 'middle' mode PWM driving signal is applied to a corresponding switching tube in a current loop with the absolute value of the instantaneous value being the next highest, and a 'high' mode PWM driving signal with the same duty ratio is applied to the rest switching tubes, so that a two-phase power supply with a higher instantaneous value forms a current path. Meanwhile, boost energy storage or partial pressure energy storage is formed on the inductor of the energy storage follow current unit, after the 'middle' mode PWM driving signal is turned off, the switching tube of the other two phases of the original applied 'high' mode PWM driving signal can provide a follow current path for the inductor to conduct continuously, if the boost mode is adopted, after the seventh switching tube Q7 is turned off, the electromotive force of the inductor is reversely connected with the input voltage in series, and then the energy release follow current mode is achieved. The specific duty ratio of the PWM driving signals in the high mode and the medium mode is determined by the real-time control operation result of the controller. When the driving of all the switching transistors is turned off, the inductance electromotive force is reversed, and the inductance current is routed by the nineteenth diode D19. In general, the time of each phase input on-current is in a relative relationship with the instantaneous value of the phase voltage, i.e. the higher the instantaneous value, the longer the current on-time, the larger the duty cycle, the current on-time of the phase with the largest instantaneous value is equal to the sum of the current on-times of the other two phases with the relatively lower instantaneous value and is less than the total time of the switching cycle, and the related waveform driving is as shown in fig. 15.

Through the control method, the three phases are effectively guaranteed to have current flowing in each switching period, meanwhile, the duty ratio of the PWM driving signal is modulated according to real-time control, so that the current waveform and the voltage waveform are consistent in follow-up, a higher PF value can be obtained, and the PFC correction function is realized. In the occasion of high power density, the advantages are very obvious, and the requirements of high-precision products can be met.

Example 2:

as shown in fig. 16, embodiment 2 proposes a modified embodiment based on embodiment 1, which includes at least two non-isolated three-phase buck-boost rectifier converters as described in embodiment 1, each of the non-isolated three-phase buck-boost rectifier converters being connected in parallel, and the operating phases of the first to sixth switching tubes Q1 to Q6 of each of the non-isolated three-phase buck-boost rectifier converters being staggered by 1/N of the high frequency switching cycles, where N is the total number of non-isolated three-phase buck-boost rectifier converters connected in parallel.

The control method of embodiment 2 is the same as that of embodiment 1, and by using the "high" and "medium" mode PWM driving signal control method of embodiment 1, N parallel-connected non-isolated three-phase buck-boost rectifier converters can be controlled, and the working phases of the first to sixth switching tubes Q1 to Q6 of the N parallel-connected three-phase rectifier converters are staggered according to 1/N high frequency switching cycles, so that the current at the ac input end can form a multiphase staggered parallel connection, thereby improving the defect of intermittent input current of the buck power supply.

The above description is only of the preferred embodiments of the present invention and is not intended to limit the present invention, but various modifications and variations can be made to the present invention by those skilled in the art. Any modification, equivalent replacement, improvement, etc. made within the spirit and principle of the present invention should be included in the protection scope of the present invention.

Claims

1. The non-isolated three-phase buck-boost rectifier converter is characterized by comprising an input rectifier bridge group, a buck switch unit and an energy storage freewheel unit; the input rectifier bridge group comprises first to third rectifier bridges, each of the first to third rectifier bridges comprises four diodes, the four diodes are respectively connected in parallel in pairs to form two bridge arm groups with the same function, the two bridge arm groups are connected in parallel to form two alternating current input ports, namely the midpoints of the two diodes in the bridge arm groups connected in series, one rectifier output positive end, namely the cathode of the bridge arm group, and one rectifier output negative end, namely the anode of the bridge arm group; the step-down switching unit comprises thirteenth to eighteenth diodes and first to sixth switching tubes, and the energy storage freewheeling unit comprises first to second freewheeling inductors, a seventh switching tube, nineteenth to twentieth diodes and first to fourth filter capacitors;

2. The non-isolated three-phase buck-boost rectifier converter of claim 1, wherein the thirteenth to eighteenth diodes are high frequency diodes, the first to seventh switching tubes are semiconductor devices that control on and off by a high frequency drive signal, and the first to seventh switching tubes are provided with anti-parallel diodes that are integrated diodes, parasitic diodes, or external diodes; the first to third filter capacitors are nonpolar capacitors, and the fourth filter capacitor is a nonpolar capacitor or a polar capacitor; the positive electrode of the capacitor with polarity is connected with the cathode of the twentieth diode, and the negative electrode of the capacitor with polarity is connected with the source electrode of the seventh switching tube.

3. The non-isolated three-phase buck-boost rectifier converter of claim 1 further including an input filter coupled between the three-phase three-wire power supply and the input rectifier bridge bank.

4. A three-phase rectifier transformer, characterized by comprising at least two non-isolated three-phase buck-boost rectifier transformers according to any one of claims 1-3, wherein each non-isolated three-phase buck-boost rectifier transformer is connected in parallel, and the working phases of the first to sixth switching tubes of each non-isolated three-phase buck-boost rectifier transformer are staggered according to 1/N high frequency switching cycles, wherein N is the total number of non-isolated three-phase buck-boost rectifier transformers connected in parallel.

5. A control method of a non-isolated three-phase buck-boost rectifier converter, for controlling the non-isolated three-phase buck-boost rectifier converter according to any one of claims 1 to 3, comprising the steps of:

6. The method according to claim 5, wherein in steps S300 to S500, the same driving signal as the PWM driving signal in the "high" mode or the driving signal turned off simultaneously with the PWM driving signal in the "high" mode is applied to the switching transistor which does not form the current path.

7. The method of claim 5, wherein the time of conducting current of each phase is proportional to the instantaneous value of the phase voltage, and the current conducting time of the phase with the largest instantaneous value is equal to the sum of the current conducting times of the other two phases.

8. A control method of a three-phase rectifying converter for controlling a three-phase rectifying converter according to claim 4, characterized by controlling each non-isolated three-phase buck-boost rectifying converter in the three-phase rectifying converter by using the following steps:

9. The method according to claim 8, wherein in steps S300 to S500, the same driving signal as the "high" mode PWM driving signal or the driving signal turned off simultaneously with the "high" mode PWM driving signal is applied to the switching tube which does not form the current path.

10. The method of claim 8, wherein the time of conducting current per phase is proportional to the instantaneous value of the phase voltage, and the current conducting time of the largest instantaneous value phase is equal to the sum of the current conducting times of the other two phases.