CN113489398A - Built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy - Google Patents

Built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy Download PDF

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CN113489398A
CN113489398A CN202110619796.9A CN202110619796A CN113489398A CN 113489398 A CN113489398 A CN 113489398A CN 202110619796 A CN202110619796 A CN 202110619796A CN 113489398 A CN113489398 A CN 113489398A
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axis
formula
flux linkage
current
permanent magnet
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CN113489398B (en
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张航
梁文睿
张辉
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Xian University of Technology
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Xian University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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Abstract

The invention discloses a built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy, which specifically comprises the following steps: estimating the MT shaft current based on a flux linkage indirect calculation method, and replacing the estimated value with a given flux linkage, thereby improving the flux linkage-based rotor position estimation; and acquiring deviation coefficients mu and lambda, performing real-time online correction on the permanent magnet flux linkage and quadrature axis inductance parameters, and using the corrected parameters for calculating the maximum torque current ratio control stator flux linkage set value of the motor, estimating the MT axis current and calculating the load angle. The invention provides a parameter error compensation strategy aiming at the position-sensorless control of a built-in permanent magnet synchronous motor under the modulation of low switching frequency SHEPWM, and the parameter deviation of permanent magnet flux linkage and quadrature axis inductance can be corrected in real time, so that the estimation precision of the rotor position is improved, and the robustness of the system is enhanced.

Description

Built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy
Technical Field
The invention belongs to the technical field of transmission control of alternating current motors, and particularly relates to a position-sensorless parameter error compensation strategy for a built-in permanent magnet synchronous motor.
Background
With the rapid development of high-speed rail trains, permanent magnet synchronous traction motors with wide speed regulation range, high power density and low energy consumption become research hotspots in the current rail traffic field, and the existing motor train unit train traction systems all adopt mechanical position/speed sensors to acquire position or rotating speed signals of the motors. In the actual running process of a high-speed train, the electromagnetic environment is complex, vibration is severe, failure of a mechanical sensor is easily caused, a traction system is caused to break down, large torque impact is caused, critical components such as a bearing, a gear and a motor are damaged in serious conditions, and the running safety of the train is damaged. The driving technology without the position sensor can fundamentally eliminate the potential safety hazard and has the advantages of strong anti-jamming capability, high integration level, long service cycle and the like.
For a high-speed rail traction system, the switching frequency of the IGBT is usually about 500Hz due to the limitation of switching loss and heat dissipation. In order to obtain better current and voltage performance in a full speed range, reduce switching loss, prolong the service life of a high-power switching tube, obtain good inverter voltage output characteristics and fully utilize bus voltage, asynchronous modulation is generally used at zero low speed, segmented synchronous modulation is used at medium and high speed, and square wave modulation is used above rated rotation speed.
When a modulation mode of a traction inverter is actually determined, a switching rotating speed is usually determined according to a carrier ratio, at present, an optimized synchronous modulation mode such as Specific Harmonic Elimination (SHEPWM) is usually used when the carrier ratio is smaller than 10, the running time of a corresponding mode accounts for about half of the running time of a train, and at the moment, the corresponding rotating speed range is usually estimated by using a back electromotive force or a flux linkage model excited by a fundamental frequency. Compared with a zero-low-speed high-frequency signal injection method based on saliency tracking, the method does not need to consider rotor saliency, does not need extra harmonic signal injection, is simple in digital implementation, and is mature in application in the industrial field. For the rotor position observer based on the motor model, the construction is based on a voltage, current or flux linkage equation, so the rotor position estimation precision and the system robustness are greatly determined by the accuracy of the command voltage, the feedback current and the motor parameters. Flux linkage based position sensorless control is typically combined with direct torque or stator field orientation control. At this time, the rotor position is usually calculated by a rotor or stator flux linkage vector, and mainly includes two ways: an "effective flux linkage" method based on an extended rotor flux linkage model and an indirect calculation method based on a load angle and a stator flux linkage angle. The rotor position estimation precision and the closed loop robustness of the two modes are determined by the precision of flux linkage amplitude and angle, and are easily influenced by flux linkage, inductance parameter change and system delay. Therefore, in the permanent magnet traction system, parameter variation and system delay characteristics under optimized synchronous modulation are combined, permanent magnet flux linkage and quadrature axis inductance parameter deviation are corrected, and the method has important practical significance for improving the control performance of a position-sensorless control system.
Disclosure of Invention
The invention aims to provide a position sensorless parameter error compensation strategy for a built-in permanent magnet synchronous motor, which corrects parameter error compensation of permanent magnet flux linkage and quadrature axis inductance in real time by solving a deviation coefficient, thereby improving the position estimation precision.
The technical scheme adopted by the invention is that a built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy is implemented according to the following steps:
step 1, estimating the current of an MT shaft based on a flux linkage indirect calculation method, and replacing an estimated value with a given flux linkage, thereby improving the rotor position estimation based on the flux linkage;
and 2, acquiring deviation coefficients mu and lambda, performing real-time online correction on the permanent magnet flux linkage and quadrature axis inductance parameters, and using the corrected parameters for calculating the maximum torque current ratio control stator flux linkage given value of the motor, estimating the MT axis current and calculating the load angle.
The present invention is also characterized in that,
in the step 1, the method specifically comprises the following steps:
step 1.1, for three-phase current iA、iB、iCSampling, performing Clarke transformation to obtain alpha-axis and beta-axis currents, and calculating stator magnetic chain angle estimation value
Figure BDA0003099101310000031
And an electromagnetic torque estimate
Figure BDA0003099101310000032
As shown in formula (1) and formula (2);
Figure BDA0003099101310000033
Figure BDA0003099101310000034
in the formula, #αIs an alpha axis stator flux linkage; psiβIs a beta axis stator flux linkage; p is a radical ofnIs the number of pole pairs; i.e. iαIs the α axis current; i.e. iβIs the beta axis current;
step 1.2, calculating the stator flux linkage given value
Figure BDA0003099101310000035
Based on the electromagnetic torque estimation value, as shown in equation (3)
Figure BDA0003099101310000036
Calculating an M-axis estimated current
Figure BDA0003099101310000037
And T-axis estimated current
Figure BDA0003099101310000038
As shown in formula (4) and formula (5); q-axis set current controlled by maximum torque current ratio
Figure BDA0003099101310000039
With d-axis set current
Figure BDA00030991013100000310
Represented by formula (6) and formula (7);
Figure BDA00030991013100000311
Figure BDA00030991013100000312
Figure BDA00030991013100000313
Figure BDA0003099101310000041
Figure BDA0003099101310000042
in the formula, #fIs a permanent magnet flux linkage; i.e. isIs the stator current; l isdIs a d-axis inductance; l isqIs a q-axis inductance; a ═ psifLd/l；
Figure BDA0003099101310000043
c＝1/l；
Figure BDA0003099101310000045
Median value
Figure BDA0003099101310000046
pnIs the number of pole pairs;
step 1.3, calculating the load angle
Figure BDA0003099101310000047
Based on the stator flux angle estimate, as shown in equation (8)
Figure BDA0003099101310000048
Calculating rotor position estimates
Figure BDA0003099101310000049
As shown in formula (9);
Figure BDA00030991013100000410
Figure BDA00030991013100000411
in the formula, #sA stator flux linkage;
Figure BDA00030991013100000412
is the M-axis estimated current;
Figure BDA00030991013100000416
is the T-axis estimated current;
step 1.4, calculating the compensated rotor position
Figure BDA00030991013100000413
As shown in formula (10);
Figure BDA00030991013100000414
in the formula (I), the compound is shown in the specification,
Figure BDA00030991013100000415
is the angular speed of the rotor, trUpdating the time for PWM; t is tsThe most recent sample time before the update.
In the step 2, the concrete steps are as follows;
step 2.1, actual voltage u based on d axisd-motAnd d-axis command voltage ud-calCalculating q-axis inductance deviation coefficient mu and permanent magnet flux linkage deviation coefficient lambda as shown in formula (11) and formula (12);
Figure BDA0003099101310000051
Figure BDA0003099101310000052
in the formula (I), the compound is shown in the specification,
Figure BDA0003099101310000053
is the rotor angular velocity; i.e. idIs the d-axis current; i.e. iqIs the q-axis current;
step 2.2, calculating the actual value L of the compensated q-axis inductanceq-acAnd the actual value psi of the permanent magnet flux linkage after compensationf-acAs shown in formula (13) and formula (14), respectively;
Lq-ac＝(1+μ)Lq (13)；
ψf-ac＝(1+λ)ψf (14)；
step 2.3, after obtaining the parameter deviation coefficients in the formula (11) and the formula (12), the permanent magnet flux linkage psi is processed according to the formula (13) and the formula (14)fAnd q-axis inductance LqPerforming real-time online correction by Lq-acIn place of LqBy psif-acInstead of psifModifying the stator flux linkage set value calculation, the MT axis current estimation, the load angle calculation and the permanent magnet flux linkage and q axis inductance parameters in the MTPA control, and then rewriting the formulas (3), (4), (5), (6), (7) and (8) as shown in the formulas (15), (16), (17), (18), (19) and (20);
Figure BDA0003099101310000054
Figure BDA0003099101310000055
Figure BDA0003099101310000056
Figure BDA0003099101310000061
Figure BDA0003099101310000062
Figure BDA0003099101310000063
step 2.4, d-axis and q-axis given currents
Figure BDA0003099101310000064
And
Figure BDA0003099101310000065
obtaining d-axis and q-axis given voltages after being adjusted by a current regulator
Figure BDA0003099101310000066
And
Figure BDA0003099101310000067
calculating modulation degree M, voltage vector angle beta and compensated voltage vector angle betare-comThe switching angle alpha can be obtained by the on-line table look-up method for the modulation degree MNFrom
Figure BDA0003099101310000068
And betare-comAdding to obtain the voltage vector angle under the ABC coordinate system
Figure BDA0003099101310000069
To pair
Figure BDA00030991013100000610
And alphaNAnd performing pulse reconstruction to obtain three-phase pulse output.
In step 2.4, the modulation M, the voltage vector angle β and the compensated voltage vector angle βre-comThe calculation formulas of (a) are respectively shown as a formula (21), a formula (22) and a formula (23);
Figure BDA00030991013100000611
Figure BDA00030991013100000612
Figure BDA00030991013100000613
in the formula of UdcIs a dc voltage.
The invention has the beneficial effects that:
1) the robustness improvement of the built-in permanent magnet synchronous motor position-sensorless control under the low switching frequency SHEPWM modulation is realized;
2) improving the traditional indirect calculation method, calculating a load angle by using the estimated dq axis current and replacing an estimated value by using a flux linkage set value;
3) the parameter error compensation of the permanent magnet flux linkage and the quadrature axis inductance is corrected in real time by solving the deviation coefficient, so that the position estimation precision is improved.
Drawings
FIG. 1 is a schematic block diagram of a position sensorless parameter error compensation strategy for an interior permanent magnet synchronous motor of the present invention;
FIG. 2 is a block diagram of a hardware circuit structure of an experimental system used in a position sensorless parameter error compensation strategy of an interior permanent magnet synchronous motor according to the present invention;
FIG. 3 is a waveform diagram of a current performance test of the present invention under SHE7 pulse-optimized synchronous modulation without using the parameter error compensation strategy;
FIG. 4 is a waveform diagram of a current performance test of the present invention under SHE7 pulse-optimized synchronous modulation using the parameter error compensation strategy;
FIG. 5 is a waveform diagram of the performance test of the speed and position estimation without using the parameter error compensation strategy under SHE7 pulse optimization synchronous modulation according to the present invention;
FIG. 6 is a waveform diagram of the performance test of the speed and position estimation using the parameter error compensation strategy under SHE7 pulse optimization synchronous modulation according to the present invention;
Detailed Description
The present invention will be described in detail below with reference to the accompanying drawings and specific embodiments.
The invention discloses a built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy, which is implemented according to the following steps:
step 1, estimating the MT axis current based on a flux linkage indirect calculation method, and replacing the estimated value with a given flux linkage, thereby improving the flux linkage-based rotor position estimation, and the schematic block diagram is shown in fig. 1, and the specific steps are as follows:
step 1.1, for three-phase current iA、iB、iCSampling, performing Clarke transformation to obtain alpha-axis and beta-axis currents, and calculating stator magnetic chain angle estimation value
Figure BDA0003099101310000081
And an electromagnetic torque estimate
Figure BDA0003099101310000082
As shown in formula (1) and formula (2);
Figure BDA0003099101310000083
Figure BDA0003099101310000084
in the formula, #αIs an alpha axis stator flux linkage; psiβIs a beta axis stator flux linkage; p is a radical ofnIs the number of pole pairs; i.e. iαIs the α axis current; i.e. iβIs the beta axis current;
step 1.2, calculating the stator flux linkage given value
Figure BDA0003099101310000085
Based on the electromagnetic torque estimation value, as shown in equation (3)
Figure BDA0003099101310000086
Calculating an M-axis estimated current
Figure BDA0003099101310000087
And T-axis estimated current
Figure BDA0003099101310000088
As shown in formula (4) and formula (5); q-axis set current by maximum torque current ratio control (MTPA)
Figure BDA0003099101310000089
With d-axis set current
Figure BDA00030991013100000810
Represented by formula (6) and formula (7);
Figure BDA00030991013100000811
Figure BDA00030991013100000812
Figure BDA00030991013100000813
Figure BDA00030991013100000814
Figure BDA0003099101310000091
in the formula, #fIs a permanent magnet flux linkage; i.e. isIs the stator current; l isdIs a d-axis inductance; l isqIs a q-axis inductance; a ═ psifLd/l；
Figure BDA0003099101310000092
C＝1/l；
Figure BDA0003099101310000093
Median value
Figure BDA0003099101310000094
pnIs the number of pole pairs;
step 1.3, calculating the load angle
Figure BDA0003099101310000095
Based on the stator flux angle estimate, as shown in equation (8)
Figure BDA0003099101310000096
Calculating rotor position estimates
Figure BDA0003099101310000097
As shown in formula (9);
Figure BDA0003099101310000098
Figure BDA0003099101310000099
in the formula, #sA stator flux linkage;
Figure BDA00030991013100000910
is the M-axis estimated current;
Figure BDA00030991013100000911
is the T-axis estimated current;
step 1.4, calculating the compensated rotor position
Figure BDA00030991013100000912
As shown in formula (10);
Figure BDA00030991013100000913
in the formula，
Figure BDA00030991013100000914
Is the angular speed of the rotor, trUpdating the time for PWM; t is tsThe latest sampling moment before updating;
based on
Figure BDA00030991013100000915
Carrying out Park conversion to obtain d-axis current idAnd q-axis current iqThereby forming a current loop, pair
Figure BDA00030991013100000916
Differential calculation and simple digital low-pass filtering are adopted to carry out rotating speed estimation so as to form a rotating speed ring and control the motor;
step 2, after obtaining deviation coefficients mu and lambda in an internal permanent magnet synchronous motor driving system, carrying out real-time online correction on permanent magnet flux linkage and quadrature axis inductance parameters, and using the corrected parameters for maximum torque current ratio control (MTPA) of the motor, stator flux linkage set value calculation, MT axis current estimation and load angle calculation, wherein the specific steps are as follows;
step 2.1, actual voltage u based on d axisd-motAnd d-axis command voltage ud-calCalculating q-axis inductance deviation coefficient mu and permanent magnet flux linkage deviation coefficient lambda as shown in formula (11) and formula (12);
Figure BDA0003099101310000101
Figure BDA0003099101310000102
in the formula (I), the compound is shown in the specification,
Figure BDA0003099101310000103
is the rotor angular velocity;
Figure BDA0003099101310000104
is d-axis given current;
Figure BDA0003099101310000105
is q-axis given current; i.e. idIs the d-axis current; i.e. iqIs the q-axis current;
step 2.2, calculating the actual value L of the compensated q-axis inductanceq-acAnd the actual value psi of the permanent magnet flux linkage after compensationf-acAs shown in formula (13) and formula (14), respectively;
Lq-ac＝(1+μ)Lq (13)；
ψf-ac＝(1+λ)ψf (14)；
step 2.3, the compensated parameters are used in motor control, that is, after obtaining the parameter deviation coefficients in the equations (11) and (12), the permanent magnet flux linkage psi of the corresponding module in fig. 1 can be mapped according to the relationship between the equations (13) and (14)fAnd q-axis inductance LqPerforming real-time online correction by Lq-acIn place of LqBy psif-acInstead of psif. Modifying the given value calculation of stator flux linkage, the estimation of MT axis current, the calculation of load angle and the permanent magnet flux linkage and q axis inductance parameters in MTPA control, and rewriting the formulas (3), (4), (5), (6), (7) and (8) as shown in the formulas (15), (16), (17), (18), (19) and (20);
Figure BDA0003099101310000111
Figure BDA0003099101310000112
Figure BDA0003099101310000113
Figure BDA0003099101310000114
Figure BDA0003099101310000115
Figure BDA0003099101310000116
step 2.4, d-axis and q-axis given currents
Figure BDA0003099101310000117
And
Figure BDA0003099101310000118
obtaining d-axis and q-axis given voltages after being adjusted by a current regulator
Figure BDA0003099101310000119
And
Figure BDA00030991013100001110
the modulation degree M is calculated from the equation (21), the voltage vector angle beta is calculated from the equation (22), and the compensated voltage vector angle beta is calculatedre-comAs shown in formula (23), the switching angle α can be obtained from the modulation M by an online table look-up methodNFrom
Figure BDA00030991013100001111
And betare-comAdding to obtain the voltage vector angle under the ABC coordinate system
Figure BDA00030991013100001112
To pair
Figure BDA00030991013100001113
And alphaNAnd performing pulse reconstruction to obtain three-phase pulse output.
Figure BDA00030991013100001114
Figure BDA0003099101310000121
Figure BDA0003099101310000122
In the formula (I), the compound is shown in the specification,
Figure BDA0003099101310000123
is the angular speed of the rotor, trUpdating the time for PWM; t is tsThe latest sampling moment before updating;
Figure BDA0003099101310000124
is the estimated electromagnetic torque; l isdIs a d-axis inductor; p is a radical ofnIs the number of pole pairs;
Figure BDA0003099101310000125
is the d-axis given voltage;
Figure BDA0003099101310000126
is the q-axis given voltage; u shapedcIs a direct current voltage;
the built-in permanent magnet synchronous motor is easily influenced by parameter changes of flux linkage and inductance, so that the following problems occur: errors of rotor flux linkage and quadrature axis inductance of the permanent magnet motor can affect position estimation accuracy and closed loop robustness. In order to improve the robustness of the built-in permanent magnet synchronous motor position sensorless control, the invention provides a parameter error compensation strategy aiming at the built-in permanent magnet synchronous motor position sensorless control under the modulation of low switching frequency SHEPWM, and the parameter deviation of permanent magnet flux linkage and quadrature axis inductance can be corrected in real time, so that the rotor position estimation precision is improved, and the system robustness is enhanced.
The system hardware structure of the present invention is shown in fig. 2, and includes: the system comprises a rectification circuit, a filter circuit, a three-phase full-bridge inverter, an IPMSM (interior permanent magnet synchronous motor), an FPGA controller, an isolation driving circuit, a rotary transformer and a current acquisition circuit; the system adopts a rotary transformer to collect real position signals and compares the real position signals with an estimated position. The output end of a three-phase full-bridge inverter in the control system is connected with an IPMSM stator three-phase winding, and the IPMSM is controlled after the initial position of a rotor is estimated. Fig. 3 to 6 are graphs showing the comparison between the current performance and the rotation speed and position estimation performance of the IPMSM under SHE7 pulse-optimized synchronous modulation under the control of the hardware system shown in fig. 2 and using the parameter compensation strategy, and the comparison without using the parameter compensation strategy. The current performance waveform without this parameter compensation strategy is shown in fig. 3: fundamental wave current obviously has phase lag and amplitude attenuation, the sine degree of current waveform is also deteriorated, magnetic field orientation error causes larger alternating-direct axis current fluctuation, and harmonic energy is mainly concentrated in 11 th harmonic, 13 th harmonic and 19 th harmonic in PSD distribution. The waveform diagram of the current performance after the parameter compensation strategy is used is shown in FIG. 4: the phase lag and amplitude error of the fundamental current are eliminated, the alternating-direct axis current fluctuation is controlled within 1A, the PSD distribution is more even, and the amplitude of each subharmonic is weakened. The speed and position estimation performance profiles without the parameter compensation strategy are shown in FIG. 5: the estimated rotating speed has larger fluctuation, the average fluctuation amplitude of the electrical angular speed can reach 300r/min, namely in a high rotating speed area, the parameter error can increase the current oscillation to cause the rotating speed fluctuation, the estimated rotor position is also distorted, and the maximum position estimation error exceeds 0.3 rad. The waveform diagram of the rotation speed and position estimation performance after the parameter compensation strategy is used is shown in FIG. 6: the fluctuation of the estimated signal is eliminated and the position estimation error does not exceed 0.1 rad.

Claims (4)

1. A built-in permanent magnet synchronous motor position sensorless parameter error compensation strategy is characterized by being implemented according to the following steps:
step 1, estimating the current of an MT shaft based on a flux linkage indirect calculation method, and replacing an estimated value with a given flux linkage, thereby improving the rotor position estimation based on the flux linkage;
and 2, acquiring deviation coefficients mu and lambda, performing real-time online correction on the permanent magnet flux linkage and quadrature axis inductance parameters, and using the corrected parameters for calculating the maximum torque current ratio control stator flux linkage given value of the motor, estimating the MT axis current and calculating the load angle.
2. The strategy for compensating the position sensorless parameter error of the interior permanent magnet synchronous motor according to claim 1, wherein in the step 1, specifically:
step 1.1, for three-phase current iA、iB、iCSampling, performing Clarke transformation to obtain alpha-axis and beta-axis currents, and calculating stator magnetic chain angle estimation value
Figure FDA0003099101300000011
And an electromagnetic torque estimate
Figure FDA0003099101300000012
As shown in formula (1) and formula (2);
Figure FDA0003099101300000013
Figure FDA0003099101300000014
in the formula, #αIs an alpha axis stator flux linkage; psiβIs a beta axis stator flux linkage; p is a radical ofnIs the number of pole pairs; i.e. iαIs the α axis current; i.e. iβIs the beta axis current;
step 1.2, calculating the stator flux linkage given value
Figure FDA0003099101300000015
Based on the electromagnetic torque estimation value, as shown in equation (3)
Figure FDA0003099101300000016
Calculating an M-axis estimated current
Figure FDA0003099101300000017
And T-axis estimated current
Figure FDA0003099101300000018
As shown in formula (4) and formula (5); q-axis set current controlled by maximum torque current ratio
Figure FDA0003099101300000019
With d-axis set current
Figure FDA00030991013000000110
Represented by formula (6) and formula (7);
Figure FDA0003099101300000021
Figure FDA0003099101300000022
Figure FDA0003099101300000023
Figure FDA0003099101300000024
Figure FDA0003099101300000025
in the formula, #fIs a permanent magnet flux linkage; i.e. isIs the stator current; l isdIs a d-axis inductance; l isqIs a q-axis inductance; a ═ psifLd/l；
Figure FDA0003099101300000026
C＝1/l；
Figure FDA0003099101300000027
Median value
Figure FDA0003099101300000028
pnIs the number of pole pairs;
step 1.3, calculating the load angle
Figure FDA0003099101300000029
Based on the stator flux angle estimate, as shown in equation (8)
Figure FDA00030991013000000210
Calculating rotor position estimates
Figure FDA00030991013000000211
As shown in formula (9);
Figure FDA00030991013000000212
Figure FDA00030991013000000213
in the formula, #sA stator flux linkage;
Figure FDA00030991013000000214
is the M-axis estimated current;
Figure FDA00030991013000000215
is the T-axis estimated current;
step 1.4, calculating the compensated rotor position
Figure FDA00030991013000000216
As shown in formula (10);
Figure FDA00030991013000000217
in the formula (I), the compound is shown in the specification,
Figure FDA00030991013000000218
is the angular speed of the rotor, trUpdating the time for PWM; t is tsThe most recent sample time before the update.
3. The strategy for compensating the position-sensorless parameter error of the interior permanent magnet synchronous motor according to claim 2, wherein in the step 2, the specific steps are as follows;
step 2.1, actual voltage u based on d axisd-motAnd d-axis command voltage ud-calCalculating q-axis inductance deviation coefficient mu and permanent magnet flux linkage deviation coefficient lambda as shown in formula (11) and formula (12);
Figure FDA0003099101300000031
Figure FDA0003099101300000032
in the formula (I), the compound is shown in the specification,
Figure FDA0003099101300000033
is the rotor angular velocity; i.e. idIs the d-axis current; i.e. iqIs the q-axis current;
step 2.2, calculating the actual value L of the compensated q-axis inductanceq-acAnd the actual value psi of the permanent magnet flux linkage after compensationf-acAs shown in formula (13) and formula (14), respectively;
Lq-ac＝(1+μ)Lq (13)；
ψf-ac＝(1+λ)ψf (14)；
step 2.3, after obtaining the parameter deviation coefficients in the formula (11) and the formula (12), the permanent magnet flux linkage psi is processed according to the formula (13) and the formula (14)fAnd q-axis inductance LqPerforming real-time online correction by Lq-acIn place of LqBy psif_acInstead of psifModifying the stator flux linkage set value calculation, the MT axis current estimation, the load angle calculation and the permanent magnet flux linkage and q axis inductance parameters in the MTPA control, and then rewriting the formulas (3), (4), (5), (6), (7) and (8) as shown in the formulas (15), (16), (17), (18), (19) and (20);
Figure FDA0003099101300000041
Figure FDA0003099101300000042
Figure FDA0003099101300000043
Figure FDA0003099101300000044
Figure FDA0003099101300000045
Figure FDA0003099101300000046
step 2.4, d-axis and q-axis given currents
Figure FDA0003099101300000047
And
Figure FDA0003099101300000048
obtaining d-axis and q-axis given voltages after being adjusted by a current regulator
Figure FDA0003099101300000049
And
Figure FDA00030991013000000410
calculating modulation degree M, voltage vector angle beta and compensated voltage vector angle betare_comThe switching angle alpha can be obtained by the on-line table look-up method for the modulation degree MNFrom
Figure FDA00030991013000000411
And betare_comAdding to obtain the voltage vector angle under the ABC coordinate system
Figure FDA00030991013000000412
To pair
Figure FDA00030991013000000413
And alphaNAnd performing pulse reconstruction to obtain three-phase pulse output.
4. The PSSM position sensorless parameter error compensation strategy according to claim 3, wherein in step 2.4, the modulation M, the voltage vector angle β and the compensated voltage vector angle βre_comThe calculation formulas of (a) are respectively shown as a formula (21), a formula (22) and a formula (23);
Figure FDA0003099101300000051
Figure FDA0003099101300000052
Figure FDA0003099101300000053
in the formula of UdcIs a dc voltage.
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