CN113285598A - Hybrid control structure of interleaved parallel Boost converters, hybrid control method of hybrid control structure, and coupling inductance optimization design method - Google Patents
Hybrid control structure of interleaved parallel Boost converters, hybrid control method of hybrid control structure, and coupling inductance optimization design method Download PDFInfo
- Publication number
- CN113285598A CN113285598A CN202110545514.5A CN202110545514A CN113285598A CN 113285598 A CN113285598 A CN 113285598A CN 202110545514 A CN202110545514 A CN 202110545514A CN 113285598 A CN113285598 A CN 113285598A
- Authority
- CN
- China
- Prior art keywords
- hybrid control
- power mosfet
- inductor
- input current
- ripple
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1582—Buck-boost converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/14—Arrangements for reducing ripples from dc input or output
Abstract
The invention discloses a hybrid control structure of interleaved parallel Boost converters, a hybrid control method of the hybrid control structure and a coupling inductance optimization design method of the hybrid control structure. The upper and lower tubes of the two branch pipelines are in complementary conduction; first power MOSFET-S1And a third power MOSFET-S3The duty ratio of (D) is represented by D, and the output voltage is stabilized by pulse width modulation PWM; when the input current has a low effective value, a control strategy that two Boost circuits have a phase difference of 180 degrees is adopted; when the input current has high effective value, two paths of phase-shifting alpha are adopted2And T. The invention is designed aiming at the requirements of wide-range input and high efficiency of the converter, can reduce the loss of the converter and improve the efficiency and power density of the system.
Description
Technical Field
The invention belongs to the field of power electronics; in particular to a hybrid control structure of an interleaved parallel Boost converter, a hybrid control method thereof and a coupling inductance optimization design method.
Background
The Boost converter is a commonly used topology in the field of power electronics, and is often used as a first-stage converter for pre-voltage stabilization in a module power supply application occasion with wide input range. The multiphase interleaving Boost converter has the advantages of low input current ripple, low output voltage ripple and the like, and is widely applied to high-power occasions.
The interleaved parallel Boost converter adds a magnetic element and works in a hard switching state, which brings challenges to high frequency and high power density. Therefore, the multiphase interleaving Boost converter based on the coupling inductor is produced, a plurality of magnetic elements can be integrated on one magnetic core by utilizing the magnetic integration technology, and the advantages of reducing input current ripple and improving EMI characteristics are achieved while the power density of a system is improved.
The coupling inductor of the traditional interleaved Boost converter is mainly reverse coupled, and the soft switching characteristic of a main switching tube cannot be realized. In order to reduce the loss of the switching tube and improve the efficiency of the converter, it is desirable that the switching tube can realize soft switching characteristics, and at the same time, reduce the conduction loss of the switching tube, i.e. reduce the input current ripple of the switching tube.
The coupling inductance is used as an important element of the interleaved parallel Boost converter, and the parameters of the coupling inductance play a decisive role in the overall performance of the converter. In order to achieve excellent steady-state and dynamic performance of the converter, reduce loss, and increase power density, a standardized design of the coupling inductor is required.
Disclosure of Invention
The invention provides a hybrid control structure of an interleaved parallel Boost converter, a hybrid control method of the interleaved parallel Boost converter, and an optimal design method of coupling inductance, which are designed aiming at the requirements of wide-range input and high efficiency of the converter, can reduce the loss of the converter, and improve the efficiency and power density of a system.
The invention is realized by the following technical scheme:
a hybrid control structure for interleaved parallel Boost converters, the structure comprising a first power MOSFET-S1A second power MOSFET-S2A third power MOSFET-S3Fourth power MOSFET-S4Coupled inductor L1Coupled inductor L2Capacitor C0And a resistance RL(ii) a Input terminal VinRespectively with a coupling inductor L1End of same name and coupling inductance L2Are connected with the same name end of the coupling inductor L1End of synonymRespectively with the first power MOSFET-S1And a second power MOSFET-S2Is connected with the source of the coupling inductor2The different name terminal is respectively connected with the third power MOSFET-S3And a fourth power MOSFET-S4Is connected to the source of the second power MOSFET-S2Respectively with a fourth power MOSFET-S4Drain electrode of (1), capacitor C0And a resistor RLIs connected with the negative electrode of the input end respectively with the first power MOSFET-S1Source of the third power MOSFET-S3Source electrode and capacitor C0Another terminal of (1) and a resistor RLThe other ends of the two are connected.
A hybrid control method of a hybrid control structure of an interleaved parallel Boost converter is characterized in that upper tubes and lower tubes of two branches are conducted complementarily;
first power MOSFET-S1And a third power MOSFET-S3The duty ratio of (D) is represented by D, and the output voltage is stabilized by pulse width modulation PWM;
when the input current has a low effective value, a control strategy that two Boost circuits have a phase difference of 180 degrees is adopted;
when the input current has high effective value, two paths of phase-shifting alpha are adopted2And T.
Furthermore, the control strategy of the two Boost circuits with the phase difference of 180 degrees is that in one period, eight modes exist, namely a switch mode 1 t0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]And obtaining expressions of the inductive current ripple and the input current ripple based on the eight modes:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
Further, the two paths of phase shifting are alpha2The control strategy for T is specifically that in one cycle, eight modes exist, namely a switch mode 1[ T [ ]0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]When S is3Phase shift alpha2，α2T＝t3-T/2>DT-0.5T, and the inductance current ripple is calculated as follows:
the input current ripple is:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
Further, when D>At 0.5, α2T>D-0.5。
A coupling inductance optimization design method of a hybrid control structure of an interleaved parallel Boost converter is characterized in that parameters can meet the requirements of input current ripple ratio and reverse zero crossing of inductor current under phasing and phase-shifting control, and the parameters are obtained through constraint conditions.
Further, the constraint conditions specifically include that the ripple coefficient of the input current meets the requirement of the ripple ratio; the inductive current has reverse moment to ensure ZVS turn-on characteristic of the main switching tube; under the phase-shift control strategy of hybrid control, the topology still ensures that the ripple coefficient of the input current meets the ripple ratio requirement and the inductive current has reverse time so as to ensure the ZVS switching-on characteristic of the main switching tube.
Further, the expression of the constraint condition is,
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, λ is the input current ripple ratio requirement, IinFor effective value of input current, Δ ILFor inductor current ripple, P is the converter output power.
Further, the expression of the constraint condition is that when D <0.5, the corresponding inequality constraint conditions are similar.
Further, the expression of the constraint condition is that a parameter range of the coupling inductor is obtained through the constraint condition, a group of parameters is selected from the parameter range of the coupling inductor and is checked, and if the check result is correct, the group of parameters are parameters capable of meeting the requirements of the ripple ratio of the input current and the reverse zero crossing of the inductor current; if the verification result is incorrect, another group of parameters is selected for verification until the verification result is correct.
The invention has the beneficial effects that:
1. according to the invention, the coupling inductor is used for replacing a discrete inductor, so that the system volume is reduced, and the power density is improved;
2. the invention adopts a hybrid control strategy for control, can ensure the soft switching of a main switching tube, simultaneously meets the requirement of input current ripple ratio as much as possible, and realizes the optimal realization effect of two control purposes.
3. The invention provides the constraint condition of the coupling inductance parameter and the normalized design flow, and is convenient for the parameter design of the coupling inductance.
Drawings
Fig. 1 is a schematic diagram of an interleaved parallel Boost converter according to the present invention.
FIG. 2 is a schematic diagram of the structure of the decoupled equivalent interleaved parallel Boost converter of the invention.
FIG. 3 is a diagram of the switching modes of the interleaved parallel Boost converters of the present invention in one cycle, wherein (a) the switching mode 1[ t ]0～t1]Mode diagram of (b) switching mode 2[ t ]1～t2](c) switching mode 3[ t ]2～t3]Mode diagram of (d) switch mode 4[ t ]3～t4]Mode diagram of (e) switching mode 5[ t ]4～t3](f) switching mode 6[ t ]4～t3](g) switch mode 7[ t ]2～t1](h) switch mode 1[ t ]1～t0]The mode shape diagram of (1).
FIG. 4 is a typical waveform diagram of the present invention under the control of phasing and two different phase shifts for the topology with duty cycle D >0.5, wherein (a) the waveform diagram under the control of phasing, (b) the waveform diagram that should be avoided under the control strategy of phase shifting, and (c) the waveform diagram under the control strategy of phase shifting.
Fig. 5 is a three-dimensional curved graph of the ratio of the inductor current ripple to the inductor current ripple for phasing control according to the present invention as a function of D and k at a phase shift ratio α 2 of D-0.5.
FIG. 6 is a comparison of current waveforms at an input voltage of 75V in accordance with the present invention.
FIG. 7 is a graph comparing current waveforms at an input voltage of 225V according to the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the accompanying drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Example 1
Referring to fig. 1, a hybrid control architecture for an interleaved parallel Boost converter includes a first power MOSFET-S1A second power MOSFET-S2A third power MOSFET-S3Fourth power MOSFET-S4Coupled inductor L1Coupled inductor L2Capacitor C0And a resistance RL(ii) a Input terminal VinRespectively with a coupling inductor L1End of same name and coupling inductance L2Are connected with the same name end of the coupling inductor L1The different name terminal of the first power MOSFET-S is respectively connected with the first power MOSFET-S1And a second power MOSFET-S2Is connected with the source of the coupling inductor2The different name terminal is respectively connected with the third power MOSFET-S3And a fourth power MOSFET-S4Is connected to the source of the second power MOSFET-S2Respectively with a fourth power MOSFET-S4Drain electrode of (1), capacitor C0And a resistor RLIs connected with the negative electrode of the input end respectively with the first power MOSFET-S1Source of the third power MOSFET-S3Source electrode and capacitor C0Another terminal of (1) and a resistor RLThe other ends of the two are connected.
A hybrid control method of a hybrid control structure of an interleaved parallel Boost converter is characterized in that upper tubes and lower tubes of two branches are conducted complementarily; said L1、S1、S2First branch of composition and L2、S3、S4And a second branch.
First power MOSFET-S1And a third power MOSFET-S3The duty ratio of (D) is represented by D, and the output voltage is stabilized by pulse width modulation PWM;
when the input current has a low effective value, a control strategy that two Boost circuits have a phase difference of 180 degrees is adopted;
when the input current has high effective value, the inductive current ripple is also high, and in order to avoid the inductive current ripple being too large, generating interference to system EMI and generating too high loss, two paths of phase-shifting alpha are adopted2And T.
Further, the two boosThe control strategy for the phasing of the st circuit by 180 DEG is embodied in that, within one cycle, eight modes are present, namely the switching mode 1[ t [ ]0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]And obtaining expressions of the inductive current ripple and the input current ripple based on the eight modes:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
Further, the two paths of phase shifting are alpha2The control strategy for T is specifically that in one cycle, eight modes exist, namely a switch mode 1[ T [ ]0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]When S is3Phase shift alpha2，α2T＝t3-T/2>DT-0.5T, and the inductance current ripple is calculated as follows:
the input current ripple is:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
Further, when D>At 0.5, α2T>D-0.5。
A coupling inductance optimization design method of a hybrid control structure of an interleaved parallel Boost converter is characterized in that parameters can meet the requirements of input current ripple ratio and reverse zero crossing of inductor current under phasing and phase-shifting control, and the parameters are obtained through constraint conditions.
Further, the constraint conditions specifically include that the ripple coefficient of the input current meets the requirement of the ripple ratio; the inductive current has reverse moment to ensure ZVS turn-on characteristic of the main switching tube; under the phase-shift control strategy of hybrid control, the topology still ensures that the ripple coefficient of the input current meets the ripple ratio requirement and the inductive current has reverse time so as to ensure the ZVS switching-on characteristic of the main switching tube.
Further, the expression of the constraint condition is,
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, λ is the input current ripple ratio requirement, IinFor effective value of input current, Δ ILFor inductor current ripple, P is the converter output power.
Further, the expression of the constraint condition is that when D <0.5, the corresponding inequality constraint conditions are similar, that is, the following conditions also need to be satisfied: the ripple factor of the input current meets the requirement of the ripple ratio; the inductive current has reverse moment to ensure ZVS turn-on characteristic of the main switching tube; under the phase-shift control strategy of hybrid control, the topology still ensures that the ripple coefficient of the input current meets the ripple ratio requirement and the inductive current has reverse time so as to ensure the ZVS switching-on characteristic of the main switching tube.
Further, the expression of the constraint condition is that a parameter range of the coupling inductor is obtained through the constraint condition, a group of parameters is selected from the parameter range of the coupling inductor and is checked, and if the check result is correct, the group of parameters are parameters capable of meeting the requirements of the ripple ratio of the input current and the reverse zero crossing of the inductor current; if the verification result is incorrect, another group of parameters is selected for verification until the verification result is correct.
Example 2
Referring to fig. 1, a hybrid control architecture for an interleaved parallel Boost converter includes a first power MOSFET-S1A second power MOSFET-S2A third power MOSFET-S3Fourth power MOSFET-S4Coupled inductor L1Coupled inductor L2Capacitor C0And a resistance RL(ii) a Input terminal VinRespectively with a coupling inductor L1End of same name and coupling inductance L2Are connected with the same name end of the coupling inductor L1The different name terminal of the first power MOSFET-S is respectively connected with the first power MOSFET-S1And a second power MOSFET-S2Is connected with the source of the coupling inductor2The different name terminal is respectively connected with the third power MOSFET-S3And a fourth power MOSFET-S4Is connected to the source of the second power MOSFET-S2Respectively with a fourth power MOSFET-S4Drain electrode of (1), capacitor C0And a resistor RLIs connected with the negative electrode of the input end respectively with the first power MOSFET-S1Source of the third power MOSFET-S3Source electrode and capacitor C0Another terminal of (1) and a resistor RLThe other ends of the two are connected.
According to fig. 2, the forward coupled inductor can be decoupled into three inductors, where kL, La, Lb, (1-k) L, k is the coupling coefficient of the coupled inductor, and L is the inductance of the discrete inductor.
According to the circuit model of fig. 2, four operating modes of the topology are obtained. Mode 1: switches S1, S3 are turned on simultaneously; mode 2: the switch S1 is turned on, and the switch S3 is turned off; mode 3: the switch S1 is turned off, and the switch S3 is turned on; mode 4: the switches S1, S3 are turned off simultaneously. According to KVL and KCL theorem, the change rate of single-circuit inductive current and input inductive current ripple under different modes can be obtained:
according to the table, the ripple waves of the inductive current and the input current under different switching periods can be calculated, so that a theoretical basis is provided for subsequent inductor design. The coupled inductance condition during soft switching is further analyzed in conjunction with the mode diagram.
In one cycle, taking D >0.5 as an example, the following eight modes are divided. The mode diagram is shown in figure 3.
Switched mode 1[ t ]0～t1]In this mode, the main switch tubes are all turned on, the auxiliary switch tubes are turned off, and the input power passes through S1And S3The inductor is charged and the inductor current rises linearly.
Switched mode 2[ t ]1～t2]This mode is a dead zone. In this mode, the main switch tube S1And the other switching tubes are switched on and off. Power supply through S1Charging the inductor i1And (4) increasing linearly. Due to the inductor current i2Cannot mutate, i2To give S3While C is chargedoBy S4The junction capacitance of (1) is discharged, in this interval, if S4The junction capacitance of S4 is fully discharged, the body diode of S4 is conducted, and the next step S can be realized4Is turned on.
Switching mode 3[ t ]2～t3]In this mode, the main switch tube S1And an auxiliary switching tube S4And the other switching tubes are switched on and off. From the last modality, S is then4Achieving ZVS conduction. At this time, the power supply passes through S1Charging the inductor i1Linearly rises, and i2Still in a discharged state. S3The junction capacitance voltage is clamped to the output voltage.
Switch mode 4[ t ]3～t4]In this mode, the main switch tube S1And the other switching tubes are switched on and off. Power supply through S1Charging the inductor i1Linear rise, i2Still in a discharged state. At this time if i2In the reverse direction, then S3Can pass through LbDischarge, ready for next modality ZVS.
The switching modes 5-8 are symmetrical modes 1-4, which are not described herein.
The ZVS conduction condition of the main switching tube can be obtained through the analysis: the inductive current must have a reverse moment to provide a discharge loop for the junction capacitor of the main switching tube; the junction capacitance of the main switching tube can be completely discharged successfully in the dead time. And the ZVS of the auxiliary switching tube only needs to satisfy item 2 above.
In order to realize soft switching of the interleaved parallel Boost circuit, the inductor is processed in a same-direction coupling mode, input current ripples are reduced, and meanwhile current ripples of a single inductor are increased, so that the current of the inductor reversely crosses zero, and soft switching is easy to realize. In this case, however, there is a phase in which the inductor current is in phase with the voltage, during which time the delivered power is of opposite polarity to the total average power and the power flows back into the power supply. When the power is transmitted, the larger backflow power can lead to larger forward power transmission amount, and further larger circulation current and current stress are generated, so that the loss of the power device and the magnetic element is increased, and the efficiency of the converter is reduced. Therefore, a control strategy combining phasing and phase shifting of two Boost circuits is adopted.
Fig. 4 is a typical waveform diagram for a topology with phasing control and two different phase shifting controls when the duty cycle D > 0.5.
Fig. 4(a) is a waveform under phasing control, and expressions for inductor current ripple and input current ripple can be derived from [0036 ].
FIG. 4(b) Is one of two phase-shift control strategies, S3 phase-shift alpha1，α1T＝t2-T/2<DT-0.5T, inductor current ripple size unchanged from the above analysis, same as (a); and the input current ripple magnitude is:
is phased controlled
And (4) doubling. Therefore, the phase shift control strategy cannot reduce the inductor current ripple, but rather increases the input current ripple, and should avoid entering this operating state.
FIG. 4(c) shows another of the two phase-shift control strategies, S3 phase-shifting alpha2，α2T＝t3-T/2>DT-0.5T, and the inductive current ripple is calculated to be
The input current ripple is:
given the ratio of the inductor current ripple to the inductor current ripple controlled by phasing in this case, the phase shift ratio alpha2The three-dimensional curved surface graph which changes with D and k when the curve is D-0.5 is shown in figure 5, and according to the curved surface graph, no matter what the coupling coefficient and the duty ratio of the coupling inductor take, the phase-shifting control strategy can effectively reduce the branch inductor current ripple, and when the value is alpha2>And D-0.5, the phase-shifting strategy can further reduce the inductor current ripple. Similarly, the input current ripple also increases at this time. Therefore, under the control strategy, the phase-shift ratio needs to be reasonably set, and the best control effect is achieved.
With D ═ 0.5 as the axis of symmetry, in the range of D >0.5, the basic principle of the hybrid control strategy can be derived from the above analysis:
(1) when the effective value of the input current is lower, the inductor current ripple is also lower. At the moment, the more important working purpose of the coupling inductor is to reduce input current ripples, pursue more excellent dynamic response performance and adopt a control strategy that two Boost circuits have a phase difference of 180 degrees.
(2) When the effective value of the input current is higher, the inductive current ripple is also higher. In order to avoid the interference to system EMI and the over-high loss caused by the overlarge ripple of the inductive current, two paths of phase-shifting alpha are adopted2T＝t3-T/2>The DT-0.5T control strategy sacrifices a portion of the input current with a small ripple, and therefore requires strict control of the phase shift ratio.
Based on the above principle, the constraint conditions to be met by the coupling inductor design can be obtained:
(1) the input current ripple factor meets the ripple ratio requirement.
(2) The inductor current has reverse moment to ensure the ZVS turn-on characteristic of the main switching tube.
(3) Under the phase-shifting control strategy of hybrid control, the topology still ensures the characteristics of (1) and (2)
In combination with the above constraints, the following inequalities are listed:
take a set of parameters as an example. The working range of the known interleaved parallel Boost circuit is: the input voltage is 50-300V, the output voltage is 350V, the output voltage is 800W, the switching frequency is 250kHz, and the input current ripple ratio is 0.5. For the purpose of implementing a hybrid control strategy, at duty cycle D>When the input voltage is 50-175V, the phase shift control is adopted when the input voltage is 50-100V, and the phase shift ratio alpha is defined2>D-0.5, preferably 0.36; and when the input voltage is 100-175V, phased control is adopted.
Then, under phased conditions, i.e. according to
The maximum value is calculated according to the inequalities of the first two phasing controls, and the maximum value can be obtained through simplification
According to the simplified constraint conditions, a group of k is 0.9, and L is 50 μ H, and the conditions are satisfied. The inequality of phase shift control is brought in to meet the condition. I.e. the set of parameters meets the requirements. The parameters of the group are substituted into PLECS power electronic simulation software for simulation, and simulation results shown in the attached figures 6-7 can be obtained. FIG. 6 is a current waveform of phasing control and phase shift control at an input voltage of 75V; fig. 7 shows a comparison of the current waveforms of the phasing control and the phase-shifting control under the input voltage 225V, it can be seen that the phase-shifting control can make the input current ripple ratio not meet the requirement, i.e. the phase-shifting control should not be adopted, and this result also corresponds to "when the effective value of the input current is lower, the inductor current ripple is also lower. At this time, the more important work purpose of the coupling inductor is to reduce input current ripples, pursue more excellent dynamic response performance, and adopt the requirement of a control strategy that two Boost circuits have a phase difference of 180 degrees.
Claims (10)
1. A hybrid control architecture for interleaved parallel Boost converters, said architecture comprising a first power MOSFET-S1A second power MOSFET-S2A third power MOSFET-S3Fourth power MOSFET-S4Coupled inductor L1Coupled inductor L2Capacitor C0And a resistance RL(ii) a Input terminal VinRespectively with a coupling inductor L1End of same name and coupling inductance L2Are connected with the same name end of the coupling inductor L1The different name terminal of the first power MOSFET-S is respectively connected with the first power MOSFET-S1And a second power MOSFET-S2Is connected to the source electrode ofL of combined inductor2The different name terminal is respectively connected with the third power MOSFET-S3And a fourth power MOSFET-S4Is connected to the source of the second power MOSFET-S2Respectively with a fourth power MOSFET-S4Drain electrode of (1), capacitor C0And a resistor RLIs connected with the negative electrode of the input end respectively with the first power MOSFET-S1Source of the third power MOSFET-S3Source electrode and capacitor C0Another terminal of (1) and a resistor RLThe other ends of the two are connected.
2. The hybrid control method of the hybrid control structure of the interleaved Boost converters according to claim 1, wherein the hybrid control method is specifically that the upper and lower tubes of the two branches are complementarily conducted;
first power MOSFET-S1And a third power MOSFET-S3The duty ratio of (D) is represented by D, and the output voltage is stabilized by pulse width modulation PWM;
when the input current has a low effective value, a control strategy that two Boost circuits have a phase difference of 180 degrees is adopted;
when the input current has high effective value, two paths of phase-shifting alpha are adopted2And T.
3. The hybrid control method of the hybrid control structure of the interleaved parallel Boost converters according to claim 2, wherein the control strategy of phasing difference of 180 ° between the two Boost circuits is that in one cycle, eight modes exist, namely, a switch mode 1[ t ] t0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]And obtaining expressions of the inductive current ripple and the input current ripple based on the eight modes:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
4. The hybrid control method of the hybrid control structure of the interleaved parallel Boost converter according to claim 2, wherein the two-way phase shift α is2The control strategy for T is specifically that in one cycle, eight modes exist, namely a switch mode 1[ T [ ]0～t1]Switched mode 2 t1～t2]Switched mode 3 t2～t3]Switched mode 4 t3～t4]Switched mode 5t4～t3]Switched mode 6 t4～t3]Switched mode 7 t2～t1]And switching mode 1[ t ]1～t0]When S is3Phase shift alpha2，α2T＝t3-T/2>DT-0.5T, and the inductance current ripple is calculated as follows:
the input current ripple is:
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, k is the coupling coefficient of the coupling inductor, Δ iLFor inductor current ripple, Δ i is input current ripple.
5. Hybrid control method of hybrid control structure of interleaved parallel Boost converters according to claim 2Method characterized by the fact that when D>At 0.5, α2T>D-0.5。
6. The method for optimally designing the coupling inductance of the hybrid control method of the hybrid control structure of the interleaved parallel Boost converter according to claim 2 is characterized in that parameters can meet the requirement of the ripple ratio of input current and the requirement of reverse zero crossing of inductor current under the control of phasing and phase shifting, and the parameters are obtained through constraint conditions.
7. The coupling inductance optimal design method according to claim 6, wherein the constraint conditions specifically include that an input current ripple coefficient meets a ripple ratio requirement; the inductive current has reverse moment to ensure ZVS turn-on characteristic of the main switching tube; under the phase-shift control strategy of hybrid control, the topology still ensures that the ripple coefficient of the input current meets the ripple ratio requirement and the inductive current has reverse time so as to ensure the ZVS switching-on characteristic of the main switching tube.
8. The method according to claim 7, wherein the constraint condition is expressed as,
where D is the duty cycle of the converter, VoFor the output voltage, T is the switching period, λ is the input current ripple ratio requirement, IinFor effective value of input current, Δ ILFor inductor current ripple, P is the converter output power.
9. The method according to claim 8, wherein the constraint conditions are expressed in such a way that when D <0.5, the corresponding inequality constraint conditions are similar.
10. The coupling inductance optimal design method according to claim 9, wherein the expression of the constraint condition is that a parameter range of the coupling inductance is obtained through the constraint condition, a group of parameters is selected from the parameter range of the coupling inductance and is checked, and if the check result is correct, the group of parameters is parameters capable of meeting the requirements of the ripple ratio of the input current and the reverse zero crossing of the inductor current; if the verification result is incorrect, another group of parameters is selected for verification until the verification result is correct.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110545514.5A CN113285598B (en) | 2021-05-19 | 2021-05-19 | Hybrid control structure and method for interleaved Boost converters |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110545514.5A CN113285598B (en) | 2021-05-19 | 2021-05-19 | Hybrid control structure and method for interleaved Boost converters |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN113285598A true CN113285598A (en) | 2021-08-20 |
| CN113285598B CN113285598B (en) | 2022-11-25 |
Family
ID=77279835
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202110545514.5A Active CN113285598B (en) | 2021-05-19 | 2021-05-19 | Hybrid control structure and method for interleaved Boost converters |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN113285598B (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN113992006A (en) * | 2021-10-27 | 2022-01-28 | 福州大学 | Control strategy and parameter calculation method of high-power three-level DC-DC converter |
Citations (19)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102122885A (en) * | 2011-03-11 | 2011-07-13 | 浙江大学 | Method for controlling alternated phase shift PWM (pulse width modulation) wave |
| US20120049894A1 (en) * | 2010-04-20 | 2012-03-01 | Rf Micro Devices, Inc. | Dc-dc converter current sensing |
| CN103944397A (en) * | 2014-04-11 | 2014-07-23 | 燕山大学 | Boost type isolated DC/DC converter and control method thereof |
| CN104506040A (en) * | 2014-09-22 | 2015-04-08 | 北京理工大学 | Dual-PWM and phase-shifting control method with the same duty ratio |
| CN104539164A (en) * | 2014-12-31 | 2015-04-22 | 北京理工大学 | Current type bidirectional DC-DC converter unequal-width PWM and double phase-shifting control method |
| CN105305829A (en) * | 2015-09-25 | 2016-02-03 | 北京理工大学 | Current type one-way DC-DC converter and symmetrical double PWM plus phase-shift control method |
| CN206211844U (en) * | 2016-12-06 | 2017-05-31 | 安徽工程大学 | The new two-way DC/DC converters of crisscross parallel |
| CN106849681A (en) * | 2017-04-11 | 2017-06-13 | 厦门大学 | A kind of high-gain isolated active clamping Sofe Switch DC DC converters |
| CN107370386A (en) * | 2017-08-04 | 2017-11-21 | 北京理工大学 | The optimal dutycycle voltage of the two-way DC DC converters of current mode mismatches control method |
| CN107395015A (en) * | 2017-08-08 | 2017-11-24 | 哈尔滨工业大学 | A kind of low ripple Sofe Switch synchronous rectification Buck converters based on coupling inductance |
| CN107517003A (en) * | 2017-08-31 | 2017-12-26 | 江苏大学 | One kind output inputs high-gain Boost translation circuits and switching method in parallel floatingly |
| CN208820683U (en) * | 2018-09-25 | 2019-05-03 | 河北科技大学 | Fuel cell crisscross parallel DC-DC converter |
| CN109980908A (en) * | 2019-03-22 | 2019-07-05 | 天津大学 | Method for multi-inverter parallel harmonics restraint under decentralised control |
| CN110445392A (en) * | 2019-08-09 | 2019-11-12 | 东北大学 | A kind of novel interleaving shunt-wound two-transistor forward power converter and its modulation strategy |
| CN110729899A (en) * | 2019-11-01 | 2020-01-24 | 山东大学 | Wide-input wide-output three-phase high-gain direct current converter and control method |
| CN110912407A (en) * | 2019-11-19 | 2020-03-24 | 中国船舶重工集团公司第七一九研究所 | Novel wide-range high-frequency direct current conversion device |
| CN111541373A (en) * | 2020-05-18 | 2020-08-14 | 哈尔滨工业大学 | Control method of two-phase parallel synchronous rectification Boost converter based on forward coupling inductor |
| US20200287461A1 (en) * | 2017-12-29 | 2020-09-10 | Huazhong University Of Science And Technology | Dc-dc converter |
| CN111725993A (en) * | 2020-06-17 | 2020-09-29 | 辽宁工程技术大学 | High-efficiency Sepic soft switch converter and control method thereof |
-
2021
- 2021-05-19 CN CN202110545514.5A patent/CN113285598B/en active Active
Patent Citations (19)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US20120049894A1 (en) * | 2010-04-20 | 2012-03-01 | Rf Micro Devices, Inc. | Dc-dc converter current sensing |
| CN102122885A (en) * | 2011-03-11 | 2011-07-13 | 浙江大学 | Method for controlling alternated phase shift PWM (pulse width modulation) wave |
| CN103944397A (en) * | 2014-04-11 | 2014-07-23 | 燕山大学 | Boost type isolated DC/DC converter and control method thereof |
| CN104506040A (en) * | 2014-09-22 | 2015-04-08 | 北京理工大学 | Dual-PWM and phase-shifting control method with the same duty ratio |
| CN104539164A (en) * | 2014-12-31 | 2015-04-22 | 北京理工大学 | Current type bidirectional DC-DC converter unequal-width PWM and double phase-shifting control method |
| CN105305829A (en) * | 2015-09-25 | 2016-02-03 | 北京理工大学 | Current type one-way DC-DC converter and symmetrical double PWM plus phase-shift control method |
| CN206211844U (en) * | 2016-12-06 | 2017-05-31 | 安徽工程大学 | The new two-way DC/DC converters of crisscross parallel |
| CN106849681A (en) * | 2017-04-11 | 2017-06-13 | 厦门大学 | A kind of high-gain isolated active clamping Sofe Switch DC DC converters |
| CN107370386A (en) * | 2017-08-04 | 2017-11-21 | 北京理工大学 | The optimal dutycycle voltage of the two-way DC DC converters of current mode mismatches control method |
| CN107395015A (en) * | 2017-08-08 | 2017-11-24 | 哈尔滨工业大学 | A kind of low ripple Sofe Switch synchronous rectification Buck converters based on coupling inductance |
| CN107517003A (en) * | 2017-08-31 | 2017-12-26 | 江苏大学 | One kind output inputs high-gain Boost translation circuits and switching method in parallel floatingly |
| US20200287461A1 (en) * | 2017-12-29 | 2020-09-10 | Huazhong University Of Science And Technology | Dc-dc converter |
| CN208820683U (en) * | 2018-09-25 | 2019-05-03 | 河北科技大学 | Fuel cell crisscross parallel DC-DC converter |
| CN109980908A (en) * | 2019-03-22 | 2019-07-05 | 天津大学 | Method for multi-inverter parallel harmonics restraint under decentralised control |
| CN110445392A (en) * | 2019-08-09 | 2019-11-12 | 东北大学 | A kind of novel interleaving shunt-wound two-transistor forward power converter and its modulation strategy |
| CN110729899A (en) * | 2019-11-01 | 2020-01-24 | 山东大学 | Wide-input wide-output three-phase high-gain direct current converter and control method |
| CN110912407A (en) * | 2019-11-19 | 2020-03-24 | 中国船舶重工集团公司第七一九研究所 | Novel wide-range high-frequency direct current conversion device |
| CN111541373A (en) * | 2020-05-18 | 2020-08-14 | 哈尔滨工业大学 | Control method of two-phase parallel synchronous rectification Boost converter based on forward coupling inductor |
| CN111725993A (en) * | 2020-06-17 | 2020-09-29 | 辽宁工程技术大学 | High-efficiency Sepic soft switch converter and control method thereof |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN113992006A (en) * | 2021-10-27 | 2022-01-28 | 福州大学 | Control strategy and parameter calculation method of high-power three-level DC-DC converter |
| CN113992006B (en) * | 2021-10-27 | 2023-11-17 | 福州大学 | Control method of high-power three-level DC-DC converter |
Also Published As
| Publication number | Publication date |
|---|---|
| CN113285598B (en) | 2022-11-25 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN110932557B (en) | High-gain quasi-resonant DC-DC converter based on voltage doubling rectifying circuit | |
| Li et al. | A family of interleaved DC–DC converters deducedfrom a basic cell with winding-cross-coupledinductors (WCCIs) for high step-upor step-down conversions | |
| CN110707931A (en) | LLC resonant converter and control method | |
| CN111541373B (en) | Control method of two-phase parallel synchronous rectification Boost converter based on forward coupling inductor | |
| CN109450260B (en) | Capacitor series connection type interleaving parallel flyback circuit | |
| CN111224553A (en) | Improved bidirectional half-bridge three-level LLC direct-current converter and synchronous control method thereof | |
| CN113346750A (en) | Soft switching in-phase buck-boost converter based on coupling inductor and control method | |
| Gao et al. | A novel DCM soft-switched SEPIC-based high-frequency converter with high step-up capacity | |
| CN113285598B (en) | Hybrid control structure and method for interleaved Boost converters | |
| CN114825968A (en) | Asymmetric duty ratio and internal phase shift hybrid control method | |
| CN109980931B (en) | Method for eliminating dead zone of non-reverse Buck-Boost converter operation | |
| CN111987918B (en) | Bidirectional DC-DC soft switch control method | |
| CN106130357A (en) | A kind of novel LCCL controlled resonant converter | |
| CN112003474B (en) | Cascade Buck-Boost high-gain converter | |
| CN113992024B (en) | Power segmentation modulation method based on bridge arm multiplexing type isolation type DC-DC converter | |
| CN113746324B (en) | High-gain soft switch Buck-Boost converter | |
| CN113691123B (en) | Zero-voltage turn-off zero-current turn-on high-gain Zeta converter | |
| CN202353464U (en) | Three-level boost soft switching converter circuit | |
| CN108631584A (en) | A kind of DC-DC power converters | |
| CN114142729A (en) | Staggered high-voltage-reduction-ratio converter with coupling inductor | |
| CN114070037A (en) | High-gain CUK converter and magnetic integration method thereof | |
| CN208522652U (en) | A kind of DC-DC converter | |
| Yao et al. | Zero input current ripple DC-DC converter with high step-up and soft-switching characteristics | |
| Ortatepe et al. | Coupled inductor based buck PFC converter for elimination of dead zones | |
| CN113691124B (en) | Zero-voltage turn-off zero-current turn-on high-gain Cuk converter |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |