CN113141121A - Current source type high-frequency isolation matrix type cascade converter and control method - Google Patents

Current source type high-frequency isolation matrix type cascade converter and control method Download PDF

Info

Publication number
CN113141121A
CN113141121A CN202110437779.3A CN202110437779A CN113141121A CN 113141121 A CN113141121 A CN 113141121A CN 202110437779 A CN202110437779 A CN 202110437779A CN 113141121 A CN113141121 A CN 113141121A
Authority
CN
China
Prior art keywords
current
current source
matrix converter
frequency
voltage
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
CN202110437779.3A
Other languages
Chinese (zh)
Other versions
CN113141121B (en
Inventor
王政
吴佳丽
徐阳
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Southeast University
Original Assignee
Southeast University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Southeast University filed Critical Southeast University
Priority to CN202110437779.3A priority Critical patent/CN113141121B/en
Publication of CN113141121A publication Critical patent/CN113141121A/en
Application granted granted Critical
Publication of CN113141121B publication Critical patent/CN113141121B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
    • H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
    • H02M5/22—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/275—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/293—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
    • H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
    • H02M5/22—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/225—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode comprising two stages of AC-AC conversion, e.g. having a high frequency intermediate link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/20—Climate change mitigation technologies for sector-wide applications using renewable energy

Abstract

The invention discloses a current source type high-frequency isolation matrix type cascade converter and a control method, comprising the following steps: after being filtered by LC, the three-phase AC power grid is respectively connected with two current source type matrix converters to transmit electric energy; the input end of the primary side of the high-frequency isolation transformer is connected with the matrix converter in series, the output end of the secondary side of the high-frequency isolation transformer is connected with the uncontrolled rectifier bridge, the alternating current sides of the two sets of current source type high-frequency isolation matrix type cascade converters are connected in parallel and share the LC filter, and the direct current output sides of the two sets of current source type high-frequency isolation matrix type cascade converters are connected with the resistance-inductance load in series. In the system, power frequency alternating current is converted into a high-frequency alternating current form through a three-phase/single-phase direct matrix converter, a direct current bus capacitor is removed, the system fault rate is reduced, the power conversion stage number is reduced, and the system volume is greatly reduced. The control technical scheme of the invention enables the switch of the matrix converter device to be flexible, inhibits the problem of electromagnetic interference caused by high-frequency operation of the switch device, meets the industrial requirements of offshore wind power generation, electric vehicle charging and the like, and has wide application prospect.

Description

Current source type high-frequency isolation matrix type cascade converter and control method
Technical Field
The invention relates to the field of wind power generation, in particular to a current source type high-frequency isolation matrix type cascade converter and a control method.
Background
Wind energy (onshore and offshore) is one of the most important energy sources in renewable energy power generation systems, and at present, with the continuous development of offshore wind power generation, the power of an electric energy transmission system is continuously increased, and the size of a corresponding converter is also continuously enlarged. A power frequency transformer in a wind energy system of a traditional permanent magnet synchronous generator is large in size, and the traditional generator in the wind energy system mainly comprises a fan impeller, a three-level gear box, a synchronous generator, a bidirectional PWM rectifier and the like. However, the conventional transformer in such a system has a large weight and volume, a small power capacity, and poor power transmission controllability, and because the transformer has a solid iron core and copper windings, the maintenance cost is high, and the manufacturing cost is expensive, so that the modular production and assembly of the wind power system are not facilitated. The transmission power of the wind power system is continuously increased, and the requirements on the power electronic converter system are higher and higher. When a certain phase in the power circuit breaks down, the transmission of other bridge arms is influenced, and the normal operation of the whole system is further influenced. For the above reasons, research on power electronic converters has been a hot spot in offshore wind power generation high voltage direct current transmission systems.
The cascade structure is a promising offshore wind farm configuration and can replace expensive and huge offshore substations. The cascade structure based on the modular direct matrix converter is very suitable for a high-voltage direct-current transmission system with a current source converter due to the characteristics of high power output and high dynamic response. A high-frequency isolation transformer is adopted to replace a low-frequency input transformer, and an electrolytic capacitor which is large in size and easy to damage in a wind energy conversion system is eliminated. Therefore, the power density, reliability and transmission efficiency of the system can be improved. In addition, the direct matrix converter can also realize single-stage power conversion and soft switching, and further reduce the loss of the system. The modular design of the direct matrix converter facilitates the installation and maintenance of the system, which is of great importance to offshore wind farms, and the cost reduction is beneficial to the large-scale application of offshore wind farms. Finally, the series configuration eliminates an offshore substation and facilitates the pooling of multiple low voltage wind power generation systems into a medium voltage grid. And a proper power device can be selected, and the advantages of high reliability, controllable power factor, high waveform quality and the like are realized.
A great part of loss of the high-frequency isolation matrix type cascade converter comes from switching loss of a switching device, and the application of the current conversion soft switching technology of the bidirectional switch of the matrix converter can greatly reduce the switching loss of the device and improve the transmission efficiency of the converter. The traditional matrix converter soft switching technology is mostly specific to a voltage source type converter, the current source type soft switching technology is not researched much, and the defect of large loss of the current source type matrix converter can be effectively overcome by adopting the soft switching technology. An active damping harmonic suppression scheme is adopted in a control strategy, so that low-order harmonics in a cascade system are effectively suppressed, and harmonic loss of the system is reduced.
Disclosure of Invention
In order to solve the defects mentioned in the background technology, the invention aims to provide a current source type high-frequency isolation matrix type cascade converter and a control method, the input sides of two current source type high-frequency isolation matrix converters are connected in parallel and the output sides of the two current source type high-frequency isolation matrix converters are connected in series, the power conversion capacity of a system is improved, the reliability of the system is enhanced, and the volume of the system is reduced; by applying the soft switching technology, the switching loss of the current source type high-frequency isolation matrix converter is reduced, and the efficiency of the system is improved.
The purpose of the invention can be realized by the following technical scheme:
a current source type high-frequency isolation matrix type cascade converter comprises a three-phase power grid, an LC filter, a first current source type matrix converter, a second current source type matrix converter, a first leakage inductor, a second leakage inductor, a first high-frequency isolation transformer, a second high-frequency isolation transformer, a first uncontrolled rectifier bridge, a second uncontrolled rectifier bridge, a first inductance resistance load and a second inductance resistance load, wherein the three-phase power grid is connected with the LC filter; the LC filter is respectively connected with the first current source type matrix converter and the second current source type matrix converter in parallel; the alternating current output side of the first current source type matrix converter is connected with the first high-frequency isolation transformer in series, and the alternating current output side of the second current source type matrix converter is connected with the second high-frequency isolation transformer in series; the secondary output side of the first high-frequency isolation transformer is connected with the first uncontrolled rectifier bridge in series, and the secondary output side of the second high-frequency isolation transformer is connected with the second uncontrolled rectifier bridge in series, so that energy is transferred through a magnetic field; the first uncontrolled rectifier bridge output end and the second uncontrolled rectifier bridge output end are connected in series through a first resistance-inductance load and a second resistance-inductance load to form a closed loop.
Further, the power direction and the power magnitude of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the first current source type matrix converter and the second current source type matrix converter; the first current source type matrix converter and the second current source type matrix converter are respectively connected with ports of the LC filter circuit and receive electric energy transmitted by the three-phase power grid; the current of the first resistance-inductance load and the current of the direct current bus inductor corresponding to the second resistance-inductance load output by the direct current side are controlled by a current loop.
Further, the control method adopted by the control module of the first current source type matrix converter comprises the following processes:
1) capacitor voltage U passing through three-phase LC filterabcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining a capacitance voltage d-axis component U of the LC filter through coordinate transformationdAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqObtaining the low-frequency component of the capacitor voltage through a low-pass filter, the electrical angle thetaeObtaining the electrical angular velocity omega after differentiationeAnd calculating to obtain the low-frequency capacitance current of the filter capacitor
Figure BDA0003033822870000021
And
Figure BDA0003033822870000022
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is obtained by a D-axis direct current component through a PI controller
Figure BDA0003033822870000023
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgObtaining the q-axis current component after calculation
Figure BDA0003033822870000024
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpvAnd kpvObtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000031
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000032
And
Figure BDA0003033822870000033
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion1dc *And phase angle thetaα1；
6) Phase current fundamental wave peak value I1dc *Divided by a given value of DC current Idc *Obtaining a modulation ratio m of the first current source type matrix converter1iPhase angle thetaα1Plus the electrical angle theta measured by the phase-locked loopeObtaining the switching pulse phase angle theta of the matrix converter1ωUsing the modulation ratio m1iAnd angle theta1ωAnd the switching period Ts generates twelve switching pulses of the first current source matrix converter.
Further, the control method adopted by the control module of the second current source type matrix converter comprises the following processes:
1) capacitor voltage U passing through three-phase LC filterabcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining LC filter by coordinate transformationCapacitor voltage d-axis component U of wave filterdAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqObtaining the low-frequency component of the capacitor voltage through a low-pass filter, the electrical angle thetaeObtaining the electrical angular velocity omega after differentiationeThe low-frequency capacitance current of the filter capacitor can be obtained after calculation
Figure BDA0003033822870000034
And
Figure BDA0003033822870000035
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is obtained by a D-axis direct current component through a PI controller
Figure BDA0003033822870000036
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgObtaining the q-axis current component after calculation
Figure BDA0003033822870000037
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpvAnd kpvObtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000038
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000039
And
Figure BDA00030338228700000310
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion2dc *And phase angle thetaα2；
6) Phase current fundamental wave peak value I2dc *Divided by a given value of DC current Idc *Obtaining a modulation ratio m of the second current source type matrix converter2iPhase angle thetaα2Plus the electrical angle theta measured by the phase-locked loopeObtaining the switching pulse phase angle theta of the matrix converter2ωUsing the modulation ratio m2iAnd angle theta2ωAnd the switching period Ts generates twelve switching pulses of the second current source matrix converter.
Further, the modulation method adopted by the first current source type matrix converter and the second current source type matrix converter comprises the following steps:
three current vectors acting on the first current source type matrix converter in one switching period are respectively I11，I12，I10The output voltage of the corresponding first current source type matrix converter is U11，U12，U10By changing the order of action of the current vectors, U is made11>U12>U10(ii) a Similarly, the three current vectors acting on the second current source type matrix converter in one switching period are respectively I21，I22，I20Corresponding to the input voltage of the second current source type matrix converter as U21，U22，U20By changing the order of action of the current vectors, U is made21>U22>U20(ii) a Because the first current source matrix converter and the second current source matrix converter are connected in parallel, the output voltage waveforms of the matrix converters in a switching period are consistent, and the working states are consistent, taking the working state of the first current source matrix converter as an example, the specific working process of the soft switch in a switching period is as follows,
1) state 0: primary side commutation
When a switching period begins, the current vectors corresponding to the first current source type matrix converter and the second current source type matrix converter are zero vectors I7Output voltage v of matrix converterpIs reduced to 0;
2) state 1: conduction time of switch tube
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer are equal, and the primary side current of the first high-frequency isolation transformer flows through a switching tube S of the first current source type matrix converter11、S22And S14、S24The current of the secondary side rectifier bridge of the first high-frequency isolation transformer flows through D1、D4Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S11And S14Conducting at zero voltage;
3) state 2: conduction time of switch tube
Zero vector I of first current source type matrix converter7End of action, current vector I1In operation, the inductive current of the first uncontrolled rectifier bridge flows through the diode D1、D4According to the four-step commutation, the switching tube S of the first current source type matrix converter11And S14Is turned off because the voltage u is nowabGreater than 0, current to S16And S26The output capacitor of the first current source type matrix converter16And S26Zero voltage conduction is carried out, and the voltage of the input side of the first high-frequency isolation transformer is equal to uabEnergy flows from the grid;
4) state 3: conduction time of switch tube
Similar to the operating state of state 2, the effective current vector I of the first current source matrix converter1End of action, current vector I2In operation, the inductive current of the first uncontrolled rectifier bridge flows through the diode D1、D4According to the four-step commutation, the power switch tube S of the first current source type matrix converter16And S26Closing, S12And S22Zero voltage conduction is carried out, and the voltage of the input side of the first high-frequency isolation transformer is equal to uacEnergy flows from the grid;
5) and 4: commutation of uncontrolled rectifier bridge
The rectification of the first uncontrolled rectifier bridge is completed with the aid of the first current source matrix converter, at t3At the moment, the diode D of the first uncontrolled rectifier bridge2、D3Zero current conduction. The current flowing through the inductor in the first uncontrolled rectifier bridge decreases linearly through D2、D3Through D1、D4Is linearly decreased at the same rate, the commutation overlap time TdSelecting 100ns, and commutating current on the first uncontrolled rectifier bridge inductor before the mode is finished;
6) and state 5: conduction time of switch tube
Commutation overlap time TdAfter that, the power switch tube S of the first current source type matrix converter12And S22Off, S14And S24Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer is 0, and no direct current energy is transmitted under the mode;
7) and 6: conduction time of switch tube
First current source type matrix converter current vector I2End of action, zero vector I0In operation, a direct current flows through the diode D of the first uncontrolled rectifier bridge on the load side2And D3While the input side current of the first high-frequency isolation transformer flows through the power switch tube S of the first current source type matrix converter11、S21And S24、S14And the three-phase capacitor in the LC filter provides a current channel for the three-phase inductor.
The invention has the beneficial effects that:
(1) the direct conversion from power frequency alternating current to high frequency alternating current is realized through the current source type matrix converter, the direct current capacitor is removed, the failure rate of the converter is reduced, the power conversion stage number is reduced, and the reliability and the power density of system operation are greatly improved;
(2) a high-frequency isolation transformer is adopted to replace a low-frequency input transformer, so that the system volume is reduced;
(3) the capacity of the power converter is increased through the cascade structure of the plurality of direct matrix converters, the output voltage of the system is improved, and the collection of medium-high voltage direct current transmission is facilitated;
(4) the control strategy of the matrix converter can realize unit power factor operation, the soft switching technology reduces the loss of the system, the modularized matrix converter is easy to install and maintain, and the cost of the offshore wind power generation system is reduced.
Drawings
The invention will be further described with reference to the accompanying drawings.
FIG. 1 is an overall architecture diagram of the present invention;
FIG. 2 is a schematic block diagram of a control method for two current source matrix converters according to the present invention;
FIG. 3(a) is a schematic diagram of the output voltage corresponding to the current vector of the first current source type matrix converter according to the present invention (v)ab＞vac)；
FIG. 3(b) is a schematic diagram of the output voltage corresponding to the current vector of the second current source type matrix converter according to the present invention (v)ac＞vab)；
Fig. 4 is a schematic diagram of current flow paths of two current source matrix converters based on the space current vector modulation method in the operating mode of the present invention, wherein fig. 4(a) -fig. 4(f) are schematic diagrams of current flow paths in the operating modes of state 1-state 6, respectively;
FIG. 5.1(a) shows the input voltage V of the current source matrix converter of the present inventionpAnd an output voltage VsAn experimental oscillogram;
FIG. 5.1(b) is a diagram showing an input voltage V of the current source matrix converter of the present inventionpAnd current ipAn experimental oscillogram;
fig. 5.2 is a simulation waveform diagram of the current source type high frequency isolation matrix type cascaded converter of the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
As shown in fig. 1, a current source type high frequency isolation matrix type cascaded converter includes a three-phase power grid 1.1, an LC filter 1.2, two sets of matrix converters (1.3, 1.4), leakage inductances (1.5, 1.6), high frequency isolation transformers (1.7, 1.8), two sets of uncontrolled rectifier bridges (1.9, 1.10), and resistance-inductance loads (1.11, 1.12).
The three-phase power grid 1.1 is connected with the LC filter 1.2;
the LC filter 1.2 is respectively connected with the first matrix converter 1.3 and the second matrix converter 1.4 in parallel; the matrix converters (1.3 and 1.4) are respectively connected with high-frequency isolation transformers (1.7 and 1.8); the secondary sides of the high-frequency isolation transformers (1.7 and 1.8) are connected with uncontrolled rectifier bridges (1.9 and 1.10) and transmit energy through magnetic fields; the uncontrolled rectifier bridges (1.9, 1.10) are respectively connected with a resistance-inductance load 1.11 and a resistance-inductance load 1.12; the inductance-resistance load 1.11 and the inductance-resistance load 1.12 on the direct current output side are mutually connected in series to form a closed loop;
the matrix converters 1.3 and 1.4 at the input side of the high-frequency isolation transformer are connected in parallel;
direct current bus inductors 1.11 and 1.12 at the output side of the high-frequency isolation transformer are mutually connected in series;
the power direction and the power magnitude of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the matrix converters 1.3 and 1.4;
the current of the direct current side output resistance-inductance loads 1.11 and 1.12 corresponding to the direct current bus inductor is controlled by a current loop.
The current source type matrix converter comprises a first current source type matrix converter 1.3 and a second current source type matrix converter 1.4, the high-frequency isolation transformer comprises a first high-frequency isolation transformer 1.7 and a second high-frequency isolation transformer 1.8, the rectifying circuit comprises a first uncontrolled rectifier bridge 1.9 and a second uncontrolled rectifier bridge 1.10, wherein:
the alternating current output side of the first current source type matrix converter 1.3 is connected with a first high-frequency isolation transformer 1.7 in series;
the alternating current output side of the second current source type matrix converter 1.4 is connected with a second high-frequency isolation transformer 1.8 in series;
the alternating current input side of the first current source type matrix converter 1.3 is connected with the alternating current input side of the second current source type matrix converter 1.4 in parallel;
the output end of the secondary side of the first high-frequency isolation transformer 1.7 is connected with a first uncontrolled rectifier bridge 1.9 in series;
the output end of the secondary side of the second high-frequency isolation transformer 1.9 is connected with a second uncontrolled rectifier bridge 1.10 in series;
the output side of the first uncontrolled rectifier bridge 1.9 and the output side of the second uncontrolled rectifier bridge 1.10 are connected in series through two sets of inductance-resistance loads 1.11 and 1.12 to form a closed loop;
the first current source type matrix converter 1.3 and the second current source type matrix converter 1.4 are respectively connected with the port of the three-phase LC filter circuit 1.2 and receive electric energy transmitted by a three-phase power grid.
As shown in fig. 2, 5.1 and 5.2, the current closed-loop control method adopted by the control strategy of the first current source type matrix converter 1.3 comprises the following steps:
1) the capacitor voltage U passing through the three-phase LC filter 1.21abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.11dAnd q-axis component U1q；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter 2.2eObtaining the electrical angular velocity omega after differentiating by 2.3eAnd 2.6, the low-frequency capacitance current of the filter capacitor can be obtained
Figure BDA0003033822870000071
And
Figure BDA0003033822870000072
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by the PI controller 2.5 to obtain the d-axis direct current component
Figure BDA0003033822870000073
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after 2.4 calculation
Figure BDA0003033822870000074
Is zero;
4) the dq axis component of the capacitor voltage passes through a high-pass filter 2.7 to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.8) obtaining dq axis component values of the virtual current (2.9), wherein the fifth harmonic and the seventh harmonic of the system current can be eliminated through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000075
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000076
And
Figure BDA0003033822870000077
obtaining the given value of the final current at the input side of the matrix converter through operation, and obtaining the phase current fundamental wave peak value I after the conversion of polar coordinates 2.171dc *And phase angle thetaα1；
6) Phase current fundamental wave peak value I1dc *Divided by a given value of DC current Idc *2.18 obtaining modulation ratio m of current source type matrix converter1iPhase angle thetaα1Plus the electrical angle theta measured by the phase-locked loope2.19 obtaining the switching pulse phase angle theta of the matrix converter1ωUsing the modulation ratio m1iAnd angle theta1ωAnd a switching period TsTwelve switching pulses of the first current source matrix converter 1.3 are generated.
As shown in fig. 2, the dc current control method adopted by the control strategy of the second current source type matrix converter 1.4 includes the following steps:
1) the capacitor voltage U passing through the three-phase LC filter 1.2abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.10dAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter 2.12eObtaining the electrical angular velocity omega after differentiating by 2.11eAnd 2.13 calculation is carried out to obtain the low-frequency capacitance current of the filter capacitor
Figure BDA0003033822870000078
And
Figure BDA0003033822870000079
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by the PI controller 2.5 to obtain the d-axis direct current component
Figure BDA00030338228700000710
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after 2.4 calculation
Figure BDA00030338228700000711
Is zero;
4) the dq axis component of the capacitor voltage passes through a high pass filter 2.14 to obtain capacitor electricityHigh frequency component of voltage UhdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.15) (2.16) obtaining dq axis component values of the virtual current, wherein the fifth harmonic and the seventh harmonic of the system current can be eliminated through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000081
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000082
And
Figure BDA0003033822870000083
obtaining the given value of the final current at the input side of the matrix converter through operation, and obtaining the phase current fundamental wave peak value I after the conversion of polar coordinates 2.202dc *And phase angle thetaα2；
6) Phase current fundamental wave peak value I2dc *Divided by a given value of DC current Idc *2.21 obtaining modulation ratio m of Current Source type matrix converter2iPhase angle thetaα2Plus the electrical angle theta measured by the phase-locked loope2.22 obtaining the switching pulse phase angle theta of the matrix converter2ωUsing the modulation ratio m2iAnd angle theta2ωAnd a switching period TsTwelve switching pulses of the second current source matrix converter 1.4 are generated.
The space vector modulation method of the matrix converters 1.3 and 1.4 comprises the following steps:
for simplicity of analysis, the three current vectors acting on the first current source matrix converter 1.3 in a switching cycle are each I11，I12，I10Corresponding to the output voltage of the first current source type matrix converter 1.3 as U11，U12，U10By changing the order of action of the current vectors, U is made11>U12>U10(ii) a In the same way, the method for preparing the composite material,three current vectors acting on the second current source matrix converter 1.4 in one switching cycle are respectively I21，I22，I20Corresponding to the input voltage of the second current source type matrix converter 1.4 as U21，U22，U20By changing the order of action of the current vectors, U is made21>U22>U20. Since the first current source matrix converter 1.3 and the second current source matrix converter 1.4 are connected in parallel, the output voltage waveforms of the matrix converters in one switching period are consistent, and the operating states are consistent, taking the operating state of the first matrix converter as an example, the specific operating process of the soft switch in one switching period is as follows, it is not assumed that the first current source matrix converter 1.3 operates in the first sector, and at this time, the second current source matrix converter 1.4 also operates in the first sector.
As shown in fig. 3(a) (b) and fig. 4 (light color for off and dark color for on):
1) state 0: primary side commutation (t)0 -)
Just after the start of a switching cycle, the current vectors corresponding to the first current source matrix converter 1.3 and the second current source matrix converter 1.4 are both zero vectors I7Output voltage v of matrix converterpIs reduced to 0;
2) state 1 (3.1): switch tube on-time (t)0-t1)
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer 1.7 are equal, and the primary side current of the first high-frequency isolation transformer 1.7 flows through the switching tube S of the first current source type matrix converter 1.311、S22And S14、S24The current of the secondary side rectifier bridge of the first high-frequency isolation transformer 1.7 flows through D1、D4Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S11And S14Conducting at zero voltage;
3) state 2 (3.2): switch tube on-time (t)1-t2)
First current source type matrix converter 1Zero vector I of 37End of action, current vector I1In operation, the inductive current of the first uncontrolled rectifier bridge 1.9 flows through the diode D1、D4According to a four-step commutation, the switching tube S of the first current source matrix converter 1.311And S14Is turned off because the voltage u is nowabGreater than 0, current to S16And S26The output capacitor of the matrix converter is charged, and the matrix converter power switch tube S16And S26Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to uabEnergy flows from the grid;
4) state 3 (3.3): switch tube on-time (t)2-t3)
Similar to the operating state of state 2, the effective current vector I of the first current source matrix converter 1.31End of action, current vector I2In operation, the inductive current of the first uncontrolled rectifier bridge 1.9 flows through the diode D1、D4According to a four-step commutation, the switching tube S of the first current source matrix converter 1.316And S26Closing, S12And S22Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to uacEnergy flows from the grid;
5) state 4 (3.4): commutation of an uncontrolled rectifier bridge (t)3-t4)
The rectification of the first uncontrolled rectifier bridge 1.9 is completed with the aid of the first current source matrix converter 1.3, at t3At the moment, the first uncontrolled rectifier bridge 1.9 diode D2、D3Zero current conduction. The current flowing through the inductor of the first uncontrolled rectifier bridge 1.9 decreases linearly through D2、D3Through D1、D4Is linearly decreased at the same rate, commutation overlap time T in this contextdSelecting 100ns, and commutating current on an inductor 1.9 of the first uncontrolled rectifier bridge before the mode is finished;
6) state 5 (3.5): switch tube on-time (t)4-t5)
Commutation overlap time TdAfter the completion of the process, the operation,power switch tube S of first current source type matrix converter 1.312And S22Off, S14And S24Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer 1.7 is 0, and no direct current energy is transmitted under the mode;
7) state 6 (3.6): switch tube on-time (t)5-t6)
First current source matrix converter 1.3 current vector I2End of action, zero vector I0In operation, a direct current flows on the load side through the diode D of the first uncontrolled rectifier bridge 1.92And D3While the input side current of the first high-frequency isolation transformer 1.7 flows through the switching tube S of the first current source type matrix converter 1.311、S21And S24、S14And the three-phase capacitor in the LC filter 1.2 provides a current channel for the three-phase inductor.
The foregoing shows and describes the general principles, essential features, and advantages of the invention. It will be understood by those skilled in the art that the present invention is not limited to the embodiments described above, which are described in the specification and illustrated only to illustrate the principle of the present invention, but that various changes and modifications may be made therein without departing from the spirit and scope of the present invention, which fall within the scope of the invention as claimed.

Claims (5)

1. A current source type high-frequency isolation matrix type cascade converter comprises a three-phase power grid (1.1), an LC filter (1.2), a first current source type matrix converter (1.3) and a second current source type matrix converter (1.4), a first leakage inductor (1.5) and a second leakage inductor (1.6), a first high-frequency isolation transformer (1.7) and a second high-frequency isolation transformer (1.8), a first uncontrolled rectifier bridge (1.9) and a second uncontrolled rectifier bridge (1.10), a first inductance resistance load (1.11) and a second inductance resistance load (1.12), and is characterized in that the three-phase power grid (1.1) is connected with the LC filter (1.2); the LC filter (1.2) is respectively connected with the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) in parallel; the alternating current output side of the first current source type matrix converter (1.3) is connected with the first high-frequency isolation transformer (1.7) in series, and the alternating current output side of the second current source type matrix converter (1.4) is connected with the second high-frequency isolation transformer (1.8) in series; the secondary output side of the first high-frequency isolation transformer (1.7) is connected with the first uncontrolled rectifier bridge (1.9) in series, the secondary output side of the second high-frequency isolation transformer (1.8) is connected with the second uncontrolled rectifier bridge (1.10) in series, and energy is transferred through a magnetic field; the output end of the first uncontrolled rectifier bridge (1.9) and the output end of the second uncontrolled rectifier bridge (1.10) are connected in series through a first inductance resistance load (1.11) and a second inductance resistance load (1.12) to form a closed loop.
2. A current source high frequency isolated matrix type cascaded converter as claimed in claim 1, wherein the power direction and power magnitude of the current source high frequency isolated matrix type cascaded converter is determined by the control module of the first current source matrix converter (1.3) and the second current source matrix converter (1.4); the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) are respectively connected with ports of the LC filter circuit (1.2) and receive electric energy transmitted by a three-phase power grid; the current of the first resistance-inductance load (1.11) and the second resistance-inductance load (1.12) corresponding to the direct current bus inductor output by the direct current side is controlled by a current loop.
3. A current source high frequency isolated matrix type cascaded converter as claimed in claim 2, characterized in that said control module of said first current source matrix converter (1.3) adopts a control method comprising the following steps:
1) a capacitor voltage U passing through the three-phase LC filter (1.2)abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining a capacitance voltage d-axis component U of the LC filter (1.2) through coordinate transformation (2.1)dAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter (2.2)eObtaining the electrical angular velocity omega after differentiation (2.3)eAnd (2.6) calculating to obtain the low-frequency capacitance current of the filter capacitor
Figure FDA0003033822860000011
And
Figure FDA0003033822860000012
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by a PI controller (2.5) to obtain a d-axis direct current component
Figure FDA0003033822860000013
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after the calculation of (2.4)
Figure FDA0003033822860000014
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter (2.7) to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.8) and kpv(2.9) obtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure FDA0003033822860000021
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure FDA0003033822860000022
And
Figure FDA0003033822860000023
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after conversion of polar coordinates (2.17)1dc *And phase angle thetaα1；
6) Phase current fundamental wave peak value I1dc *Divided by a given value of DC current Idc *(2.18) obtaining a modulation ratio m of the first current source type matrix converter (1.3)1iPhase angle thetaα1Plus the electrical angle theta measured by the phase-locked loope(2.19) obtaining the switching pulse phase angle theta of the matrix converter1ωUsing the modulation ratio m1iAnd angle theta1ωAnd a switching period Ts generates twelve switching pulses of the first current source matrix converter (1.3).
4. A current source high frequency isolated matrix type cascaded converter as claimed in claim 2, wherein said control module of said second current source matrix converter (1.4) adopts a control method comprising the following steps:
1) a capacitor voltage U passing through the three-phase LC filter (1.2)abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining a capacitance voltage d-axis component U of the LC filter (1.2) through coordinate transformation (2.10)dAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter (2.12)eObtaining the electrical angular velocity omega after differentiation (2.11)eAnd the low-frequency capacitance current of the filter capacitor can be obtained after the calculation of (2.13)
Figure FDA0003033822860000024
And
Figure FDA0003033822860000025
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by a PI controller (2.5) to obtain a d-axis direct current component
Figure FDA0003033822860000026
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after the calculation of (2.4)
Figure FDA0003033822860000027
Is zero;
4) the dq component of the capacitor voltage is processed by a high-pass filter (2.14) to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.15) and kpv(2.16) obtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure FDA0003033822860000028
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure FDA0003033822860000029
And
Figure FDA00030338228600000210
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate (2.20) conversion2dc *And phase angle thetaα2；
6) Phase current fundamental wave peak value I2dc *Divided by a given value of DC current Idc *(2.21) obtaining a modulation ratio m of the second current source type matrix converter (1.4)2iPhase angle thetaα2Plus the electrical angle theta measured by the phase-locked loope(2.22) obtaining the switching pulse phase angle theta of the matrix converter2ωUsing the modulation ratio m2iAnd angle theta2ωAnd a switching period Ts generates twelve switching pulses of the second current source matrix converter (1.4).
5. A current source high frequency isolated matrix type cascaded converter as claimed in claim 1, wherein said first current source matrix converter (1.3) and said second current source matrix converter (1.4) employ a modulation method comprising the steps of:
three current vectors acting on the first current source matrix converter (1.3) in one switching cycle are respectively I11，I12，I10Corresponding to the output voltage of the first current source type matrix converter (1.3) being U11，U12，U10By changing the order of action of the current vectors, U is made11>U12>U10(ii) a Similarly, three current vectors acting on the second current source type matrix converter (1.4) in one switching period are respectively I21，I22，I20Corresponding to the input voltage of the second current source type matrix converter (1.4) being U21，U22，U20By changing the order of action of the current vectors, U is made21>U22>U20(ii) a Because the first current source matrix converter (1.3) and the second current source matrix converter (1.4) are connected in parallel, the output voltage waveforms of the matrix converters in a switching period are consistent, the working states are consistent, taking the working state of the first current source matrix converter (1.3) as an example, the specific working process of the soft switch in one switching period is as follows,
1) state 0: primary side commutation (t)0 -)
Just after the beginning of a switching period, the current vectors corresponding to the first current source matrix converter (1.3) and the second current source matrix converter (1.4) are all zero vectors I7Output voltage v of matrix converterpIs reduced to 0;
2) state 1 (3.1): switch tube on-time (t)0-t1)
Zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer (1.7) are equal, and the primary side current of the first high-frequency isolation transformer (1.7) flows through a switch tube S of the first current source type matrix converter (1.3)11、S22And S14、S24The current of the secondary side rectifier bridge of the first high-frequency isolation transformer (1.7) flows through D1、D4Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S11And S14Conducting at zero voltage;
3) state 2 (3.2): switch tube on-time (t)1-t2)
Zero vector I of a first current source matrix converter (1.3)7End of action, current vector I1In the beginning, the inductive current of the first uncontrolled rectifier bridge (1.9) flows through the diode D1、D4According to a four-step commutation, the switching tube S of the first current source matrix converter (1.3)11And S14Is turned off because the voltage u is nowabGreater than 0, current to S16And S26The output capacitor of the first current source type matrix converter (1.3)16And S26Zero voltage conduction is carried out when the input side voltage of the first high-frequency isolation transformer (1.7) is equal to uabEnergy flows from the grid;
4) state 3 (3.3): switch tube on-time (t)2-t3)
The effective current vector I of the first current source matrix converter (1.3) is similar to the operating state of state 21End of action, current vector I2In the beginning, the inductive current of the first uncontrolled rectifier bridge (1.9) flows through the diode D1、D4According to a four-step commutation, the power switch S of the first current source matrix converter (1.3)16And S26Closing, S12And S22Zero voltage conduction is carried out when the input side voltage of the first high-frequency isolation transformer (1.7) is equal to uacEnergy flows from the grid;
5) state 4 (3.4): commutation of an uncontrolled rectifier bridge (t)3-t4)
The rectification of the first uncontrolled rectifier bridge (1.9) is completed with the aid of the first current source matrix converter (1.3), at t3At the moment, the diode D of the first uncontrolled rectifier bridge (1.9)2、D3Zero current conduction. First uncontrolledThe current flowing through the inductor in the rectifier bridge (1.9) decreases linearly through D2、D3Through D1、D4Is linearly decreased at the same rate, the commutation overlap time TdSelecting 100ns, and commutating current on an inductor of the first uncontrolled rectifier bridge (1.9) before the mode is finished;
6) state 5 (3.5): switch tube on-time (t)4-t5)
Commutation overlap time TdAfter that, the power switch tube S of the first current source type matrix converter (1.3)12And S22Off, S14And S24Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer (1.7) is 0, and no direct current energy is transmitted under the mode;
7) state 6 (3.6): switch tube on-time (t)5-t6)
First current source type matrix converter (1.3) current vector I2End of action, zero vector I0In operation, a load-side direct current flows through the diode D of the first uncontrolled rectifier bridge (1.9)2And D3Meanwhile, the current at the input side of the first high-frequency isolation transformer (1.7) flows through the power switch tube S of the first current source type matrix converter (1.3)11、S21And S24、S14And the three-phase capacitor in the LC filter (1.2) provides a current channel for the three-phase inductor.
CN202110437779.3A 2021-04-22 2021-04-22 Current source type high-frequency isolation matrix type cascade converter and control method Active CN113141121B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202110437779.3A CN113141121B (en) 2021-04-22 2021-04-22 Current source type high-frequency isolation matrix type cascade converter and control method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202110437779.3A CN113141121B (en) 2021-04-22 2021-04-22 Current source type high-frequency isolation matrix type cascade converter and control method

Publications (2)

Publication Number Publication Date
CN113141121A true CN113141121A (en) 2021-07-20
CN113141121B CN113141121B (en) 2022-07-26

Family

ID=76813631

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202110437779.3A Active CN113141121B (en) 2021-04-22 2021-04-22 Current source type high-frequency isolation matrix type cascade converter and control method

Country Status (1)

Country Link
CN (1) CN113141121B (en)

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2023197547A1 (en) * 2022-04-14 2023-10-19 山东大学 Reliable parallel control system and method for efficient bidirectional power conversion module

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106953570A (en) * 2017-04-25 2017-07-14 南京福致通电气自动化有限公司 Energy feedback type elevator traction drive system control method based on matrix converter
CN107612386A (en) * 2017-09-01 2018-01-19 中国科学院近代物理研究所 Current mode Simplify matrix converter and its control method for coordinating
CN109039205A (en) * 2018-08-20 2018-12-18 东南大学 SiC device current source type double three-phase permanent-magnetic Synchromous machine drive system and method
CN109768729A (en) * 2019-03-25 2019-05-17 哈工大（张家口）工业技术研究院 Single-phase high frequency isolated form variable topology AC-DC converter

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106953570A (en) * 2017-04-25 2017-07-14 南京福致通电气自动化有限公司 Energy feedback type elevator traction drive system control method based on matrix converter
CN107612386A (en) * 2017-09-01 2018-01-19 中国科学院近代物理研究所 Current mode Simplify matrix converter and its control method for coordinating
CN109039205A (en) * 2018-08-20 2018-12-18 东南大学 SiC device current source type double three-phase permanent-magnetic Synchromous machine drive system and method
CN109768729A (en) * 2019-03-25 2019-05-17 哈工大（张家口）工业技术研究院 Single-phase high frequency isolated form variable topology AC-DC converter

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2023197547A1 (en) * 2022-04-14 2023-10-19 山东大学 Reliable parallel control system and method for efficient bidirectional power conversion module

Also Published As

Publication number Publication date
CN113141121B (en) 2022-07-26

Similar Documents

Publication Publication Date Title
CN101316074B (en) Back-to-back three-power level midpoint clamping current transformer of wind power generation system
CN203405559U (en) Electronic load of electric energy feedback type
CN102223090B (en) High-power simplified electrolytic and electroplating high-frequency switch power supply and control method thereof
CN105162350B (en) The three-phase micro inverter and its control method of a kind of wide loading range of high efficiency
CN110611435B (en) Topological structure of cascade flexible alternating current chain converter
CN104052079A (en) Electric energy feedback type electronic load
CN111277159A (en) Modular three-phase photovoltaic inverter and topological system thereof
CN104993713A (en) Control method for double PWM solid-state transformer
Meng et al. Optimal input and output power quality control of single-phase AC–DC–DC converter with significant DC-link voltage ripple
CN111682787A (en) Single-stage three-phase AC/DC converter based on isolation converter module and method
CN107579666B (en) Multifunctional hybrid power electronic transformer based on MMC matrix converter and control method
CN107017691A (en) Many charger parallel Operation Control methods of transformer station
CN202085085U (en) Large power efficient energy-consuming high frequency switch power supply
Zhang et al. Direct power flow controller—A new concept in power transmission
Cheng et al. A novel unidirectional three-phase multilevel rectifier composed of star-connected three single-phase topology based on five-level flying capacitor DC–DC converter
Feng et al. Solid-state transformer-based new traction drive system and control
CN113141121B (en) Current source type high-frequency isolation matrix type cascade converter and control method
CN116260348B (en) MMC-based high-capacity electrolytic hydrogen production hybrid rectifier and control method
CN113078829A (en) MMC topology with interconnected upper bridge arm sub-modules and high-frequency chain and control method
Wang et al. Research on loss reduction of dual active bridge converter over wide load range for solid state transformer application
CN116632902A (en) Multi-port flexible direct current transmission system based on PWM current source converter
Xiao et al. A novel multistation cooperative negative sequence current compensation method for high-speed railway
CN113241962A (en) Power electronic transformer based on three-phase four-leg MMC and control method thereof
CN104836465B (en) LC serial-type three-phase PWM rectifier current iterative learning control method
Wu et al. A novel comprehensive control scheme of modular multilevel converter-based power electronic transformer

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant