CN112737337B - Control method of LLC resonant converter - Google Patents
Control method of LLC resonant converter Download PDFInfo
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- CN112737337B CN112737337B CN201911031735.XA CN201911031735A CN112737337B CN 112737337 B CN112737337 B CN 112737337B CN 201911031735 A CN201911031735 A CN 201911031735A CN 112737337 B CN112737337 B CN 112737337B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
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Abstract
The invention provides a control method of an LLC resonant converter, wherein a secondary side modulation wave is a square wave with fixed voltage and amplitude regulation voltage in each period; and adjusting the amplitude adjusting voltage according to the modulation frequency of the secondary side modulation wave so that the switching devices of the secondary side bridge circuit are selectively switched on and/or switched off synchronously with the switching devices of the primary side bridge circuit. By adjusting the alignment mode of the secondary side pulse, the circuit gain under various working conditions can be finely adjusted, and the circuit gain is prevented from being changed greatly under the condition of forward and reverse flow of energy. After stable operation, the LLC resonant converter can be fixed at the optimal switching frequency, so that the problem of overcurrent caused by saturation of a high-frequency transformer due to large-range frequency modulation is avoided.
Description
Technical Field
The invention relates to the field of power electronics, in particular to a control method of an LLC resonant converter.
Background
In recent years, high-frequency power electronic technology has been developed, and research on novel power conversion technology in the fields of renewable energy, electric vehicles, electric power traction, and the like has attracted much attention. In the field of power grids, new concepts such as alternating-current and direct-current hybrid transmission and distribution networks, direct-current micro-grids, power electronic transformers, energy routers and the like are continuously proposed, and in the research hotspots and application occasions, DC/DC converters with energy flowing bidirectionally are needed to serve as interfaces among direct-current buses with different voltage levels. Meanwhile, in order to ensure high efficiency and high power density of the DC converter, the DC/DC converter is required to have a certain soft switching capability and an operating frequency as high as possible, and also have an electrical isolation capability. Based on the requirements, the LLC resonant converter serving as a high-efficiency isolated soft switching DC/DC converter has extremely high application potential and value in a DC conversion occasion.
The existing bidirectional control method comprises the following steps:
the method comprises the following steps: when energy flows in the positive direction, the primary side bridge circuit sends square waves to act, and the secondary side is blocked; when the energy flows reversely, the secondary side bridge circuit sends square wave action, and the primary side is blocked. By selecting a proper gain point, an open-loop fixed-frequency square wave mode is adopted.
The gain difference between the forward and reverse operation in this fixed frequency open loop mode is large. That is, the forward operation mode and the reverse operation mode of the LLC bidirectional resonant converter are not consistent, the LLC resonance mode is the LLC resonance in the forward operation mode of the LLC bidirectional resonant converter, the reverse operation mode is the LC resonance, the gain curves of the two are not consistent, it is difficult to consider the gain consistency in the forward and reverse operation modes, which causes the output voltage of the two modes in steady state operation to be inconsistent, and it is not guaranteed that the output voltage maintains a constant value.
Meanwhile, the change of the power flow direction can feed back the direct-current voltage at the output side, namely the abnormal fluctuation of the output voltage is easily caused, even overvoltage faults are caused, the stable output voltage under the condition of working condition switching cannot be realized, and the reliability and the service life of the converter can be influenced. Therefore, in the prior art, through reasonably setting a direct-current voltage threshold for mode switching, hysteresis control is introduced, which results in that the direct-current voltage is not obviously changed in the direction of power current under low power on one hand, and on the other hand, the continuous step change of the output voltage is inevitably caused by the control mode of the direct-current upper and lower hysteresis loops, and finally the steady state is restricted within the hysteresis range, so that the requirement of rapid power current direction switching cannot be met.
In addition, although the method can realize the bidirectional energy flow, the energy flow direction needs to be detected, the energy flow direction is inaccurate or frequently changed, the abnormal fluctuation of the output voltage is easily caused, and the stable operation under the condition switching cannot be realized; frequent conversion requires that converter pulses on the primary side and the secondary side are triggered back and forth, which affects the reliability and the service life of the converter.
The second method comprises the following steps: according to the difference value between the output voltage and the target value, the target frequency of the output is adjusted through an analog or digital controller (such as a PI controller, a fuzzy controller and the like), and the gain of the whole circuit is adjusted by adjusting the operating frequency of the converter, so that the output voltage is stabilized and the power flow control is realized. However, the frequency modulation in an excessively large range easily causes the saturation of the high-frequency transformer, and the normal operation of the resonant converter is affected. The rated operation frequency of the high-frequency transformer is determined when LLC parameters are designed, the deviation from the rated frequency is more, the iron core of the transformer is saturated, a current magnetic field curve is changed, the normal steady-state operation working point of the transformer is changed, the magnetic circuit is saturated, the excitation inductance is rapidly reduced due to demagnetization, overcurrent spikes are caused to cause overcurrent, and the normal operation of the LLC resonant converter is influenced due to the fact that the overcurrent spikes cause overcurrent.
The third method comprises the following steps: and a double-Active-full-Bridge DAB (DAB) control method is adopted to adjust the phase shift angle (including single phase shift, double phase shift and triple phase shift) of the square waves emitted by the converters at two sides, and the output voltage and the transmission power are dynamically adjusted.
The third method is, for example, to adopt the phase shift control of the H-bridge on both sides of the LLC resonant converter, when the direction of the power flow changes, the phase shift angles on both sides are automatically adjusted, to realize the bidirectional control of the power flow, but in the method, because the switching device is turned off at the current maximum, a large loss is generated, which is not conducive to improving the overall efficiency of the LLC resonant converter, increasing the design difficulty of the radiator; when the input and output voltage amplitudes are not matched, the power circulation and the current stress of the converter are greatly increased, so that the loss of a power device and a magnetic element is increased, and the efficiency of the converter is reduced. The double phase shifting and the triple phase shifting can be optimized to a certain degree, but the soft switching within the full load range cannot be guaranteed, and due to the fact that control variables are added, the control algorithm is complex, the current conversion modes of the circuit are numerous, the dynamic performance of closed-loop control is difficult to guarantee, and the engineering implementation is not convenient.
Disclosure of Invention
Aiming at the problems, the invention provides a control method of an LLC resonant converter, which does not relate to the detection of the energy flow direction, has no restriction on the precision of a sensor and the frequency of working condition switching, and realizes the forward and reverse flow of energy and automatic switching.
The invention relates to a control method of an LLC resonant converter, wherein the LLC resonant converter comprises a transformer with a primary side and a secondary side, a primary side bridge circuit and an LLC resonant circuit which are connected to the primary side of the transformer, and a secondary side bridge circuit which is connected to the secondary side of the transformer; the control method comprises the following steps: the same triangular carrier is modulated by a primary side modulation wave and a secondary side modulation wave to obtain a primary side pulse driving signal and a secondary side pulse driving signal, wherein: the primary side modulation wave has a fixed voltage, and each switching device of the primary side bridge circuit driven by the primary side pulse driving signal realizes zero voltage switching-on/switching-off; the secondary side modulation wave is a square wave with the fixed voltage and the amplitude adjusting voltage in each period; and: according to the modulation frequency f of the secondary side modulation wavexAnd adjusting the amplitude adjusting voltage to enable the switching devices of the secondary side bridge circuit to be selectively switched on and/or switched off synchronously with the switching devices of the primary side bridge circuit.
Preferably, by changing only the modulation frequency, it is possible to make each switching device of the secondary bridge circuit driven by the secondary pulse drive signal have a minimum off-current.
Preferably, the modulation frequency is adjusted by acquiring an actual dc output voltage of the LLC resonant converter and performing closed-loop regulation with respect to a target dc output voltage.
Preferably, adjusting the modulation frequency comprises the steps of: generating and storing a mapping set of a plurality of switching frequencies and a plurality of preset modulation frequencies in advance, wherein the mapping relationship of the mapping set is as follows: when the primary side pulse driving signal has any one of the plurality of switching frequencies, each switching device of the secondary side bridge circuit has the minimum turn-off current corresponding to a preset modulation frequency; acquiring preset modulation frequencies corresponding to the current switching frequencies of the switching devices of the primary side bridge circuit by inquiring the mapping set; generating a modulation frequency fine adjustment quantity through the closed-loop adjustment; and superposing the corresponding preset modulation frequency and the modulation frequency fine adjustment quantity to serve as the modulation frequency.
Preferably, the relationship between the amplitude adjustment voltage and the modulation frequency is:
wherein: u. ofs21For amplitude adjustment of voltage, fxTo modulate frequency, udgThe voltage amplitude of the triangular carrier wave is obtained; f. ofswThe switching frequency of each switching device of the primary side bridge circuit; t isdminAnd the switching protection time of each switching device of the secondary side bridge circuit is obtained.
Preferably, the voltage value of the secondary-side modulation wave is switched between the fixed voltage and the amplitude adjustment voltage by comparing the voltage value of the triangular carrier wave with the fixed voltage and the amplitude adjustment voltage; wherein when the triangular carrier wave is at a falling edge and a voltage value of the triangular carrier wave is reduced to be less than the amplitude adjustment voltage, the voltage of the secondary side modulation wave is switched from the fixed voltage to the amplitude adjustment voltage; when the triangular carrier wave is at a rising edge and the voltage value of the triangular carrier wave rises to be greater than the fixed voltage, the voltage of the secondary side modulation wave is switched from the amplitude adjustment voltage to the fixed voltage.
Preferably, the LLC resonant converter is selectively operable in a positive-going mode of operation or a negative-going mode of operation; in the forward operation mode, the primary side of the transformer transmits power to the secondary side of the transformer; in the negative-going mode of operation, the secondary side of the transformer transmits power to the primary side of the transformer.
The invention has the following advantages:
on one hand, the invention realizes that the pulse edge of a secondary side bridge circuit follows (synchronously turns on, synchronously acts and synchronously turns off) a primary side bridge circuit through the setting parameters of online dynamic adjustment, thereby realizing the synchronous control of pulses on two sides; the method does not need to detect the magnitude and the direction of the energy flow at the direct current output side, realizes an energy bidirectional flow path, and can ensure the stability of output voltage under no-load and working condition switching; because frequent triggering of pulses on two sides is not involved, the overall operation control mode of the converter is simplified, and the overall reliability and usability of the converter are improved; the gain change in the bidirectional control mode is very small, and the output voltage can be maintained in a small fluctuation range under various working conditions; according to the method, large-range frequency modulation is not needed, the switching frequency can be fixed after stable operation, the problem of transformer saturation overcurrent caused by wide-range frequency modulation is avoided, and the design difficulty of the transformer is reduced; the low-current turn-off of the primary zero-voltage switch and the secondary can be realized under forward and reverse operation, the loss of the LLC resonant converter is between open-loop control and shift control, the direct-current voltage is stabilized, the efficiency of the LLC resonant converter is considered, and the engineering realization is facilitated.
Drawings
The foregoing summary, as well as the following detailed description of the invention, will be better understood when read in conjunction with the appended drawings. It is to be noted that the appended drawings are intended as examples of the claimed invention. In the drawings, like reference characters designate the same or similar elements.
Fig. 1 shows a circuit topology of an LLC-type resonant converter of the invention;
fig. 2 shows a schematic diagram of a control method of an LLC-type resonant converter in accordance with the invention;
FIG. 3 is a schematic diagram showing the manner of generating a primary side pulse driving signal and a secondary side pulse driving signal of the LLC resonant converter of the invention;
FIG. 4 shows waveforms of a primary side pulsed driving signal and a secondary side pulsed driving signal of the LLC resonant converter of the invention;
FIG. 5 shows LL of the present inventionPrimary side pulse driving signal, secondary side pulse signal and resonant current i of C resonant converter in forward operation modeLrSecondary side current iDAnd an excitation current iLmA relationship diagram of (a);
FIG. 6 shows a primary side pulse driving signal, a secondary side pulse signal, a resonant current i in a negative-going operation mode of an LLC resonant converter of the inventionLrSecondary side current iDWith excitation current iLmSchematic diagram of the relationship of (1).
Detailed Description
The detailed features and advantages of the present invention are described in detail in the detailed description which follows, and will be sufficient for anyone skilled in the art to understand the technical content of the present invention and to implement the present invention, and the related objects and advantages of the present invention will be easily understood by those skilled in the art from the description, claims and drawings disclosed in the present specification.
Referring to fig. 1, the present invention is applied to an LLC resonant converter 100, where the LLC resonant converter 100 includes a transformer 110, a primary bridge circuit 120, an LLC resonant circuit 130, and a secondary bridge circuit 140. The transformer 110 has a primary side and a secondary side. The primary bridge circuit 120 and the LLC resonant circuit 130 are connected to the primary side of the transformer 110, and the secondary bridge circuit 140 is connected to the secondary side of the transformer 110. Resonant capacitor Cr, high frequency transformer leakage inductanceLrAnd an excitation inductanceLmConstituting an LLC resonant circuit 130.
Preferably, the LLC resonant converter 100 is selectively operable in a positive-going mode of operation or a negative-going mode of operation; wherein in a positive-going mode of operation the primary side of the transformer 110 transfers power to the secondary side of the transformer 110, and in a negative-going mode of operation the secondary side of the transformer 110 transfers power to the primary side of the transformer 110.
In order to realize the bidirectional flow of energy, each switching device S of the primary side bridge circuit 120 of the present embodiment1-S4And respective switching devices V of the secondary side bridge circuit 1401-V4All the fully-controlled power devices IGBT are used, however, the devices used in the LLC resonant converter 100 of the present invention are not limited, and may be selected from MOSFET, IGBT, IGCT, IPM, or other power semiconductor devices.
The control method of the LLC resonant converter of the present invention will be described below with reference to fig. 2 and 3.
In the control method, the same triangular carrier wave WTRI needs to be modulated by the primary side modulation wave WT and the secondary side modulation wave WV to obtain a primary side pulse driving signal and a secondary side pulse driving signal, and the primary side pulse driving signal is used for driving each switching device S of the primary side bridge circuit 1201-S4. The secondary pulse driving signal is used for driving each switching device V of the secondary bridge circuit 1201-V4. In which two switching devices S on one leg of the primary bridge circuit 1201,S4The same pulse driving signal is used for the other two switching devices S2, S3, and the same pulse driving signal is used for the switching device S1,S4Are 180 degrees out of phase. Respective switching devices V of the secondary side bridge circuit 1401-V4For the same reason, it is not described herein.
Fig. 3 shows a method for modulating the same triangular carrier wave WTRI by the primary-side modulated wave WT and the secondary-side modulated wave WV according to the present invention. The upper diagram shows the pulse comparison generation principle of the primary bridge circuit 120 of the present invention, and the lower diagram shows the pulse comparison generation principle of the secondary bridge circuit 140 of the present invention.
Primary side pulse driving signal
The voltage amplitude of the triangular carrier WTRI is udgFrequency and switching frequency fswAnd the consistency is maintained. Having a fixed voltage amplitude u of the primary modulated wave WTs1And the switching devices of the primary bridge circuit 120 driven by the primary pulse drive signal are turned on/off at zero voltage using any known method.
By changing us1I.e. by u in FIG. 3s1And the linear primary side pulse driving circuit can generate a square wave primary side pulse driving signal with any duty ratio. Taking into account the need for individual switching devices S of the primary bridge circuit 1201-S4Reserving a certain off-time, usually adding a protection time Tdmin,TdminFor arbitrarily set guard timeLength.
Secondary side pulse driving signal
The same voltage amplitude as the primary side is udgThe triangular carrier WTRI. The secondary side modulation wave WV has a fixed voltage u in each periods22And amplitude regulating voltage us21Wherein the fixed voltage u of the secondary-side modulated wave WVs22Constant voltage u to primary modulated wave WTs1The same is true. That is, the voltage amplitudes of the secondary modulation wave WV are respectively the fixed voltage u of the primary modulation wave WTs1And amplitude regulating voltage us21The two parts are spliced and built.
Amplitude regulation voltage us21The determination is as follows.
As shown in fig. 3, the upper and lower pulse signals are the primary pulse driving signal and the secondary pulse driving signal respectively, and the amplitude of the triangular carrier is udgIf the height of the peak and the trough of the triangular carrier is tri 2udg。
The pulse width of the secondary side pulse driving signal is d, and the corresponding frequency is fxThen:
switching frequency of primary side of fswDue to the fact that
Then:
the triangle similarity principle is utilized to obtain:
and
substituting the formula (2) into the above formula to obtain the amplitude adjustment voltage us21And a modulation frequency fxThe relationship of (1) is:
wherein: u. ofdgThe voltage amplitude of the triangular carrier WTRI is obtained; f. ofswThe switching frequency of the switching devices of the primary side bridge circuit 120; t isdminSwitching device S for primary side bridge circuit 1201-S4Switching device V connected to secondary bridge circuit 1401-V4Of the switching device of (1) is a minimum value of the switching protection time of the switching device.
xModulation frequency f
The modulation frequency f in the present invention will be described belowxAnd (4) obtaining the method.
First, in STEP0 of fig. 2, a mapping set of a plurality of switching frequencies and a plurality of preset modulation frequencies is generated and stored in advance, and the mapping relationship of the mapping set is as follows: when the primary side pulse driving signal has any one of a plurality of switching frequencies, each switching device of the secondary side bridge circuit has the theoretical minimum turn-off current corresponding to a preset modulation frequency. That is, the switching device S of the primary bridge circuit 120 is connected to the primary bridge circuit in advance through off-line simulation and theoretical calculation1-S4Modulation frequency f corresponding to switching frequency within a certain rangexThe theoretical optimum value of (b) is stored.
When the LLC resonant converter 100 is actually operated, STEP1 in fig. 2 is executed to obtain each switching device S in the primary bridge circuit 120 by querying such a mapping set1-S4Of the currently employed switching frequency of the switching devicex0. Can adjust the presetSystem frequency fx0I.e. as the actual modulation frequency fx。
Preferably, in order to cope with various factors such as the deviation of LLC parameters, the fluctuation of output voltage, and the dynamic change of the operation mode of the LLC resonant converter 100, STEP2 in fig. 2 is optionally executed to collect the dc output voltage Vout, perform closed-loop adjustment on the target value of the dc output voltage and the actual collected value of the dc output voltage, and generate the modulation frequency fine adjustment amount Δ f of the bidirectional synchronous controlxAnd for Δ fxThe amplitude limiting process is performed and the preset modulation frequency f is set in STEP3x0And the modulation frequency fine adjustment amount delta fxSuperimposed as modulation frequency fx. Therefore, even if the parameters of the resonant inductor Lr, the resonant capacitor Cr and the excitation inductor Lm of the high-frequency transformer have errors and/or real-time fluctuation of the direct-current voltage, the modulation frequency f is adjusted by introducing a closed loop to the direct-current output voltage VoutxAnd certain correction and fine adjustment are carried out, the adaptability of the overall control strategy is improved, and the dynamic change of the load and the energy flow working condition is convenient to deal with. The loss of the LLC resonant converter 100 is maintained at a lower level, and the overall efficiency of the LLC resonant converter is improved.
Secondary side modulated wave WV
At a determined modulation frequency fxThen, STEP4 in fig. 2 is executed, according to the above equation (3) based on the modulation frequency fxRegulating amplitude regulating voltage us21。
From the above formula, the switching frequency f of the switching devices of the primary side bridge circuit 120 isswVoltage amplitude u of triangular carrier WTRIdgAnd minimum value T of switch protection timedminIs kept fixed during the operation of the LLC resonant converter, so that the modulation frequency fxCan be adjusted with the amplitude adjustment voltage us21The numerical values of (a) establish a one-to-one correspondence relationship. In other words, by setting the modulation frequency f onlyxTo obtain the required amplitude adjustment voltage us21。
After obtaining the amplitude regulation voltage us21Then, STEP STEP5 is executed to make the voltage amplitudes be fixed voltages us21And amplitude adjustmentVoltage us22The secondary side modulation wave WV can be formed by splicing the two parts.
Specifically, the triglag is set to 0 at the rising stage of the triangular carrier WTRIxAA and is set to trifllag 0 in its lower stagexThe rule for splicing the voltage amplitude us2 of the side-modulation wave WV is defined as follows:
the triglag is a slope direction flag bit of a triangular carrier, represents the ascending/descending stage of the carrier, and can be set arbitrarily according to actual needs, and the value of the triglag can be distinguished into any value.
In other words, the voltage u is adjusted by adjusting the voltage value and amplitude of the triangular carrier WTRIs2Comparing the voltage of the secondary side modulation wave WV with a fixed voltage us22And amplitude regulating voltage us21To switch between. When the triangular carrier wave WTRI is at the falling edge and the voltage value of the triangular carrier wave WTRI is reduced to be less than the amplitude regulation voltage us21While the voltage of the secondary side modulation wave WV is switched to the amplitude regulation voltage us21(ii) a When the triangular carrier wave WTRI is at the rising edge and the voltage value of the triangular carrier wave WTRI rises to be greater than the fixed voltage us1(equal to a fixed voltage us21) While the voltage of the secondary side modulation wave WV is switched to a fixed voltage us21Otherwise, switching to us22。
Then, the same triangular carrier wave can be modulated by the secondary modulation wave and the secondary modulation wave in STEP6 of fig. 2 to obtain the primary side pulse driving signal and the secondary side pulse driving signal, and each switching device S of the primary side bridge circuit is driven by the primary side pulse driving signal in STEP 71-S4Driving each switching device V of the secondary bridge circuit with a secondary pulse driving signal1-V4。
The following describes a process of achieving bidirectional operation and reducing the overall loss of the LLC resonant converter by using the secondary-side modulation wave WV generated by the control method of the present invention, with reference to fig. 4 to 6.
As already mentioned above, the modulation frequency f has been determinedxUniquely determine the amplitude regulation voltage us22That is, the square waveform of the sub-side modulation wave WV is determined, the duty ratio, the rising edge, and the falling edge of the sub-side pulse drive signal obtained by the sub-side modulation wave WV on the triangular carrier WTRI can be determined accordingly. I.e. different modulation frequencies fxAnd synchronously turning off (see figure 4a) and synchronously turning on (see figure 4b) corresponding to different pulse widths of the secondary side pulse driving signal and different alignment modes of the primary side pulse driving signal and the secondary side pulse driving signal. Accordingly, the secondary side pulse driving signal can be modulated by setting a single parameter modulation frequency fxThree edge alignment modes of synchronous turn-on (see figure 5a), synchronous turn-on and turn-off (see figure 5b) and synchronous turn-off (see the lowest figure of figure 5) are realized, so that the method is suitable for primary side pulse driving signals which are flexible and changeable and have any duty ratio, and the realization mode is simple.
In the case of first ensuring that the primary bridge circuit 120 realizes the zero-voltage switch ZVS, the main loss of the LLC resonant converter 100 is represented by the turn-off loss of the secondary bridge circuit 140. Thus, by setting the modulation frequency fxDifferent pulse widths of the secondary side pulse driving signal, different alignment modes of the primary side pulse driving signal and the secondary side pulse driving signal can enable each switching device V of the secondary side bridge circuit to be capable of1-V4And the circuit is turned off by the minimum turn-off current, so that the loss of the secondary side bridge circuit 140 is minimized, and the overall efficiency of the LLC resonant converter is improved.
I.e. according to the modulation frequency f of the secondary side modulation wave WVxRegulating amplitude regulating voltage us21So that the respective switching devices V of the secondary bridge circuit 120 driven by the secondary pulse drive signal1-V4With a minimum off current. The primary side pulse driving signal and the secondary side pulse driving signal generated in the above process are respectively applied to the respective switching devices of the primary side bridge circuit 120 and the secondary side bridge circuit 140, so that the primary side pulse driving signal, the secondary side pulse signal, and the resonant current iLrSecondary side current iDWith excitation current iLmAs shown in the figureFig. 5 and 6 show the case of the positive running mode, and fig. 5 shows the case of the negative running mode.
Specifically, as shown in FIG. 5, t0Time of day, switching device S1,S4,V1,V4Open, resonant inductorLrAnd a resonance capacitor CrForward resonance, resonant current iLrIs negative, secondary side current iDFrom zero, the primary bridge circuit 120 and the secondary bridge circuit 140 respectively flow through the respective freewheeling diodes, and the respective switching devices S of the primary bridge circuit 1201-S4Realizing zero voltage turn-on and current at t0～t1LC resonance in time period, voltage at two ends of the excitation inductor is clamped by the secondary bridge circuit 140, and the excitation current iLmLinear increase;
t1time of day, switching device S1,S4,V1,V4Off, the current resonates to 0, but the resonant period is not over, the resonant current iLrRe-lifting after zero crossing point, secondary side switch device V1,V4Small current turn-off, secondary side switch device S2,S3The anti-parallel diode of (1) is turned on;
t2at the moment, the LC resonance period is over, and the resonance current i is forcedLrAnd an excitation current iLmDraw to unity, at which point LLC resonance begins, due to Lm＞＞LrThe current is approximate to a linear straight line, and the secondary side current returns to zero at the stage; according to the principle of symmetry, t3～t6Procedure and t0～t2And the two bridge arms are completely consistent, but the switching devices of the other bridge arm complete corresponding processes of LC resonance, zero voltage switching-on of the primary side, LLC resonance, small current switching-off of the secondary side and the like.
Because the invention adopts the modulation wave and the triangular carrier wave to compare and generate the pulse driving signal of the primary side and the secondary side synchronous control to realize the bilateral two-way control of the LLC resonant converter, the detection of the energy flow direction is not involved, the precision of the sensor and the frequency of the working condition switching are not restricted, the forward and the reverse flow of the energy and the automatic switching are realized, the problems of the middle voltage fluctuation under the switching of the operation modes and the small load fluctuation are solved, the automatic smooth transition control when the power direction changes can be realized, the switching of the forward operation mode and the reverse operation mode is avoided, and the LLC resonant converter integrally operates only in one two-way operation mode.
The control method of the invention has the following advantages:
by adjusting the alignment mode of the secondary side pulse, the circuit gain under various working conditions can be finely adjusted to a certain extent, and the circuit gain is ensured not to be greatly changed under the condition of forward and reverse flow of energy.
The invention does not need to adjust the operation frequency of the converter in a large range, and the conversion output regulating quantity is only used as a fine adjustment quantity to compensate the influence caused by factors such as parameter deviation, sudden change of working conditions and the like, and the switching frequency does not need to be adjusted in a large range under a steady state. After stable operation, the LLC resonant converter can be fixed at the optimal switching frequency, so that the problem of overcurrent caused by saturation of a high-frequency transformer due to large-range frequency modulation is avoided.
The method has the advantages that fine adjustment is carried out to a certain degree under the condition that the minimum turn-off current of the secondary side current is ensured, the full-working-condition zero-voltage switch of the primary side bridge circuit and the small-current turn-off of the secondary side bridge circuit are achieved, the current stress of a switch device is greatly reduced, the loss of the LLC resonant converter is reduced, the heat dissipation design difficulty of the LLC resonant converter is reduced, and the whole efficiency of the LLC resonant converter is improved.
Generating a set parameter fine adjustment quantity delta f of a pulse synchronization mode by performing closed-loop adjustment on the output voltagexThe modulation frequency obtained by off-line simulation and theoretical calculation is subjected to certain fine adjustment, the adaptability of the whole algorithm is improved, the accuracy dependence on LLC parameters is reduced, and the dynamic response capability of the LLC resonant converter is further improved.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.
For example, the novel bidirectional synchronous control strategy is applicable to all LLC resonant converters, is not limited to a specific application occasion, and can simplify and adopt the same control strategy implementation method to regulate and control the temperature of the original secondary bridge circuit on line if energy only needs to flow in a single direction, so as to ensure the healthy and stable operation of the high-frequency resonant converter.
The novel bidirectional synchronous control strategy set forth by the invention can be simply changed and popularized to be applied to circuit topologies of various forms such as three-level, multi-level, full-bridge circuit, half-bridge circuit and the like, and the novel bidirectional synchronous control strategy is included in the protection scope of the invention.
The circuit topology, the voltage grade and the power adopted by the LLC resonant converter are not limited, the topology of the converter can adopt various application topologies or occasions such as two-level, three-level, H-bridge cascade, chain, MMC, full bridge, half bridge, full wave, half wave rectification and the like, and the converter can be selectively provided with a filter.
The terms and expressions which have been employed herein are used as terms of description and not of limitation. The use of such terms and expressions is not intended to exclude any equivalents of the features shown and described (or portions thereof), and it is recognized that various modifications may be made within the scope of the claims. Other modifications, variations, and alternatives are also possible. Accordingly, the claims are to be regarded as covering all such equivalents.
Certain terms are used throughout this document to indicate specific LLC resonant converter components. As one skilled in the art will recognize, identical components may often be referred to by different names, and thus this document does not intend to distinguish between components that differ in name but not function. In this document, the terms "including", "comprising" and "having" are used in an open-ended fashion, and thus should be interpreted to mean "including, but not limited to …".
Also, it should be noted that although the present invention has been described with reference to the current specific embodiments, it should be understood by those skilled in the art that the above embodiments are merely illustrative of the present invention, and various equivalent changes or substitutions may be made without departing from the spirit of the present invention, and therefore, it is intended that all changes and modifications to the above embodiments be included within the scope of the claims of the present application.
Claims (6)
1. A control method of an LLC resonant converter, the LLC resonant converter comprises a transformer with a primary side and a secondary side, a primary side bridge circuit and an LLC resonant circuit which are connected to the primary side of the transformer, and a secondary side bridge circuit which is connected to the secondary side of the transformer;
the control method comprises the following steps: the same triangular carrier is modulated by a primary side modulation wave and a secondary side modulation wave to obtain a primary side pulse driving signal and a secondary side pulse driving signal, wherein:
the primary side modulation wave has a fixed voltage, and each switching device of the primary side bridge circuit driven by the primary side pulse driving signal realizes zero voltage switching-on/switching-off;
the secondary side modulation wave is a square wave with the fixed voltage and the amplitude adjusting voltage in each period; switching a voltage value of the secondary-side modulation wave between the fixed voltage and the amplitude adjustment voltage by comparing the voltage value of the triangular carrier with the fixed voltage and the amplitude adjustment voltage; wherein
When the triangular carrier wave is at a falling edge and the voltage value of the triangular carrier wave is reduced to be smaller than the amplitude adjusting voltage, the voltage of the secondary side modulation wave is switched from the fixed voltage to the amplitude adjusting voltage;
when the triangular carrier wave is at a rising edge and the voltage value of the triangular carrier wave rises to be greater than the fixed voltage, the voltage of the secondary side modulation wave is switched from the amplitude regulation voltage to the fixed voltage; and:
according to the modulation frequency f of the secondary side modulation wavexAnd adjusting the amplitude adjusting voltage to enable the switching devices of the secondary side bridge circuit to be selectively switched on and/or switched off synchronously with the switching devices of the primary side bridge circuit.
2. The control method according to claim 1, characterized in that:
by changing only the modulation frequency, it is possible to make each switching device of the secondary bridge circuit driven by the secondary pulse drive signal have a minimum off-current.
3. The control method according to claim 1, characterized in that:
and adjusting the modulation frequency by acquiring the actual direct current output voltage of the LLC resonant converter and performing closed-loop adjustment relative to the target direct current output voltage.
4. The control method according to claim 3, characterized in that:
adjusting the modulation frequency comprises the steps of:
generating and storing a mapping set of a plurality of switching frequencies and a plurality of preset modulation frequencies in advance, wherein the mapping relationship of the mapping set is as follows: when the primary side pulse driving signal has any one of the plurality of switching frequencies, each switching device of the secondary side bridge circuit has the minimum turn-off current corresponding to a preset modulation frequency;
acquiring preset modulation frequencies corresponding to the current switching frequencies of the switching devices of the primary side bridge circuit by inquiring the mapping set;
generating a modulation frequency fine adjustment quantity through the closed-loop adjustment;
and superposing the corresponding preset modulation frequency and the modulation frequency fine adjustment quantity to serve as the modulation frequency.
5. The control method according to claim 1, characterized in that:
the relationship between the amplitude regulating voltage and the modulation frequency is as follows:
wherein:us21for amplitude regulation of voltage, fxTo modulate frequency, udgThe voltage amplitude of the triangular carrier wave is obtained; f. ofswThe switching frequency of each switching device of the primary side bridge circuit; t isdminAnd the switching protection time of each switching device of the secondary side bridge circuit is obtained.
6. The control method according to claim 1, characterized in that:
the LLC resonant converter can be selectively operated in a positive-going mode of operation or a negative-going mode of operation; in the forward operation mode, the primary side of the transformer transmits power to the secondary side of the transformer; in the negative operating mode, the secondary side of the transformer transfers power to the primary side of the transformer.
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| TWI285467B (en) * | 2005-10-20 | 2007-08-11 | Delta Electronics Inc | Adaptive synchronous rectification control circuit and method thereof |
| US9559602B2 (en) * | 2015-02-26 | 2017-01-31 | Infineon Technologies Austria Ag | Magnetizing current based control of resonant converters |
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| CN106877676A (en) * | 2017-04-06 | 2017-06-20 | 珠海英搏尔电气股份有限公司 | A kind of two-way resonance translation circuit, converter and its control method |
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