CN112688611B - Method for inhibiting low-speed noise of single-resistor sampling permanent magnet synchronous motor - Google Patents
Method for inhibiting low-speed noise of single-resistor sampling permanent magnet synchronous motor Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
- H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
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Abstract
The invention discloses a method for inhibiting low-speed noise of a single-resistor sampling permanent magnet synchronous motor, which comprises the following steps of: (1) According to the parameters of the single-resistance sampling permanent magnet synchronous motor driving system, the minimum sampling time T is determined min (ii) a (2) For the lowest set speed n set_min The working condition of no-load operation of the motor is processed, and the time T is calculated 1 、T 2 The variation curve of (d); (3) Judging whether node n is running at a given low speed set_min Whether a low modulation unobservable region exists; (4) If no unobservable region exists, directly switching to the next step (5), otherwise, returning to the step (1) for recalculation; (5) According to T 1 、T 2 Determining T in each vector frequency conversion control iterative operation process 1 、T 2 Varying maximum amplitude for T during operation 1 、T 2 The change of (2) is subjected to clipping processing. The invention reduces T 1 、T 2 The waveform burr reduces the decibel value of low-speed noise of the single-resistor sampling scheme.
Description
Technical Field
The invention relates to the field of motor control, in particular to a method for inhibiting low-speed noise of a single-resistor sampling permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the advantages of simple structure, wide speed regulation range, high power density, high motor efficiency and the like, and is widely applied to the fields of household appliances, fan and pump products and the like at present. The medium and small power permanent magnet synchronous motor driving system has higher requirement on product cost performance, the proportion of a current sampling circuit in the hardware cost of the whole machine cannot be ignored, and a common current sampling method adopts two current sensors to measure phase current, but the method has higher cost. The system structure can be simplified by using resistance sampling, the cost of a hardware circuit is reduced, and the current scheme which is more commonly used is a double-resistance sampling scheme. The single-resistor sampling scheme of the motor phase current can further simplify the layout of a hardware circuit and reduce the cost of the hardware circuit, and in recent years, the scheme is more and more emphasized by manufacturers, but the single-resistor sampling needs to reconstruct the three-phase current of the bus current of a driving system, and a blind area, namely an unobservable area, exists in the current reconstruction process, and can be generally divided into a low-modulation unobservable area, a sector transition unobservable area and a high-modulation unobservable area.
The basic principle of single-resistor sampling permanent magnet synchronous motor three-phase current reconstruction is to sample bus current at different moments in a PWM period and obtain each phase current through phase current reconstruction, as shown in fig. 1, which is a schematic diagram of single-resistor sampling permanent magnet synchronous motor three-phase current reconstruction. The motor controller is controlled by adopting SVPWM modulation mode, as shown in FIG. 2, the space voltage vector diagram corresponding to 7-segment SVPWM of permanent magnet synchronous motor vector control has 8 switch working states including six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 Which divides the voltage space plane into 6 sectors, at each period T s Within each sector, an arbitrary target voltage vector V ref The non-zero voltage vector and the zero vector of the sector can be synthesized together, and the analysis of the current reconstruction unobservable region firstly needs to calculate the action time T of the non-zero voltage vector corresponding to each sector 1 、T 2 。
The relation between the DC bus current and the three-phase current isThe determination of the state of the instantaneous switching value, taking sector 4 as an example, is shown in FIG. 3, which shows the two-phase current sampled in one PWM cycle, where the reference voltage vector in FIG. 3 is decomposed into a basic voltage vector V 1 (001) And V 3 (011) At a voltage vector V 1 Bus current I sampled during action dc Corresponding to W phase current at voltage vector V 3 Bus current I sampled during action dc Corresponding to the U-phase current.
In practical systems, considering that the sampling of the bus current requires a sufficient sampling window, this requires that the non-zero voltage vector must last for a minimum sampling time T min . When the output voltage vector is in a low modulation region or near a non-zero voltage vector, the time that the non-zero voltage vector possibly acts in one PWM period is less than T min This makes the sampled bus current meaningless, as shown in fig. 4, which is the shortest sampling time T in the actual situation min Schematic representation. T in FIG. 4 min The medicine consists of three parts: dead time T d Bus current establishing time T set And AD conversion time T conv ，T min The size is generally 3. Mu.s to 5. Mu.s.
From FIG. 3, it can be known that it is necessary to ensure that the non-zero voltage vector is in the first half sampling period T to complete the single-resistor sampling current reconstruction s Internal action time greater than T min That is, it is required to satisfy:
the region in which different phase currents cannot be sampled in one PWM period is called an unobservable region, and in practical application, for convenience of processing, the space voltage vector hexagon is specifically divided into an observable region, a low modulation unobservable region, a sector transition unobservable region and a high modulation unobservable region.
In the practical process of early production, for a fan product for indoor air purification, the noise of a single-resistor sampling scheme in a low-speed area (500-800 rpm) is obviously higher than that of a double-resistor sampling scheme, and the single-resistor sampling scheme also has abnormal noise. Further contrast research shows that when single-resistor sampling current reconstruction is carried out in a low-modulation unobserved area, the action time of a certain basic voltage vector state is short, the pulse needs to be translated in a PWM period to be staggered so as to obtain enough single-resistor sampling time, namely pulse phase shifting needs to be carried out, the PWM output waveform is asymmetric, phase current harmonic components are increased, and the pulse phase shifting is a main reason of abnormal sound existing in a single-resistor sampling scheme. On the other hand, the load is very light when the low-speed operation is carried out, the phase current is very small (such as 0.1A), the sampling error is not negligible relative to the amplitude of the phase current, the phase current is subjected to waveform distortion caused by the sampling error and is accompanied by more burrs, and the noise of the single-resistor sampling scheme is obviously larger than that of the double-resistor sampling scheme. The current permanent magnet synchronous motor sampling current waveform distortion and the low-speed noise caused by the distortion limit the popularization and application of a single-resistor sampling scheme.
At present, researches on restraining low-speed noise of a single-resistor sampling permanent magnet synchronous motor are mainly focused on reconstruction of single-resistor sampling current, yellow Keyuan and the like provide a single-resistor sampling permanent magnet synchronous motor phase current reconstruction strategy on a power system and an automatic journal thereof (VOL.30, NO. 9), measurement pulses are inserted into a low-modulation unobservable area and a medium-modulation unobservable area, and a voltage vector approximation method is adopted in a high-modulation unobservable area, but the method has the defects of long algorithm execution time and complex implementation process; saritha B et al, IEEE Trans on Industrial Electronics (VOL.54, no. 5), used a sine curve fitting observer for the unobservable zone to bring the estimated current close to the reference sinusoidal three-phase current, but this method relied on motor parameters and did not solve the low speed error problem.
The existing scheme for improving the single-resistor sampling precision is complex in algorithm and long in algorithm execution time of a processor, and the scheme for inhibiting the low-speed noise of the permanent magnet synchronous motor by optimizing the single-resistor sampling current reconstruction algorithm is difficult to implement in a low-cost microprocessor control system.
Disclosure of Invention
In order to solve the technical problems, the invention provides a method which is simple in algorithm, high in efficiency and capable of effectively suppressing low-speed noise of a single-resistor sampling permanent magnet synchronous motor.
In order to achieve the purpose, the technical scheme adopted by the invention is as follows: a method for suppressing low-speed noise of a single-resistor sampling permanent magnet synchronous motor comprises the following steps:
step 1: determining the minimum sampling time T of single resistance sampling according to the parameters of the single resistance sampling permanent magnet synchronous motor driving system min ；
And 2, step: at the lowest set speed n set_min Calculating the change curves of two non-zero voltage vector action times T1 and T2 corresponding to each sector by vector frequency conversion control, and calculating the intersection point T of T1 and T2 cross The size of (d);
and 3, step 3: judging the lowest set rotating speed n set_min Whether a low modulation unobservable region reconstructed by single-resistor three-phase current exists or not is judged;
when in use
When the reference voltage space vector is determined to be outside the low modulation unobservable region reconstructed by the single resistance current;
when in use
When the single-resistance current reconstruction exists, judging that a low modulation unobservable region exists;
and 4, step 4: if the single-resistor three-phase current reconstruction does not have a low modulation unobservable region, directly turning to the step 5;
otherwise, increasing the number of turns of the motor winding and correspondingly reducing the wire diameter of the winding, increasing the back electromotive force coefficient of the permanent magnet synchronous motor under the condition of maintaining the slot filling rate of the motor to be basically unchanged, and returning to the step 1 to recalculate the T min ；
And 5: and determining the maximum amplitude of the change of the T1 and the T2 in the iterative operation process of each vector frequency conversion control according to the change curves of the T1 and the T2, and carrying out amplitude limiting treatment on the change of the T1 and the T2 in the operation process.
Preferably, in step 1: t is min ＝T d +T set +T conv Where Td, tset, and Tconv are system parameters.
Preferably, the PMSM vector control comprises six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 Six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 The voltage space plane is divided into 6 sectors, and in each period T s Within, an arbitrary target voltage vector V in each sector ref Each generated by the joint combination of two non-zero voltage vectors and a zero vector of the sector, wherein T of the 1 st sector 1 、T 2 Obtained from the following equation:
T s ＝T 0 +T 1 +T 2 (2)
said T is cross Is T 1 、T 2 The intersection of the curve functions;
the Ts is a vector control operation period; v4 and V6 are non-zero voltage vectors corresponding to the 1 st sector; t0 is the sum of action time of the zero voltage vectors V0 and V7; vdc is the bus voltage; theta ref Is the angle between the voltage vector Vref and the non-zero voltage vector V4.
Preferably, the specific steps of determining the maximum amplitude of the change of T1 and T2 in each vector frequency conversion control iterative operation process in step 5 are as follows:
step 5-1: according to the lowest set speed n set_min Calculating the electrical period of each phase current
Wherein p is n Each phase current period comprises 6 sectors for the number of pole pairs of the motor;
further calculating the number of control iterative operations of each sector
Wherein T is s Is a sampling period;
step 5-2: calculating the maximum amplitude of the change of each vector frequency conversion control iterative operation T1 and T2:
wherein T is 1_max 、T 2_max Maximum values of T1 and T2, respectively, T 1_min 、T 2_min Is the minimum value of T1 and T2 respectively, and xi is the amplification factor.
Preferably, the xi takes the value as follows: ξ =1.3.
Preferably, in the step 4, the specific steps of performing amplitude limiting processing on the changes of T1 and T2 in the operation process are as follows:
step 5-3: according to the condition that T1 and T2 are in the rising period or the falling period, carrying out the amplitude limiting processing on T1 and T2:
if T1 is in the rise phase: t is 1 (n)＜T 1 (n-1)+ΔT 1_max ；
If T1 is in the decline phase: t is 1 (n)＞T 1 (n-1)-ΔT 1_max ；
If T2 is in the rise phase: t is 2 (n)＜T 2 (n-1)+ΔT 2_max ；
If T2 is in the decline phase: t is 2 (n)＞T 2 (n-1)-ΔT 2_max 。
The invention has the beneficial technical effects that: according to the method, by researching the law that the action time T1 and T2 curves of two non-zero voltage vectors corresponding to each sector are controlled through vector frequency conversion, on one hand, single-resistor sampling three-phase current reconstruction is ensured not to enter a low-modulation unobservable region through motor back electromotive force coefficient optimization, the number of pulse shift in the current sampling process is reduced, and therefore abnormal sound existing in a single-resistor sampling scheme is eliminated; on the other hand, burrs of T1 and T2 waveforms are reduced through an optimization algorithm, and the influence of sampling errors and the distortion of motor phase current waveforms are reduced, so that the decibel value of low-speed noise of a single-resistor sampling scheme is reduced.
Drawings
FIG. 1 is a schematic diagram of a three-phase current reconstruction of a single-resistor sampling permanent magnet synchronous motor;
FIG. 2 is a space voltage vector diagram of a permanent magnet synchronous motor;
FIG. 3 is a PWM waveform diagram of a three-phase current reconstruction observable region;
FIG. 4 is a schematic diagram of a three-phase current reconstruction shortest sampling time Tmin;
FIG. 5-1 is a schematic diagram of a sector transition unobservable region in a three-phase current reconstruction unobservable region;
FIG. 5-2 is a schematic diagram of a low modulation unobservable region in a three-phase current reconstruction unobservable region;
FIG. 6 is a waveform diagram of the simulation of curves T1 and T2;
FIG. 7 is a control block diagram of a single-resistor sampling permanent magnet synchronous motor system;
FIG. 8-1 is a waveform diagram of T1 and T2 before Ke optimization;
FIG. 8-2 is a waveform diagram of T1 and T2 without clipping after Ke optimization;
FIGS. 8-3 are waveforms of T1 and T2 after Ke optimization and clipping;
FIG. 9-1 is a phase current waveform diagram before optimization;
FIG. 9-2 is a graph of phase current waveforms after optimization;
FIG. 10-1 shows the motor at n before optimization set_min Noise spectrogram at 500 rpm;
FIG. 10-2 shows the optimized motor at n set_min Noise spectrogram at 500 rpm.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is further described in detail with reference to the following embodiments, but the scope of the present invention is not limited to the following embodiments.
As shown in figures 1-2, the invention controls the action time T of two non-zero voltage vectors corresponding to each sector by researching vector frequency conversion 1 、T 2 On one hand, the rule of the curve ensures that the single-resistor sampling three-phase current reconstruction does not enter a low-modulation unobservable region through the optimization of the back electromotive force coefficient of the motor, and reduces the times of pulse shift in the current sampling process, thereby eliminating abnormal sound existing in the single-resistor sampling scheme; on the other hand, by optimizing the algorithm, T is reduced 1 、T 2 The burr of wave form reduces the influence of sampling error and the distortion of motor phase current wave form to reduce the decibel value of single resistance sampling scheme low-speed noise, specifically include the following step:
step 1: determining the minimum sampling time T of single resistance sampling according to the parameters of the single resistance sampling permanent magnet synchronous motor driving system min ；
T min The size is generally 3 mus to 5 mus, which is related as follows:
T min ＝T d +T set +T conv (5) Td, tset and Tconv are system parameters;
from FIG. 3, it can be known that it is necessary to ensure that the non-zero voltage vector is in the first half sampling period T to complete the single-resistor sampling current reconstruction s Internal action time greater than T min That is, it is required to satisfy:
step 2: at a given minimum set speed n set_min Calculating vector frequency conversion to control action time T of two non-zero voltage vectors corresponding to each sector 1 、T 2 And calculating T 1 And T 2 Cross point of (A) T cross The size of (d);
specifically, the PMSM vector control comprises six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 Six ofNon-zero voltage vector V 1 ～V 6 And two zero voltage vectors V 0 、V 7 The voltage space plane is divided into 6 sectors, and in each period T s Within, an arbitrary target voltage vector V in each sector ref Each generated by the joint combination of two non-zero voltage vectors and a zero vector of the sector, wherein T of the 1 st sector 1 、T 2 Obtained from the following equation:
T s ＝T 0 +T 1 +T 2 (2)
other T1 and T2 of the 2 nd to 6 th regions are formed by non-zero voltage vectors V of corresponding intervals 1 ～V 6 And two zero voltage vectors V 0 、V 7 And the vector control operation period Ts is available.
The T is cross Is T 1 、T 2 The intersection of the curve functions;
the Ts is a vector control operation period; v4 and V6 are non-zero voltage vectors corresponding to the 1 st sector; t0 is the sum of action time of zero voltage vectors V0 and V7; vdc is the bus voltage; theta ref Is the included angle between the voltage vector Vref and the non-zero voltage vector V4;
and step 3: at the lowest set speed n set_min And (3) judging whether a low modulation unobservable region reconstructed by single resistance current exists or not when the motor runs in no-load mode: as can be seen from FIGS. 5-1 and 5-2, when
When the space vector of the reference voltage is atOutside the low modulation unobservable region of single resistance current reconstruction, no longer will ever appear->
While being less than T min I.e. single resistance current reconstruction, there is no low modulation unobservable region.
When in use
At some point in time, it may happen>
While being less than T min The case of (1), i.e., single resistance current reconstruction, exists in the low modulation unobservable region.
And 4, step 4: if the single resistance current reconstruction does not enter the low modulation unobservable region, directly turning to the step 5, otherwise, increasing the back electromotive force coefficient of the permanent magnet synchronous motor under the condition of keeping the full rate of the motor groove basically unchanged by increasing the number of turns of the motor winding and correspondingly reducing the wire diameter of the winding, returning to the step 1 to recalculate the T min 。
The voltage vector Vref for calculating T1, T2 in equations (5), (6) can be calculated from the direct-axis voltage Vd and the quadrature-axis voltage Vq:
and the steady state voltage equation of the permanent magnet synchronous motor:
wherein i d 、i q D and q axis currents respectively; r is a stator resistor; l is d 、L q D-axis and q-axis inductors respectively; omega is the electrical angular velocity; k e Is the counter electromotive force coefficient of the permanent magnet synchronous motor.
V can be derived from the above equations (7), (8) ref And K e Is (2)Comprises the following steps:
|V ref |∝K e (9)
further, the following equations (3) and (4) can be derived:
T 1 ∝K e ，T 2 ∝K e (10)
t1, T2 and the back emf K e Is proportional to the magnitude of, and T cross Is the intersection of T1 and T2, therefore T cross Also with K e In direct proportion, increase K e Can achieve the purpose of
The purpose of (1).
According to the design principle of the motor, the back electromotive force coefficient K of the permanent magnet synchronous motor can be increased by increasing the number of turns of the motor winding and correspondingly reducing the wire diameter of the winding e In actual operation, in order to ensure that the ampere-turns of the motor are not changed, the phase current of the motor can be properly increased, and the problem that the motor generates heat can not be caused by properly increasing the phase current under the conditions of low-speed operation and light motor load.
And 5: by preceding processing at n set_min When the motor is in no-load, the single-resistor current reconstruction does not enter a low-modulation unobservable region, so that the running speed n of the motor is increased set ＞n set_min During the process, the single-resistor current reconstruction can not enter a low-modulation unobservable region but only has a sector transition region unobservable region, so that the T1 and the T2 can be further subjected to amplitude limiting treatment to reduce burrs of waveforms of the T1 and the T2, the influence of sampling errors and the distortion of a motor phase current waveform are reduced, and the decibel value of low-speed noise of a single-resistor sampling scheme is reduced.
It can be seen from fig. 6 that the variation curves of T1 and T2 are substantially sawtooth-shaped, so that the amplitude limiting process for T1 and T2 is much simpler than for the detected phase current.
1) According to the lowest set speed n set Calculating the electrical period of each phase current
Wherein p is n The number of pole pairs of the motor is shown;
each phase current period comprises 6 sectors, and the control iterative operation times of each sector are further calculated
Wherein T is s The sampling period in fig. 2.
2) Calculating the maximum amplitude of the change of each vector frequency conversion control iterative operation T1 and T2
Wherein T is 1_max 、T 2_max Maximum values of T1 and T2, respectively, T 1_min 、T 2_min Is the minimum value of T1 and T2 respectively, and xi is the amplification factor, and xi =1.3 is taken here.
3) According to the condition that T1 and T2 are in the rising period or the falling period, the T1 and T2 are subjected to amplitude limiting processing
If T1 is in the rise phase:
T 1 (n)＜T 1 (n-1)+ΔT 1_max (13)
if T1 is in the decline phase:
T 1 (n)＞T 1 (n-1)-ΔT 1_max (14)
if T2 is in the rise phase:
T 2 (n)＜T 2 (n-1)+ΔT 2_max (15)
if T2 is in the decline phase:
T 2 (n)＞T 2 (n-1)-ΔT 2_max (16)
this embodiment experiment verifies that the PMSM who adopts is an external rotor fan motor who is applied to domestic air purifier, and wherein PMSM's parameter is: rated voltage DC 24V, minimum operating speed n set_min =500 revolutions per minute (rpm), maximum operating speed n set_max =3000 rpm, number of pole pairs p n =2, stator resistance R s =1.6 Ω, stator direct axis inductance L d =1.0mH, quadrature axis inductance L q =1.2mH, counter potential coefficient k before optimization e And the frequency of the vector control PWM is 16KHz and is 3.0 Vkrpm. According to the hardware parameters of the system and the dead time setting of the power tube, the minimum sampling time T of single resistance sampling min Set to 4 μ s.
In this embodiment, a single-resistor sampling permanent magnet synchronous motor system adopts position-sensorless vector control, and as shown in fig. 7, the system control block diagram includes units such as current sampling, rotor position estimation, clarke and PARK transformation, maximum torque-to-current ratio control (MTPA), speed loop, dq-axis current loop, PARK inverse transformation, SVPWM calculation, and three-phase PWM inverter.
Counter potential coefficient k before optimization e =3.0Vkrpm, and is the waveform of T1 and T2 before Ke optimization as shown in FIG. 8-1, and the intersection point T of T1 and T2 cross Approximately 7 mus, while the minimum sampling time Tmin is 4 mus, not satisfying the condition T cross /2＞T min This means that the single resistance current reconstruction may enter both the low modulation unobservable region and the sector transition region unobservable region, and in order to obtain sufficient sampling time, the control algorithm needs to perform frequent pulse phase shifting, which results in extremely asymmetric PWM output waveforms, greatly increasing phase current harmonic components, and increasing sampling errors.
According to the scheme of the invention, the counter electromotive force coefficient of the motor is changed from k e Increasing k to 3.0Vkrpm e =4.2Vkrpm, and experimental verification shows that the motor can also be operated safely at 3000rpm, as shown in fig. 8-2, the waveforms of T1 and T2 after Ke optimization are shown, and as shown in the intersection point T of T1 and T2 cross Increases from 7 mus to 10 mus, when the condition T is satisfied cross /2＞T min The single resistance current reconstruction does not enter a low modulation non-observable region, but only enters a sector transition region non-observable region, and the frequency of pulse phase shifting can be greatly reduced through the processing.
It can be seen from fig. 8-2 that the T1, T2 waveforms also have many glitches, especially at the intersection of T1 and T2, which may result in single resistance current reconfiguration into low modulationThe method of the present invention performs amplitude limiting treatment on T1 and T2 waveforms, where T is taken in the figure 1_max ＝T 2_max ＝20μs，T 1_min ＝T 2_min =0, number of iterative operations per sector
ξ =1.3, as shown in fig. 8-3, the T1 and T2 curve waveforms after the amplitude limiting processing is performed according to the formulas (13) to (16), as can be seen by comparing the T1 and T2 waveforms before and after the amplitude limiting processing, the burrs of the T1 and T2 waveforms can be greatly reduced by the amplitude limiting processing, the influence of sampling errors shown by the actual waveform of the phase current is also greatly reduced, and the distortion of the phase current waveform of the motor is greatly reduced.
FIGS. 10-1 to 10-2 show the motor at n before and after the optimization of the present invention set_min The comparison of noise frequency spectrums at 500rpm shows that the total noise of the motor reaches 22.8dB before optimization, and because the pulse phase shift times are frequent, one noise reaches 14dB at 16KHz, and the noise with the frequency exceeding 10KHz is often regarded as abnormal noise in engineering; after the optimization treatment, the total noise of the motor is reduced to 19.6dB, the abnormal sound at 16KHz is also reduced to 4dB, and the graphs from 10-1 to 10-2 show that the invention has practical effect on inhibiting the motor noise.
Variations and modifications to the above-described embodiments may occur to those skilled in the art, which fall within the scope and spirit of the above description. Therefore, the present invention is not limited to the specific embodiments disclosed and described above, and some modifications and variations of the present invention should fall within the scope of the claims of the present invention. Furthermore, although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation.
Claims (6)
1. A method for suppressing low-speed noise of a single-resistor sampling permanent magnet synchronous motor is characterized by comprising the following steps:
step 1: determining the minimum sampling time of single resistance sampling according to the parameters of the single resistance sampling permanent magnet synchronous motor driving systemT min ；
And 2, step: at the lowest set speed n set_min Calculating vector frequency conversion to control action time T of two non-zero voltage vectors corresponding to each sector 1 、T 2 And calculating T 1 And T 2 Cross point of (A) T cross The size of (d);
and step 3: judging the lowest set rotating speed n set_min Whether a low modulation unobservable region reconstructed by single-resistor three-phase current exists or not is judged;
when the temperature is higher than the set temperature
When the reference voltage space vector is determined to be outside the low modulation unobservable region reconstructed by the single resistance current;
when the temperature is higher than the set temperature
When the single-resistance current reconstruction exists, judging that a low modulation unobservable region exists;
and 4, step 4: if the single-resistor three-phase current reconstruction does not have a low modulation unobservable region, directly turning to the step 5;
otherwise, increasing the number of turns of the motor winding and correspondingly reducing the wire diameter of the winding, increasing the back electromotive force coefficient of the permanent magnet synchronous motor under the condition of maintaining the slot filling rate of the motor to be basically unchanged, and returning to the step 1 to recalculate the T min ；
And 5: according to T 1 、T 2 Determining T in each vector frequency conversion control iterative operation process 1 、T 2 Varying maximum amplitude, during operation, for T 1 、T 2 The change of (2) is subjected to clipping processing.
2. The method for suppressing low-speed noise of a single-resistor sampling permanent magnet synchronous motor according to claim 1, wherein in the step 1: t is min ＝T d +T set +T conv Wherein Td, tset and Tconv are system parameters, T d Is dead time, T set Establishing time, T, for bus current conv Is the AD conversion time.
3. The method for suppressing low-speed noise of a single-resistor sampling permanent magnet synchronous motor according to claim 1, wherein the permanent magnet synchronous motor vector control comprises six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 Six non-zero voltage vectors V 1 ～V 6 And two zero voltage vectors V 0 、V 7 The voltage space plane is divided into 6 sectors, and in each period T s Within, an arbitrary target voltage vector V in each sector ref Each generated by the joint combination of two non-zero voltage vectors and a zero vector of the sector, wherein T of the 1 st sector 1 、T 2 Obtained from the following equation:
T s ＝T 0 +T 1 +T 2 (2)
the T is cross Is of size T 1 、T 2 The intersection of the curve functions;
the T is s Controlling an operation period for one vector; v 4 、V 6 A non-zero voltage vector corresponding to sector 1; t is 0 Is a zero voltage vector V 0 、V 7 The sum of the action times of (a); v dc Is the bus voltage; theta ref Is a voltage vector V ref And non-zero voltage vector V 4 The included angle therebetween.
4. The method for suppressing the low-speed noise of the single-resistor sampling permanent magnet synchronous motor according to claim 1, wherein the T in each iterative operation process of vector frequency conversion control is determined in the step 5 1 、T 2 The specific steps for varying the maximum amplitude are as follows:
step 5-1: according to the lowest set speed n set_min Calculating the electrical period of each phase current
Wherein p is n The number of pole pairs of the motor is;
further calculating the number of control iterative operations of each sector
Wherein T is s Is a sampling period;
step 5-2: calculating the maximum amplitude of the change of each vector frequency conversion control iterative operation T1 and T2:
wherein T is 1_max 、T 2_max Maximum values of T1 and T2, respectively, T 1_min 、T 2_min Is the minimum value of T1 and T2 respectively, and xi is the amplification factor.
5. The method for suppressing the low-speed noise of the single-resistance sampling permanent magnet synchronous motor according to claim 4,
the xi is taken as the value: ξ =1.3.
6. The method for suppressing the low-speed noise of the single-resistor sampling permanent magnet synchronous motor as claimed in claim 4, wherein the T in the step 5 is subjected to the operation process 1 、T 2 The specific steps of performing the clipping process for the change of (2) are as follows:
step 5-3: according to T 1 、T 2 In the case of a rising or falling phase, for T 1 、T 2 Carrying out amplitude limiting treatment:
if T1 is in the rise phase: t is 1 (n)＜T 1 (n-1)+ΔT 1_max ；
If T1 is in the decline phase: t is a unit of 1 (n)＞T 1 (n-1)-ΔT 1_max ；
If T2 is in the rise phase: t is 2 (n)＜T 2 (n-1)+ΔT 2_max ；
If T2 is in the decline phase: t is a unit of 2 (n)＞T 2 (n-1)-ΔT 2_max ；
Wherein n is the number of iterative operations of vector frequency conversion control, and delta T 1_max T calculated for step 5-2 1 Maximum amplitude of change, Δ T 2_max T calculated for step 5-2 2 The maximum magnitude of the change.
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| CN113224992B (en) * | 2021-04-27 | 2022-04-22 | 广东工业大学 | Method for inhibiting distortion of single-resistor sampling current of permanent magnet synchronous motor |
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| CN115347842B (en) * | 2022-10-12 | 2023-01-24 | 旋智电子科技（上海）有限公司 | Permanent magnet synchronous motor current control method based on constrained model predictive control |
| CN116015128B (en) * | 2022-11-18 | 2023-08-22 | 浙江大学 | Servo motor controller PWM control method based on single-resistance current sampling |
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| CN108900123A (en) * | 2018-07-20 | 2018-11-27 | 魏海峰 | The single resistor current reconstructing method and device of permanent magnet synchronous motor |
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