CN112653369A - Multi-mode pulse width modulation switching method - Google Patents

Multi-mode pulse width modulation switching method Download PDF

Info

Publication number
CN112653369A
CN112653369A CN202011191677.XA CN202011191677A CN112653369A CN 112653369 A CN112653369 A CN 112653369A CN 202011191677 A CN202011191677 A CN 202011191677A CN 112653369 A CN112653369 A CN 112653369A
Authority
CN
China
Prior art keywords
switching
flux linkage
phase
stator
pulse width
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
CN202011191677.XA
Other languages
Chinese (zh)
Other versions
CN112653369B (en
Inventor
苑国锋
沈阳
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
North China University of Technology
Original Assignee
North China University of Technology
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by North China University of Technology filed Critical North China University of Technology
Priority to CN202011191677.XA priority Critical patent/CN112653369B/en
Publication of CN112653369A publication Critical patent/CN112653369A/en
Application granted granted Critical
Publication of CN112653369B publication Critical patent/CN112653369B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/141—Flux estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention discloses a switching method of multi-mode pulse width modulation, which comprises the following steps: obtaining switching sequences of different modulation modes and switching sequences of different switching angles N; obtaining a pre-switching sequence P (m, N)1) And a switched switching sequence P (m, N)2) (ii) a Obtaining stator flux linkage errors caused by modulation mode switching; obtaining flux linkage error d psi under two-phase static coordinate systemαAnd d psiβ(ii) a Obtaining the switching time compensation time of each phase; correcting the switched switching sequence to obtain a new switching sequence; and generating PWM pulses according to the corrected switching sequence, driving an inverter and further driving the motor. The switching method of the multi-mode pulse width modulation provided by the invention has the advantages of not depending on specific modulation modes and motor parameters, realizing switching at any time and improving stator flux linkage errorRapidity and accuracy of the elimination, etc.

Description

Multi-mode pulse width modulation switching method
Technical Field
The invention relates to a multi-mode pulse width modulation switching method, in particular to a multi-mode pulse width modulation switching method applied to a high-power traction transmission system, and belongs to the field of motor control.
Background
The traction transmission system has the characteristics of high voltage, high power and large current, the switching frequency of the traction transmission system is limited by the heat dissipation system, and the switching frequency of the locomotive traction converter is only hundreds of hertz.
Compared with a common inverter, the lower switching frequency of the locomotive traction system puts higher requirements on a modulation algorithm of the inverter, and the modulation algorithm is specifically represented by the following two points: firstly, the speed range is wide in the operation process, which causes the carrier ratio to be greatly changed, so that a plurality of different modulation strategies need to be adopted to meet the requirements of different operation speeds; secondly, smooth switching between different modulation modes or between the same modulation mode and different carrier ratios needs to be realized, and smooth operation of the traction system is ensured.
As for the Modulation mode under the condition of low carrier ratio, a great deal of research has been carried out by domestic and foreign scholars, and various Pulse Width Modulation (PWM) theories and realization methods thereof under the condition of low carrier ratio are provided and optimized on the basis. Including synchronous SVPWM, SHEPWM (Selected Harmonic excitation PWM), and CHMPWM (Current Harmonic Minimum PWM), each modulation method has advantages and disadvantages. In practical applications, in order to ensure stable operation of the system, it is necessary to ensure smooth transition of the system between different modulation modes through a reasonable switching strategy.
In a traction drive system, the rotor can be considered as an ideal waveform for the motor due to the time constant of the rotor, and the torque ripple and the current impact generated by the motor mainly depend on the amplitude and phase angle change of the stator flux linkage. In the modulation mode switching process, if the stator current stator flux linkage before and after the modulation mode switching can be ensured to be continuous, the stable output of the motor torque in the switching process can be realized.
The currently applied and widely applied switching methods are mainly three-phase independent switching and three-phase synchronous switching. The three-phase independent switching is realized by analyzing a current transient response equation and selecting a point with zero harmonic current transient response for switching, but the harmonic current transient response analysis process used by the method is complicated, the harmonic current transient response in different switching processes is different, the universality is not strong, and when level jump is scattered, pulse disorder is easily caused by independent switching, so that the system overcurrent is caused, and the practical application significance is not sufficient.
The three-phase synchronous switching method comprises the steps of analyzing and comparing current characteristics of different switching points or stator flux linkage characteristics, and selecting an optimal switching point, wherein the current characteristics are analyzed based on current harmonic characteristics of the different switching points, the influence of harmonic current is considered, and the optimal switching point is selected, but the current harmonic analysis is relatively difficult, the difference is huge under different modulation modes, and the universality is not strong; the method can only carry out optimal selection according to the existing switching points, does not correspondingly process the deviation caused by switching, and can carry out switching only by waiting for the optimal switching point, so that the system has insufficient rapidity.
The existing switching strategy method can only select from the existing switching points, the switching can be carried out only by waiting for the arrival of the optimal switching point, and even if the optimal switching point is the optimal switching point, certain flux linkage deviation can still be caused due to the fixed deviation existing between the optimized flux linkage tracks in the modulation modes, so that current impact and torque pulsation are generated in the switching process, and the method has certain limitation.
Therefore, it is necessary to design a switching method that has strong versatility, does not need to wait for the optimal switching point, and can directly eliminate the stator flux linkage error caused by switching, so as to make fast and smooth transition between different modulation modes.
Disclosure of Invention
In order to overcome the above problems, the present inventors have conducted intensive studies and have proposed a switching method of multi-mode pulse width modulation, including the steps of:
s1, obtaining switching sequences of different modulation modes and switching sequences of different switching angle numbers N;
s2, obtaining a switching sequence P (m, N) before switching1) And a switched switching sequence P (m, N)2)；
S3, obtaining stator flux linkage errors caused by modulation mode switching;
s4, acquiring flux linkage error d psi under the two-phase static coordinate systemαAnd d psiβ；
S5, obtaining the switch time compensation time of each phase;
s6, correcting the switched switching sequence to obtain a new switching sequence;
and S7, generating PWM pulses according to the corrected switching sequence, driving the inverter and further driving the motor.
According to the invention, in step S3, according to the pre-switching sequence P (m, N)1) Switched switching sequence P (m, N)2) Reconstructing steady state stator voltage before switching with motor stator fundamental frequency omega at switching moment
Figure BDA0002752962200000031
And steady state stator voltage after switching
Figure BDA0002752962200000032
By reconstructing the steady state stator voltage
Figure BDA0002752962200000033
Integral calculation is carried out to obtain the actual stator flux linkage track at the switching moment
Figure BDA0002752962200000034
And expected stator flux linkage trajectory
Figure BDA0002752962200000035
Further, the actual stator flux linkage path at the moment of switching
Figure BDA0002752962200000036
As follows:
Figure BDA0002752962200000037
wherein, t1To initiate the transfer of the switching sequence P (m, N)1) At the moment of switching, steady state stator voltage before switching
Figure BDA0002752962200000041
Is P (m, N)1) Corresponding PWM output steady-state stator voltage psiss(t1) Is t1And (5) starting the stator flux linkage at the moment.
Further, when t is2The working condition changes at any moment, and the stator flux linkage vector expression is as follows:
Figure BDA0002752962200000042
wherein the switched steady state stator voltage
Figure BDA0002752962200000043
Is P (m, N)2) The corresponding PWM outputs a steady state stator voltage,
Figure BDA0002752962200000044
is composed of
Figure BDA0002752962200000045
At t ═ t2The flux linkage amplitude of (a).
Preferably, a stator flux linkage trajectory is desired
Figure BDA0002752962200000046
Can be expressed by the following formula:
Figure BDA0002752962200000047
wherein the content of the first and second substances,
Figure BDA0002752962200000048
is psiss (2)(t) at t ═ t2The amplitude of the flux linkage of (a),
stator flux linkage error:
Figure BDA0002752962200000049
according to the invention, in step S4, the stator flux linkage error in the three-phase stationary coordinate system is transformed into an alpha-beta two-phase stationary coordinate system through coordinate transformation, and the alpha and beta axis flux linkage error d psi is calculatedαAnd d psiβ。
In step S5, a flux linkage error correction rule is established based on the influence of the switching time adjustment of each phase on the α and β axis flux linkages, and the switching time compensation time is calculated.
Preferably, the flux linkage error correction principle includes three principles, respectively:
principle 1: when flux linkage error only exists alpha-axis component d psiα(t), the compensation time for each phase switching time is as follows:
Figure BDA0002752962200000051
principle 2: when flux linkage error is equal to alpha axis flux linkage error component d ψα(t) and the beta axis flux linkage error component d ψβ(t) when the positive and negative phases are the same, the compensation of the switching time of each phase is as follows:
Figure BDA0002752962200000052
principle 3: when the alpha axis flux linkage error component d ψα(t) and the beta axis flux linkage error component d ψβ(t) when the phases are different in positive and negative, the compensation time of the switching time of each phase is as follows:
Figure BDA0002752962200000053
wherein, UdcRepresenting the bus voltage, sa、sb、scEach represents a three-phase output voltage level change of the inverter A, B, C, and if the output level jumps from a low level to a high level, s is equal to 1; if the output level jumps from the high level to the low level, s is equal to-1; s is 0 if there is no level change within one sampling period.
In a preferred embodiment, when the sampling period in which the switching process is located does not have a switching angle, the duty cycle of each phase in the sampling period is modified according to the compensation time of the switching angle of each phase, and the modified duty cycle is represented by formula (thirteen):
Figure BDA0002752962200000061
wherein, Δ daDenotes the magnitude of the duty modification of the A phase, Δ dbRepresents the B-phase duty ratio modification size, Δ dcRepresents the B-phase duty ratio modification size, TsRepresenting the sampling period.
The invention has the following beneficial effects:
(1) the switching method of the multi-mode pulse width modulation provided by the invention does not need to select and judge the switching point, and directly corrects the dynamic flux linkage mutation caused by the switching of the modulation mode, thereby realizing the switching at any moment, being independent of a specific modulation mode and motor parameters, being applicable to all optimized modulation modes, providing a processing strategy with stronger universality and better performance for the switching process of the modulation mode of a high-power traction traditional system, and enabling the traction system to be capable of more quickly and stably transiting from a low speed to a high-speed running state;
(2) according to the switching method of the multi-mode pulse width modulation, the voltage waveform actually acting on the inverter is reconstructed and integrated, and the actual flux linkage track before and after switching is directly calculated, so that the problem of deviation between the observed flux linkage and the actual flux linkage is solved, and the accuracy of stator flux linkage error calculation is improved;
(3) according to the switching method of the multi-mode pulse width modulation, the stator flux linkage error compensation method based on the switch angle correction is used, the dynamic stator flux linkage error can be completely eliminated in a few sampling periods, the multi-mode pulse width modulation strategy is switched at any time, the problem that the response speed of the existing widely-used selection switching strategy based on the optimal switching point is low is solved, and the switching performance of the optimal switching point can be further optimized;
(4) the switching method of the multi-mode pulse width modulation provided by the invention is a three-phase synchronous switching mode for compensating flux linkage. By compensating the switched stator flux linkage, the current impact and the torque ripple caused by the switching of the modulation mode are reduced, and the limitation that the multi-mode modulation mode three-phase synchronous switching needs to wait for the optimal switching point is effectively solved.
Drawings
Fig. 1 shows a schematic diagram of a switching method of multi-mode pulse width modulation according to a preferred embodiment of the present invention;
FIG. 2 shows a schematic diagram of a multi-mode modulation voltage waveform of a preferred embodiment of the present invention;
FIG. 3 is a schematic diagram showing the results of a 9-division SHEPWM switching angle solution in accordance with a preferred embodiment of the present invention;
FIG. 4 is a schematic diagram illustrating transient flux linkage error during switching of operating modes in accordance with a preferred embodiment of the present invention;
FIG. 5 illustrates a schematic diagram of the switch angle correction for different level cases in accordance with a preferred embodiment of the present invention;
FIG. 6 shows a schematic diagram of an error correction strategy according to a preferred embodiment of the present invention;
FIG. 7 shows 5-pulse SHEPWM switching 3-pulse SHEPWM flux linkage traces without flux linkage compensation applied in example 1 of the present invention and comparative example 1;
FIG. 8 is a schematic diagram showing the flux linkage trajectory of 5-pulse SHEPWM switching 3-pulse SHEPWM after flux linkage compensation is applied in embodiment 1 of the present invention;
FIG. 9 shows simulated waveforms for 5 pulse SHEPWM switching 3 pulse SHEPWM motor line voltage, stator current and output torque without compensation applied in comparative example 1 of the present invention;
fig. 10 shows simulated waveforms for applying the compensated 5 pulse SHEPWM switching 3 pulse SHEPWM motor line voltage, stator current and output torque in embodiment 1 of the present invention.
Detailed Description
The invention is explained in more detail below with reference to the drawings and preferred embodiments. The features and advantages of the present invention will become more apparent from the description.
The invention provides a switching method of multi-mode pulse width modulation, as shown in fig. 1, comprising the following steps:
s1, obtaining switching sequences of different modulation modes and switching sequences of different switching angle numbers N;
s2, obtaining a switching sequence P (m, N) before switching1) And a switched switching sequence P (m, N)2)；
S3, obtaining stator flux linkage errors caused by modulation mode switching;
s4, acquiring flux linkage error d psi under the two-phase static coordinate systemαAnd d psiβ；
S5, obtaining the switch time compensation time of each phase;
s6, correcting the switched switching sequence to obtain a new switching sequence;
and S7, generating PWM pulses according to the corrected switching sequence, driving the inverter and further driving the motor.
In step S1, the modulation pattern generation voltage waveform has half-wave symmetry and quarter-wave symmetry, and a typical waveform thereof is shown in fig. 2, taking the SHEPWM modulation pattern as an example, the profile of the switching sequence with the modulation when N is 4 is shown in fig. 3,
the switching sequences of different modulation modes are obtained by solving voltage harmonic nonlinear equations corresponding to different modulation modes;
the switching sequences with different numbers of switching angles N are obtained by solving phase voltage equations with different times;
the specific solving process can adopt any one of the known methods, and is not described in detail in the present invention.
Further, the obtained switching sequences of different modulation modes and the switching sequences at different numbers of switching angles N are stored in a table form so as to be called when PWM pulses are subsequently generated.
In step S2, the pre-switch switching sequence P (m, N)1) And a switched switching sequence P (m, N)2) The switching sequence, the degree of modulation m at the switching instant, and the number of switching angles N before switching are obtained according to step S11And the number of switch angles N after switching2The specific calculation method can be any known method, and is not described in detail in the present invention.
In step S3, according to the pre-switching sequence P (m, N)1) Switched switching sequence P (m, N)2) Reconstructing steady state stator voltage before switching with motor stator fundamental frequency omega at switching moment
Figure BDA0002752962200000091
And steady state stator voltage after switching
Figure BDA0002752962200000092
Further, by applying the reconstructed steady-state stator voltage
Figure BDA0002752962200000093
Integral calculation is carried out, and the actual stator flux linkage track at the switching moment can be obtained
Figure BDA0002752962200000094
And expected stator flux linkage trajectory
Figure BDA0002752962200000095
To pair
Figure BDA0002752962200000097
And obtaining the flux linkage error of the stator by making difference.
In particular, the steady state voltage
Figure BDA0002752962200000098
According to a pre-switching sequence P(m,N1) Switched switching sequence P (m, N)2) And the fundamental frequency omega of the motor stator are obtained by PWM pulse voltage reconstruction,
Figure BDA0002752962200000099
by means of a steady-state voltage
Figure BDA00027529622000000910
Obtained by performing integration:
switching transient actual stator flux linkage trajectory
Figure BDA00027529622000000911
As follows:
Figure BDA00027529622000000912
wherein, t1To initiate the transfer of the switching sequence P (m, N)1) At the moment of switching, steady state stator voltage before switching
Figure BDA00027529622000000913
Is P (m, N)1) Corresponding PWM output steady-state stator voltage psiss(t1) Is t1And (5) starting the stator flux linkage at the moment.
When t is2The working condition changes at any moment, and the stator flux linkage vector expression is as follows:
Figure BDA00027529622000000914
wherein the switched steady state stator voltage
Figure BDA00027529622000000915
Is P (m, N)2) The corresponding PWM outputs a steady state stator voltage,
Figure BDA00027529622000000916
is composed of
Figure BDA00027529622000000917
At t ═ t2The flux linkage amplitude of (a).
Expected stator flux linkage trajectory
Figure BDA00027529622000000918
Can be expressed by the following formula:
Figure BDA0002752962200000101
wherein the content of the first and second substances,
Figure BDA0002752962200000102
is psiss (2)(t) at t ═ t2The flux linkage amplitude of (a).
Due to the fact that
Figure BDA0002752962200000103
So that t is t2Actual flux linkage at time
Figure BDA0002752962200000104
Is not optimizing the flux linkage track psiss (2)(t), and thus dynamic modulation errors. Taking the example of switching from divide-by-5 SHEPWM to divide-by-3 SHEPWM, the flux error between the two is schematically shown in FIG. 4, where the dotted line is the divide-by-5 SHEPWM switching sequence P (m, N)1) Corresponding to the optimized flux linkage trajectory, the solid line is a 3-division SHEPWM switching sequence P (m, N)2) Corresponding optimized flux linkage track, wherein the point O is the center of the flux linkage track, and the point A is
Figure BDA0002752962200000105
A value of (B) is
Figure BDA0002752962200000106
There is a significant deviation between the two. Therefore, the switching time t can be obtained as t2Real flux linkage and steady state optimized flux linkage
Figure BDA0002752962200000107
The difference between the two values gives the stator flux linkage error, and the expression is obtained as follows:
Figure BDA0002752962200000108
the stator flux linkage error is stored in a table form so as to be called in real time according to the modulation degree m and the phase of the fundamental wave.
The traditional strategy is to take an observed flux linkage track output by a flux linkage observer as a motor stator flux linkage before switching, and take a difference between the observed flux linkage track and an expected stator flux linkage to obtain a dynamic stator flux linkage error in a switching process, however, the stator flux linkage output by the observer is only a fundamental flux linkage, an actual flux linkage contains various harmonic flux linkages, and the observed flux linkage has a larger deviation from the actual flux linkage. In the present invention, the actual stator flux linkage path at the moment of switching
Figure BDA0002752962200000109
The method is obtained by solving PWM pulses actually acting on the inverter, harmonic flux linkage is fully considered, and the solved result is closer to the actual stator flux linkage than the result obtained by an observer. Therefore, compared with the existing stator flux linkage error calculation strategy which makes difference between the observed flux linkage and the target flux linkage, the dynamic stator flux linkage error calculation method provided by the invention is more in line with the actual switching process, and the error calculation result is more accurate.
In step S4, the stator flux linkage error in the three-phase stationary coordinate system is transformed into the two-phase stationary coordinate system (α - β coordinate system) by coordinate transformation, and the α -axis and β -axis flux linkage errors d ψ are calculatedαAnd d psiβAnd calculating the three-phase voltage sequence adjustment quantity delta tai、Δtbi、ΔtciInfluence on the magnetic linkage of alpha and beta axes.
Specifically, the switching sequence adjustment process and its physical quantity definition are shown in fig. 5. According to the flux linkage trajectory equation, it can be known that increasing the duration of the high level can increase the flux linkage, decreasing the duration of the high level can decrease the flux linkage, and the principle of correcting the low level situation is similar to this, so that the principle of correcting the switching time can be obtained, as shown in the following formula:
Figure BDA0002752962200000111
where Δ t represents the time adjustment for one switching instant: Δ t > 0 indicates an advance of the switching time, and Δ t < 0 indicates a delay of the switching time; s represents the inverter three-phase output voltage level change, sa、sb、scFor A, B, C three phases respectively, if the output level jumps from low level to high level, s is equal to 1; if the output level jumps from the high level to the low level, s is equal to-1; if the level is not changed in one sampling period, s is equal to 0; if there are multiple level changes within a sampling period, they are distinguished by subscript i.
According to the rule of the two-level inverter, the relationship between the output phase voltage of the inverter and the switching state of a bridge arm is shown as the following formula:
Figure BDA0002752962200000112
wherein, Uan、Ubn、UcnA, B, C phase voltage, UdcRepresents the bus voltage, S represents the switching state of the three-phase arms of the inverter: sA、SB、SCCorresponding to A, B, C three phases, S is 1 when the upper arm is on, and S is 0 when the lower arm is on.
Because the change of the single-phase switching moment can not only change the phase voltage waveform of the phase, but also influence the voltages of the other two phases, in order to solve the coupling problem existing in flux linkage error elimination under three-term static coordinate systems, the elimination problem of dynamic flux linkage error is discussed in an alpha-beta two-phase static coordinate system, and coordinate transformation is carried out on the phase voltages, so that the relational expression of the voltages under the two-phase static coordinate system and the bridge arm switching state is shown as the following formula:
Figure BDA0002752962200000121
wherein, Uα、UβThe voltages of the alpha and beta axes.
The influence of the change of the three-phase switching time on the alpha-axis voltage and the beta-axis voltage can be obtained by the formula (seven), the change of the A-phase switching time only affects the alpha-axis voltage, the change amounts of B relative to the alpha-axis voltage and the beta-axis voltage are just different, and the change amounts of C relative to the alpha-axis voltage and the beta-axis voltage are just the same, so that the stator flux linkage expression under the two-phase static coordinate system is as follows:
ψs＝ψα+jψβ(eight)
Wherein psisRepresenting the stator flux linkage vector and j representing the imaginary component.
The time adjustment quantity delta t of the ith switching moment in the three-phase voltage sequence in one sampling periodai、Δtbi、ΔtciChange amount of dynamic stator flux linkage error:
Figure BDA0002752962200000122
according to the formulas (seven) to (nine), according to the influence characteristics of the ABC three-phase switching sequence on the alpha and beta axis voltages, the A phase completely only influences the alpha axis flux linkage, so that the B, C phase switching time is adjusted to completely eliminate the beta axis flux linkage error, then the A phase switching time is adjusted to uniformly eliminate the alpha axis flux linkage error and the coupling flux linkage error caused by B, C phase switching sequence adjustment, and the flux linkage is completely eliminated, so that the flux linkage error d psi under a two-phase static coordinate system is obtainedαAnd d psiβ。
In step S5, a flux linkage error correction rule is established based on the influence of the switching time adjustment of each phase on the α and β axis flux linkages, and the switching time compensation time is calculated.
The flux linkage error correction principle comprises three principles, and the error adjustment process is shown in a figure (6):
principle 1: d ψ (t) in FIG. 6(1)It is shown that when the flux linkage error exists only the α -axis component d ψαAnd (t), the A-phase switching sequence is only required to be adjusted to eliminate the A-phase switching sequence.
The switching time compensation time for each phase is as follows:
Figure BDA0002752962200000131
principle 2: d ψ (t) in FIG. 6(2)As shown, the flux linkage error component d ψ is obtained when the flux linkage error is equal to the α axis flux linkage errorα(t) and the beta axis flux linkage error component d ψβ(t) when positive and negative phases are the same, the C-phase and A-phase switching sequences are adjusted simultaneously to eliminate d ψα(t) and d ψβ(t)。
The magnitude of the compensation at the switching time of each phase is as follows:
Figure BDA0002752962200000132
principle 3: d ψ (t) in FIG. 6(3)As shown, when the α -axis flux linkage error component d ψα(t) and the beta axis flux linkage error component d ψβ(t) in case of positive and negative differences, the switching sequences for the phases B and A are adjusted simultaneously to eliminate d ψα(t) and d ψβ(t)。
The switching time compensation time for each phase is as follows:
Figure BDA0002752962200000141
the expressions (ten) to (twelve) are all the conditions that the switching instant sampling period needs to correct the phase existing level, preferably, when the sampling period in which the switching process is located does not have a switching angle, the duty ratio of each phase in the sampling period is changed according to the compensation time of the switching angle of each phase obtained by the expressions (ten) to (twelve), because the sampling period does not have level jump, the duty ratio of the sampling period is 1 or 0, and the amplitude of the changed duty ratio is as shown in the expression (thirteen):
Figure BDA0002752962200000142
wherein, Δ daDenotes the magnitude of the duty modification of the A phase, Δ dbRepresents the B-phase duty ratio modification size, Δ dcRepresents the B-phase duty ratio modification size, TsRepresenting the sampling period.
In the invention, the dynamic flux linkage mutation caused by the switching of the modulation mode is directly compensated without selecting and judging the switching point, so that the switching at any time can be realized, and the method does not depend on a specific modulation mode and motor parameters and can be applied to all optimized modulation modes.
In step S6, the time Δ t is compensated for according to the switching timing obtained in step S5a、Δtb、ΔtcUpdating the original switching sequence to obtain a new switched switching sequence P (m, N)2)new。
In the invention, the stator flux linkage error in the two-phase static coordinate system (alpha-beta coordinate system) is eliminated by using a method of correcting the switching angle. Compared with the conventional method for adjusting the switch sequence in the three-phase static coordinate system, the method can completely eliminate the dynamic stator flux linkage error in as few sampling periods as possible, solves the problem that the stator flux linkage error caused by mutual coupling of three phases cannot be completely eliminated in the conventional stator flux linkage error elimination process, and improves the rapidity and the accuracy of stator flux linkage error elimination.
In step S7, the voltage fundamental wave phase is determined
Figure BDA0002752962200000153
Frequency omega and modified switching sequence P (m, N)2)newAnd generating PWM pulse to drive the inverter and further drive the motor.
Examples
Example 1
In the process of switching from the frequency-division-5 SHEPWM to the frequency-division-3 SHEPWM, as shown in fig. 7, where a dotted line is an optimized flux linkage trajectory corresponding to the frequency-division-5 SHEPWM switching sequence at the switching time, and a dotted line is an optimized flux linkage trajectory corresponding to the frequency-division-3 SHEPWM switching sequence at the switching time, the optimized flux linkage trajectory before and after switching is obtained by performing voltage waveform reconstruction on the switching sequences before and after switching.
In particular, the actual stator flux linkage path at the moment of switching
Figure BDA0002752962200000151
As follows:
Figure BDA0002752962200000152
after the working condition changes, the stator flux linkage vector expression is as follows:
Figure BDA0002752962200000161
expected stator flux linkage trajectory
Figure BDA0002752962200000162
Can be expressed by the following formula:
Figure BDA0002752962200000163
compensating for flux linkage track errors before and after switching by:
Figure BDA0002752962200000164
the alpha-axis flux linkage error component d psi can be obtainedα(t) is 0.1417Wb, and the beta-axis flux linkage error component d ψβ(t) is-0.011 Wb, and the signs are different, so that the switching time of the B phase and the A phase are adjusted to eliminate d psiα(t) and d ψβ(t)。
According to the influence of the switching time adjustment of each phase on the alpha-axis flux linkage and the beta-axis flux linkage, establishing a flux linkage error correction principle, and according to the principle 3, when the alpha-axis flux linkage error component d psiα(t) and beta axis flux linkage error scoreQuantity d psiβ(t) in case of positive and negative differences, the switching sequences for the phases B and A are adjusted simultaneously to eliminate d ψα(t) and d ψβ(t), the compensation time of each phase switching time is as follows:
Figure BDA0002752962200000165
wherein U isdcAt 540V, the switching time of the a-phase switching sequence and the switching time of the B-phase switching sequence both jump from high level to low level (s is 1), so that the specific switching time correction Δ t can be calculatedaIs-0.00038 s and Δ tbThe switching sequence is updated according to the compensation magnitude pair for 0.000035s, and PWM pulses are generated to control the inverter.
The stator flux linkage is compensated, the flux linkage track of one fundamental wave period after switching is carried out is shown as a solid line in fig. 8, after switching is finished, deviation between the actual flux linkage track and the optimized flux linkage track basically does not exist, and the center of the flux linkage track basically does not deviate. As shown in fig. 10 by line voltage, stator current and torque waveforms, the current surge and torque ripple are small, and smooth transition between different modulation modes is realized by applying a flux linkage compensation switching mode.
Comparative example 1
During the switching from 5-division SHEPWM to 3-division SHEPWM, the switching is performed when the phase of the fundamental wave is 155 degrees, and when compensation is not performed according to flux linkage error, the flux linkage locus of the next fundamental wave period after switching is shown in FIG. 7, wherein a dotted line is an optimized flux linkage locus 1 corresponding to the 5-division SHEPWM switching sequence before switching, a dotted line is an optimized flux linkage locus 2 corresponding to the 3-division SHEPWM switching sequence after switching, and a solid line is an actual flux linkage locus of the next fundamental wave period after switching. In the figure, a switching point A is positioned on an optimized flux linkage track 1, a point in an optimized flux linkage track 2, which has the same phase as the fundamental wave of the point A, is a point B, and the point B and the point A have obvious flux linkage errors; and after switching, a large deviation exists between the actual flux linkage track and the optimized flux linkage track, and the center deviation of the flux linkage track is serious.
Fig. 9 shows the line voltage, stator current and torque waveforms of the motor without compensation according to flux linkage error, where the current waveform has more obvious pulse, the torque waveform has obvious impact, and the deviation of flux linkage trajectory causes the motor to have larger current impact and torque ripple.
The invention has been described in detail with reference to the preferred embodiments and illustrative examples. It should be noted, however, that these specific embodiments are only illustrative of the present invention and do not limit the scope of the present invention in any way. Various modifications, equivalent substitutions and alterations can be made to the technical content and embodiments of the present invention without departing from the spirit and scope of the present invention, and these are within the scope of the present invention. The scope of the invention is defined by the appended claims.

Claims (9)

1. A switching method of multi-mode pulse width modulation, comprising the steps of:
s1, obtaining switching sequences of different modulation modes and switching sequences of different switching angle numbers N;
s2, obtaining a switching sequence P (m, N) before switching1) And a switched switching sequence P (m, N)2)；
S3, obtaining stator flux linkage errors caused by modulation mode switching;
s4, acquiring flux linkage error d psi under the two-phase static coordinate systemαAnd d psiβ；
S5, obtaining the switch time compensation time of each phase;
s6, correcting the switched switching sequence to obtain a new switching sequence;
and S7, generating PWM pulses according to the corrected switching sequence, driving the inverter and further driving the motor.
2. The multi-mode pulse width modulation switching method of claim 1,
in step S3, according to the pre-switching sequence P (m, N)1) Switched switching sequence P (m, N)2) Reconstructing steady state stator voltage before switching with motor stator fundamental frequency omega at switching moment
Figure FDA0002752962190000011
And steady state stator voltage after switching
Figure FDA0002752962190000012
By reconstructing the steady state stator voltage
Figure FDA0002752962190000013
Integral calculation is carried out to obtain the actual stator flux linkage track at the switching moment
Figure FDA0002752962190000014
And expected stator flux linkage trajectory
Figure FDA0002752962190000015
3. The multi-mode pulse width modulation switching method of claim 2,
switching transient actual stator flux linkage trajectory
Figure FDA0002752962190000016
As follows:
Figure FDA0002752962190000017
wherein, t1To initiate the transfer of the switching sequence P (m, N)1) At the moment of switching, steady state stator voltage before switching
Figure FDA0002752962190000018
Is P (m, N)1) Corresponding PWM output steady-state stator voltage psiss(t1) Is t1And (5) starting the stator flux linkage at the moment.
4. The multi-mode pulse width modulation switching method of claim 2,
when t is2The working condition changes at any moment, and the stator flux linkage vector expression is as follows:
Figure FDA0002752962190000021
wherein the switched steady state stator voltage
Figure FDA0002752962190000022
Is P (m, N)2) The corresponding PWM outputs a steady state stator voltage,
Figure FDA0002752962190000023
is composed of
Figure FDA0002752962190000024
At t ═ t2The flux linkage amplitude of (a).
5. The multi-mode pulse width modulation switching method of claim 2,
expected stator flux linkage trajectory
Figure FDA0002752962190000025
Can be expressed by the following formula:
Figure FDA0002752962190000026
wherein the content of the first and second substances,
Figure FDA0002752962190000027
is psiss (2)(t) at t ═ t2The amplitude of the flux linkage of (a),
stator flux linkage error:
Figure FDA0002752962190000028
6. the multi-mode pulse width modulation switching method of claim 1,
in step S4, the stator flux linkage error in the three-phase stationary coordinate system is transformed into the α - β two-phase stationary coordinate system by coordinate transformation, and the α -and β -axis flux linkage error d ψ is calculatedαAnd d psiβ。
7. The multi-mode pulse width modulation switching method of claim 1,
in step S5, a flux linkage error correction rule is established based on the influence of the switching time adjustment of each phase on the α and β axis flux linkages, and the switching time compensation time is calculated.
8. The multi-mode pulse width modulation switching method of claim 7,
the flux linkage error correction principle comprises three principles, namely:
principle 1: when flux linkage error only exists alpha-axis component d psiα(t), the compensation time for each phase switching time is as follows:
Figure FDA0002752962190000031
principle 2: when flux linkage error is equal to alpha axis flux linkage error component d ψα(t) and the beta axis flux linkage error component d ψβ(t) when the positive and negative phases are the same, the compensation of the switching time of each phase is as follows:
Figure FDA0002752962190000032
principle 3: when the alpha axis flux linkage error component d ψα(t) and the beta axis flux linkage error component d ψβ(t) when the positive and negative phases are different, each phase is switchedThe time of day compensation time is as follows:
Figure FDA0002752962190000033
wherein, UdcRepresenting the bus voltage, sa、sb、scEach represents a three-phase output voltage level change of the inverter A, B, C, and if the output level jumps from a low level to a high level, s is equal to 1; if the output level jumps from the high level to the low level, s is equal to-1; s is 0 if there is no level change within one sampling period.
9. The multi-mode pulse width modulation switching method of claim 7,
when the sampling period in which the switching process is located does not have a switching angle, changing the duty ratio of each phase in the sampling period according to the compensation time of the switching angle of each phase, wherein the amplitude of the changed duty ratio is as shown in formula (thirteen):
Figure FDA0002752962190000041
wherein, Δ daDenotes the magnitude of the duty modification of the A phase, Δ dbRepresents the B-phase duty ratio modification size, Δ dcRepresents the B-phase duty ratio modification size, TsRepresenting the sampling period.
CN202011191677.XA 2020-10-30 2020-10-30 Multi-mode pulse width modulation switching method Active CN112653369B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202011191677.XA CN112653369B (en) 2020-10-30 2020-10-30 Multi-mode pulse width modulation switching method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202011191677.XA CN112653369B (en) 2020-10-30 2020-10-30 Multi-mode pulse width modulation switching method

Publications (2)

Publication Number Publication Date
CN112653369A true CN112653369A (en) 2021-04-13
CN112653369B CN112653369B (en) 2022-04-29

Family

ID=75346823

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202011191677.XA Active CN112653369B (en) 2020-10-30 2020-10-30 Multi-mode pulse width modulation switching method

Country Status (1)

Country Link
CN (1) CN112653369B (en)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114221589A (en) * 2021-12-08 2022-03-22 北方工业大学 Multi-mode modulation switching method and device for pulse injection
CN116317767A (en) * 2022-12-27 2023-06-23 北方工业大学 Alternating current motor prediction control method with embedded optimized pulse sequence

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103210580A (en) * 2010-12-10 2013-07-17 爱信艾达株式会社 Control device
CN104935196A (en) * 2015-07-06 2015-09-23 中国矿业大学 A space vector modulation method for dual three-level inverter system zero sequence voltage elimination
CN105245159A (en) * 2015-09-08 2016-01-13 江苏大学 Five-phase permanent magnet fault-tolerant motor direct torque control method based on novel pulse width modulation

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103210580A (en) * 2010-12-10 2013-07-17 爱信艾达株式会社 Control device
CN104935196A (en) * 2015-07-06 2015-09-23 中国矿业大学 A space vector modulation method for dual three-level inverter system zero sequence voltage elimination
CN105245159A (en) * 2015-09-08 2016-01-13 江苏大学 Five-phase permanent magnet fault-tolerant motor direct torque control method based on novel pulse width modulation

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
阚京波: ""低开关频率下异步电机高性能控制技术研究"", 《中国优秀硕博士学位论文全文数据库（博士）工程科技II辑》 *

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114221589A (en) * 2021-12-08 2022-03-22 北方工业大学 Multi-mode modulation switching method and device for pulse injection
CN114221589B (en) * 2021-12-08 2023-09-29 北方工业大学 Pulse injection multi-mode modulation switching method and device
CN116317767A (en) * 2022-12-27 2023-06-23 北方工业大学 Alternating current motor prediction control method with embedded optimized pulse sequence
CN116317767B (en) * 2022-12-27 2023-10-13 北方工业大学 Alternating current motor prediction control method with embedded optimized pulse sequence

Also Published As

Publication number Publication date
CN112653369B (en) 2022-04-29

Similar Documents

Publication Publication Date Title
CN106936356B (en) Vector screening and duty ratio combined motor model prediction control system and method
CN112653369B (en) Multi-mode pulse width modulation switching method
Shen et al. Current reconstruction of three-phase voltage source inverters considering current ripple
CN110971162B (en) Model prediction torque control method of NPC three-level converter-PMSM system
CN110011359B (en) Grid-connected inverter parameter identification method under finite set model prediction control
CN113659898A (en) Double three-phase permanent magnet synchronous motor model prediction torque control method based on multi-vector continuous optimization strategy
CN110445441B (en) Permanent magnet synchronous motor predicted torque control method
CN112039409B (en) Low switching frequency control method and system for direct current bias type sine current motor system
CN108683377A (en) Five leg inverters drive double three-phase machine system Strategy of Direct Torque Control
CN108390604B (en) Zero-voltage vector optimization modulation device and method for five-bridge-arm two-permanent magnet motor system
CN110838808A (en) Diagnosis-free self-repairing method for open-circuit fault of double three-phase permanent magnet synchronous motor driving system
CN111082726B (en) Current control method of permanent magnet motor servo system
CN110350836B (en) Direct torque control method of common direct current bus type open winding permanent magnet synchronous motor system
CN110620539B (en) Direct torque control optimization method for open-winding motor under variable bus voltage working condition
CN111756287B (en) Dead zone compensation method suitable for permanent magnet motor control based on current prediction
CN112953318A (en) Nonlinear compensation method for permanent magnet synchronous motor driving system inverter
CN114759854A (en) Voltage modulation method for isolated bus type open winding permanent magnet synchronous motor
CN115913038A (en) Model prediction control method for double three-phase permanent magnet synchronous motor
CN115021636A (en) Dead-beat-based five-phase permanent magnet synchronous motor model prediction fault-tolerant control method
CN111431459B (en) Minimum copper consumption control method and device for direct current bias type sinusoidal current motor
CN113992095A (en) Low-complexity direct power control method for dual three-phase permanent magnet synchronous generator PMSG model prediction
CN113922720A (en) PMSM model prediction current control algorithm based on duty ratio control
CN111541409A (en) SVPWM control method for single-phase open-circuit fault of five-phase permanent magnet synchronous motor based on modulation function
CN113162500A (en) Inverter multi-mode modulation switching method based on flux linkage deviation vector
Dong et al. Hybrid PWM and smooth switching technology for traction PMSM drives in low switching frequency

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant