CN112398399B - Active suppression method for vibration noise of permanent magnet synchronous motor - Google Patents

Active suppression method for vibration noise of permanent magnet synchronous motor Download PDF

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CN112398399B
CN112398399B CN202011426168.0A CN202011426168A CN112398399B CN 112398399 B CN112398399 B CN 112398399B CN 202011426168 A CN202011426168 A CN 202011426168A CN 112398399 B CN112398399 B CN 112398399B
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CN112398399A (en
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杜会卿
杨森
蔡宗举
刘玉明
蔡道萌
张智杰
苗春晖
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CSSC Systems Engineering Research Institute
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/06—Rotor flux based control involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
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    • Y02T10/64—Electric machine technologies in electromobility

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Abstract

The invention discloses a method for actively suppressing vibration noise of a permanent magnet synchronous motor, which is based on a DSP + FPGA embedded control system architecture, extracts angle jump variables between adjacent sampling points on the basis of finishing sampling processing and closed-loop control by interrupting a DSP timer and obtaining modulation degree and angle real-time instructions, calculates angle step values by adopting a virtual high-frequency method, sends modulation instructions to FPGA through a high-speed data bus, equivalently linearly steps the amplitude of modulation waves in a control period to the instruction values by utilizing the advantage of high running speed of the FPGA, ensures the reliable zero clearing of a counter and a stepper cache by utilizing clock synchronization, further generates smooth modulation waves, compares the smooth modulation waves with triangular carriers, outputs PWM pulses, drives an inverter to convert a direct current bus voltage into a three-phase alternating current voltage to drive a motor to run, thereby realizing the multi-sampling fine control of key variables, the variable updating frequency and the control precision are obviously improved, the variable jumping is avoided, and the torque pulsation and the electromagnetic vibration noise of the motor are effectively improved.

Description

Active suppression method for vibration noise of permanent magnet synchronous motor
Technical Field
The invention belongs to the technical field of alternating current motor vibration control, and particularly relates to a high-frequency vibration noise suppression method for a digital controller of a ship permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the advantages of high power density, simple mechanical structure, high efficiency of a full speed regulation range, high torque output ratio and the like, and is widely applied to important equipment such as energy power, transportation, high-end manufacturing and the like. With the wide application of permanent magnet motors, the vibration noise characteristics of the permanent magnet motors are attracting more and more attention. The high-end application fields such as ship propulsion, high-precision servo and electric automobiles have very urgent demands on low-vibration noise motor systems.
The vibration noise source of the permanent magnet synchronous motor mainly comprises three parts: electromagnetic vibration sources, mechanical vibration sources, and fluid (aerodynamic) vibration sources. For a high-power medium-low speed permanent magnet synchronous motor for ship propulsion, an electromagnetic vibration source is the most main vibration noise source of the motor and is caused by radial electromagnetic force generated by the interaction of magnetic fields of a stator and a rotor of the motor. However, at present, the research on the electromagnetic vibration noise of the permanent magnet synchronous motor is generally focused on the analysis and the suppression of a low-frequency electromagnetic vibration source at home and abroad, the research on the structural vibration response of the motor under electromagnetic excitation is relatively less, and the high-frequency vibration noise is still lack of comprehensive and systematic research. The high-frequency electromagnetic vibration source is mainly caused by a switching frequency band generated by Pulse Width Modulation (PWM) power supply of a frequency converter and frequency multiplication band harmonics thereof, and mainly research is also focused on a PWM strategy level. On the other hand, the main reason for generating the high-frequency electromagnetic noise is that when the motor system performs digital discrete closed-loop control, the electromagnetic torque pulsation of the motor is generated because the magnetic field of the motor is not well controlled.
Patent document 1(CN108258947A) discloses a method for suppressing high frequency noise by an electric vehicle motor controller, which employs motor controller software to set the switching frequency of an IGBT drive module according to a received torque command and a collected motor speed signal. The whole motor control is divided into three ranges according to the rotating speed, the IGBT switching frequency is fixed at the rotating speed of 500rpm at a low rotating speed, whether the PWM frequency is fixed or random frequency conversion is carried out at the rotating speed of 500-6000 rpm according to the torque, and whether the PWM frequency is fixed or random frequency conversion is carried out according to the torque at the rotating speed of 6000rpm or above, so that the pertinence is not strong, the IGBT switching frequency mode is only random frequency conversion, and the problem of noise bulge is difficult to effectively solve.
Patent document 2(CN105429540B) discloses a vibration suppression method for an ac servo motor based on model tracking control, which adopts the concept of model tracking control algorithm (MFC), inputs the control quantity of the system into an ideal control model, considers that the output of the ideal model is equal to the actual output, when the external disturbance is applied to the actual system, the ideal model responds to the disturbance, adds the calculated control quantity for canceling the disturbance into the control quantity of the actual system, and plays the purpose of stabilizing the system and eliminating the disturbance.
Patent document 3(CN105577059B) discloses a noise suppression method for a marine asynchronous motor system, which includes calculating electromagnetic torque ripples of a three-phase motor under different loads and different rotating speeds offline, drawing q curves, calculating theoretical values of the electromagnetic torque ripples under the current rotating speed and the load when the motor is in online closed-loop control, comparing the theoretical values with actual values of the electromagnetic torque ripples of the current motor, and comparing the theoretical values with actual values of the electromagnetic torque ripples of the current motor to output a voltage compensation value through a PI regulator.
The method is characterized in that an embedded digital controller is used for discretely controlling angles, a DSP, an MCU or an FPGA is commonly used in the field of variable frequency motor driving, a TIC2000 series DSP is taken as an example, as links such as sampling, PID control and protection need to be processed, the main frequency and the switching frequency of the controller need to meet a certain proportion, and a timer interruption needs a certain execution time, and as the switching frequency is increased, the purpose that a control cycle can be completed by a plurality of switching cycles under high frequency is achieved basically, and the high-bandwidth controller is difficult to design due to digital delay caused by multi-cycle adjustment.
Disclosure of Invention
The invention aims to provide a method for suppressing high-frequency vibration noise at the control frequency of a ship permanent magnet synchronous motor on the basis of the existing DSP + FPGA digital controller architecture, and the method adopts a virtual high-frequency method to realize the multi-sampling fine control of key variables, breaks through the inherent defect of multi-period regulation, improves the data updating frequency, avoids variable jump, and improves the torque pulsation and electromagnetic vibration noise of the motor. The technical scheme for solving the technical problems is as follows:
a method for actively suppressing vibration noise of a permanent magnet synchronous motor is used for controlling a three-phase permanent magnet synchronous motor system, wherein the three-phase permanent magnet synchronous motor system comprises a three-phase full-bridge inverter, a filtering loop, the permanent magnet synchronous motor, a sampling loop and a control system, and the method comprises the following steps:
the sampling loop samples direct-current voltage input into the three-phase full-bridge inverter, samples three-phase current output by the three-phase full-bridge inverter, and samples sine and cosine signals output by a rotary transformer of the permanent magnet synchronous motor;
the control system processes and controls a closed loop of direct-current voltage, three-phase current and sine and cosine signals sampled by the sampling loop to obtain a modulation degree, an angle and an angle stepping value, generates a three-phase modulation wave according to the modulation degree, the angle and the angle stepping value, generates a three-phase PWM waveform after comparing the three-phase modulation wave with a triangular carrier wave, and controls the three-phase full-bridge inverter to convert direct-current bus voltage into three-phase alternating-current voltage to drive a motor to operate;
the modulation degree m and the angle are specifically calculated according to the following formula:
Figure GDA0003578214710000041
Figure GDA0003578214710000042
wherein u isodControlling the output voltage u for the d-axisoqControlling output voltage, V, for q-axisdcIs a sampled direct current voltage;
the method comprises the following steps of extracting angle jump variables between adjacent sampling points of a DSP (digital signal processor) by adopting a virtual high-frequency method, equally dividing the angle jump variables to obtain angle stepping values, and specifically comprising the following steps:
record last sampling time t0Angle value angle 0]And this time t1Angle value angle [1 ]]And calculating to obtain an angle jump value angle _ jv ═ angle [1 ═ angle]-angle[0]；
Equally dividing the angle jump value angle _ jv into angle step values angle _ dlt ═ angle _ jv/k by adopting a virtual high-frequency method; after the angle jump value is obtained through calculation, the method further comprises the following steps:
judging whether the absolute value of the angle jump value is larger than a set value or not;
if yes, correcting the angle jump value according to the following formula:
angle_jv’＝angle_jv+360×p×dm
wherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen the value is-1, the motor is reversely rotated, and p is an angle magnification factor.
On the basis of the technical scheme, the invention can be further improved as follows.
Further, the control system adopts a DSP + FPGA architecture to process and perform closed-loop control on the dc voltage, the three-phase current, and the sine and cosine signals sampled by the sampling circuit, so as to obtain a modulation degree, an angle, and an angle step value, and specifically includes:
the direct-current voltage, the three-phase current and the sine and cosine signals sampled by the sampling loop are processed by a three-loop control loop, namely a speed loop, a voltage loop and a current loop, so as to obtain d-axis and q-axis control output voltages, and the modulation degree m and the angle are calculated according to the d-axis and q-axis control output voltages;
and extracting an angle jump variable angle _ jv between adjacent sampling points of the DSP by adopting a virtual high-frequency method, and dividing the angle jump variable angle _ jv into k equal parts to obtain an angle step value angle _ dlt.
Further, the d-axis and q-axis control output voltages are obtained through a three-loop control loop, namely a speed loop, a voltage loop and a current loop, and the method specifically comprises the following steps:
decoding the sampled sine and cosine signals to obtain modulated position signals Pos, wherein the position signals Pos are digital signals from 0 to 65535;
multiplying the position signal Pos by a proportionality coefficient P2 pi/65536 to obtain an electric angle theta of the motoreSaid electrical angle θeZero offset angle theta between electrical zero position of superposition rotary transformer and electrical zero position of motor rotor0To obtain the true rotor angle position theta of the motorc；
For the electrical angle thetaeDifferential processing is carried out to obtain a feedback rotating speed signal omegaeFor the feedback speed signal omegaeFiltering to obtain feedback filtering rotation speed signal omegae-FLT；
Filtering the feedback filtering rotation speed signal omegae-FLTConverted into motor speed NmotorThe motor speed N is setmotorWith a given speed NrefComparing the two currents, and adjusting the difference by a speed loop PI to obtain a given current is *；
d-axis control output voltage uodAnd q-axis control of output voltage uoqThe sum of the squares of both, i.e., √ (u) as the rootod 2+uoq 2) The maximum peak value of the phase voltage allowed to be output by the system, namely the DC voltage VdcV, comparing the two results, and adjusting the difference by a voltage loop PI to obtain a voltage loop control output current idv *Wherein V isdcIs a sampled direct current voltage;
id0 *control and superposition of the voltage ring to control the output current idv *To obtain d-axis given current id *；
Given current is *D-axis set current id *The square difference of the two is open root√(is *2-id *2) To obtain a q-axis given current iq *；
Three-phase current i to be sampled by Clark conversionu、iv、iwConverting the three-phase stationary coordinate system abc coordinate system into a two-phase stationary coordinate system alpha beta coordinate system to obtain an alpha-axis current component iαBeta axis current component iβ；
Converting alpha-axis and beta-axis current components i through Park transformationα、iβTransforming the current to a two-phase rotating coordinate system dq coordinate system to obtain d-axis feedback current idAnd q-axis feedback current iq；
d-axis feedback current idD-axis set current id *Make a comparison of idAnd id *The error of the two is regulated by a d-axis current loop PI to obtain a d-axis given voltage ud *；
q-axis feedback current iqWith q-axis given current iq *Make a comparison of iqAnd iq *The error of the two is regulated by a q-axis current loop PI to obtain a q-axis given voltage uq *；
Given voltage u on d-axisd *On the basis of the sum of feedforward decoupling component-omegaeLqiqTo obtain d-axis control output voltage uodGiving a voltage u on the q-axisq *On the basis of the sum of the feedforward decoupling component omegae(ψf+Ldid) To obtain a q-axis control output voltage uoqWherein L isd、LqInductance of direct and quadrature axes, psi, of the motorfIs a motor rotor flux linkage.
Further, after obtaining the modulated position signal Pos, the method further includes:
judging whether the sampling point of the position signal Pos is a jumping point, if so, correcting the position signal Pos, and obtaining a corrected position signal Pos ═ Pos +65536 × dmWherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen-1, the motor is reversed.
Further, the feedback filtering rotation speed signal omegae-FLTThe calculation formula of (2) is as follows:
ωe-FLT＝ωe-FLT×LPF_ka+ωe×LPF_kb
LPF_ka＝1/(1+2×π×Tc×fLPF)
LPF_kb＝1-LPF_ka
wherein, TcTo control the period, fLPFThe low pass filter cut-off frequency.
Further, generating a three-phase modulation wave according to the modulation degree, the angle and the angle step value specifically includes:
the FPGA sequentially superposes the angle stepping values angle _ dlt to the angle according to an internal timer to serve as new angle given values Aa;
the clock synchronization clears the timer count value and the angle step value when receiving the next angle;
and according to the angle given value Aa, combining the modulation degree m to obtain a three-phase modulation wave.
The invention has the beneficial effects that:
1) the method optimizes the vibration noise of the permanent magnet synchronous motor, does not need to additionally increase an external circuit, does not need to modify the existing controller framework, is convenient to realize and saves the cost;
2) according to the invention, the virtual high-frequency method is adopted to realize the multi-sampling fine control of key variables, the inherent defect of multi-period regulation of a discrete digital controller is overcome, the data updating frequency is improved, and the torque pulsation and the electromagnetic vibration noise of the motor are effectively improved;
3) the invention can effectively improve the quality of the PWM modulation wave, improve the control precision and avoid the system oscillation caused by variable jump under the high-speed working condition.
Drawings
Fig. 1 is an overall architecture diagram of an application object permanent magnet synchronous motor system of the present invention;
FIG. 2 is a block diagram of a closed loop control of the active vibration noise suppression method of the present invention;
FIG. 3a is a schematic diagram of a virtual high-frequency multi-sampling refinement control scheme according to the present invention;
FIG. 3b is a schematic diagram of a virtual high-frequency multi-sampling refinement control according to the present invention;
fig. 4 is a diagram of an exemplary embedded control system architecture to which the present invention relates.
Detailed Description
The principles and features of this invention are described below in conjunction with the following drawings, which are set forth by way of illustration only and are not intended to limit the scope of the invention.
As shown in fig. 1, the invention is a typical architecture diagram of a three-phase permanent magnet synchronous motor system of an application object ship, and the whole system comprises a three-phase full-bridge inverter 1, an LC filter circuit 2, a three-phase permanent magnet synchronous motor 3, a sampling circuit 4 and a control system 5, wherein the three-phase permanent magnet synchronous motor 3 is connected with the filter circuit 2 formed by LC through three power lines, and the LC filter circuit 2 is connected with the output end of the three-phase full-bridge inverter 1; a bus of the three-phase full-bridge inverter 1 is connected with a direct-current supporting capacitor C1 in series; the sampling loop 4 mainly performs direct-current voltage sampling, three-phase current sampling and resolver sine and cosine signal sampling; the sampling signals are all connected into the control system 5, the output end of the sampling signals is connected with the three-phase full-bridge inverter 1 after a series of processing, control and modulation, pulse signals are output to the three-phase full-bridge inverter 1, power devices from Q1 to Q3 are driven to perform regular switching actions, and the control system 5 adopts a DSP + FPGA framework.
As shown in fig. 2, it is an overall closed-loop control block diagram of the active suppression method of vibration noise of the present invention, and specifically includes the following steps:
1) decoding a sine and cosine signal of a PMSM rotary transformer by adopting a 16-bit rotary variable digital converter AD2S1210, multiplying a modulated position signal Pos which is a digital signal from 0 to 65535 by a proportionality coefficient P2 pi/65536 to obtain an electric angle theta of the motoreAnd superimposing the null angle theta between the electrical null position of the resolver and the electrical null of the rotor of the electric machine0To obtain the true rotor angle position theta of the motorc；
2) To electrical angle thetaeDifferential processing is carried out to obtain a feedback rotating speed signal omegaeSince the position signal returns to zero when reaching 65536, and the differentiation causes a false jump of the speed signal, these jump points are processed before differentiation, and the method for judging the jump point of the speed signal is as follows: and when the absolute value of the difference between two adjacent sampling values is larger than a fixed value, the latter sampling point is a jumping point. After the jumping point is detected, the position signal needs to be corrected, and the correction formula is Pos ═ Pos +65536 × dmWherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen the value is-1, the motor is reversely rotated;
3) the differential signal contains real rotating speed and harmonic interference, and a digital low-pass filter LPF shown in the following formula is designed for filtering to obtain a smooth rotating speed signal omegae-FLTWherein, TcTo control the period, fLPFCut-off frequency of the low-pass filter;
ωe-FLT＝ωe-FLT×LPF_ka+ωe×LPF_kb (1)
LPF_ka＝1/(1+2×π×Tc×fLPF) (2)
LPF_kb＝1-LPF_ka (3)
4) feedback filtering speed signal omegae-FLTMultiplying by a scaling factor of 30/pi/pn(pnNumber of pole pairs of the motor, in this embodiment 4 pole pairs of the motor), converted into the motor speed NmotorAfter that, with a given rotation speed NrefComparing the two currents, and adjusting the difference by a speed loop PI to obtain a given current is *；
5) Three-phase current i through Clark conversionu、iv、iwTransforming the three-phase stationary coordinate system (abc coordinate system) to a two-phase stationary coordinate system (alpha beta coordinate system) to obtain an alpha-axis current component iαBeta axis current component iβ；
6) Converting alpha-axis and beta-axis current components i through Park transformationα、iβTransforming to a two-phase rotating coordinate system (dq coordinate system) to obtain d-axis feedback currentidAnd q-axis feedback current iq；
7) d-axis feedback current idD-axis set current idComparing, wherein id *Is idThe control output of the superposed weak magnetic and electric pressure ring is controlled as 0, specifically the control output u of the d-axis current ringodAnd q-axis current loop control output uoqSquare sum of both and root-opening number √ (u)od 2+uoq 2) And then the maximum phase voltage peak value which is the DC voltage V allowed to be output by the systemdcv/V3 are compared, the difference between the two is regulated by a weak magnetic ring PI to obtain a given d-axis current id *，idAnd id *The error of the two is regulated by a d-axis current loop PI to obtain a d-axis given voltage ud *；
8) q-axis feedback current iqWith q-axis given current iq *A comparison is made wherein iq *For a given current is *D-axis set current id *The squared difference of the two is the root of √ (i)s *2-id *2)，iqAnd iq *The error of the two is regulated by a q-axis current loop PI to obtain a q-axis given voltage uq *；
9) In order to realize the decoupling control of the d-axis current and the q-axis current, voltages u are respectively given to the d-axis current and the q-axis currentd *And uq *On the basis of (a) add a feedforward decoupling component, respectively-omegaeLqiqAnd ωe(ψf+Ldid) Wherein L isd、LqFor direct and quadrature axis inductances, psi, of the machinefFor the motor rotor flux linkage, and further obtain d-axis current loop control output uodAnd q-axis current loop control output uoq；
10) U based on control loop outputodAnd uoqCalculating to obtain input signals of a Pulse Width Modulation (PWM) module, namely modulation degree m and angle, wherein the specific calculation formula is as follows:
Figure GDA0003578214710000091
Figure GDA0003578214710000092
11) the regulation system m is basically maintained constant under the steady-state operation condition without great fluctuation, but a certain execution period is needed for the interruption of the timer, and the interruption frequency fcUsually several kHz, results in the jump of the angle between two adjacent sampling points under the high-speed operation condition of the motor, such as the control frequency f in the embodimentcAt 2.0kHz, when the motor rotates at n_motorAt 1000rpm, corresponding to the rotation frequency fm＝n_motor×pn200/3, the angular jump Δ ang between two adjacent samples is 360/(f)c/fm) 12 degrees, the jump amplitude is large;
12) by recording the last sampling instant t0Angle value, and then the current time t1Comparing the angle values, calculating the difference, and obtaining the angle jump value angle _ jv ═ angle [1 [ ]]-angle[0]In order to avoid the influence of false jump, when the absolute value of the difference between two adjacent sampling values is greater than a fixed value, the angular jump value needs to be corrected, and the correction formula is angle _ jv ═ angle _ jv +360 × p × dmWherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen the value is equal to-1, the motor is represented to rotate reversely, p is an angle amplification factor, and then a virtual high-frequency method is adopted to divide the jump value into a plurality of step values of angle _ dlt equal to angle _ jv/k, for example, when the value is equal to 10 and the rotating speed is 1000rpm, the theoretically calculated angle _ dlt is 1.2 degrees, so that a foundation is laid for the multiple sampling fine control of the angle;
13) as shown in fig. 4, the DSP sends the modulation m, the angle and the angle step _ dlt output by the closed-loop control to the FPGA through the EMIF bus;
14) as shown in fig. 3a, if the FPGA receives the angle real-time value and then directly performs table lookup to generate the modulation wave using the conventional single-cycle updating mode, then the FPGA will perform the table lookupThe pulse waveform is not changed in a plurality of switching periods and modulated wave amplitude jump exists at the updating moment, the advantages of switching frequency improvement and high FPGA operation speed cannot be fully utilized, but the invention adopts a virtual high-frequency multi-sampling updating mode as shown in figure 3b and adopts an internal timer _1 (the count value cnt is f)p/fcK) sequentially superposing angle _ dlt to angle as a new angle given value Aa, so that the amplitude of the modulation wave in a control period is stepped to the given value k times, the control precision is improved, and the jump amplitude of the modulation wave is reduced;
15) as shown in FIG. 4, the clock synchronization ensures that the timer count value and the angle step value are cleared when the next angle real-time value is received, such as f in this embodimentp＝20MHz，fcWhen k is equal to 10, the count value cnt is equal to 1000, Aa is 30 ° at a certain moment, the received angle is 31 °, the angle _ dlt is 1 °, the current counter is cleared, Aa starts to increment the angle, after 10 counter cycles, Aa is increased from 31 ° to 41 ° by a step value of 1 °, at this time, the next control cycle receiving angle is 42 °, and the next operation cycle is started;
16) the FPGA cannot perform floating point operation and adjacent angle jump values in the dynamic adjustment process have deviation, so that the deviation exists between the last angle value in the control period and the real-time value of the next period, but the precision of the angle is improved by k times before the precision of the angle in multi-sampling is improved, and the execution frequency of the FPGA program is far higher than the DSP control interrupt frequency, and the k value can be set to be more than 100, so that the deviation can be limited within 1 degree;
17) as shown in fig. 4, based on the table lookup method, a smooth three-phase modulation wave is obtained by combining the modulation degree m with the angle given value Aa obtained by the virtual high-frequency operation, and further the frequency fsThe triangular carriers are compared, corresponding three-phase PWM waveforms are generated after dead zones are superposed, and the inverter is controlled to convert the direct-current bus voltage into three-phase alternating-current voltage to drive the motor to operate.
The invention carries out targeted analysis and inhibition aiming at the problem of high-frequency vibration noise of the existing controller, adopts a virtual high-frequency method, carries out sampling and closed-loop control in a DSP based on a DSP + FPGA control architecture, calculates to obtain a real-time value and a step value of a key variable, and sends the real-time value and the step value to an FPGA Pulse Width Modulation (PWM) module through EMIF for high-frequency and linear processing, thereby fully utilizing the advantage of high running speed of the FPGA, realizing multi-sampling fine digital control, remarkably improving data updating frequency, avoiding variable jump and further effectively inhibiting vibration noise at the control frequency.
The above description is only for the purpose of illustrating the preferred embodiments of the present invention and is not to be construed as limiting the invention, and any modifications, equivalents, improvements and the like that fall within the spirit and principle of the present invention are intended to be included therein.

Claims (6)

1. The active suppression method for the vibration noise of the permanent magnet synchronous motor is used for controlling a three-phase permanent magnet synchronous motor system, wherein the three-phase permanent magnet synchronous motor system comprises a three-phase full-bridge inverter, a filtering loop, the permanent magnet synchronous motor, a sampling loop and a control system, and is characterized by comprising the following steps of:
the sampling loop samples direct-current voltage input into the three-phase full-bridge inverter, samples three-phase current output by the three-phase full-bridge inverter, and samples sine and cosine signals output by a rotary transformer of the permanent magnet synchronous motor;
the control system processes and controls closed loops of direct-current voltage, three-phase current and sine and cosine signals sampled by the sampling circuit to obtain a modulation degree, an angle and an angle stepping value, generates a three-phase modulation wave according to the modulation degree, the angle and the angle stepping value, generates a three-phase PWM waveform after comparing with a triangular carrier wave, and controls the three-phase full-bridge inverter to convert direct-current bus voltage into three-phase alternating-current voltage to drive a motor to operate;
the modulation degree m and the angle are specifically calculated according to the following formula:
Figure FDA0003578214700000011
Figure FDA0003578214700000012
wherein u isodControlling the output voltage u for the d-axisoqControlling the output voltage, V, for the q-axisdcIs a sampled direct voltage;
the method comprises the following steps of extracting angle jump variables between adjacent sampling points of a DSP (digital signal processor) by adopting a virtual high-frequency method, equally dividing the angle jump variables to obtain angle stepping values, and specifically comprising the following steps:
record last sampling time t0Angle value angle 0]And this time t1Angle value angle [1 ]]And calculating to obtain an angle jump value angle _ jv ═ angle [1 ═ angle]-angle[0]；
Equally dividing the angle jump value angle _ jv into angle step values angle _ dlt ═ angle _ jv/k by adopting a virtual high-frequency method; after the angle jump value is obtained through calculation, the method further comprises the following steps:
judging whether the absolute value of the angle jump value is larger than a set value or not;
if yes, correcting the angle jump value according to the following formula:
angle_jv’＝angle_jv+360×p×dm
wherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen the value is-1, the motor is reversely rotated, and p is an angle magnification factor.
2. The method according to claim 1, wherein the control system adopts a DSP + FPGA architecture, processes and performs closed-loop control on the dc voltage, the three-phase current, and the sine and cosine signals sampled by the sampling loop to obtain a modulation degree, an angle, and an angle step value, and specifically includes:
the direct-current voltage, the three-phase current and the sine and cosine signals sampled by the sampling loop are processed by a three-loop control loop, namely a speed loop, a voltage loop and a current loop, so as to obtain d-axis and q-axis control output voltages, and the modulation degree m and the angle are calculated according to the d-axis and q-axis control output voltages;
and extracting an angle jump variable angle _ jv between adjacent sampling points of the DSP by adopting a virtual high-frequency method, and dividing the angle jump variable angle _ jv into k equal parts to obtain an angle step value angle _ dlt.
3. The method of claim 2, wherein the d-and q-axis control output voltages are obtained through a three-loop control loop, namely a speed loop, a voltage loop and a current loop, and specifically comprises:
decoding the sampled sine and cosine signals to obtain modulated position signals Pos, wherein the position signals Pos are digital signals from 0 to 65535;
multiplying the position signal Pos by a proportionality coefficient P2 pi/65536 to obtain an electric angle theta of the motoreThe electrical angle θeZero offset angle theta between electrical zero position of superposition rotary transformer and electrical zero position of motor rotor0To obtain the true rotor angle position theta of the motorc；
For the electrical angle thetaeDifferential processing is carried out to obtain a feedback rotating speed signal omegaeFor the feedback speed signal omegaeFiltering to obtain feedback filtering rotation speed signal omegae-FLT；
Filtering the feedback filtering rotation speed signal omegae-FLTConverted into motor speed NmotorThe motor speed N is setmotorWith a given speed NrefComparing the two currents, and adjusting the difference by a speed loop PI to obtain a given current is *；
d-axis control output voltage uodAnd q-axis control of output voltage uoqSquare sum of both and root-opening number √ (u)od 2+uoq 2) The maximum peak value of the phase voltage allowed to be output by the system, namely the DC voltage VdcV, comparing the two results, and adjusting the difference by a voltage loop PI to obtain a voltage loop control output current idv *Wherein V isdcIs a sampled direct current voltage;
id0 *controlling and superposing the voltage ring to control the output current i as 0dv *To obtain d-axis given current id *；
Given current is *D-axis set current id *The squared difference of the two is the root of √ (i)s *2-id *2) To obtain a q-axis given current iq *；
Three-phase current i to be sampled by Clark conversionu、iv、iwConverting the three-phase stationary coordinate system abc coordinate system into a two-phase stationary coordinate system alpha beta coordinate system to obtain an alpha-axis current component iαBeta axis current component iβ；
Converting alpha-axis and beta-axis current components i through Park transformationα、iβTransforming the current to a two-phase rotating coordinate system dq coordinate system to obtain d-axis feedback current idAnd q-axis feedback current iq；
d-axis feedback current idD-axis set current id *Make a comparison of idAnd id *The error of the two is regulated by a d-axis current loop PI to obtain a d-axis given voltage ud *；
q-axis feedback current iqWith q-axis given current iq *Make a comparison of iqAnd iq *The error of the two is regulated by a q-axis current loop PI to obtain a q-axis given voltage uq *；
Given voltage u on d-axisd *On the basis of the sum of feedforward decoupling component-omegaeLqiqTo obtain d-axis control output voltage uodGiving a voltage u on the q-axisq *On the basis of the sum of the feedforward decoupling component omegae(ψf+Ldid) To obtain a q-axis control output voltage uoqWherein L isd、LqInductance of direct and quadrature axes, psi, of the motorfIs a motor rotor flux linkage.
4. The method of claim 3, further comprising, after obtaining the modulated position signal Pos:
judging whether the sampling point of the position signal Pos is a jumping point, if so, correcting the position signal Pos, and obtaining a corrected position signal Pos ═ Pos +65536 × dmWherein d ismRepresenting motor steering when dmWhen 1, the motor rotates in the forward direction, and when dmWhen-1, the motor is reversed.
5. A method according to claim 3, wherein the feedback filtered speed signal ωe-FLTThe calculation formula of (2) is as follows:
ωe-FLT＝ωe-FLT×LPF_ka+ωe×LPF_kb
LPF_ka＝1/(1+2×π×Tc×fLPF)
LPF_kb＝1-LPF_ka
wherein, TcTo control the period, fLPFThe low pass filter cut-off frequency.
6. The method according to any one of claims 2 to 5, wherein generating a three-phase modulated wave from the modulation degree, the angle and the angle step value specifically comprises:
the FPGA sequentially superposes the angle stepping values angle _ dlt to the angle according to an internal timer to serve as new angle given values Aa;
the clock synchronization clears the timer count value and the angle step value when receiving the next angle;
and according to the angle given value Aa, combining the modulation degree m to obtain a three-phase modulation wave.
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