CN112117915B - Method for suppressing capacitor voltage fluctuation of series H-bridge type frequency converter - Google Patents

Method for suppressing capacitor voltage fluctuation of series H-bridge type frequency converter Download PDF

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CN112117915B
CN112117915B CN202010762815.9A CN202010762815A CN112117915B CN 112117915 B CN112117915 B CN 112117915B CN 202010762815 A CN202010762815 A CN 202010762815A CN 112117915 B CN112117915 B CN 112117915B
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CN112117915A (en
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刘进军
赵世锋
杜思行
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Xi'an Singularity Energy Co ltd
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Xian Jiaotong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
    • H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
    • H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M5/4585—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only having a rectifier with controlled elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/14—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation with three or more levels of voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0038—Circuits or arrangements for suppressing, e.g. by masking incorrect turn-on or turn-off signals, e.g. due to current spikes in current mode control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/01—Asynchronous machines

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention discloses a method for suppressing the voltage fluctuation of a capacitor of a series H-bridge type frequency converter, which comprises the following steps: calculating an input side current instruction of the rectifier, and controlling an input side current value according to the input side current instruction so that input power is equal to power generated by inversion side fundamental voltage and fundamental current; wherein the input power of the rectifier is equal to the voltage source input power minus the total instantaneous power across the three inductors. The input power of the sub-module defined by the invention is not pure voltage source input power any more, and the total power of the three inductors is considered, and the difference between the two is the total input power. The power generated by the inversion side fundamental wave voltage and the fundamental wave current is equivalent to the power generated by the inversion side fundamental wave voltage and the fundamental wave current, and the minimization of the capacitor voltage ripple can be realized. The voltage fluctuation suppression strategy of the invention does not need additional elements and sensors, and has good fluctuation suppression effect under different working conditions of the medium-voltage high-power asynchronous motor only by algorithm optimization.

Description

Method for suppressing capacitor voltage fluctuation of series H-bridge type frequency converter
Technical Field
The invention belongs to the technical field of frequency converters, and particularly relates to a method for suppressing voltage fluctuation of a capacitor of a series H-bridge frequency converter.
Background
The series H-bridge multi-level Converter (CHB) has the advantages of high output voltage sine degree, multiple output levels, modularization, easiness in maintenance and the like, so that the series H-bridge multi-level converter is widely applied to occasions where medium-voltage high-power motors drag. The whole system consists of a multi-winding step-down transformer, three phases of 3n sub-modules (n per phase) and a three-phase asynchronous motor. The submodule comprises a three-phase PWM rectifier as an input end, a capacitor and an H-bridge inverter as an output end to complete AC-DC-AC conversion. The output ends of the sub-modules of each phase are connected in series and are superposed to generate single-phase total output voltage. Because the instantaneous power of the input end and the output end of the submodule is unequal, the power difference is absorbed by the capacitor, voltage fluctuation on the capacitor is caused, and the fluctuation is increased along with the reduction of the rotating speed, so that the system stability is threatened.
By using a large capacitor, voltage fluctuation due to power fluctuation is reduced, which has problems in that the cost and volume of the equipment are increased, and the electrolytic capacitor having a large capacitance value has a short life. With the additional energy storage element and the switching device, power fluctuation on the capacitor is transferred to the energy storage element, with the problems that additional cost is added and the control becomes complicated. The power feedforward is provided by researching that the power of an output end of an inversion side is fed forward to a rectification side, and the essence is to inject currents with two frequencies into the rectification side, so that the input power of a rectifier voltage source is basically equal to the output power of the inversion side, the power difference between the input end and the output end is reduced, and the power fluctuation and the voltage fluctuation on a capacitor are reduced accordingly. The disadvantages are as follows: new power fluctuations are introduced at the input side, resulting in large power and voltage fluctuations still existing on the capacitor.
If an effective voltage ripple suppression method is adopted, the conventional electrolytic capacitor can be replaced by the thin-film capacitor with smaller capacitance value and volume and longer service life under the condition of meeting the same voltage fluctuation allowable index, so that the cost is greatly saved, the equipment volume is reduced, and the reliability is improved.
Disclosure of Invention
Aiming at the problem of large capacitor voltage ripple in a CHB motor dragging system, the invention provides a method for suppressing capacitor voltage fluctuation of a series H-bridge type frequency converter, which is characterized in that no additional component is added, new injection current is designed on the input side again, the power (including three inductance instantaneous powers) on the rectification side is equal to the output power, no new power fluctuation is introduced, and a simplified generation link of the injection current is further designed, so that the design can be greatly simplified, and the purpose of reducing the capacitor in a submodule is achieved.
In order to achieve the purpose, the invention adopts the following technical scheme:
a method for suppressing voltage fluctuation of a capacitor of a series H-bridge type frequency converter comprises the following steps:
calculating an input side current instruction of the rectifier, and controlling an input side current value according to the input side current instruction so that input power is equal to power generated by inversion side fundamental voltage and fundamental current;
wherein the input power of the rectifier is equal to the voltage source input power minus the instantaneous power across the three inductors.
As a further improvement of the invention, the series H-bridge type frequency converter comprises a multi-winding step-down transformer, three phases of 3n sub-modules and a three-phase asynchronous motor;
the input side of the sub-module is connected to the secondary side of the multi-winding transformer, and the output sides of the n sub-modules of each phase are connected in series; the three-phase H-bridge type frequency converter adopts a star connection method, one end of the three-phase H-bridge type frequency converter is connected into a middle point, and the other end of the three-phase H-bridge type frequency converter is connected with a three-phase asynchronous motor; each submodule sequentially comprises a front-end controllable rectifier bridge, a middle capacitor and a rear-end H-bridge inverter circuit.
As a further improvement of the invention, the method specifically comprises the following control processes:
d, q axis current instruction value for feeding forward power
Figure BDA0002613554610000021
And
Figure BDA0002613554610000022
carrying out dq/abc conversion to obtain three-phase input current under an abc coordinate system
Iin＝[ia ib ic]T＝I1+I2+I3 (1)
Wherein, I1 I2Is a positive sequence current, I3Is a set of negative-sequence currents; u shapeoFor output of voltage fundamental amplitude, I, of the submoduleoFor outputting the current amplitude, omegasFor side angular frequency, omega, of the gridoOutputting the side angular frequency for the submodule;
Figure BDA0002613554610000023
for the power angle of the motor
Figure BDA0002613554610000024
Figure BDA0002613554610000025
Figure BDA0002613554610000026
Calculating to obtain the input power of the voltage source
Figure BDA0002613554610000027
Namely the input power of the voltage source compensates the output power;
the power applied to the capacitor is the input power p of the voltage sourcesSubtracting the instantaneous power p across the three inductorsLMinus the output power po，pLThe calculation formula is as follows:
Figure BDA0002613554610000031
l is an inductor; instantaneous total power p of three inductorsLThe calculation result is as follows:
Figure BDA0002613554610000032
the input power is the power generated by the power supply to reduce the instantaneous power of the inductor, i.e.
pin＝ps-pL (8)
As a further improvement of the present invention, the input side current command acquisition method is as follows:
let the input current comprise two positive sequences and one negative sequence, namely:
Figure BDA0002613554610000033
Figure BDA0002613554610000034
Figure BDA0002613554610000035
the amplitude, the phase, the frequency and the like are all to-be-solved quantities; further obtaining an input power of
Figure BDA0002613554610000036
pinContains a total of 6 items, each of which is denoted as pin1，pin2..inAnd poTo obtain the input current:
Figure BDA0002613554610000041
substituting the above parameters into (12), pin5Is 4 frequency doubling component. Due to poDoes not contain the frequency component, therefore, let I2Is 0, the input power is obtained
Figure BDA0002613554610000042
Wherein,
Figure BDA0002613554610000043
for the sub-module, the output voltage and the current fundamental wave are respectively
uo＝Uosin(ωot) (16)
Figure BDA0002613554610000044
Wherein
Figure BDA0002613554610000045
Is the voltage and current phase difference of the submodules, which is equal to the power angle and the output power of the motor
Figure BDA0002613554610000046
In order to make the input power exactly equal to the output power, there are:
Figure BDA0002613554610000047
simplification of
Figure BDA0002613554610000048
To obtain
Figure BDA0002613554610000049
Wherein, UmidAnd α is an intermediate variable which satisfies:
Figure BDA00026135546100000410
Figure BDA0002613554610000051
comparing formula (18) with formula (20) to obtain
Figure BDA0002613554610000052
Figure BDA0002613554610000053
Therefore, the amplitude and the phase of each component of the input current command value are determined, and the amplitude and the phase information contained in the input current command value are the formulas (13), (23) and (24);
because the control is carried out under the dq coordinate system, the input current instruction value under the abc coordinate system is subjected to abc/dq conversion to obtain the current instruction value under the dq coordinate system
Figure BDA0002613554610000054
Wherein [ id1 iq1]T[id3 iq3]TThe results of coordinate transformation are (9) and (11), respectively.
As a further improvement of the present invention, the input side current command is further optimized and calculated as follows:
will output a voltage uoAnd an output current ioThe phase is increased by 90 °, resulting in:
uod＝Uosin(ωot+δ+90°)＝Uocos(ωot+δ) (26)
Figure BDA0002613554610000055
further obtaining:
Figure BDA0002613554610000056
Figure BDA0002613554610000057
Figure BDA0002613554610000058
thus, each component of the current command value is obtained as
Figure BDA0002613554610000059
iq1＝0 (32)
Figure BDA0002613554610000061
Figure BDA0002613554610000062
The current command in the dq coordinate system is finally obtained as follows:
Figure BDA0002613554610000063
Figure BDA0002613554610000064
in the case of a 6kV, 710kW motor, a three-phase 15-submodule motor drive, the capacitance value of the capacitor is 100 μ F.
Compared with the prior art, the invention has the following advantages:
the submodule input power defined by the invention is not pure voltage source input power any more, and the total power of the three inductors is counted, and the difference between the two is the total input power. The power generated by the inversion side fundamental wave voltage and the fundamental wave current is equivalent to the power generated by the inversion side fundamental wave voltage and the fundamental wave current, and the minimization of the capacitor voltage ripple can be realized. Based on the above thought, in order to realize the equivalent offset of the inductive power and the output power under the condition that the inductive power is counted into the total input power, the input current is designed, and the derivation of the input current is completed. And aiming at the problems of more parameters and complex calculation required by input current calculation, a simplified current calculation method is further provided, various parameters required by real-time monitoring and calculation are not required, and simple combination and calculation are performed only through existing signals in Flux oriented control of the motor rotor, so that the engineering realization of the algorithm provided by the invention is facilitated. The voltage fluctuation suppression strategy of the invention does not need additional elements and sensors, and has good fluctuation suppression effect under different working conditions of the medium-voltage high-power asynchronous motor only by the optimization of the algorithm.
Furthermore, the invention can solve the problem that the capacitor voltage fluctuation range is large under the conditions of low speed and large torque of the CHB-based medium-voltage high-power motor dragging system, so that a smaller capacitor can be adopted within the same voltage allowable fluctuation range (such as within +/-5% of a steady state value). Taking the working conditions of 6KV and 710kW motors and 5 submodules per phase as examples, if the voltage in the full rotating speed range is to be suppressed within the allowable range, the traditional power feedforward method needs an electrolytic capacitor of 1mF, but the algorithm provided by the invention can reduce the capacitor to 100 muF, so that the traditional method uses 10% of the capacitance value of the capacitor, and the thin film capacitor with longer service life and smaller volume can be used, thereby greatly reducing the total equipment volume, saving the cost and prolonging the equipment life. And no additional sensor and device are added, and only algorithm optimization is carried out, so that the method is simple and reliable to implement.
Drawings
FIG. 1 is a block diagram of a CHB-based medium voltage high power motor drive system;
FIG. 2 is a schematic diagram of the internal structure of each submodule of the CHB;
FIG. 3 is a rectifier control block diagram;
FIG. 4 is a schematic diagram of a conventional rotor flux linkage orientation control;
FIG. 5 illustrates a calculation step of the current command according to the present invention;
FIG. 6 illustrates a control element of the command current calculated in the present invention;
FIG. 7 shows a capacitor voltage ripple with a capacitance of 1mF in the conventional power feed-forward method;
fig. 8 shows the capacitor voltage ripple when the capacitance is 1mF in the power cancellation algorithm according to the present invention;
fig. 9 shows the simulation result for a capacitance of 100 muf in the case of the proposed algorithm.
Detailed Description
A Cascaded H-bridge (CHB) is widely applied to the dragging occasions of medium-voltage high-power motors due to the advantages of high modularization degree, high input voltage and power grade and the like, but the capacitor in a submodule of the structure has a serious voltage fluctuation problem, and the problem is more prominent particularly under the working condition of low speed and large torque. The problem can be improved to a certain extent through power feedforward under the condition of not adding external equipment and devices, but voltage fluctuation with larger amplitude still exists.
Fig. 1 is a structural diagram of a CHB-based medium-voltage high-power motor drive system, in which sub-modules have input sides connected to secondary sides of a multi-winding transformer, and n sub-modules of each phase have output sides connected in series. The three-phase CHB star connection method has one end connected to a midpoint and the other end connected to a three-phase asynchronous motor. The internal structure of each sub-module is shown in fig. 2, and comprises a front-end controllable rectifier bridge, a middle capacitor and a rear-end H-bridge inverter circuit.
Rectifier control block diagram as shown in FIG. 3, id、ia、ed、eqThe quantities of three-phase input voltage and input current after abc/dq conversion are indicated by the asterisk in the superscript as command values. And respectively controlling d-axis current and q-axis current by adopting a power grid voltage orientation and feedforward decoupling control strategy, and generating PWM (pulse-width modulation) waves to be transmitted to a rectifier switching tube. Because given the
Figure BDA0002613554610000071
Equal to 0, so the input current and the input voltage are in phase.
Let the three-phase input voltage be
Figure BDA0002613554610000072
Three-phase input current of
Figure BDA0002613554610000081
UsAnd IsRespectively input voltage and current amplitude. OmegasIs the angular frequency.
Then the input power is
Figure BDA0002613554610000082
The output side voltage and current only consider the fundamental wave, respectively
uo＝Uosin(ωot) (4)
Figure BDA0002613554610000083
Wherein,
Figure BDA0002613554610000084
is the motor power angle, omegaoFor output side angular frequency, UoAnd IoRespectively, the output voltage and the current fundamental wave amplitude. Hereinafter, frequency doubling, frequency quadrupling, and the like are referred to as output frequencies. The output side power can be obtained
Figure BDA0002613554610000085
In steady state conditions, the average power on the input side and the output side is equal, i.e.
Figure BDA0002613554610000086
Therefore, the instantaneous values of the two are different and absorbed by the capacitor, resulting in voltage fluctuation. The difference in power absorbed by the capacitor is
Figure BDA0002613554610000087
Traditional power feedforward, based on the instantaneous power theory, rectifier inputs active and reactive power p, q as:
Figure BDA0002613554610000088
Figure BDA0002613554610000089
ed eqis the component of the secondary voltage in the dq coordinate system. i.e. id iqIs the component of the input current in the dq coordinate system. Therefore, the rectifier input power p can be made equal to the output side power poI.e. by
Figure BDA00026135546100000810
Wherein, under the directional control of the network voltage, edEqual to the input voltage amplitude, i.e. Us. Obtaining d and q axis current instruction values
Figure BDA00026135546100000811
Figure BDA0002613554610000091
The traditional power feedforward method can make the input power of the rectifier equal to the output power by changing the current instruction values of the d and q axes, and reduce the voltage ripple of the capacitor.
However, this method has a neglected problem, which is a drawback of this method, and limits the further reduction of the capacitor voltage ripple, which is analyzed below and the solution of the present invention to this problem is proposed.
Subjecting the above obtained
Figure BDA0002613554610000092
And
Figure BDA0002613554610000093
carrying out dq/abc conversion to obtain three-phase input current under an abc coordinate system
Iin＝[ia ib ic]T＝I1+I2+I3 (13)
Wherein, I1 I2Is a positive sequence current, I3Is a set of negative sequence currents
Figure BDA0002613554610000094
Figure BDA0002613554610000095
Figure BDA0002613554610000096
The input power of the voltage source is obtained by calculation
Figure BDA0002613554610000097
I.e. the voltage source input power is equally compensated for the output power. However, in the research process, the double-frequency and quadruple-frequency power fluctuation with larger components exists on the capacitor, and corresponding voltage fluctuation is caused. This phenomenon occurs because conventional studies defaulted the input power to the voltage source input power. But for the capacitor, the power actually applied to it is the voltage source input power psSubtracting the instantaneous power p across the three inductorsLMinus the output power po，pLThe calculation formula is as follows:
Figure BDA0002613554610000098
not paying attention to p in the pastLThis is because the total instantaneous power of the three inductors is 0 in the case where the three-term input current is a symmetrical sine wave as shown in equation (2). However, in the case of power feed forward, the current becomes equation (13), the instantaneous total power p of the three inductorsLThe calculation result is as follows:
Figure BDA0002613554610000101
it can be seen from equation (19) that, since the three-phase current is not a three-phase sinusoidal symmetric current, but a superposition of the respective components, a frequency-doubled and frequency-quadrupled power fluctuation is generated on the inductor. Even under the condition that the input power of the voltage source is equal to the output power, the capacitor can be considered to no longer buffer the power fluctuation caused by the power difference between the capacitor and the inductor, but the power exchange is carried out between the capacitor and the inductor. This is where the previous power feed forward approach has not been concerned and is a problem that it introduces. To solve this problem, the input current needs to be redesigned by taking the inductive power into account. And, redefined as the power supply emits power with respect to input power requirements, i.e. instantaneous power of the sense
pin＝ps-pL (20)
The definition of the output power is not changed and is still po. In order to redesign the input current, starting from the input current of the conventional power feed forward, it is still assumed that the input current comprises two positive sequences and one negative sequence, namely:
Figure BDA0002613554610000102
Figure BDA0002613554610000103
Figure BDA0002613554610000104
combining (1), (18), (20), (21) - (23) yields an input power of
Figure BDA0002613554610000105
Comparison of pinAnd poThe value of a certain quantity in the input current can be obtained
Figure BDA0002613554610000111
Substituting (25) for formula (24), the fifth term is 4 times frequency, so can directly make I2Or I3Equal to 0 to eliminate the quadruple frequency component. Here, the set of currents with higher frequency should be made equal to 0, i.e. I20, thus obtaining input power
Figure BDA0002613554610000112
Wherein,
Figure BDA0002613554610000113
therefore, if the input power under the new definition is exactly equal to the output power, there are:
Figure BDA0002613554610000114
simplification of
Figure BDA0002613554610000115
Can obtain
Figure BDA0002613554610000116
Wherein, UmidAnd alpha is convenientCalculating defined intermediate variables which satisfy:
Figure BDA0002613554610000117
Figure BDA0002613554610000118
by comparing the formula (29) with the formula (6), the compound (I) can be obtained
Figure BDA0002613554610000119
Figure BDA00026135546100001110
Therefore, the amplitude and the phase of the input current command value are determined, and the amplitude and the phase information contained in the input current command value are the formulas (25) (32) (33), and the input current command value can be realized by reasonably designing a controller. The equivalent cancellation of the input and output power can be realized under the condition of considering the inductive instantaneous power, so that the voltage fluctuation on the capacitor can be greatly reduced.
The input current command values are given by the equations (21) and (23), respectively, because the control of the present invention is performed in dq coordinate system, and the current command values in dq coordinate system can be obtained by performing abc/dq conversion on the same
Figure BDA0002613554610000121
Wherein [ id1 iq1]T[id3 iq3]TThe results of coordinate transformation (21) and (23) are obtained.
The target current contains more quantities to be measured, such as output voltage, current effective value, motor power angle and the like. In order to avoid as much as possible of the newly added measurement variables and to make more use of the existing variables, a simplified target current calculation procedure is as follows:
will output a voltage uoAnd an output current ioThe phase is increased by 90 °, resulting in:
uod＝Uosin(ωot+δ+90°)＝Uocos(ωot+δ) (35)
Figure BDA0002613554610000122
further, it is possible to obtain:
Figure BDA0002613554610000123
Figure BDA0002613554610000124
Figure BDA0002613554610000125
thus, it can be seen that in the present algorithm, each component of the current command value is
Figure BDA0002613554610000126
iq1＝0 (41)
Figure BDA0002613554610000127
Figure BDA0002613554610000128
The current command under the dq coordinate system is finally obtained as
Figure BDA0002613554610000131
Figure BDA0002613554610000132
That is, various amplitude and phase parameters required for calculating the injection current are contained in the output voltage and the output current, and the amplitude and phase information is not required to be extracted respectively by a complex algorithm or a sensor and then calculated, and the output voltage and the output current are directly used for calculation.
The capacitor voltage ripple suppression algorithm based on power cancellation is provided, the problem of inductance instantaneous power fluctuation introduced by the conventional method is analyzed, the inductance power is counted in the total input power, the input current is designed, the current instruction under a dq coordinate system is obtained by a simplified method, and the purpose of reducing the voltage ripple is finally achieved.
The principle of the invention is as follows:
the method is characterized in that problems of the conventional method are analyzed from inductance power fluctuation which is ignored in the previous research, and input current at a rectifying side is designed based on the inductance power fluctuation, so that the total input power including inductance instantaneous power is balanced with output power of a submodule, voltage ripples on a capacitor are greatly reduced, and a smaller and longer-life film capacitor can be adopted to replace an electrolytic capacitor with a large capacitance value under the same ripple requirement, so that the equipment cost is saved, the equipment volume is reduced, the power density is improved, the system stability is improved, and the method has a high application value.
The present invention is further described in detail by the following embodiments, but it should not be understood that the scope of the present invention is limited to the following examples, and it will be apparent to those skilled in the art that the present invention can be easily replaced or changed without departing from the spirit of the present invention.
The invention can solve the problem that the capacitor voltage fluctuation range is large under the conditions of low speed and large torque of the CHB-based medium-voltage high-power motor dragging system, so that a smaller capacitor can be adopted within the same voltage allowable fluctuation range (such as within +/-5% of a steady-state value).
Taking the working conditions of 6KV and 710kW motors and 5 submodules per phase as examples, if the voltage in the full rotating speed range is to be suppressed within the allowable range, the traditional power feedforward method needs an electrolytic capacitor of 1mF, but the algorithm provided by the invention can reduce the capacitor to 100 muF, so that the traditional method uses 10% of the capacitance value of the capacitor, and the thin film capacitor with longer service life and smaller volume can be used, thereby greatly reducing the total equipment volume, saving the cost and prolonging the equipment life. And no additional sensor and device are added, and only algorithm optimization is carried out, so that the method is simple and reliable to implement.
The conventional voltage ripple suppression strategy enables the input power of a voltage source to be equal to the output power of an inverter side by adjusting the current instruction value of the rectifier side, so as to achieve the purpose of reducing the voltage ripple on the intermediate capacitor. However, the current on the rectifying side is not a three-phase symmetrical sine quantity any more, but the superposition of three components, and the conventional power feedforward method ignores a problem that the total instantaneous power on three input side inductors is not 0 any more due to the change of the current, and even if the input power of a voltage source is equal to the output power of an inversion side, the power on a capacitor is not 0, but power exchange exists between the capacitor and the inductor.
Therefore, in order to solve this problem, the sub-module input power should be redefined to be equal to the voltage source input power minus the instantaneous power on the three inductors, so as to be equal to the output power, and thus the power fluctuation and the voltage fluctuation on the capacitor can be minimized, and the purpose of reducing the ripple and using the capacitor with small capacitance value can be achieved. Starting from the control target, the input current when the input power and the output power are completely offset is designed, and aiming at the problem that the amount to be solved in the solved input current formula is large, the final formula of the input side instruction current of the rectifier is obtained by reasonably converting the output voltage and the output current and simply combining the output voltage and the output current before conversion and performing algebraic operation, so that the additional measurement link is avoided, and the design is simplified. The effectiveness and correctness of the proposed algorithm are verified by MATLAB/Simulink simulation.
Fig. 4 shows the rotor flux linkage orientation control commonly used in engineering, and the principle is not specifically described here, but only the symbols therein are described:
ψris the rotor flux linkage value, ismAs stator current excitation component, usx(x ═ A, B, C) is motor stator voltage fundamental wave, and the division by the number of each phase submodule is the output voltage fundamental wave of each phase submodule, i.e. uox(x＝A，B，C)。ma mb mcThe wave is modulated for a three-phase H-bridge. T iseIs the electromagnetic torque of the motor. i.e. ioxAnd (x ═ a, B, C) is the three-phase stator current and is also the output current of each submodule. OmegarIs the motor rotor angular velocity. Theta is the rotor flux linkage space angle. 2r/3s is a conversion link from a two-phase rotating coordinate system to a three-phase stationary coordinate system, and the other way is that 3s/2r is opposite.
FIG. 5 is a view that the conversion angle of the output voltage in the coordinate conversion link of "2 r/3 s" is increased by 90 degrees, so as to achieve the effect of obtaining the phase shift of 90 degrees of the output voltage fundamental wave and the output current of the sub-module. Then, the dq current command value of the phase module is calculated by the formula (44) (45) in the previous section. By controlling the input current of the rectifier to follow the command, the equivalent offset of the input power and the output power can be realized, and the ripple value is greatly reduced. And a smaller capacitor can be adopted under the condition of meeting the requirement of the same voltage ripple.
Fig. 6 is a directional control of the grid voltage of a rectifier commonly used in engineering, the algorithm provided by the present invention changes the command value of dq current to achieve equivalent cancellation of power, and the calculation of the command value is shown in fig. 2 and is not described again. By controlling the input current in the dq coordinate system, a better tracking effect can be achieved. e.g. of the typed eqThe value of the input side voltage in the dq coordinate system is made. Because the directional vector control of the grid voltage is mature, the directional vector control is only a control implementation means of the current obtained by the algorithm of the invention, and therefore, the detailed description is omitted.
Fig. 7, fig. 8, and fig. 9 are both capacitor voltage ripple simulation waveforms under different capacitance values, and correspond to a left subgraph and a right subgraph in each case, where the left subgraph is the case of the rated torque and the rated rotation speed of the motor, and the right subgraph is the case of the rated torque and the zero rotation speed of the motor. And (4) observing the two extreme working conditions, and if the two extreme working conditions meet the requirement of the ripple range, meeting the requirement under other working conditions. Fig. 7 to 9 will be explained below.
Fig. 7 shows a conventional power feed-forward method, in which the capacitance value of the capacitor is 1mF, and the ripple of the capacitor is about 90V at the rated torque and the rated rotational speed, and occupies about 7.34% of the set voltage value (1225V).
Fig. 8 shows the capacitor voltage ripple under the power cancellation algorithm, which still keeps the capacitance value of the capacitor at 1mF, and it can be seen that under the rated working condition, the capacitor voltage is greatly reduced, which is about 15V, and is 1.2% of the set value of the capacitor voltage. The validity of the proposed algorithm is verified.
Fig. 9 is a simulation result in the case of the proposed algorithm of the present invention, limiting the voltage ripple to around 90V as shown in fig. 7. Under the same ripple index requirement, the capacitance adopted in fig. 9 is only 100 muf, and compared with the case of 1mF in fig. 7, the capacitance is reduced to 10% of the original capacitance, so that a film capacitor can be adopted, the cost is greatly saved, the service life of equipment is prolonged, and the volume of the equipment is reduced.
The terms "first," "second," "third," "fourth," and the like in the description of the application and the above-described figures, if any, are used for distinguishing between similar elements and not necessarily for describing a particular sequential or chronological order. It is to be understood that the data so used is interchangeable under appropriate circumstances such that the embodiments of the application described herein are, for example, capable of operation in sequences other than those illustrated or otherwise described herein. Furthermore, the terms "comprises," "comprising," and "having," and any variations thereof, are intended to cover a non-exclusive inclusion, such that a process, method, system, article, or apparatus that comprises a list of steps or elements is not necessarily limited to those steps or elements expressly listed, but may include other steps or elements not expressly listed or inherent to such process, method, article, or apparatus.
It should be understood that in the present application, "at least one" means one or more, "a plurality" means two or more. "and/or" for describing an association relationship of associated objects, indicating that there may be three relationships, e.g., "a and/or B" may indicate: only A, only B and both A and B are present, wherein A and B may be singular or plural. The character "/" generally indicates that the former and latter associated objects are in an "or" relationship. "at least one of the following" or similar expressions refer to any combination of these items, including any combination of single item(s) or plural items. For example, at least one (one) of a, B, or C, may represent: a, B, C, "A and B", "A and C", "B and C", or "A and B and C", wherein A, B, C may be single or plural.
The above embodiments are only used for illustrating the technical solutions of the present application, and not for limiting the same; although the present application has been described in detail with reference to the foregoing embodiments, it should be understood by those of ordinary skill in the art that: the technical solutions described in the foregoing embodiments may still be modified, or some technical features may be equivalently replaced; and such modifications or substitutions do not depart from the spirit and scope of the corresponding technical solutions in the embodiments of the present application.

Claims (4)

1. A method for suppressing voltage fluctuation of a capacitor of a series H-bridge type frequency converter is characterized by comprising the following steps:
calculating an input side current instruction of the rectifier, and controlling an input side current value according to the input side current instruction so that input power is equal to power generated by inversion side fundamental voltage and fundamental current;
wherein the input power of the rectifier is equal to the voltage source input power minus the instantaneous power across the three inductors;
the method specifically comprises the following control processes:
d, q axis current instruction value for feeding forward power
Figure FDA0003108284730000011
And
Figure FDA0003108284730000012
carrying out dq/abc conversion to obtain three-phase input current under an abc coordinate system
Iin＝[ia ib ic]T＝I1+I2+I3 (1)
Wherein, I1 I2Is a positive sequence current, I3Is a set of negative-sequence currents; u shapeoFor output of voltage fundamental amplitude, I, of the submoduleoFor outputting the current amplitude, omegasFor side angular frequency, omega, of the gridoOutputting the side angular frequency for the submodule;
Figure FDA0003108284730000013
for the power angle of the motor
Figure FDA0003108284730000014
Figure FDA0003108284730000015
Figure FDA0003108284730000016
Calculating to obtain the input power of the voltage source
Figure FDA0003108284730000017
Namely the input power of the voltage source compensates the output power;
the power applied to the capacitor is the input power p of the voltage sourcesSubtracting the instantaneous power p across the three inductorsLMinus the output power po，pLThe calculation formula is as follows:
Figure FDA0003108284730000018
l is an inductor; instantaneous total power p of three inductorsLThe calculation result is as follows:
Figure FDA0003108284730000021
the input power is the power generated by the power supply to reduce the instantaneous power of the inductor, i.e.
pin＝ps-pL (8)。
2. The method for suppressing the capacitor voltage fluctuation of the series H-bridge type frequency converter according to claim 1, wherein the input side current instruction obtaining method is as follows:
let the input current comprise two positive sequences and one negative sequence, namely:
Figure FDA0003108284730000022
Figure FDA0003108284730000023
Figure FDA0003108284730000024
the amplitude, the phase, the frequency and the like are all to-be-solved quantities; further obtaining an input power of
Figure FDA0003108284730000025
pinContains a total of 6 items, each of which is denoted as pin1，pin2… … comparison of pinAnd poTo obtain the input current:
Figure FDA0003108284730000026
substituting the above parameters into (12), pin54 frequency doubling component; due to poDoes not contain the frequency component, therefore, let I2Is 0, the input power is obtained
Figure FDA0003108284730000031
Wherein,
Figure FDA0003108284730000032
for the sub-module, the output voltage and the current fundamental wave are respectively
uo＝Uosin(ωot) (16)
Figure FDA0003108284730000033
Wherein
Figure FDA0003108284730000034
Is the voltage and current phase difference of the submodules, which is equal to the power angle and the output power of the motor
Figure FDA0003108284730000035
In order to make the input power exactly equal to the output power, there are:
Figure FDA0003108284730000036
simplification of
Figure FDA0003108284730000037
To obtain
Figure FDA0003108284730000038
Wherein, UmidAnd α is an intermediate variable which satisfies:
Figure FDA0003108284730000039
Figure FDA00031082847300000310
comparing formula (18) with formula (20) to obtain
Figure FDA00031082847300000311
Figure FDA00031082847300000312
Therefore, the amplitude and the phase of each component of the input current command value are determined, and the amplitude and the phase information contained in the input current command value are the formulas (13), (23) and (24);
because the control is carried out under the dq coordinate system, the input current instruction value under the abc coordinate system is subjected to abc/dq conversion to obtain the current instruction value under the dq coordinate system
Figure FDA0003108284730000041
Wherein [ id1 iq1]T[id3 iq3]TThe results of coordinate transformation are (9) and (11), respectively.
3. The method for suppressing capacitor voltage fluctuation of a series H-bridge type frequency converter according to claim 1, wherein the input side current command is further optimized by the following calculation process:
will output a voltage uoAnd an output current ioThe phase is increased by 90 °, resulting in:
uod＝Uosin(ωot+δ+90°)＝Uocos(ωot+δ) (26)
Figure FDA0003108284730000042
further obtaining:
Figure FDA0003108284730000043
Figure FDA0003108284730000044
Figure FDA0003108284730000045
thus, each component of the current command value is obtained as
Figure FDA0003108284730000046
iq1＝0 (32)
Figure FDA0003108284730000047
Figure FDA0003108284730000048
The current command in the dq coordinate system is finally obtained as follows:
Figure FDA0003108284730000049
Figure FDA00031082847300000410
4. the method for suppressing capacitor voltage ripple of a series H-bridge type frequency converter according to claim 1, wherein the series H-bridge type frequency converter comprises a multi-winding step-down transformer, three phases of 3n sub-modules in total, and a three-phase asynchronous motor;
the input side of the sub-module is connected to the secondary side of the multi-winding transformer, and the output sides of the n sub-modules of each phase are connected in series; the three-phase H-bridge type frequency converter adopts a star connection method, one end of the three-phase H-bridge type frequency converter is connected into a middle point, and the other end of the three-phase H-bridge type frequency converter is connected with a three-phase asynchronous motor; each submodule sequentially comprises a front-end controllable rectifier bridge, a middle capacitor and a rear-end H-bridge inverter circuit.
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