CN112003518B

CN112003518B - High-speed doubly salient starting generator controller and control method thereof

Info

Publication number: CN112003518B
Application number: CN202010587450.0A
Authority: CN
Inventors: 甘忠文; 陈强; 薛开昶; 黄效维; 罗宗鑫
Original assignee: Guizhou Aerospace Linquan Motor Co Ltd
Current assignee: Guizhou Aerospace Linquan Motor Co Ltd
Priority date: 2020-06-24
Filing date: 2020-06-24
Publication date: 2023-07-28
Anticipated expiration: 2040-06-24
Also published as: CN112003518A

Abstract

The invention provides a high-speed doubly salient starter generator controller and a control method thereof, which are characterized in that: the device comprises a generator, a generator controller, a three-phase full-bridge circuit, a phase current comparison circuit, an exciting current control circuit, an FPGA control circuit, a DSP control circuit and an output capacitor C; compared with the three-phase full-bridge topology of Si IGBT devices, the three-phase full-bridge topology of SiC MOS device overcomes the defect that the conduction voltage drop of the IGBT cannot be lower than a certain fixed value when a switch tube is conducted, and reduces the conduction loss; the three-phase current comparison circuit controlled by the FPGA duty ratio is simple and easy to operate; compared with a method for judging the phase sequence and the zero position by the armature voltage after dragging the electro-magnetic doubly-salient motor, the novel phase sequence detection and zero position calibration method for the electro-magnetic doubly-salient motor is simpler, more convenient and more efficient.

Description

High-speed doubly salient starting generator controller and control method thereof

Technical Field

The invention relates to a controller of a high-speed electro-magnetic doubly salient starter generator and a control method thereof.

Background

The starter generator controller is used for driving the starter generator to work, and in a starting stage, the starter generator is used as a motor, energy flows to the starter generator from the starter generator controller, the engine is dragged from rest to an ignition rotating speed, and the engine is boosted after ignition. In the power generation stage, the starter generator is used as a generator, the engine inputs mechanical energy to the generator, the generator converts the mechanical energy into variable-voltage alternating-current electric energy, and the starter generator controller converts the variable-voltage alternating-current electric energy into constant-voltage direct-current electric energy and outputs the constant-voltage direct-current electric energy. The starter generator system formed by the starter generator controller and the starter generator has bidirectional energy fluidity, has two functions of starting and generating electricity, and can effectively improve the utilization rate of equipment.

The aviation field has stricter volume and weight requirements to airborne equipment, and simultaneously, the aviation field generally uses turbine engine, and turbine engine usually work rotational speed is higher, in order to lighten the weight of starter generator system and match with high-speed turbine engine, usually needs starter generator work rotational speed higher, reaches 20000r/min-70000r/min. At the same output voltage, the relation between the number of turns N of an armature winding of the starter generator and the rotating speed N is N1/N, and the relation between the self inductance L of the armature winding and the number of turns N of the winding is L1N ² Therefore, as the operating speed of the starter generator increases, the number of turns of the armature winding of the starter generator decreases, thereby causing a significant decrease in the inductance of the armature winding itself. Besides the small self inductance of the high-speed starter generator, the output electric frequency of the high-speed starter generator is also high, and the high-frequency and small inductance characteristics of the high-speed starter generator obviously increase the design difficulty of the starter generator controller.

Aviation power supply systems include Low Voltage Direct Current (LVDC), constant Speed and Constant Frequency (CSCF) alternating current, variable Speed and Constant Frequency (VSCF), variable Frequency Alternating Current (VFAC) and High Voltage Direct Current (HVDC), with the development of aircraft 'multi-electricity' and 'full electricity', the onboard setting electricity consumption is continuously increased, and the advantages of HVDC are more and more obvious, representing the main stream direction of future development.

For controllers of electro-magnetic doubly salient generators, the prior art is generally implemented in three ways:

1) The three-phase full bridge is adopted as the hardware of the starting controller, and the three-phase rectifier bridge is adopted as the hardware of the generating controller. The starting and generating processes respectively use independent hardware structures, when in starting, the three-phase full-bridge output variable-voltage variable-frequency alternating-current driving motor formed by Si IGBT works, the armature current is regulated as a main component, the exciting current is regulated as an auxiliary component, the torque, the rotating speed and the power are controlled during starting, the starting control mode comprises three-phase three-state control, three-phase six-state control, three-phase alpha plus beta control and three-phase nine-state control, and the voltage regulation is realized through the upper tube or lower tube chopping mode. During power generation, the three-phase rectifier bridge formed by the Si fast recovery diode converts alternating current into direct current, and the power generation voltage stabilization is realized by adjusting exciting current.

2) The front-stage converter and the three-phase full bridge are adopted as starting controller hardware, and the three-phase rectifier bridge is adopted as power generation controller hardware. The starting and generating processes still use separate hardware structures, respectively. When starting, as the motor speed increases, the realization difficulty of voltage regulation by chopping through the three-phase full bridge is higher, so the voltage regulation function is realized through the pre-stage converter, and the three-phase full bridge is correspondingly switched only according to the electric frequency of the armature winding. The pre-stage converter is typically a Buck converter and the push-pull forward converter is disclosed in CN 106357164B. The principle of operation is the same as in scheme 1) when generating electricity.

3) Three-phase full bridge is adopted as the starting/generating multiplexing hardware. And a three-phase full bridge formed by the Si IGBT modules is adopted. During power generation, the three-phase rectifier bridge is formed by utilizing diodes integrated by the Si IGBT module. Hardware multiplexing at the time of starting and generating power is realized. The operating principle at start-up and power generation is the same as in scheme 1).

The main disadvantage of solution 1) is that the armature current ripple of the high-speed starter generator at start-up is large, and the non-multiplexed start/generation hardware results in a large controller volume and weight. The scheme 2) solves the problem that the armature current ripple of the high-speed starter generator is large when the scheme 1) is started, but the hardware is more complicated. The main disadvantage of scheme 3) is that the upper limit frequency of the Si IGBT is limited, and the armature current ripple of the high-speed starter generator is large during starting; during high-speed power generation, the conduction voltage drop of the integrated fast recovery diode is high, reverse recovery is not negligible, and the conduction loss and the reverse recovery loss are high.

Disclosure of Invention

In order to solve the technical problems, the invention provides a controller of a high-speed electro-magnetic doubly salient starting generator and a control method thereof.

The invention is realized by the following technical scheme.

The invention provides a controller of a high-speed electro-magnetic doubly salient starter generator and a control method thereof, which are characterized in that: the device comprises a generator, a generator controller, a three-phase full-bridge circuit, a phase current comparison circuit, an exciting current control circuit, an FPGA control circuit, a DSP control circuit and an output capacitor C;

the DSP control circuit is connected with the generator armature group position signal theta through the rotary transformer RT through the rotary transformer decoding chip _raw And a rotating speed signal n, connected to the generator DC bus voltage V _dc The exciting current I is connected to the exciting control circuit _f Accessing an interrupt request signal INT of an FPGA control circuit through a PB bus;

the DSP control circuit outputs an excitation control signal PWMf to the excitation control circuit;

the FPGA control circuit is connected with a comparison current signal output by the three-phase current and phase current comparison circuit of the generator armature, and outputs a maximum current duty ratio signal I to the phase current comparison circuit _maxD ；

The FPGA control circuit also outputs PWM signals to the three-phase full-bridge circuit;

the exciting current control circuit is also connected with an exciting winding of the generator;

The phase current comparison circuit is also connected with the armature phase current of the generator;

one end of the three-phase full-bridge circuit is connected with the generator armature, and the other end of the three-phase full-bridge circuit is connected with the output capacitor C.

The phase current comparison circuit comprises a 1# phase circuit comparison circuit and a 2# phase current comparison circuit which are respectively connected with a 1# armature and a 2# armature of the generator;

the three-phase full-bridge circuit comprises a 1# three-phase full-bridge circuit and a 2# three-phase full-bridge circuit which are respectively connected with a 1# armature and a 2# armature of the generator.

The FPGA control circuit adopts alpha+beta controlled unipolar frequency multiplication PWM to drive the three-phase full-bridge circuit.

A control method of a high-speed electro-magnetic doubly salient starter generator controller comprises the following steps:

A. determining the phase sequence of an electric winding and acquiring zero offset between a zero position of a motor and a zero position of a rotary transformer;

B. the engine is dragged from rest to the ignition rotating speed through starting control, and then the engine is boosted;

C. the engine drags the starting generator to the power generation rotating speed;

D. and the output energy of the starter generator is converted into voltage-stabilizing direct current output through power generation control.

The determining mode of the phase sequence of the electric winding in the step A is as follows:

the phase sequence of the No. 1 armature is determined in the following way:

A ₁₁ setting the three-phase output lines of the No. 1 armature as L respectively ₁₁ 、L ₁₂ 、L ₁₃ Three-phase windings are sequentially set to be A respectively by motor steering ₁ 、B ₁ 、C ₁ Setting L ₁₁ Correspond to A ₁ And assume L ₁₂ 、L ₁₃ Corresponds to B in turn ₁ 、C ₁ ；

A ₁₂ The external power supply leads the current to flow from A ₁ End into motor, from C ₁ A motor is output;

A ₁₃ applying a power supply to make the current flow from B ₁ The end enters the motor from A ₁ A motor is output;

A ₁₄ applying a power supply to make current flow from C _{Zedoary turmeric} End into motor, from B ₁ A motor is output;

A ₁₅ observe the A ₁₂ To A. Sup. Th ₁₄ The rotation direction of the motor during the step operation is L ₁₂ Correspond to B ₁ And L ₁₃ Correspond to C ₁ If the rotation direction of the motor is opposite to the defined direction, L ₁₂ Correspond to C ₁ And L ₁₃ Correspond to B ₁ ；

Determining the three-phase winding A of the No. 2 armature in the same manner ₂ ’、B ₂ ' and C ₂ ' phase sequence.

The zero offset obtaining step in the step A is as follows:

A ₂₁ optionally selecting one exciting current direction, and introducing direct exciting current I _f ；

A ₂₂ Applying a current source to the No. 1 armature and sequentially applying the current from C ₁ The end entering motor is arranged at the end B1 and the end A1, and the reading angle theta of the rotary transformer is recorded in sequence _f1 、θ _f2 According to the formula delta theta _f |＝|θ _f1 -θ _f2 The direction of the dc excitation current If in step a21 is adjusted to be Δθ _f |<30°；

A23, determining the relation between the 1# armature and the 2# armature winding:

a231, external power supply to make current flow from A ₁ Into the motor from C ₁ End-out motor for recording readout angle theta of rotary transformer ₁ ；

A232, external power supply to make current flow from A ₂ ' into the motor, from C ₂ ' end out motor, record resolver read angle θ ₂ ；

A233, will be delta theta ₂₁ ＝θ ₂ -θ ₁ The result is calculated to be within 0-360 DEG, and if delta theta is calculated for the doubly salient motor with two stators aligned with the same teeth and two rotors staggered at equal intervals ₂₁ =180°, the final 2# armature winding a ₂ ＝A ₂ ’、B ₂ ＝B ₂ ' and C ₂ ＝C ₂ ' if delta theta ₂₁ =300°, the final 2# armature winding a ₂ ＝C ₂ ’、 B ₂ ＝A ₂ ' and C ₂ ＝B ₂ ' if delta theta ₂₁ =60°, the final 2# armature winding a ₂ ＝B ₂ ’、B ₂ ＝C ₂ ' and C ₂ ＝A ₂ ’；

A24, applying current source to make current flow from C ₁ End into motor, from B ₁ Terminating the motor, and after the motor is stationary, turning on the power A ₁ End and B ₁ The ends are connected in parallel, and a current source is externally added to enable current to flow from C ₁ The end enters the motor from A ₁ And B is connected with ₁ The parallel end outputs a motor, so that the 1# armature winding is locked at the 0-degree electric angle position, and the output position signal of the rotary transformer is recorded as a zero offset signal delta theta at the moment;

the starting control step in the step B is as follows:

b1, generating an interrupt request signal INT once in each switching period Ts in the FPGA, and sending the interrupt request signal INT to the DSP so that the DSP executes one-time calculation;

b2, reading a position signal thetaw and a rotating speed signal n which are output by the rotary-variable decoding chip through the DSP,

b3 by position signal θ _raw Calculating the electrical angle θ of the 1# and 2# armature windings _raw1 And theta _raw2 ，

θ _raw1 ＝θ _raw -△θ；

θ _raw2 ＝θ _raw1 +180°；

B4, determining the exciting current parameter value I in the DSP according to the rotating speed signal n _fref Advance angles alpha and beta, armature current parameter value I _ref Back emf E _q Armature winding electrical frequency f _e ；

B4 according to the exciting current parameter value I _fref And exciting current I _f In DSP, exciting current parameter value I _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _{Zedoary turmeric} ；

B5, according to the armature winding electrical frequency f _e Calculating the frequency compensation signal f of the PLL angle encryption module in a DSP _scl ；

B6, through parallel bus PB, the position signal θ acquired by DSP _raw1 And theta _raw2 Frequency compensation signal f _scl Advance angles alpha and beta, armature current parameter value I _ref Back emf E _q Sending the data to an FPGA;

b7, according to the position signal θ _raw1 And a frequency compensation signal f _scl Outputting an encryption angle theta through a PLL angle encryption module in the FPGA;

b8, according to armature current 1#, i _a1-c1 According to I in FPGA _feed1 ＝|i _a1 |+|i _b1 |+|i _c1 Computing armature current feedback signal I _feed1 ；

B9, according to the armature current parameter value I _ref Armature current feedback signal I _feed1 And counter potential E _q Current parameter value I in FPGA _ref Feedback signal I with armature current _feed1 Performing PI operation to output V after difference _1PI Then press D= (V) _1PI +E _q )/V _{Zedoary turmeric} Calculating a duty ratio D and outputting the duty ratio D;

b10, according to the advance angles Russian and beta, the encryption angle theta and the duty ratio D, the single-polarity frequency multiplication PWM controlled by alpha+beta is realized through a driving pulse generator module in the FPGA, and a driving signal PWM of the SiC MOS full bridge 1 is generated _11-16 ；

B11, generating a driving signal PWM of the SiC MOS full bridge 2 according to the method of the steps B1-B10 _21-26 。

The step D is characterized in that the power generation control step comprises the following steps:

d1, generating an interrupt request signal INT once in each switching period Ts in the FPGA and sending the interrupt request signal INT to the DSP so that the DSP executes one-time calculation;

d2 according to DC voltage V _dc And exciting current I _f In DSP, the reference voltage V will be output _ref And the PI operation is carried out after the difference with the DC voltage Vdc to output the exciting current parameter value I _fref In DSP, exciting current parameter value I _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _fref To make the DC voltage V _dc ＝V _ref The power generation and voltage stabilization are realized;

d3, A according to armature winding # 1 ₁ Phase current i _a1 Will i in FGPA _a1 And-i _th 、i _th Comparison, i _th For a number slightly greater than 0, if i _a1 >i _th Make SiC MOS full bridge 1 middle power tube Q ₁₁ Is controlled by the driving signal PWM ₁₁ Set high, if i _a1 <i _th Make SiC MOS full bridge 1 middle power tube Q ₁₁ Is controlled by the driving signal PWM ₁₁ If i is lower than the height _a1 >-i _th Make SiC MOS full bridge 1 middle power tube Q ₁₂ Is controlled by the driving signal PWM ₁₂ Put low, if i _a1 <-i _th Make SiC MOS full bridge 1 middle power tube Q ₁₂ Is controlled by the driving signal PWM ₁₂ Setting low and high;

d4, generating power tube Q in SiC MOS full bridge 1 according to the mode of step D3 _13-16 Is controlled by the driving signal PWM _13-16 And in the SiC MOS full bridge 2Power tube Q _21-26 Is controlled by the driving signal PWM _21-26 。

The calculation process of the angle encryption signal θ of the angle encryption module in the step B5 is as follows:

b51, position signal θ _raw1 And the encryption angle signal theta is 0-2 ^x Representing 0 ° -360 °, each FPGA executing frequency f _clk Triggering the angle encryption module to perform one-time operation;

b52, position signal θ _raw1 The signal is limited after the difference with the encryption angle signal theta to obtain an error signal e _rr ；

B53 to convert error signal e _rr And a frequency compensation signal f _scl Carrying out summation operation to obtain a frequency signal f;

b54, sending the frequency signal f to an (x+y) bit accumulator for accumulation;

and B55, right shifting the accumulator value of the binary representation by y bits to obtain an encryption angle signal theta.

The step of generating the PWM driving signal of the SiC MOS full bridge 1 by the FPGA driving pulse generator in the step B10 is:

b101, execution frequency f of each FPGA _clk Performing one-time increment or one-time decrement on the triangular waveform counter vtri, generating a triangular waveform in a mode of firstly increment to N and then decrement to 0, enabling one switching period Ts to correspond to 2N FPGA counting periods, and enabling a direction signal D when the triangular waveform counter vtri is respectively increment and decrement _ir1 Setting 0 and 1 respectively;

b102, according to the duty ratio D of the output data, according to N _ca1 Calculate the comparison value N =n/2- (1+d) and nca2=n/2- (1-D) _ca1 And N _ca2 ；

B103, when D _ir1 When vtri=n, if=0 _ca2 Upper tube driving signal G _u1 From 0 to 1, when D _ir1 If v when oxa=1 _tri ＝N _ca2 Or I _shut1 =0, then go up tube driving signal G _u1 From 1 to 0;

b104, when D _ir1 When=1, if v _tri ＝N _ca1 Then down tube driving signal G _d1 From 0 to 1, when D _ir1 When=0, if v _tri ＝N _ca1 Or I _shut1 =0, then the down tube driving signal G _d1 From 0 to 1;

b105, encrypting angle signal theta, advance angles alpha and beta according to No. 1 armature winding, and setting an electric frequency f according to theta= (-beta 0-120 deg. beta 1), (120 deg. beta 2-120 deg. beta 3), (120 deg. beta-240 deg. alpha), (240 deg. alpha-240 deg. beta), (240 deg. beta-360 deg. alpha), (360 deg. alpha-360 deg. beta) _e The corresponding period is divided into working areas 1-6;

b106, according to the working areas 1-6, the upper tube driving signal G _u1 And a down tube driving signal G _d1 In the working area 1, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₆ Is controlled by the driving signal PWM ₁₁ And PWM ₁₆ In the working area 2, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₆ Is controlled by the driving signal PWM ₁₃ And PWM ₁₆ In the working area 3, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₂ Is controlled by the driving signal PWM ₁₃ And PWM ₁₂ In the working area 4, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₂ Is controlled by the driving signal PWM ₁₅ And PWM ₁₂ In the working area 5, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₄ Is controlled by the driving signal PWM ₁₅ And PWM ₁₄ In the working area 6, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₄ Is controlled by the driving signal PWM ₁₁ And PWM ₁₄ ；

The invention has the beneficial effects that:

1) Compared with the three-phase full-bridge topology of Si IGBT devices, the three-phase full-bridge topology of SiC MOS devices overcomes the defect that the conduction voltage drop of the IGBT cannot be lower than a certain fixed value when the switching tube is conducted, so that the conduction loss is reduced;

2) The three-phase current comparison circuit controlled by the FPGA duty ratio is simple and easy to operate;

3) Compared with a method for judging the phase sequence and the zero position by the armature voltage after dragging the electro-magnetic doubly-salient motor, the novel phase sequence detection and zero position calibration method of the electro-magnetic doubly-salient motor is simpler, more convenient and more efficient;

4) The method for encrypting the original output angle of the rotary transformer based on the FPGA is provided, so that the accuracy of starting control can be effectively improved;

5) The method for combining unipolar frequency multiplication PWM and alpha+beta control is provided, and under the condition of not increasing the control switching frequency, the equivalent switching frequency is doubled, so that the current ripple wave of an armature winding is effectively reduced, and the method can be better suitable for the winding low inductance characteristic of a high-speed electro-magnetic doubly salient motor;

6) Compared with the traditional single peak current control, the control is more stable and reliable, and the electromagnetic compatibility characteristic is better;

7) The power generation synchronous rectification control method based on the SiC MOS can overcome the defect that the conduction voltage drop of the Si fast recovery diode cannot be lower than that of a PN junction, so that the conduction loss during power generation is obviously reduced; meanwhile, the reverse recovery problem of the Si fast recovery diode is also solved, so that the reverse recovery loss during power generation is obviously reduced.

Drawings

FIG. 1 is a block diagram of a high-speed electro-magnetic doubly salient generator controller of the present invention;

FIG. 2 is a schematic diagram of a SiC MOS full bridge circuit of the present invention;

FIG. 3 is a schematic diagram of a phase current comparison circuit of the present invention;

FIG. 4 is a schematic diagram of the startup control operation of the present invention;

FIG. 5 is a schematic diagram of the power generation control operation of the present invention;

FIG. 6 is a schematic diagram of the operation of the phase-locked loop encryption module of the present invention;

FIG. 7 is a schematic diagram of the unipolar frequency doubling PWM control of the present invention;

FIG. 8 is a diagram of a unipolar frequency doubling PWM control operating mode of the present invention;

FIG. 9 is a schematic illustration of the α+β control operating region of the present invention;

fig. 10 is a diagram comparing the present invention with a conventional start control mode.

Detailed Description

The technical solution of the present invention is further described below, but the scope of the claimed invention is not limited to the above.

A controller of a high-speed electro-magnetic doubly salient starter generator and a control method thereof are characterized in that: the device comprises a generator, a generator controller, a three-phase full-bridge circuit, a phase current comparison circuit, an exciting current control circuit, an FPGA control circuit, a DSP control circuit and an output capacitor C;

the FPGA control circuit is connected with a comparison current signal output by the three-phase current and phase current comparison circuit of the generator armature, and outputs a maximum current duty ratio signal ImaxD to the phase current comparison circuit;

the phase sequence of the No. 1 armature is determined in the following way:

a11, setting the three-phase output lines of the No. 1 armature as L respectively ₁₁ 、L ₁₂ 、L ₁₃ Three-phase windings are sequentially set to be A respectively by motor steering ₁ 、B ₁ 、C ₁ Setting L ₁₁ Correspond to A ₁ And assume L ₁₂ 、L ₁₃ Corresponds to B in turn ₁ 、C ₁ ；

A12, an external power supply leads the current to flow from A ₁ End into motor, from C ₁ A motor is output;

a13, externally-applied power supply to enable current to flow from B ₁ The end enters the motor from A ₁ A motor is output;

a14, external power supply to make current flow from C ₁ End into motor, from B ₁ A motor is output;

a15, observing the rotation direction of the motor in the operation of the steps A12 to A14, and if the rotation direction of the motor is the same as the defined positive direction, L ₁₂ Correspond to B ₁ And L ₁₃ Correspond to C ₁ If the rotation direction of the motor is opposite to the defined direction, L ₁₂ Correspond to C ₁ And L ₁₃ Correspond to B ₁ ；

The zero offset obtaining step in the step A is as follows:

a21, randomly selecting one exciting current direction, and introducing direct-current exciting current I _f ；

A22, applying a current source to the No. 1 armature and enabling the current to enter the motor from the C1 end and to exit from the B1 end and the A1 end in sequence, and recording the readout angle of the rotary transformer as theta in sequence _f1 、θ _f2 According to the formula delta theta _f |＝|θ _f1 -θ _f2 I, the direct current exciting current I in the step A21 is adjusted _f Is oriented such that delta theta _f |< 30°；

A233, will be delta theta ₂₁ For a doubly salient motor with two stators aligned with the same teeth and two rotors staggered equidistantly, if Δθ, the value of =θ2- θ1 is calculated to be within 0-360 DEG ₂₁ =180°, the final 2# armature winding a ₂ ＝A ₂ ’、B ₂ ＝B ₂ ' and C ₂ ＝C ₂ ' if delta theta ₂₁ =300°, the final 2# armature winding a ₂ ＝ C ₂ ’、B ₂ ＝A ₂ ' and C ₂ ＝B ₂ ' if delta theta ₂₁ =60°, the final 2# armature winding a ₂ ＝B ₂ ’、B ₂ ＝C ₂ ' and C ₂ ＝A ₂ ’；

A24, applying current source to make current flow from C ₁ End into motor, from B ₁ Terminating the motor, and after the motor is stationary, turning on the power A ₁ End and B ₁ The ends are connected in parallel, and a current source is externally added to enable current to flow from C ₁ The end enters the motor from A ₁ And B is connected with ₁ The parallel end-out motor locks the No. 1 armature winding at the 0-degree electric angle position, records the output position signal of the rotary transformer as zero offsetA signal Δθ;

the starting control step in the step B is as follows:

b2, reading the position signal theta output by the rotary decoding chip through the DSP _raw And a rotational speed signal n;

b3 by position signal θ _raw Calculating the electrical angles theta raw1 and theta raw2 of the armature windings 1# and 2 #;

θ _raw1 ＝θ _raw -△θ；

θ _raw2 ＝θ _raw1 +180°；

B4 according to the exciting current parameter value I _fref And the exciting current If, the exciting current parameter value I is set in DSP _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _fref ；

B9, according to the armature current parameter value I _ref Armature current feedback signal I _feed1 And counter potential E _q Current parameter value I in FPGA _ref Feedback signal I with armature current _feed1 After the difference is made, PI operation is carried out to output V1PI, and then D= (V1 PI+E) _q )/V _dc Calculating a duty ratio D and outputting the duty ratio D;

B10, according to the advance angles alpha and beta, the encryption angle theta and the duty ratio D, a single-polarity frequency multiplication PWM controlled by alpha+beta is realized through a driving pulse generator module in the FPGA, and driving signals PWM11-16 of the SiC MOS full bridge 1 are generated;

d3, A according to armature winding # 1 ₁ Phase current i _a1 Will i in FGPA _a1 Comparing with-ith, i _th For a number slightly greater than 0, if i _a1 >i _th Make SiC MOS full bridge 1 middle power tube Q ₁₁ Is controlled by the driving signal PWM ₁₁ Set high, if i _a1 <i _th Make SiC MOS full bridge 1 middle power tube Q ₁₁ Is controlled by the driving signal PWM ₁₁ If i is lower than the height _a1 >-i _th Make SiC MOS full bridge 1 middle power tube Q ₁₂ Is controlled by the driving signal PWM ₁₂ Put low, if i _a1 <-i _th Make SiC MOS full bridge 1 middle power tube Q ₁₂ Is controlled by the driving signal PWM ₁₂ Setting low and high;

d4, generating power tube Q in SiC MOS full bridge 1 according to the mode of step D3 _13-16 Is controlled by the driving signal PWM _13-16 And SiC MOS full bridge 2 medium power tube Q _21-26 Is controlled by the driving signal PWM _21-26 。

b101, execution frequency f of each FPGA _clk For triangle waveform counter v _tri Performing one-time increment or one-time decrement counting, generating triangular waveforms in a mode of firstly increasing N and then decreasing to 0, enabling one switching period Ts to correspond to 2N FPGA counting periods, and enabling a direction signal D when a triangular waveform counter vtri is respectively increased and decreased _ir1 Setting 0 and 1 respectively;

b102, according to the duty ratio D of the output data, according to N _ca1 =n/2· (1+d) and N _ca2 Calculate the comparison value N =n/2· (1-D) _ca1 And N _ca2 ；

B103, when D _ir1 When=0, if v _tri ＝N _ca2 Upper tube driving signal G _u1 From 0 to 1, when D _ir1 When=1, if v _tri ＝N _ca2 Or I _shut1 =0, then go up tube driving signal G _u1 From 1 to 0;

b105, according to the encrypting angle signal theta of the No. 1 armature winding and the advance angles alpha and beta, dividing a corresponding period of the electric frequency fe into working areas 1-6 according to the angle theta= (-beta-120 deg. alpha), (120 deg. -alpha 120 deg. -beta), (120 deg. -beta-240 deg. alpha), (240 deg. -alpha-240 deg. -beta), (240 deg. -beta 360 deg. -alpha), (360 deg. -alpha) -360 deg. -beta);

Examples:

the high-speed generator adopts a 12/8 electro-magnetic doubly salient starter generator, adopts a structure that two stators are aligned with the same teeth and two rotors are staggered at equal intervals to reduce torque fluctuation, and the electric angles of a 1# armature winding and a 2# armature winding are 180 degrees different. The starting rotation speed is 0-35000r/min, 30 N.m constant power starting is carried out at 0-10000r/min, and 30kW constant power starting is carried out at 10000r/min-35000 r/min. The rated rotation speed of power generation is 65000r/min, and 270V and 150kW are required to be output within the rotation speed range of 90-105% of the rated generator. The application background is aviation Auxiliary Power Units (APUs), which employ HVDC power systems.

Fig. 1 shows a structural block diagram of a proposed high-speed electro-magnetic doubly salient starter generator controller, which comprises a SiC MOS full bridge 1, a SiC MOS full bridge 2, an output capacitor C, a rotary decoding chip, a phase current comparison circuit 1, a phase current comparison circuit 2, a DSP and an FPGA; the motor has armature No. 1 connected via SiC MOS full bridge 1 to output capacitor C, armature No. 2 connected via SiC MOS full bridge 2 to output capacitor C, output capacitor C connected to DC bus, exciting current control circuit connected to exciting winding of the motor, and rotary transformer RT outputting position signal theta via rotary decoding chip _raw The rotating speed signal n is connected with the DSP, and the three-phase current i of the No. 1 armature is equal to the three-phase current i of the No. 1 armature _a1 、 i _b1 And i _c1 Respectively connected with the phase current comparison circuit 1 and the FPGA, and the No. 2 armature three-phase current i _a2 、 i _b2 And i _c2 Respectively connected with the phase current comparison circuit 2 and the FPGA, and the FPGA outputs a maximum current duty ratio signal I _maxD Are connected with a phase current comparison circuit 1 and a phase current comparison circuit 2 respectively, and the phase current comparison circuit 1 outputs I _shut1 Connected with FPGA, the phase current comparison circuit 2 outputs I _shut2 Is connected with FPGA, DC bus voltage V _dc And exciting current I _f Is connected with the DSP, the FPGA is connected with the DSP through a parallel bus PB, the FPGA outputs an interrupt request signal INT to be connected with the DSP, and the DSP outputs an excitation control signal PWM _f Is connected with an excitation current control circuit, and an FPGA outputs a power tube driving signal PWM _11-16 Is connected with the SiC MOS full bridge 1, and the FPGA outputs a power tube driving signal PWM _21-26 Is connected with the SiC MOS full bridge 2.

The structure of the SiC MOS full bridge 1 and the SiC MOS full bridge 2 in FIG. 1 is the same, and are all three-phase full bridge circuits formed by SiC MOS devices, and FIG. 2 shows the proposed SiC MOS full bridge 1 circuitIn the figure, the SiC MOS full bridge 1 includes a SiC MOS device Q ₁₁ -Q ₁₆ Is composed of a DC side positive electrode and Q ₁₁ Is D, Q of (C) ₁₃ Drains D and Q of (2) ₁₅ Is connected with the drain electrode D of the transistor, the negative electrode on the direct current side is respectively connected with the Q ₁₂ Source S, Q of (2) ₁₂ Drains S and Q of (2) ₁₆ Is connected with drain S of A ₁ Phase alternating current armature winding is respectively connected with Q ₁₁ Sources S and Q of (2) ₁₂ Drain D of B ₁ Phase alternating current armature winding is respectively connected with Q ₁₃ Sources S and Q of (2) ₁₄ Is connected with the drain electrode D, C ₁ Phase alternating current armature winding is respectively connected with Q ₁₅ Sources S and Q of (2) ₁₆ Is connected to the drain electrode D. The three-phase full-bridge topology based on the SiC MOS can realize hardware multiplexing of starting and power generation, and compared with the three-phase full-bridge topology of an Si IGBT device, the switching tube overcomes the defect that the conduction voltage drop of the IGBT cannot be lower than a certain fixed value when being conducted, so that the conduction loss is reduced; the switching loss of the SiC MOS is obviously lower than that of the Si IGBT, so that the switching loss is obviously reduced, the SiC MOS can work at higher frequency, and the SiC MOS is more suitable for high-frequency and low-inductance high-speed starting generator control application; the integrated diode in the SiC MOS module is a Schottky diode, and compared with the Si fast recovery diode, the integrated diode has no reverse recovery phenomenon, and the reverse loss of the diode is obviously reduced when the power tube is started to switch and when the power generation diode is switched.

In fig. 1, the phase current comparing circuit 1 has the same structure as the phase current comparing circuit 2, fig. 3 shows a circuit diagram of the proposed phase current comparing circuit 1, and the phase current comparing circuit 1 includes a comparator C _a1 Comparator C _b1 Comparator C _c1 Resistance R ₁ Resistance R ₂ And capacitor C ₁ Is made up of a maximum current duty cycle signal I _maxD Through resistor R ₁ Respectively with comparator C _a1 Comparator C _b1 And comparator C _c1 Is connected with the non-inverting input terminal of the comparator C _a1 Comparator C _b1 And comparator C _c1 Through capacitor C ₁ Connected to power supply ground, phase current i _a1 And comparator C _a1 Is connected to the inverting input terminal of (1), phase current i _b1 And comparator C _b1 Is connected to the inverting input terminal of (1), phase current i _c1 And comparator C _c1 Is connected with the inverting input terminal of the comparator C _a1 Comparator C _b1 And comparator C _c1 The output ends of the phase current comparison circuit 1 are respectively connected with the output end I of the phase current comparison circuit _shut1 Connection, I _shut1 Through resistor R ₂ And the positive electrode V of the power supply _cc And the connection is formed. The three-phase current comparison circuit controlled by the FPGA duty ratio is simple and easy to operate.

The controller control method of the high-speed electro-magnetic doubly salient starter generator comprises the following steps:

A. determining the phase sequence of an electric winding, and obtaining zero offset between the zero position of the motor and the zero position of the rotary transformer through zero calibration;

The step A of determining the phase sequence and zero calibration of the electric winding comprises the following steps:

a. optionally selecting one square exciting current direction, and introducing direct exciting current I _f ；

b. Three-phase output line L for No. 1 armature winding ₁₁ 、L ₁₂ And L ₁₃ Let L ₁₁ ＝A ₁ Let L ₁₂ ＝B ₁ And L ₁₃ ＝C ₁ ；

c. Applying a current source to make the current flow from A ₁ End into motor, from C ₁ A motor is output;

d. applying a current source to make the current flow from B ₁ The end enters the motor from A ₁ A motor is output;

e. applying a current source to make the current flow from C ₁ End entry motor slave B ₁ A motor is arranged at the end;

f. observing the rotation direction of the motor in the operation of the c-th to e-th steps, and if the rotation direction of the motor is the same as the defined positive direction, L ₁₂ ＝B ₁ And L ₁₃ ＝C ₁ If the rotation direction of the motor is opposite to the defined direction, L ₁₂ ＝C ₁ And L ₁₃ ＝B ₁ ；

g. Applying a current source to make the current flow from C ₁ End entry motor slave B ₁ End-out motor for recording readout angle theta of rotary transformer _f1 The method comprises the steps of carrying out a first treatment on the surface of the Applying a current source to make the current flow from C ₁ End entry motor slave A ₁ End-out motor for recording readout angle theta of rotary transformer _f2 The method comprises the steps of carrying out a first treatment on the surface of the Let delta theta _f |＝ |θ _f1 -θ _f2 I, if I delta theta _f |<30 degrees, the exciting current direction selected in the step a is the expected direction, if delta theta _f C, the I is more than or equal to 30 degrees, and the direct current excitation current I selected in the step a _f Not in the expected direction, the direct current exciting current I needs to be changed _f And then executing the subsequent steps;

h. determining A of the 2# armature winding according to the method described in steps b to f ₂ ’、B ₂ ' and C ₂ ’；

i. Applying a current source to make the current flow from A ₁ Into the motor from C ₁ End-out motor for recording readout angle theta of rotary transformer ₁ ；

j. Applying a current source to make the current flow from A ₂ ' into the motor, from C ₂ ' end out motor, record resolver read angle θ ₂ ；

k. Will be delta theta ₂₁ ＝θ ₂ -θ ₁ The result is calculated to be within 0-360 DEG, and if delta theta is calculated for the doubly salient motor with two stators aligned with the same teeth and two rotors staggered at equal intervals ₂₁ =180°, the final 2# armature winding a ₂ ＝A ₂ ’、B ₂ ＝B ₂ ' and C ₂ ＝C ₂ ' if delta theta ₂₁ =300°, the final 2# armature winding a ₂ ＝C ₂ ’、B ₂ ＝A ₂ ' and C ₂ ＝B ₂ ' if delta theta ₂₁ =60°, the final 2# armature winding a ₂ ＝B ₂ ’、B ₂ ＝C ₂ ' and C ₂ ＝A ₂ ’；

l, applying a current source to make the current flow from C ₁ End into motor, from B ₁ Terminating the motor, and after the motor is stationary, turning on the power A ₁ End and B ₁ The ends are connected in parallel, and a current source is externally added to enable current to flow from C ₁ The end enters the motor from A ₁ And B is connected with ₁ The parallel end outputs a motor, so that the 1# armature winding is locked at the 0-degree electric angle position, and the output position signal delta theta of the rotary transformer is recorded;

m, let resolver output position signal be θ _raw Angle θ of 1# armature winding _raw1 ＝θ _raw Angle θ of- Δθ,2# armature winding _raw2 ＝θ _raw1 +180°。

Compared with the method for judging the phase sequence and the zero position by the armature voltage after dragging the electro-magnetic doubly-salient motor, the novel phase sequence detection and zero position calibration method for the electro-magnetic doubly-salient motor is simpler, more convenient and more efficient.

In connection with the starting control operation schematic diagram of fig. 4, the starting control described for step B includes the following steps:

a. each switching period T in the FPGA _s Generating a primary interrupt request signal INT and sending the primary interrupt request signal INT to the DSP so that the DSP executes primary calculation;

b. reading position signal theta output by rotary decoding chip through DSP _raw And a rotation speed signal n, and calculates an electrical angle θ of the 1# and 2# armature windings _raw1 And theta _raw2 ；

c. Determining exciting current parameter I in DSP according to rotation speed n _fref Advance angles alpha and beta, armature current parameter value I _ref Back emf E _q Armature winding electrical frequency f _e ；

d. According to exciting current parameter value I _fref And exciting current I _f In DSP, exciting current parameter value I _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝ I _fref ；

e. According to the armature winding electrical frequency f _e Calculating the frequency compensation signal f of the PLL angle encryption module in a DSP _scl ；

f. Through parallel bus PB, DSP obtains position signal θ _raw1 And theta _raw2 Frequency compensation signal f _scl Advance angles alpha and beta, armature current parameter value I _ref Back emf E _q Sending the data to an FPGA;

g. according to the position signal theta _raw1 And a frequency compensation signal f _scl Outputting an encryption angle theta through a PLL angle encryption module in the FPGA;

h. According to armature current i # 1 _a1-c1 According to I in FPGA _feed1 ＝|i _a1 |+ |i _b1 |+|i _c1 Computing armature current feedback signal I _feed1 ；

i. According to armature current parameter value I _ref Armature current feedback signal I _feed1 And counter potential E _q Current parameter value I in FPGA _ref Feedback signal I with armature current _feed1 Performing PI operation to output V after difference _1PI Then press D= (V) _1PI +E _q )/V _dc Calculating a duty ratio D and outputting the duty ratio D;

j. according to the advance angles alpha and beta, the encryption angle theta and the duty ratio D, the single-polarity frequency multiplication PWM controlled by alpha+beta is realized through a driving pulse generator module in the FPGA, and a driving signal PWM of the SiC MOS full bridge 1 is generated _11-16 Drive signal PWM _11-16 Q respectively with SiC MOS full bridge 1 _11-16 One-to-one correspondence;

k. generating a driving signal PWM of the SiC MOS full bridge 2 according to the method of the step g-j _21-26 。

In combination with the power generation control operation schematic diagram of fig. 5, the power generation control described in step D includes the following steps:

b. according to DC voltage V _dc And exciting current I _f In DSP, the reference voltage V will be output _ref With DC voltage V _dc After making difference, PI operation is carried out to output exciting current parameter value I _fref In DSP, exciting current parameter value I _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _fref To make the DC voltage V _dc ＝V _ref The power generation and voltage stabilization are realized;

c. a according to armature winding # 1 ₁ Phase current i _a1 Will i in FGPA _a1 And-i _th 、i _th Comparison, i _th For numbers slightly greater than 0, according to i _a1 >i _th Or i _a1 <i _th Make SiC MOS full bridge 1 middle power tube Q ₁₁ Is controlled by the driving signal PWM ₁₁ Set high or low according to i _a1 >-i _th Or i _a1 <-i _th Make SiC MOS full bridge 1 middle power tube Q ₁₂ Is controlled by the driving signal PWM ₁₂ Setting low or high;

d. generating the power tube Q in the SiC MOS full bridge 1 in the manner described in the step c) _13-16 Is controlled by the driving signal PWM _13-16 And SiC MOS full bridge 2 medium power tube Q _21-26 Is controlled by the driving signal PWM _21-26 。

The power generation synchronous rectification control method based on the SiC MOS can overcome the defect that the conduction voltage drop of the Si fast recovery diode cannot be lower than that of a PN junction, so that the conduction loss during power generation is obviously reduced; meanwhile, the reverse recovery problem of the Si fast recovery diode is also solved, so that the reverse recovery loss during power generation is obviously reduced.

In combination with the working schematic diagram of the phase-locked loop encryption module shown in fig. 6, the angle encryption module described in step g in step B includes the following steps:

a) Position signal theta _raw1 And the encryption angle signal theta is 0-2 ^x Representing 0 ° -360 °, each FPGA executing frequency f _clk Triggering the angle encryption module to perform one-time operation;

b) For position signal theta _raw1 The signal is limited after the difference with the encryption angle signal theta to obtain an error signal e _rr ；

c) Error signal e _rr And a frequency compensation signal f _scl Carrying out summation operation to obtain a frequency signal f;

d) Sending the frequency signal f to an (x+y) bit accumulator for accumulation;

e) The binary-represented accumulator value is shifted right by y bits to obtain the encryption angle signal θ.

FIG. 6a shows the original angle signal θ output by the resolver _raw1 The method has the advantages of larger discreteness and lower precision, and the method for encrypting the original output angle of the rotary transformer based on the FPGA can effectively improve the precision of starting control.

Combining the polar frequency multiplication PWM control schematic diagram shown in fig. 7, the unipolar frequency multiplication PWM control working mode diagram shown in fig. 8 and the alpha+beta control working area schematic diagram shown in fig. 9; the driving pulse generator module described in step j in step B includes the following steps:

a) Frequency f of execution per FPGA _clk For triangle waveform counter v _tri Performing one-time increment or one-time decrement to generate triangular waveform in a mode of firstly increasing N and then decreasing to 0 so as to lead the switching period T to be _s Corresponding to 2N FPGA counting periods, the three-waveform counter v _tri The direction signal D is caused to be increased and decreased respectively _ir1 Setting 0 and 1 respectively;

b) According to the output duty ratio D, according to N _ca1 =n/2· (1+d) and N _ca2 Calculate the comparison value N =n/2· (1-D) _ca1 And N _ca2 ；

c) When D is _ir1 When=0, if v _tri ＝N _ca2 Upper tube driving signal G _u1 From 0 to 1, when D _ir1 When=1, if v _tri ＝N _ca2 Or I _shut1 =0, then go up tube driving signal G _u1 From 1 to 0;

d) When D is _ir1 When=1, if v _tri ＝N _ca1 Then down tube driving signal G _d1 From 0 to 1, when D _ir1 When=0, if v _tri ＝N _ca1 Or I _shut1 =0, then the down tube driving signal G _d1 From 0 to 1;

e) According to the encrypting angle signal theta of the No. 1 armature winding, the advance angles alpha and beta, the method comprises the steps of theta= (-beta 0-120-beta 1), (120-alpha-120-beta), (120-beta-240-alpha), (240-alpha 240-beta), (240-beta-360-alpha), and the like,(360-alpha-360-beta) to be an electric frequency f _e The corresponding period is divided into working areas 1-6;

f) According to the working areas 1-6, the upper tube driving signal G _u1 And a down tube driving signal G _d1 In the working area 1, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₆ Is controlled by the driving signal PWM ₁₁ And PWM ₁₆ In the working area 2, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₆ Is controlled by the driving signal PWM ₁₃ And PWM ₁₆ In the working area 3, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₂ Is controlled by the driving signal PWM ₁₃ And PWM ₁₂ In the working area 4, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₂ Is controlled by the driving signal PWM ₁₅ And PWM ₁₂ In the working area 5, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₄ Is controlled by the driving signal PWM ₁₅ And PWM ₁₄ In the working area 6, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₄ Is controlled by the driving signal PWM ₁₁ And PWM ₁₄ ；

g) The driving signals of the power tubes in the SiC MOS full bridge 2 are generated in the manner described in steps a) to f).

As can be seen from FIG. 8, in one control cycle, the upper tube Q _u1 And down tube Q _d1 Respectively acts once, and outputs voltage v relative to the traditional mode of switching the upper pipe or the lower pipe once in one control period _ud1 Is doubled.

Fig. 10 is a diagram showing a comparison of the actuation control method proposed by the present patent and a conventional actuation control method. The method for combining unipolar frequency multiplication PWM with alpha+beta control ensures that the equivalent switching frequency is doubled without increasing the control switching frequency, effectively reduces current ripple of an armature winding, and can better adapt to the winding low-inductance characteristic of the high-speed electro-magnetic doubly-salient motor. Compared with the traditional single peak current control, the starting control adopts a duty ratio control strategy with peak value limitation, and has more stable and reliable control and better electromagnetic compatibility.

Claims

1. The control method of the high-speed doubly salient starting generator controller comprises a generator, a generator controller, a three-phase full-bridge circuit, a phase current comparison circuit, an exciting current control circuit, an FPGA control circuit, a DSP control circuit and an output capacitor C;

the DSP control circuit outputs an excitation control signal PWM _f To the excitation control circuit;

one end of the three-phase full-bridge circuit is connected with the generator armature, and the other end of the three-phase full-bridge circuit is connected with the output capacitor C;

the method is characterized in that:

the control method of the high-speed doubly salient starter generator controller comprises the following steps:

D. converting the output energy of the starting generator into voltage-stabilizing direct current output through power generation control;

the method for determining the phase sequence of the armature winding in the step A is as follows:

the phase sequence of the No. 1 armature is determined in the following way:

a11, direct current exciting current I in any direction is introduced _f ；

A12, setting the three-phase output lines of the No. 1 armature as L respectively ₁₁ 、L ₁₂ 、L ₁₃ Three-phase windings are sequentially set to be A respectively by motor steering ₁ 、B ₁ 、C ₁ Setting L ₁₁ Correspond to A ₁ And assume L ₁₂ 、L ₁₃ Corresponds to B in turn ₁ 、C ₁ ；

A13, externally applied power source leads current to flow from A ₁ End into motor, from C ₁ A motor is output;

a14, external power supply to make current flow from B ₁ The end enters the motor from A ₁ A motor is output;

a15, external power supply to make current flow from C ₁ End into motor, from B ₁ A motor is output;

a16, observing the rotation direction of the motor in the operation of the steps A12 to A14, and if the rotation direction of the motor is the same as the defined positive direction, L ₁₂ Correspond to B ₁ And L ₁₃ Correspond to C ₁ If the rotation direction of the motor is opposite to the defined direction, L ₁₂ Correspond to C ₁ And L ₁₃ Correspond to B ₁ ；

2. The control method of a high-speed doubly salient generator controller as claimed in claim 1, wherein: the phase current comparison circuit comprises a 1# phase circuit comparison circuit and a 2# phase current comparison circuit which are respectively connected with a 1# armature and a 2# armature of the generator;

3. The control method of a high-speed doubly salient generator controller as claimed in claim 1, wherein: the FPGA control circuit adopts alpha+beta controlled unipolar frequency multiplication PWM to drive the three-phase full-bridge circuit.

4. The control method of a high-speed doubly salient generator controller as claimed in claim 1, wherein: the zero offset obtaining step in the step A is as follows:

a21, direct current exciting current I in any direction is introduced _f ；

A22, applying current source to the No. 1 armature and leading the current to flow from C in turn ₁ End entry motor slave B ₁ End and A ₁ Terminating, sequentially recording the readout angle theta of the rotary transformer _f1 、θ _f2 According to the formula delta theta _f |＝|θ _f1 -θ _f2 I, the direct current exciting current I in the step A21 is adjusted _f Is oriented such that delta theta _f |<30°；

A233, will be delta theta ₂₁ ＝θ ₂ -θ ₁ The result is calculated to be within 0-360 DEG, and if delta theta is calculated for the doubly salient motor with two stators aligned with the same teeth and two rotors staggered at equal intervals ₂₁ =180°, the final 2# armature winding a ₂ ＝A ₂ ’、B ₂ ＝B ₂ ' and C ₂ ＝C ₂ ' if delta theta ₂₁ =300°, the final 2# armature winding a ₂ ＝C ₂ ’、B ₂ ＝A ₂ ' and C ₂ ＝B ₂ ' if delta theta ₂₁ =60°, the final 2# armature winding a ₂ ＝B ₂ ’、B ₂ ＝C ₂ ' and C ₂ ＝A ₂ ’；

A24, applying current source to make current flow from C ₁ End into motor, from B ₁ Terminating the motor, and after the motor is stationary, turning on the power A ₁ End and B ₁ The ends are connected in parallel, and a current source is externally added to enable current to flow from C ₁ The end enters the motor from A ₁ And B is connected with ₁ And (3) connecting the motors in parallel to lock the No. 1 armature winding at the 0-degree electric angle position, and recording that the output position signal of the rotary transformer is a zero offset signal delta theta.

5. The control method of a high-speed doubly salient generator controller as claimed in claim 1, wherein: the starting control step in the step B is as follows:

b1, each switching period T in FPGA _s Generating a primary interrupt request signal INT and sending the primary interrupt request signal INT to the DSP so that the DSP executes primary calculation;

b2, reading the position signal theta output by the rotary decoding chip through the DSP _raw And a rotational speed signal n,

b3 by position signal θ _raw Calculating the electrical angle θ of the 1# and 2# armature windings _raw1 And theta _raw2 ，θ _raw1 ＝θ _raw -△θ；

θ _raw2 ＝θ _raw1 +180°；

B4, according to the rotating speed signal n, determining an exciting current parameter value I corresponding to the rotating speed in the DSP program _fref Advance angles alpha and beta, armature current parameter value I _ref Back emf E _q Armature winding electrical frequency f _e ；

B4 according to the exciting current parameter value I _fref And exciting current I _f In DSP, exciting current parameter value I _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _fref ；

b8, according to armature current 1#, i _a1c1 According to I in FPGA _feed1 ＝|i _a1 |+|i _b1 |+|i _c1 Computing armature current feedback signal I _feed1 ；

B9, according to the armature current parameter value I _ref Armature current feedback signal I _feed1 And counter potential E _q Current parameter value I in FPGA _ref Feedback signal I with armature current _feed1 Performing PI operation to output V after difference _1PI Then press D= (V) _1PI +E _q )/V _dc Calculating a duty ratio D and outputting the duty ratio D;

b10, according to the advance angles alpha and beta, the encryption angle theta and the duty ratio D, the single-polarity frequency multiplication PWM controlled by alpha+beta is realized through a driving pulse generator module in the FPGA, and a driving signal PWM of the SiC MOS full bridge 1 is generated _11-16 ；

6. The control method of a high-speed doubly salient generator controller as claimed in claim 1, wherein: the step D is characterized in that the power generation control step comprises the following steps:

d1, each switching period T in FPGA _s Generating a primary interrupt request signal INT and sending the primary interrupt request signal INT to the DSP so that the DSP executes primary calculation;

d2 according to DC voltage V _dc And exciting current I _f In DSP, the reference voltage V will be output _ref With DC voltage V _dc After making difference, PI operation is carried out to output exciting current parameter value I _fref In DSP, exciting current parameter valueI _fref And exciting current I _f After making the difference, PI operation is carried out to output an excitation control signal PWM _f The exciting current control circuit outputs exciting current I _f ＝I _fref To make the DC voltage V _dc ＝V _ref The power generation and voltage stabilization are realized;

7. The control method of the high-speed doubly salient generator controller as claimed in claim 5, wherein: the calculation process of the angle encryption signal θ of the angle encryption module in the step B5 is as follows:

8. The control method of the high-speed doubly salient generator controller as claimed in claim 5, wherein: the step of generating the PWM driving signal of the SiC MOS full bridge 1 by the FPGA driving pulse generator in the step B10 is:

b101, execution frequency f of each FPGA _clk For triangle waveform counter v _tri Performing one-time increment or one-time decrement to generate triangular waveform in a mode of firstly increasing N and then decreasing to 0 so as to lead the switching period T to be _s Corresponding to 2N FPGA counting periods, the three-waveform counter v _tri The direction signal D is caused to be increased and decreased respectively _ir1 Setting 0 and 1 respectively;

b106, according to the working areas 1-6, the upper tube driving signal G _u1 And a down tube driving signal G _d1 In operationZone 1, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₆ Is controlled by the driving signal PWM ₁₁ And PWM ₁₆ In the working area 2, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₆ Is controlled by the driving signal PWM ₁₃ And PWM ₁₆ In the working area 3, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₃ And Q ₁₂ Is controlled by the driving signal PWM ₁₃ And PWM ₁₂ In the working area 4, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₂ Is controlled by the driving signal PWM ₁₅ And PWM ₁₂ In the working area 5, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₅ And Q ₁₄ Is controlled by the driving signal PWM ₁₅ And PWM ₁₄ In the working area 6, G _u1 And G _d1 Respectively as Q in SiC MOS full bridge 1 ₁₁ And Q ₁₄ Is controlled by the driving signal PWM ₁₁ And PWM ₁₄ 。