CN111953434B - IEEE802-11ax signal high-precision demodulation test method - Google Patents
IEEE802-11ax signal high-precision demodulation test method Download PDFInfo
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Abstract
The invention discloses a high-precision demodulation test method for IEEE802-11ax signals, which solves the problem that a signal analyzer with only a single receiving channel cannot acquire MIMO data by utilizing the parallel connection of a plurality of signal analyzers, and simultaneously can carry out multi-path simultaneous demodulation test, multi-path simultaneous capture and multi-path simultaneous analysis on the 802-11ax signals in the MIMO form; in addition, the invention adopts the phase tracking technology to realize the high-precision demodulation of 802-11ax signals, so that the algorithm can be applied to high-precision measuring instruments such as a signal analyzer and the like.
Description
Technical Field
The invention relates to the technical field of high-precision measurement of signal analyzers and the like, in particular to a high-precision demodulation test method for IEEE802-11ax signals.
Background
As an upgrade and improvement of the previous wifi standard IEEE802-11ac, the IEEE802-11ax standard employs 1024QAM high-order modulation, OFDMA technology for multi-user resource allocation, subcarrier spacing reduction and other measures to greatly improve network efficiency and throughput. Signal analyzers are increasingly used as a measuring instrument with integrated measuring functions, and the traditional single spectrum analysis is transited to various complex modulation signal test analysis. The 802-11ax signal demodulation and analysis function is configured in the signal analyzer, so that the radio frequency consistency test, the fault diagnosis and the like of the 802-11ax signal transmitting equipment can be realized. As the signal analyzer only has one radio frequency receiving channel, the existing 802-11ax signal demodulation technology only demodulates one path of 802-11ax signal, and the technical scheme is as shown in figure 1, firstly, data acquisition is carried out, then, a training sequence is used for synchronization, then, frequency offset is eliminated, then, long training symbols of a leader sequence are used for channel estimation to obtain amplitude-frequency response of a channel, the amplitude-frequency response is used for carrying out uniform balance on a signal domain, data part modulation information is obtained by analyzing the signal domain, and the data part is balanced by using the modulation information and combining the channel response, so that the demodulation of the data part can be realized. The prior art has the defects that single-path data is captured, and only one path of data can be analyzed; secondly, the realization cost is high and the program transportability is poor because a programmable logic device is mostly adopted for realization; and thirdly, the demodulation precision is not high, and the method is difficult to be applied to high-precision measuring instruments such as signal analyzers.
Disclosure of Invention
The invention mainly aims to design an 802-11ax signal demodulation test scheme, which can demodulate and analyze MIMO-OFDM signals based on 802-11ax protocol standards, has high demodulation precision, is realized by pure software, has low realization cost, is suitable for being configured on high-precision measuring instruments such as a signal analyzer and the like, and can meet the requirements of modulation characteristic test analysis, auxiliary fault diagnosis and the like of 802.11ax signal transmitting equipment.
The technical scheme of the invention is as follows: a high-precision demodulation test method for IEEE802-11ax signals specifically comprises the following steps:
step 1: capturing data; the method is realized by adopting a mode of connecting a plurality of signal analyzers in parallel, one of the signal analyzers is a main signal analyzer and is used for analyzing captured data, other accessory signal analyzers are only responsible for data acquisition, and the 802-11ax signal demodulation and analysis function is only deployed in the main signal analyzer; the main signal analyzer trigger output is connected with the auxiliary signal analyzer trigger input to ensure that data acquisition is carried out among a plurality of signal analyzers simultaneously, and meanwhile, the main signal analyzer is connected with other signal analyzers through a network cable to realize data transmission from the auxiliary signal analyzer to the main signal analyzer;
step 2: searching for pulses; firstly, searching the maximum value and the minimum value of the amplitude of the acquired data, and determining the range of the pulse amplitude according to the maximum value and the minimum value; dividing the range of pulse amplitude at equal intervals, counting the probability that the pulse amplitude falls into each amplitude interval, wherein the two intervals with the highest probability are the pulse bottom and the pulse top respectively, and automatically setting a pulse detection threshold value by utilizing the pulse bottom and the pulse top so as to realize the extraction of the pulse;
and step 3: sampling rate conversion; carrying out sampling rate conversion according to different signal bandwidth types to convert the signals to 802.11ax specific sampling rate;
and 4, step 4: frame synchronization; the conventional short training sequence circular correlation operation is adopted for synchronization, the window length is half of the short training sequence, and only one correlation result appearsPeak, setting the received signal as r, sliding window length as L, and frame synchronization detection of the received signal by conjugating multiplication and accumulation modulo of delayed D sampling values; cnThe cross-correlation coefficient of the received signal and the delay thereof at the time point of n is calculated, and is shown as the formula (1):
where r denotes the received signal, r*Denotes the conjugate of r, i denotes the cyclic variable; pnRepresenting the energy of the received signal during the window when the cross-correlation coefficient is calculated shifted to time n, as shown in equation (2), for normalization of the decision statistics;
and finally, carrying out statistical judgment through a formula (3):
wherein M isnRepresenting a normalized value for the decision statistic;
and 5: carrying out carrier synchronization; let Δ f be the residual frequency offset of the receiver and the transmitter, and the correlation operation of the conventional long training sequence after frame synchronization is as shown in formula (4):
where z is the correlation value of the received training sequence, rnIndicating that a sample at time n of the training sequence was received,
representing the conjugate of the sample at time n + D of the received training sequence, LLSIndicates the conventional long training sequence length, SnRepresenting ideal trainingSequence of n time samples, Sn+DRepresenting the sample at time n + D of an ideal training sequence, TsIn order to be the sampling period of time,
denotes Sn+DConjugation of (1);
the frequency offset estimation of equation (4) is, equation (5):
wherein,
representing a frequency offset estimate;
step 6: OFDM demodulation; firstly, removing guard intervals among OFDM symbols, and then realizing demodulation of the OFDM symbols through FFT;
and 7: estimating a conventional preamble channel; the long training symbol of the conventional leader sequence is used for channel estimation of single-path data, and the conventional long training sequence R is received at the k number subcarrierLTF,kExpressed as, equation (6):
RLTF,k＝LTFk·Hk+Wk (6)
wherein HkIndicating the channel response, LTF, of sub-carriers kkFor the long training symbol, W, for sub-carrier kkRepresenting the noise of k subcarriers after the FFT transformation of the long training symbol; the channel response estimate at the k-sub-carrier
To, formula (7):
and 8: conventional leading channel equalization; let a be the sampling value of the symbol received on the k number subcarrierkThen balance the result
In order, equation (8):
and step 9: analyzing a signal domain; acquiring modulation parameters of the data part from a signal domain before demodulating the data part; decoding the signal domain through the steps of demapping, deinterleaving and deconvolution, and recovering relevant modulation parameters of the data part by contrasting with a protocol;
step 10: efficient preamble channel estimation; two spatial streams and two receiving channels, and the multi-channel reception of the high-efficiency pilot length training sequence is expressed as formula (9):
wherein
Indicating that the receiving channel 1 is at t1The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 1 is at t2The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 2 is at t1The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 2 is at t2Value at subcarrier k of HELTF received at a time, HELTFkFor transmitting the value, Delta, at k number of subcarriers of the efficient preamble training sequenceFWhich indicates the spacing between the sub-carriers,
representing the cyclic shift value of the transmit path 1,
representing the cyclic shift value of the transmit path 2,
representing the channel response of the transmit antenna 1 to the receive antenna 1,
representing the channel response from transmit antenna 2 to receive antenna 1,
representing the channel response of transmit antenna 1 to receive antenna 2,
representing the channel response from the transmitting antenna 2 to the receiving antenna 2, the channel response at the k number of sub-carriers obtained by solving the formula (9) is shown as the formula (10):
wherein
Are respectively as
An estimated value of (d);
step 11: data part channel equalization; the data portion reception is expressed as, equation (11):
wherein r is1 kIndicating that the 1 st received data is at k number sub-carrierThe value of (a) is,
indicating the value of the received data of path 2 at subcarrier k,
represents the value of the 1 st transmission data at the k number sub-carrier,
indicating the value of the 2 nd transmitted data at sub-carrier k, as specified in the standard
Will estimate the value
Substituting formula (11) and solving formula (11) to obtain formula (12) and formula (13):
wherein
The values of the 1 st path data and the 2 nd path data at the k number sub-carrier after equalization of the received signal, namely the transmitted data
And
an estimated value of (d);
step 12: phase tracking; phase of received signal at k subcarriers of symbol l after FFT demodulation and channel equalization
Expressed as, formula (14):
wherein
In order to transmit the ideal phase of the data,
common to all sub-carriers at symbol l,
for phase drift of timing at k subcarriers of symbol l, equation (15) and equation (16):
Δfrestthe residual frequency deviation after the coarse frequency deviation compensation is shown as xi is the clock deviation of the crystal oscillator, and d gamma islIs the phase jitter of the ith OFDM symbol, T represents the OFDM symbol period; pilot signal for phase tracking using pilot signal
For known parameters, a cost function is established by combining the pilot subcarriers of all OFDM symbols of the data part according to formula (14), and formula (17):
wherein nof _ symbols indexAccording to the number of partial OFDM symbols, pilot _ sub guides frequency subcarriers, the pilot symbols are substituted into the cost function, and delta f is obtained by utilizing a maximum likelihood estimation methodrest、ξ、 dγlIs estimated value of
Substituting the estimated value into a formula (14) to correct the phase of the demodulation signal to obtain a measurement signal Meas;
step 13: generating a reference signal; the measurement signal obtained by demodulation is used for judging an ideal constellation point to generate an ideal signal, namely a reference signal;
step 14: outputting an error parameter; substituting the measurement signal and the reference signal into a corresponding error parameter calculation formula to obtain parameters such as error vector amplitude, amplitude error, phase error and the like, wherein the specific calculation method is shown in formulas (18), (19) and (20), wherein Evm represents the error vector amplitude, amplierr represents the amplitude error, PhaseErr represents the phase error, Imeas represents a measurement signal path I, Qmeas represents a measurement signal path Q, Iref represents a reference signal path I, Qref represents a reference signal path Q, and arg represents phase taking; at this point, the modulation characteristic measurements for the 802-11ax signals are completed.
PhaseErr＝arg(Qmeas,Imeas)-arg(Qref,Iref) (20)
In the above, the step 1 further includes setting an IP address of the auxiliary signal analyzer at the end of the main signal analyzer, and implementing program control on the auxiliary signal analyzer through the program control command, thereby implementing simultaneous capture of multiple channels of data by the main signal analyzer.
Compared with the prior art, the invention adopts an 802-11ax signal capturing method, solves the problem that the signal analyzer with only a single receiving channel can not acquire MIMO data by utilizing the parallel connection of a plurality of signal analyzers, and simultaneously can carry out multichannel simultaneous demodulation test, multichannel simultaneous capturing and multichannel simultaneous analysis on the 802-11ax signals in the MIMO form; in addition, the invention adopts the phase tracking technology to realize the high-precision demodulation of 802-11ax signals, so that the algorithm can be applied to high-precision measuring instruments such as a signal analyzer and the like.
Drawings
FIG. 1 is a schematic diagram of a prior art signal demodulation scheme
FIG. 2 is a schematic diagram of the high-precision demodulation test of 802-11ax signals according to the present invention.
Fig. 3 is a schematic diagram of MIMO data capture for a signal analyzer platform according to the present invention.
FIG. 4 is a diagram illustrating the simulation effect of the conventional short training sequence circular correlation operation on the synchronization signal in the embodiment of the present invention.
Detailed Description
In order to facilitate an understanding of the invention, the invention is described in more detail below with reference to the accompanying drawings and specific examples. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. The terminology used in the description of the invention herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. As used herein, the term "and/or" includes any and all combinations of one or more of the associated listed items.
An embodiment of the invention provides a high-precision demodulation test method for 802-11ax signals, which can realize the demodulation test of 802-11ax signals in an MIMO form by using a signal analyzer with only one receiving channel, the demodulation test scheme of the invention has the principle as shown in figure 2, a radio frequency signal is input into the signal analyzer, the signal analyzer is used for acquisition and capture, then pulse search is carried out through a software algorithm, a frame of complete 802.11ax signals is extracted, the acquired signals are resampled according to the 802.11ax signal bandwidth, the preparation work of demodulation analysis is completed at the moment, then frame synchronization is carried out to determine the initial position of the frame, then carrier estimation is carried out to eliminate residual frequency offset, and then conventional leading channel user frequency offset and equalization are carried out, so as to realize the analysis of a high-efficiency leading signal domain, parameters required by subsequent demodulation of the modulation type, the number of symbols, resource allocation and the like of the data part can be obtained by analyzing the high-efficiency preamble signal field, then channel estimation and equalization of an MIMO channel are carried out to obtain a constellation diagram, a demodulation measurement signal is obtained by correcting the phase of the demodulation signal through phase tracking to realize high-precision demodulation, finally an ideal reference signal is generated from the measurement signal, and the measurement signal and the reference signal are compared to realize analysis of modulation errors. This is explained in detail below.
Step 1: capturing data; when 802-11ax transmits using multiple antennas, its data reception also needs to be implemented using multiple antennas. However, one signal analyzer only includes one receiving channel, and in order to solve the receiving and testing of the MIMO-OFDM signal, a plurality of signal analyzers may be implemented in parallel, and the specific connection is as shown in fig. 3, one of the plurality of signal analyzers is a main signal analyzer for analyzing the captured data, and the other accessory signal analyzers are only responsible for data acquisition, so that the 802-11ax signal demodulation and analysis function only needs to be deployed in the main signal analyzer. The main signal analyzer trigger output is connected with the auxiliary signal analyzer trigger input, so that data acquisition can be simultaneously carried out among a plurality of signal analyzers, and meanwhile, the main signal analyzer is connected with other signal analyzers through a network cable, so that data transmission from the auxiliary signal analyzer to the main signal analyzer is realized. And setting an IP address of the auxiliary signal analyzer at the main signal analyzer end, and realizing program control on the auxiliary signal analyzer through a program control command, thereby realizing simultaneous capture of multi-channel data by the main signal analyzer.
Step 2: searching for pulses; since the 802-11ax signal is in the form of bursts, a pulse acquisition operation is performed first before it is demodulated, and a complete pulse, i.e., a complete frame, is extracted. The pulse search can roughly judge the beginning and the end of the signals of one frame 802-11ax, so that the search overhead of a correlation algorithm in the synchronization of subsequent frames can be effectively saved. The pulse search may be implemented using statistical methods. Firstly, searching the maximum value and the minimum value of the amplitude of the acquired data, and determining the range of the pulse amplitude according to the maximum value and the minimum value; the pulse amplitude range is divided at equal intervals, the probability that the pulse amplitude falls into each amplitude interval is counted, the two intervals with the maximum probability are the pulse bottom and the pulse top respectively, and the pulse detection threshold can be automatically set by utilizing the pulse bottom and the pulse top, so that the extraction of the pulse is realized.
And step 3: sampling rate conversion; the 802-11ax signal has a fixed bandwidth and should be sample rate converted to a sample rate specific to 802.11ax according to different signal bandwidth types before being demodulated and analyzed.
And 4, step 4: frame synchronization; the purpose of frame synchronization is to find the start of a frame 802-11ax signal, and only the start position of the frame is determined, so that the position of each OFDM symbol in a frame can be determined. The 802-11ax signal is compatible with the 802-11a// n standard and comprises a conventional preamble and an efficient preamble of its own. Training sequences can be used for synchronization, but different training sequences and different window lengths have different synchronization effects, for example, when conventional long training sequences are used for synchronization, a platform phenomenon may exist in a related result, which is not favorable for extraction of a synchronization position. If the efficient preamble short training sequence is adopted, the extraction of the synchronous position is also influenced due to the uncertain position of the efficient preamble short training sequence. If a conventional short training sequence is used for synchronization, multiple peak values will appear when the window length is selected to be inappropriate for correlation operation, and the extraction of the synchronization position will be affected. The invention adopts the conventional short training sequence cyclic correlation operation for synchronization, the window length is half of the short training sequence, only one peak appears in the correlation result, and the simulation effect is as shown in figure 4.
Setting the received signal as r and the sliding window length as L, and performing frame synchronization detection on the received signal by conjugating, multiplying, accumulating and modulus-taking the delayed D sampling values;
Cnrepresents the calculation of the cross-correlation coefficient of the received signal and its delay at time n, as shown in equation (1):
where r denotes the received signal, r*Denotes the conjugate of r, i denotes the cyclic variable;
Pnrepresenting the energy of the received signal during the window when the cross-correlation coefficients are calculated shifted to time n, as shown in equation (2), for normalization of the decision statistics;
and finally, carrying out statistical judgment through a formula (3):
wherein M isnRepresenting a normalized value for the decision statistic;
and 5: and (5) carrying wave synchronization.
After frame synchronization, the position of the regular long training sequence can be determined. The frequency offset is caused by the factor of instability of the first-time double-frequency local oscillation or Doppler frequency shift, and the orthogonality among the subcarriers of the OFDM symbols can be influenced if the frequency offset is not eliminated. Let Δ f be the residual frequency offset of the receiver and the transmitter, and the correlation operation of the conventional long training sequence after frame synchronization is as shown in formula (4):
where z is the correlation value of the received training sequence, rnRepresenting the sample value at time n of the received training sequence,
representing the conjugate of the sample at time n + D of the received training sequence, LLSRepresents the length of the conventional long training sequence, SnRepresenting samples of the ideal training sequence at time n, Sn+DRepresenting the time sample value, T, of the ideal training sequence n + DsIs a time period of the sampling, and,
denotes Sn+DConjugation of (1);
the frequency offset estimation of equation (4) is, equation (5):
wherein,
representing a frequency offset estimate
Step 6: and OFDM demodulation. The OFDM symbols can be demodulated after carrier frequency offset is eliminated, firstly, the guard interval between the OFDM symbols is removed, and then the demodulation of the OFDM symbols can be realized through FFT.
And 7: and (4) estimating a conventional preamble channel. The long training symbols of the conventional preamble sequence may be used for channel estimation for single-channel data. Receiving conventional long training sequence R at k number subcarrierLTF,kCan be expressed as, formula (6):
RLTF,k＝LTFk·Hk+Wk (6)
wherein HkIndicating the channel response, LTF, of sub-carriers kkFor the length of the training symbol at sub-carrier number k, WkRepresenting the noise at the k sub-carrier after the FFT transformation of the long training symbol. Then the channel response estimate at sub-carrier k
To, formula (7):
and 8: conventional preamble channel equalization.
The channel response obtained by using the long training symbol of the conventional preamble sequence can be used for equalizing only one-way data, so that the channel response can be used for equalizing the signal field of the conventional preamble and the signal field of the efficient preamble. Under the condition that the channel characteristics are known, the channel equalization can realize equalization by frequency domain division;
let a be the sampling value of the symbol received on the k number subcarrierkThen balance the result
In order, equation (8):
and step 9: and (4) signal domain analysis.
The signal domain includes a normal preamble domain and an efficient preamble domain. The high-efficiency preamble signal field contains information such as modulation type, symbol number, resource allocation and the like of the data part. The modulation parameters of the data part should be obtained from the signal domain first before the demodulation of the data part. The decoding of the signal domain can be realized through the steps of demapping, deinterleaving, deconvolution and the like, and the relevant modulation parameters of the data part can be recovered according to the protocol.
Step 10: efficient preamble channel estimation. When the 802.11ax signal is communicated by adopting the MIMO system, the frame structure data part of any path of receiving signal is the mixture of the data parts of each path of transmitting signal, so the channel response obtained by the long training symbol cannot equalize the part. For this purpose, the 802.11ax standard specifies that high efficiency preamble Length Training Sequences (HELTFs) can be used for channel estimation of MIMO data. Taking two spatial streams and two receiving channels as an example, the multipath reception of the high-efficiency preamble training sequence can be expressed as formula (9):
wherein
Indicating that the receiving channel 1 is at t1The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 1 is at t2The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 2 is at t1The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 2 is at t2Value at subcarrier k of HELTF received at a time, HELTFkFor transmitting the value, Delta, at k number of subcarriers of the efficient preamble training sequenceFWhich indicates the spacing between the sub-carriers,
indicating the cyclic shift value of the transmit path 1,
indicating the cyclic shift value of the transmit path 2,
representing the channel response of the transmit antenna 1 to the receive antenna 1,
representing the channel response from transmit antenna 2 to receive antenna 1,
representing the channel response of transmit antenna 1 to receive antenna 2,
indicating the channel response from the transmit antenna 2 to the receive antenna 2. The channel response at the k subcarrier can be obtained by solving the above equation as shown in formula (10):
wherein
Are respectively as
An estimate of (d).
Step 11: and equalizing the data part channel. Since the data portion is a mixture of the data portions of the respective transmit signals, equalization of the data portions needs to be handled in conjunction with the channel responses of the respective channels. The data portion reception situation can be expressed as equation (11):
wherein r is1 kIndicating the value of the 1 st received data at subcarrier k,
indicating the value of the received data of path 2 at subcarrier k,
represents the value of the 1 st transmission data at the k number sub-carrier,
indicating the value of the 2 nd transmitted data at sub-carrier k, as specified in the standard
Will estimate the value
Substituting formula (11) into formula (11) and solving formula (11) to obtain formula(12) And formula (13):
wherein
The values of the 1 st path data and the 2 nd path data at the k number sub-carrier after equalization of the received signal, namely the transmitted data
And
an estimate of (d).
Step 12: and (4) phase tracking. When burst OFDM signal demodulation is carried out, the equalization response calculated by utilizing the preamble training sequence is often not ideal enough, and particularly, along with the increase of OFDM pulse, the equalization effect of the equalization response obtained by utilizing the preamble is reduced, so that extra measures are needed to be adopted for improving demodulation quality of a signal analyzer with high demodulation quality.
Phase of received signal at k subcarriers of symbol l after FFT demodulation and channel equalization
Can be expressed as, formula (14):
wherein
In order to transmit the ideal phase of the data,
common phase drift for all subcarriers at symbol l,
for phase drift of timing at k subcarriers of symbol l, equation (15) and equation (16):
Δfrestthe residual frequency deviation after the coarse frequency deviation compensation is shown as xi is the clock deviation of the crystal oscillator, and d gamma islIs the phase jitter of the ith OFDM symbol, T represents the OFDM symbol period. Since the transmitted pilot signal is a known signal, when phase tracking is performed using the pilot signal
For known parameters, a cost function can be established by combining the pilot subcarriers of all OFDM symbols of the data part by equation (14), equation (17):
nof _ symbols indicates the number of OFDM symbols in data part, pilot _ sub indicates frequency subcarrier, pilot symbols are substituted into the cost function, and Δ f is obtained by maximum likelihood estimation methodrest、ξ、dγlIs estimated value of
The phase of the demodulated signal can be corrected by substituting the estimated value into the formula (14), and the measurement signal Meas can be obtained.
Step 13: and generating a reference signal. The above steps complete the entire demodulation operation, and if the demodulation operation is applied to a measurement instrument such as a signal analyzer, which is called a modulation measurement function, error parameter calculation related to modulation characteristics is also required. The measurement signal obtained by demodulation is used for judging an ideal constellation point to generate an ideal signal, namely a reference signal Ref.
Step 14: and outputting the error parameters. After the measurement signal and the reference signal are obtained, the two are substituted into a corresponding error parameter calculation formula to obtain parameters such as error vector amplitude, amplitude error, phase error and the like, and the specific calculation method is shown in formulas (18), (19) and (20), wherein Evm represents error vector amplitude, amplierr represents amplitude error, PhaseErr represents phase error, Imeas represents a measurement signal path I, Qmeas represents a measurement signal path Q, Iref represents a reference signal path I, Qref represents a reference signal path Q, and arg represents phase taking. At this point, the modulation characteristic measurements for the 802-11ax signals are completed.
PhaseErr＝arg(Qmeas,Imeas)-arg(Qref,Iref) (20)
Compared with the prior art, the invention adopts an 802-11ax signal capturing method, solves the problem that the signal analyzer with only a single receiving channel can not acquire MIMO data by utilizing the parallel connection of a plurality of signal analyzers, and simultaneously can carry out multichannel simultaneous demodulation test, multichannel simultaneous capturing and multichannel simultaneous analysis on the 802-11ax signals in the MIMO form; in addition, the invention adopts the phase tracking technology to realize the high-precision demodulation of 802-11ax signals, so that the algorithm can be applied to high-precision measuring instruments such as a signal analyzer and the like.
The technical features mentioned above are combined with each other to form various embodiments which are not listed above, and all of them are regarded as the scope of the present invention described in the specification; also, modifications and variations may be suggested to those skilled in the art in light of the above teachings, and it is intended to cover all such modifications and variations as fall within the true spirit and scope of the invention as defined by the appended claims.
Claims (2)
1. A high-precision demodulation test method for IEEE802-11ax signals is characterized by comprising the following steps:
step 1: capturing data; the method is realized by adopting a mode of connecting a plurality of signal analyzers in parallel, wherein one of the plurality of signal analyzers is a main signal analyzer and is used for analyzing captured data, other accessory signal analyzers are only responsible for data acquisition, and the 802-11ax signal demodulation and analysis function is only deployed in the main signal analyzer; the main signal analyzer trigger output is connected with the auxiliary signal analyzer trigger input to ensure that data acquisition is carried out among a plurality of signal analyzers simultaneously, and meanwhile, the main signal analyzer is connected with other signal analyzers through a network cable to realize data transmission from the auxiliary signal analyzer to the main signal analyzer;
step 2: searching for pulses; firstly, searching the maximum value and the minimum value of the amplitude of the acquired data, and determining the range of the pulse amplitude according to the maximum value and the minimum value; dividing the range of pulse amplitude at equal intervals, counting the probability that the pulse amplitude falls into each amplitude interval, wherein the two intervals with the highest probability are respectively the pulse bottom and the pulse top, and automatically setting a pulse detection threshold value by utilizing the pulse bottom and the pulse top so as to realize the extraction of the pulse;
and step 3: sampling rate conversion; carrying out sampling rate conversion according to different signal bandwidth types to convert the signals to 802.11ax specific sampling rate;
and 4, step 4: frame synchronization; performing synchronization by adopting conventional short training sequence cyclic correlation operation, wherein the window length is half of the short training sequence, only one peak appears in a correlation result, a received signal is set as r, the length of a sliding window is set as L, and the received signal is subjected to frame synchronization detection by performing conjugate multiplication and accumulation modulo multiplication on a sampling value delayed by the received signal by D sampling values; cnRepresents the calculation of the cross-correlation coefficient of the received signal and its delay at time n, as shown in equation (1):
where r denotes the received signal, r*Denotes the conjugate of r, i denotes the cyclic variable; pnRepresenting the energy of the received signal during the window when the cross-correlation coefficient is calculated shifted to time n, as shown in equation (2), for normalization of the decision statistics;
and finally, carrying out statistical judgment through a formula (3):
wherein M isnRepresenting a normalized value for the decision statistic;
and 5: carrying out carrier synchronization; let Δ f be the residual frequency offset of the receiver and the transmitter, and the correlation operation of the conventional long training sequence after frame synchronization is as shown in formula (4):
where z is the correlation value of the received training sequence, rnIndicating that a sample at time n of the training sequence was received,
representing the conjugate of the sample at time n + D of the received training sequence, LLSRepresents the length of the conventional long training sequence, SnRepresenting samples of the ideal training sequence at time n, Sn+DRepresenting the sample at time n + D of the ideal training sequence, TsIn order to be the sampling period of time,
denotes Sn+DConjugation of (1);
the frequency offset estimation of equation (4) is, equation (5):
wherein,
representing a frequency offset estimate;
step 6: OFDM demodulation; firstly, removing guard intervals among OFDM symbols, and then realizing demodulation of the OFDM symbols through FFT;
and 7: estimating a conventional preamble channel; the long training symbol of the conventional leader sequence is used for channel estimation of single-path data, and the conventional long training sequence R is received at the k number subcarrierLTF,kExpressed as, equation (6):
RLTF,k＝LTFk·Hk+Wk (6)
wherein HkIndicating the channel response, LTF, of sub-carriers kkFor the length of the training symbol at sub-carrier number k, WkRepresenting the noise of k subcarriers after the FFT transformation of the long training symbol; then the channel response estimate at sub-carrier k
To, formula (7):
and 8: conventional leading channel equalization; let a be the sampling value of the symbol received on the k number subcarrierkThen balance the result
In order, equation (8):
and step 9: analyzing a signal domain; the modulation parameters of the data part are acquired by the signal domain before the demodulation of the data part is carried out; decoding the signal domain through the steps of demapping, deinterleaving and deconvolution, and recovering relevant modulation parameters of the data part by contrasting with a protocol;
step 10: efficient preamble channel estimation; two spatial streams and two receiving channels, and the multipath reception of the high-efficiency preamble long training sequence is expressed as formula (9):
wherein
Indicating that the receiving channel 1 is at t1The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 1 is at t2The value at subcarrier k of the HELTF received at time instant,
indicating that the receiving channel 2 is at t1The value at subcarrier k of the HELTF received at that time,
indicating that the receiving channel 2 is at t2Value at subcarrier k of HELTF received at a time, HELTFkTo send the value at k number subcarrier of the efficient preamble long training sequence, ΔFWhich indicates the spacing between the sub-carriers,
representing the cyclic shift value of the transmit path 1,
representing the cyclic shift value of the transmit path 2,
representing the channel response of the transmit antenna 1 to the receive antenna 1,
representing the channel response from transmit antenna 2 to receive antenna 1,
representing the channel response of transmit antenna 1 to receive antenna 2,
representing the channel response from the transmitting antenna 2 to the receiving antenna 2, the channel response at the k number of sub-carriers obtained by solving the formula (9) is shown as the formula (10):
wherein
Are respectively as
An estimated value of (d);
step 11: data part channel equalization; the data portion reception is expressed as, equation (11):
wherein r is1 kIndicating the value of the 1 st received data at subcarrier k,
indicating the value of the received data of path 2 at subcarrier k,
indicating the value of the 1 st transmitted data at the k number sub-carrier,
indicating the value of the 2 nd transmitted data at sub-carrier k, as specified in the standard
Will estimate the value
Substituting formula (11) and solving formula (11) to obtain formula (12) and formula (13):
wherein
The values of the 1 st path data and the 2 nd path data at the k number sub-carrier after equalization of the received signal, namely the transmitted data
And
an estimated value of (d);
step 12: phase (C)Bit tracking; phase of received signal at k subcarriers of symbol l after FFT demodulation and channel equalization
Expressed as, formula (14):
wherein
In order to transmit the ideal phase of the data,
common phase drift for all subcarriers at symbol l,
for phase drift of timing at k subcarriers of symbol l, equation (15) and equation (16):
△frestthe residual frequency deviation after the coarse frequency deviation compensation is shown as xi is the clock deviation of the crystal oscillator, and d gamma islIs the phase jitter of the ith OFDM symbol, T represents the OFDM symbol period; pilot signal for phase tracking using pilot signal
For known parameters, a cost function is established by combining the pilot subcarriers of all OFDM symbols of the data part by equation (14), equation (17):
nof _ symbols indicates the number of OFDM symbols in data part, pilot _ sub indicates frequency subcarrier, pilot symbols are substituted into the cost function, and the maximum likelihood estimation method is used to obtain delta frest、ξ、dγlIs estimated value of
Substituting the estimated value into a formula (14) to correct the phase of the demodulation signal to obtain a measurement signal Meas;
step 13: generating a reference signal; the measurement signal obtained by demodulation is used for judging an ideal constellation point to generate an ideal signal, namely a reference signal;
step 14: outputting an error parameter; substituting the measurement signal and the reference signal into a corresponding error parameter calculation formula to obtain parameters such as error vector amplitude, amplitude error, phase error and the like, wherein the specific calculation method is shown in formulas (18), (19) and (20), wherein Evm represents the error vector amplitude, Amplerr represents the amplitude error, PhaseErr represents the phase error, Imeas represents a measurement signal path I, Qmeas represents a measurement signal path Q, Iref represents a reference signal path I, Qref represents a reference signal path Q, and arg represents phase taking; at this time, the modulation characteristic measurement of 802-11ax signals is completed;
PhaseErr＝arg(Qmeas,Imeas)-arg(Qref,Iref) (20)。
2. the debugging and testing method of claim 1, wherein the step 1 further comprises setting an IP address of the auxiliary signal analyzer at the end of the main signal analyzer, and implementing program control on the auxiliary signal analyzer through a program control command, thereby implementing simultaneous capture of multiple channels of data by the main signal analyzer.
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