CN111934589B - Traction motor speed-sensorless control method based on improved q-type phase-locked loop - Google Patents

Traction motor speed-sensorless control method based on improved q-type phase-locked loop Download PDF

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CN111934589B
CN111934589B CN202010830257.5A CN202010830257A CN111934589B CN 111934589 B CN111934589 B CN 111934589B CN 202010830257 A CN202010830257 A CN 202010830257A CN 111934589 B CN111934589 B CN 111934589B
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speed
rotor
flux linkage
pll
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CN111934589A (en
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葛兴来
左运
冯晓云
宋文胜
苟斌
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Southwest Jiaotong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/26—Rotor flux based control

Abstract

The invention discloses a traction motor speed-sensorless control method based on an improved q-type phase-locked loop, which specifically comprises the following steps: 1. constructing a closed-loop rotor flux linkage observer to observe the rotor flux linkage; 2. building an improved q-type phase-locked loop; 3. inputting the detected rotor flux linkage into an improved q-type phase-locked loop to obtain the motor rotating speed and the rotor phase angle; 4. and inputting the calculated motor rotation speed and rotor phase angle into a traction motor vector control system to complete speed sensorless control of the traction motor. The invention can effectively make up the speed identification error and rotor phase angle identification error of the traditional q-type phase-locked loop when the traction motor rises and falls; the order of the system is not increased, so that the dynamic performance and stability margin of the system are not adversely affected; the invention can realize good estimation performance under multiple working conditions, has simple structure and is easy to realize.

Description

Traction motor speed-sensorless control method based on improved q-type phase-locked loop
Technical Field
The invention belongs to the technical field of electric traction alternating current transmission, and relates to a speed estimation method in a traction motor speed sensorless control system; in particular to a traction motor speed-sensorless control method based on an improved q-type phase-locked loop.
Background
With the development of economic construction and the continuous promotion of urban mass transit, the total mileage of urban mass transit construction in China is in the front of the world, and meanwhile, the traction transmission system serving as the core of the railway vehicle is receiving more and more attention. In urban rail train traction transmission systems, mechanical speed sensors are generally used for speed detection to realize closed-loop control, but the existence of the speed sensors can significantly increase system cost, reduce system reliability, and are susceptible to problems such as assembly, signal transmission, environmental condition limitation and the like. To improve the reliability of traction drive systems, sensorless technology of induction motors is of great interest.
For the sensorless technology, two general categories are available, namely, a method based on high-frequency signal injection and rotor slot harmonic extraction and a method based on a motor model. The former, while being able to identify rotational speed more accurately and independently of motor parameter variations, increases system complexity and therefore speed estimation schemes based on motor models are more widely used.
The speed estimation scheme based on the motor model is a popular speed-sensor-free control technology due to the characteristics of easiness in implementation, good universality and the like. Including model reference adaptation methods, adaptive full-order state observers, extended kalman filters, sliding mode observers, etc.
Another interesting approach in motor model based speed estimation schemes is the phase-locked loop (PLL) based speed identification approach. The PLL is used as an important link in a grid-connected system, and can realize the synchronization of the amplitude, the frequency and the phase of an output signal and an input signal. Based on this, PLL is extended to a speed sensorless control system and is favored for its simple structure and good performance. Among these, a q-type PLL (q-type PLL) based speed estimation scheme is attracting attention, but q-PLL generally employs a PI regulator as a loop filter, based on which a conventional PLL based speed estimation scheme can be simplified to a type 2 control system. And the type 2 control system cannot realize accurate tracking of the input signal in the frequency slope. This will result in a significant identification error in the conventional q-PLL based speed estimation scheme when the motor is running at a ramp up and down speed (at which time the frequency is ramped).
Disclosure of Invention
In order to solve the problems, the invention provides a traction motor speed sensorless control method based on an improved q-type phase-locked loop.
The invention relates to a traction motor speed-sensorless control method based on an improved q-type phase-locked loop, which comprises the following steps of:
step 1: and constructing a closed-loop rotor flux linkage observer to observe the rotor flux linkage.
The traditional voltage type flux linkage observer is an open loop observation model in nature, is easily influenced by stator resistance and an integrator, has poor robustness and is obvious at low speed; the current model is also an open loop observation model, and the model contains more rotor side motor parameters, so that the robustness is poor. Therefore, the invention adopts a closed-loop rotor flux linkage observer, combines the two, adopts a current model with more obvious advantages at low speed, adopts a voltage model with more obvious advantages at high speed, and compensates errors caused by integration and stator resistance voltage drop.
Step 2: and constructing an improved q-type phase-locked loop.
In a two-phase fixed coordinate system, when the imaginary part u is to be output from the phase detector dq (t) when used as a feedback error signal, a q-PLL is obtained, and:
Figure GDA0004218826350000021
wherein u is 1α ，u 1β In order to input a signal to the device,
Figure GDA0004218826350000022
u 2α ，u 2β for outputting signals +.>
Figure GDA0004218826350000023
U 1 ，ω 1 ，/>
Figure GDA0004218826350000024
For the amplitude, angular frequency and initial phase of the input signal, U 2 ，ω 2 ，/>
Figure GDA0004218826350000025
For the amplitude, angular frequency and initial phase of the output signal.
Real part of phase detector outputu dp (t) is:
Figure GDA0004218826350000026
the q-PLL scheme has an estimation error when tracking the frequency ramp signal; if the input is considered as an ideal signal with normalized amplitude, the closed loop transfer function of the q-PLL is expressed as follows:
Figure GDA0004218826350000027
wherein k is p ，k i Is the loop filter gain.
The error transfer function is obtained from equation (3):
Figure GDA0004218826350000028
when the input signals are respectively frequency-ramped:
Figure GDA0004218826350000029
wherein: θ i3 (s) is the transfer function of the different input signals, k is the gain of the frequency ramp.
Further, the error transfer function is:
Figure GDA00042188263500000210
wherein:
Figure GDA0004218826350000031
is the phase error transfer function at the frequency ramp input signal.
And obtaining steady-state errors under different input signals by using a final value theorem:
Figure GDA0004218826350000032
wherein:
Figure GDA0004218826350000033
is the steady state error at the frequency ramp input signal.
Knowing from equation (7) that the steady state error of the q-PLL scheme is not zero when tracking the frequency ramp signal; when introducing the p-component in the q-PLL, the angular error can be calculated from equations (1) and (2):
Figure GDA0004218826350000034
the compensated angles and frequency outputs are respectively:
Figure GDA0004218826350000035
Figure GDA0004218826350000036
step 3: inputting the detected rotor flux linkage into an improved q-type phase-locked loop to obtain the motor rotating speed and the rotor phase angle;
the speed identification based on the improved q-PLL mainly comprises a closed-loop rotor flux linkage observation model and the improved q-PLL, the rotor flux linkage observed by a closed-loop rotor flux linkage observer is input into the improved q-PLL, and the synchronous rotating speed of the rotor flux linkage is estimated
Figure GDA0004218826350000038
Subtracting the slip frequency omega sl The motor rotation speed can be obtained, and the slip frequency is as follows:
Figure GDA0004218826350000037
in the middle of：ψ r I is the rotor flux linkage vector sd L is the component of the stator current in the d-axis m Is mutual inductance T r Is the rotor time constant, T r ＝L r /R r ，R r Is rotor resistance, L r Is rotor inductance.
Step 4: and inputting the calculated motor rotation speed and rotor phase angle into a traction motor vector control system to complete speed sensorless control of the traction motor.
The beneficial technical effects of the invention are as follows:
(1) The invention can effectively make up the speed identification error and rotor phase angle identification error of the traditional q-PLL scheme when the traction motor is in speed increasing and decreasing;
(2) The improved q-PLL scheme of the present invention does not increase the order of the system, thus not adversely affecting the dynamic performance and stability margin of the system;
(3) The invention can realize good estimation performance under multiple working conditions, has simple structure and is easy to realize.
Drawings
FIG. 1 is a schematic block diagram of a closed-loop rotor flux observer employed in an embodiment of the invention.
Fig. 2 is a waveform diagram of a rotor flux linkage of a closed-loop rotor flux linkage observer used in an embodiment of the present invention after an integrator adds a direct current in a MATLAB/Simulink environment.
Fig. 3 is a diagram of an improved q-PLL structure in an embodiment of the invention.
Fig. 4 is a schematic diagram of a speed identification scheme based on an improved q-PLL in an embodiment of the invention.
Fig. 5 is a block diagram of a traction motor sensorless control system in an embodiment of the invention.
FIG. 6 shows a traction motor sensorless control system setting a constant load torque to T in a MATLAB/Simulink environment according to an embodiment of the present invention L =100 n·m, rotational speed N: the actual speed and estimated speed waveform diagram under the working condition of 500r/min suddenly changes to-500 r/min.
FIG. 7 shows a traction motor sensorless control system in MATLAB/Simulink environment according to an embodiment of the present inventionSetting a constant load torque to T L =100 n·m, rotational speed N: a waveform diagram of actual speed and estimated speed error under the working condition of 500r/min suddenly changes to-500 r/min.
FIG. 8 is a schematic diagram showing a traction motor sensorless control system setting a constant load torque to T in a MATLAB/Simulink environment according to an embodiment of the present invention L =100 n·m, rotational speed N: an actual observer rotor phase angle and estimated rotor phase angle waveform diagram under the working condition of 500r/min suddenly changes to-500 r/min.
Fig. 9 shows a traction motor speed sensorless control system setting a constant rotation speed of n=800 r/min and a load torque T in a MATLAB/Simulink environment according to an embodiment of the present invention L : actual speed and estimated speed waveforms for 100 N.m abrupt change to 700 N.m and then 100 N.m.
Fig. 10 shows a traction motor speed sensorless control system setting a constant rotation speed of n=800 r/min and a load torque T in a MATLAB/Simulink environment according to an embodiment of the present invention L : waveform diagram of actual speed and estimated speed error in case of 100n·m abrupt change to 700n·m and then 100n·m abrupt change.
Detailed Description
The invention will be described in further detail with reference to the drawings and examples.
According to the specific implementation method of the traction motor speed sensorless control method based on the improved q-type phase-locked loop, a simulation environment is set as MATLAB/Simulink. The method comprises the following steps:
step 1: and constructing a closed-loop flux linkage observer to accurately observe the flux linkage of the rotor.
The traditional voltage type flux linkage observer is an open loop observation model in nature, is easily influenced by stator resistance and an integrator, has poor robustness and is obvious at low speed; the current model is also an open loop observation model, and the model contains more rotor side motor parameters, so that the robustness is poor. Therefore, the invention adopts a closed-loop rotor flux linkage observer, combines the two, adopts a current model with more obvious advantages at low speed, adopts a voltage model with more obvious advantages at high speed, and compensates errors caused by integration and stator resistance voltage drop.
A block diagram of a closed loop flux linkage observer implementation is shown in fig. 1, in which: psi phi type s ＝[ψ sα ψ sβ ] T ，ψ r ＝[ψ rα ψ rβ ] T ，u s ＝[u sα u sβ ] T ，i s ＝[i sα i sβ ] T ，ψ s Is stator flux linkage vector, ψ sα 、ψ sβ For the components of the stator flux linkage in the alpha and beta axes, ψ r As a rotor flux linkage vector, ψ rα Sum phi rβ U, the component of the rotor flux linkage in the alpha and beta axes s U is the stator voltage vector sα 、u sβ I is the component of the stator voltage in the alpha and beta axes s I is the stator current vector sα 、i sβ Is the component of the stator current in the alpha and beta axes. Psi phi type rd I is the component of the rotor flux linkage in the d-axis sd L is the component of the stator current in the d-axis m Is mutual inductance T r Is the rotor time constant, T r ＝L r /R r ，R r Is rotor resistance, L r For rotor inductance, R s Is the stator resistance. k (k) p1 ＝ω 1 +ω 2 ，k i1 ＝ω 1 ω 2 ，ω 1 And omega 2 Two poles of the transfer function of the closed loop flux linkage observer for voltage and current.
Fig. 2 is a waveform diagram of a rotor flux linkage of a closed-loop rotor flux linkage observer adopted in an embodiment of the present invention after an integrator adds a direct current in a MATLAB/Simulink environment, so that it can be seen that when a drift error occurs in the integrator, a closed-loop rotor flux linkage model can effectively compensate, thereby avoiding a direct current bias in a rotor flux linkage.
Step 2: and constructing an improved q-type phase-locked loop.
In a two-phase fixed coordinate system, when the imaginary part u is to be output from the phase detector dq (t) when used as a feedback error signal, a q-PLL can be obtained and:
u dq (t)＝u 1β u 2α -u 1α u 2β
＝U 1 U 2 sinθ 1 cosθ 2 -U 1 U 2 cosθ 1 sinθ 2
＝U 1 U 2 sin(θ 1 -θ 2 )
the real part u of the phase detector output dp (t) is:
u dp (t)＝u 1α u 2α +u 1β u 2β
＝U 1 U 2 cosθ 1 cosθ 2 +U 1 U 2 sinθ 1 sinθ 2
＝U 1 U 2 cos(θ 1 -θ 2 )
the q-PLL scheme may have an estimation error when tracking the frequency ramp signal.
If the input is considered as an ideal signal with normalized amplitude, the closed loop transfer function of the q-PLL is expressed as follows:
Figure GDA0004218826350000051
the error transfer function is as follows:
Figure GDA0004218826350000052
when the input signals are respectively frequency-ramped:
Figure GDA0004218826350000053
wherein: θ i3 (s) is the transfer function of the different input signals, k is the gain of the frequency ramp.
Further, the error transfer function is:
Figure GDA0004218826350000061
wherein:
Figure GDA0004218826350000062
is the phase error transfer function at the frequency ramp input signal.
By using the final value theorem, steady state errors under different input signals can be obtained:
Figure GDA0004218826350000063
wherein:
Figure GDA0004218826350000064
for steady state error at the frequency ramp input signal, it is known that the steady state error of the q-PLL scheme is not zero when tracking the frequency ramp signal.
When the p-component is introduced in the q-PLL, an angular error is calculated:
Figure GDA0004218826350000065
compensated angle and frequency output:
Figure GDA0004218826350000066
Figure GDA0004218826350000067
the structure of the improved q-PLL implementation is shown in fig. 3.
Step 3: and inputting the detected rotor flux linkage into an improved q-type phase-locked loop to obtain the motor rotating speed and the rotor phase angle.
The speed recognition scheme based on the improved q-PLL is shown in FIG. 5, and mainly comprises a closed-loop rotor flux linkage observation model and the improved q-PLL, and the rotor flux linkage observed by a closed-loop rotor flux linkage observer is input into the improved q-PLLEstimating synchronous rotation speed of rotor flux linkage
Figure GDA0004218826350000068
Subtracting the slip frequency omega sl The motor rotation speed can be obtained, and the slip frequency is as follows:
Figure GDA0004218826350000069
wherein: psi phi type r I is the rotor flux linkage vector sd L is the component of the stator current in the d-axis m Is mutual inductance T r Is the rotor time constant, T r ＝L r /R r ，R r Is rotor resistance, L r Is rotor inductance.
A specific schematic diagram of a speed identification scheme based on a modified q-PLL is shown in fig. 4.
Step 4: and inputting the calculated motor rotation speed and rotor phase angle into a traction motor vector control system to complete the speed-sensorless control of the traction motor, wherein a specific traction motor speed-sensorless control system block diagram is shown in fig. 5.
In the speed-sensor-free control of the traction motor under the condition of MATLAB/Simulink, the parameters of the traction motor are as follows: stator resistor R s =0.147 Ω, rotor resistance R r =0.075Ω, excitation inductance L m = 31.291mH, stator leakage inductance L ls =0.956 mH, rotor leakage inductance L lr = 1.129mH. Simulation results of speed estimation under different conditions of sensorless control of the traction motor are shown in fig. 6-10.
FIG. 6 shows a traction motor sensorless control system setting a constant load torque to T in a MATLAB/Simulink environment according to an embodiment of the present invention L =100 n·m, rotational speed N: the waveform diagram of the actual speed and the estimated speed under the working condition from 500r/min to-500 r/min is that the estimated rotating speed can always track the actual rotating speed in the dynamic process that the rotating speed is suddenly changed from 500r/min to-500 r/min in the constant load condition of the traction motor according to the test result of the speed suddenly changed.
FIG. 7 is a schematic illustration of an embodiment of the present inventionIn the example, a traction motor speed sensorless control system sets constant load torque as T under MATLAB/Simulink environment L =100 n·m, rotational speed N: the 500r/min jump to a waveform of actual speed versus estimated speed error at-500 r/min conditions, the estimated error remains within an acceptable range.
FIG. 8 is a schematic diagram showing a traction motor sensorless control system setting a constant load torque to T in a MATLAB/Simulink environment according to an embodiment of the present invention L =100 n·m, rotational speed N: the actual observer rotor phase angle and the estimated rotor phase angle waveform diagram under the working condition that 500r/min suddenly changes to-500 r/min, the estimated rotor phase angle can be always fit with the actual observer rotor phase angle in the dynamic process that the rotating speed suddenly changes from 500r/min to-500 r/min in the constant load condition of the traction motor according to the test result of speed suddenly changes.
Fig. 9 shows a traction motor speed sensorless control system setting a constant rotation speed of n=800 r/min and a load torque T in a MATLAB/Simulink environment according to an embodiment of the present invention L : the waveform diagram of the actual speed and the estimated speed under the working condition that 100 N.m suddenly changes to 700 N.m and then to 100 N.m, the estimated rotating speed can always track the actual rotating speed in the dynamic process that the load torque suddenly changes from 100 N.m to 700 N.m and then drops to 100 N.m under the condition that the load suddenly changes under the constant rotating speed of the improved q-PLL.
Fig. 10 shows a traction motor speed sensorless control system setting a constant rotation speed of n=800 r/min and a load torque T in a MATLAB/Simulink environment according to an embodiment of the present invention L : the waveform diagram of the actual speed and the estimated speed error in the case of 100 N.m abrupt change to 700 N.m and then abrupt change to 100 N.m, the estimated error is kept in a small range.

Claims (1)

1. The traction motor speed-sensorless control method based on the improved q-type phase-locked loop is characterized by comprising the following steps of:
step 1: constructing a closed-loop rotor flux linkage observer to observe the rotor flux linkage;
a closed-loop rotor flux linkage observer is adopted, wherein a current model is adopted at low speed, a voltage model is adopted at high speed, and errors caused by integration and stator resistance voltage drop are compensated;
step 2: building an improved q-type phase-locked loop, namely a q-PLL;
in a two-phase fixed coordinate system, when the imaginary part u is to be output from the phase detector dq (t) when used as a feedback error signal, a q-PLL is obtained, and:
Figure FDA0004189550570000011
wherein u is 1α ，u 1β In order to input a signal to the device,
Figure FDA0004189550570000012
u 2α ，u 2β in order to output the signal,
Figure FDA0004189550570000013
U 1 ，ω 1 ，/>
Figure FDA0004189550570000014
for the amplitude, angular frequency and initial phase of the input signal, U 2 ，ω 2 ，/>
Figure FDA0004189550570000015
The amplitude, angular frequency and initial phase of the output signal;
the real part u of the phase detector output dp (t) is:
Figure FDA0004189550570000016
the q-PLL scheme has an estimation error when tracking the frequency ramp signal; if the input is considered as an ideal signal with normalized amplitude, the closed loop transfer function of the q-PLL is expressed as follows:
Figure FDA0004189550570000017
wherein k is p ，k i Gain for the loop filter;
the error transfer function is obtained from equation (3):
Figure FDA0004189550570000018
when the input signals are respectively frequency-ramped:
Figure FDA0004189550570000019
in θ i3 (s) is a transfer function of different input signals, k is the gain of the frequency ramp;
further, the error transfer function is:
Figure FDA00041895505700000110
in the method, in the process of the invention,
Figure FDA0004189550570000021
a phase error transfer function under a frequency ramp input signal;
and obtaining steady-state errors under different input signals by using a final value theorem:
Figure FDA0004189550570000022
in the method, in the process of the invention,
Figure FDA0004189550570000023
steady state error at the frequency ramp input signal;
knowing from equation (7) that the steady state error of the q-PLL scheme is not zero when tracking the frequency ramp signal; when introducing the p-component in the q-PLL, the angular error can be calculated from equations (1) and (2):
Figure FDA0004189550570000024
compensated angle and frequency output:
Figure FDA0004189550570000025
Figure FDA0004189550570000026
step 3: inputting the detected rotor flux linkage into an improved q-type phase-locked loop to obtain the motor rotating speed and the rotor phase angle;
the speed identification based on the improved q-PLL mainly comprises a closed-loop rotor flux linkage observation model and the improved q-PLL, the rotor flux linkage observed by a closed-loop rotor flux linkage observer is input into the improved q-PLL, and the synchronous rotating speed of the rotor flux linkage is estimated
Figure FDA0004189550570000028
Subtracting the slip frequency omega sl The motor rotation speed can be obtained, and the slip frequency is as follows:
Figure FDA0004189550570000027
in the psi- r I is the rotor flux linkage vector sq L is the component of the stator current in the q-axis m Is mutual inductance T r Is the rotor time constant, T r ＝L r /R r ，R r Is rotor resistance, L r The rotor inductance;
step 4: and inputting the calculated motor rotation speed and rotor phase angle into a traction motor vector control system to complete speed sensorless control of the traction motor.
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