CN111884535A - High-frequency pulse alternating-current link inverter hybrid modulation strategy - Google Patents

High-frequency pulse alternating-current link inverter hybrid modulation strategy Download PDF

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CN111884535A
CN111884535A CN202010613935.2A CN202010613935A CN111884535A CN 111884535 A CN111884535 A CN 111884535A CN 202010613935 A CN202010613935 A CN 202010613935A CN 111884535 A CN111884535 A CN 111884535A
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node
switching tube
transformer
modulation
strategy
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CN111884535B (en
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张晓斌
孙振东
王婷
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Northwestern Polytechnical University
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Northwestern Polytechnical University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer

Abstract

The invention relates to a high-frequency pulse alternating-current link inverter hybrid modulation strategy which comprises an SPWM (sinusoidal pulse width modulation) strategy and an SPWPM (sinusoidal pulse width modulation) strategy, wherein the judgment is carried out according to a phase phi, when the absolute value of the phase phi is larger, the hybrid modulation strategy is degenerated into a single SPWPM modulation strategy when the phase phi is 0 degrees, and the hybrid modulation strategy is degenerated into a single SPWM modulation strategy when the phase phi is 90 degrees. When 0 ° < Φ <90 °, the two modulation strategies alternate and the two modulation strategies can be seamlessly joined. Compared with the traditional SPWPM modulation strategy, the voltage spike can be inhibited, the reliability and the service life of the full-bridge full-wave high-frequency pulse alternating current link inverter are improved, compared with the full-bridge full-wave high-frequency pulse alternating current link inverter with the source clamping circuit, the circuit structure is simplified, the power loss is reduced, and the circuit expenditure is saved.

Description

High-frequency pulse alternating-current link inverter hybrid modulation strategy
Technical Field
The invention relates to a modulation strategy of a high-frequency pulse alternating current link inverter, and belongs to the technical field of isolated direct current-alternating current (DC-AC) converters.
Background
Because fossil energy has a great influence on the environment and the cost is gradually increased, the application of new energy is gradually valued by people. The new energy sources are applied with great attention paid to volume, cost and efficiency, which puts more strict requirements on the research of the power converter. Compared with a traditional SPWM DC-AC converter, the traditional DC-AC converter has a large-size capacitor and a heavy power frequency transformer, and the high-frequency pulse AC link inverter realizes current isolation through the high-frequency transformer, so that the power density of the inverter is improved, and the power density of the inverter is reduced. In recent years, high-frequency pulse alternating current link inverters are widely applied to occasions requiring high efficiency and high power density, such as uninterruptible power systems, renewable energy transmission modules, electric vehicle battery charging devices, audio power amplifiers, submarines, airplanes and the like.
Referring to fig. 1, the single-phase full-bridge full-wave high-frequency pulse alternating current link inverter topology is composed of a high-frequency inverter, a high-frequency transformer, a cycle converter and a filter. Switch Q1、Q2、Q3、Q4A high-frequency inverter is formed to convert the direct current into high-frequency alternating current; the transformer T forms a high-frequency transformer and isolates the primary side and the secondary side of the transformer; switch S1、S2、S3、S4A cycle converter is formed, and high-frequency alternating current is converted into power-frequency alternating current, so that energy flows in both input and output directions; the filter is composed of a filter inductor L1And a filter capacitor C1The filter is used for filtering the higher harmonic voltage.
Referring to fig. 2, the modulation of the conventional high-frequency pulse ac link inverter adopts a unipolar spwpm (sinusoidal pulse Width Phase modulation) modulation strategy. Under the SPWPM modulation strategy, the primary side voltage u of the transformer(3)(4)The voltage-second products of the adjacent positive and negative pulses are mutually offset, so that the magnetic flux density of the transformer is always kept at a relatively low value, and the sectional area of the transformer is small when the transformer works in a high-frequency state. However, the SPWPM modulation strategy also causes a series of problems, the leakage inductance current of the transformer is interrupted, and the voltage u is increased and decreased at the tenth node 10 and the twelfth node 12(10)(12)The high voltage spike is generated, so that the on-time of the cycle converter switch can be increased to inhibit the generation of high voltage. However, when the circuit operates in the first region u as shown in FIG. 2(10)(12)>0,
Figure BDA0002563104750000021
Or a third region u(10)(12)<0,
Figure BDA0002563104750000022
In the process, due to the leakage inductance of the transformer and the parasitic capacitance of the switch tube, the cycle converter switch is switched off, the leakage inductance current of the transformer is interrupted, and the secondary side of the transformer generates a voltage spike, which generally causes the avalanche breakdown of the MOS tube. An Active voltage clamp (ACC) may be used to suppress voltage spikes, but the Active clamp also makes the converter more complex, increasing the cost of the circuit and increasing the size of the circuit.
Therefore, a modulation strategy which can reduce the circuit cost and restrain the voltage spike is needed to be searched, the economic cost and the output electric energy quality are balanced as much as possible, and the application of new energy sources is promoted.
Disclosure of Invention
Technical problem to be solved
The method aims to overcome the defect that the leakage inductance current of the transformer is interrupted to generate a high voltage peak due to the disconnection of a cycle converter switch in the conventional SPWPM modulation strategy, reduce the circuit cost and improve the power density of a high-frequency pulse alternating current link inverter. The invention provides a high-frequency pulse alternating current link inverter hybrid modulation strategy.
Technical scheme
A mixed modulation strategy of a high-frequency pulse alternating current link inverter comprises a first switching tube Q1A second switch tube Q2And a third switching tube Q3And a fourth switching tube Q4The fifth switch tube S1The sixth switching tube S2Seventh switching tube S3The eighth switching tube S4A first inductor L1A first capacitor C1Load R1And a transformer, a first switch tube Q1Drain electrode of (1), third switching tube Q3And a power supply anode connected to the first node, a first switch tube Q1Source electrode of and the second switch tube Q2The drain electrode and the primary side dotted terminal of the transformer are connected to a third node, and a second switching tube Q2Source electrode and fourth switching tube Q4The source electrode of the power supply and the negative electrode of the power supply are connected to a second node, and a third switching tube Q3Source electrode and fourth switching tube Q4The drain electrode of the first switching tube S is connected with the primary different name end of the transformer at a fourth node1The drain electrode of the first switch tube is connected with the dotted terminal of the first winding on the secondary side of the transformer at a fifth node, and a sixth switch tube S2Source electrode of and fifth switching tube S1Is connected to the eighth node, and a seventh switching tube S3The drain electrode of the first switch tube is connected with the synonym end of the secondary side second winding of the transformer at a seventh node, and a seventh switch tube S3Source electrode and eighth switching tube S4The source electrode of the first switch tube is connected with the ninth node and the sixth switch tube S2Drain electrode of and the eighth switching tube S4And the first inductor L1One end of which is connected to the tenth node; first inductance L1And the other end of the first capacitor C1And a load R1Is connected to the eleventh node, the first capacitor C1Another end of (1) and a load R1The other end of the capacitor is connected with the other end of the load and the other end of the load, and is characterized in that: when in use
Figure BDA0002563104750000031
Then, SPWM modulation is adopted, where u(10)(12)Voltage u at the tenth node and the twelfth node(10)(12)，
Figure BDA0002563104750000032
Is a first inductance L1The current of (a); when in use
Figure BDA0002563104750000033
Then, an SPWPM modulation mode is adopted; i.e. calculating the phase
Figure BDA0002563104750000034
Get phi0Phi is more than or equal to phi, and the ratio of:
Figure BDA0002563104750000035
wherein T is u(10)(12)The working period of (a), one working period T is divided into the following 5 phases: when 0 is present<t<taWhen the method is adopted, an SPWM (sinusoidal pulse Width modulation) strategy is adopted, and when t isa<t<tbWhen the SPWPM modulation strategy is adopted, when t isb<t<tcThen, SPWM modulation is adopted, and when t isc<t<tdThen, the SPWPM modulation strategy is adopted, and finally when t isd<t<And T, selecting an SPWM modulation strategy.
Advantageous effects
Compared with the traditional SPWPM (pulse-modulated wave modulation) strategy, the hybrid modulation strategy of the high-frequency pulse alternating-current link inverter provided by the invention can inhibit the generation of voltage spikes, improve the reliability and the service life of the full-bridge full-wave high-frequency pulse alternating-current link inverter, simplify the circuit structure, reduce the power loss and save the circuit expenditure compared with the full-bridge full-wave high-frequency pulse alternating-current link inverter with the source clamping circuit.
Drawings
FIG. 1 is a single-phase full-bridge full-wave high-frequency pulse AC link inverter topology
FIG. 2 is a diagram of main waveforms of a conventional SPWPM modulation strategy
FIG. 3 is a diagram of the main waveforms of the novel hybrid modulation strategy
FIG. 4 is a graph of the change of the magnetic flux density of a transformer with a single SPWPM modulation strategy
FIG. 5 is a graph of the magnetic flux density variation of a single SPWM modulation strategy transformer
FIG. 6 is a graph of transformer flux density variation in SPWM and SPWPM modulation strategies
FIG. 7 is a diagram of main waveforms of a circuit operating in SPWM modulation strategy under hybrid modulation strategy
FIG. 8 is a schematic diagram of main circuit mode 0 operation
FIG. 9 is a schematic diagram of main circuit mode 1 operation
FIG. 10 is a schematic diagram of main circuit mode 2 operation
FIG. 11 is a schematic diagram of main circuit mode 3 operation
FIG. 12 is a diagram of main waveforms of a circuit operating in a SPWPM modulation strategy under a hybrid modulation strategy
FIG. 13 is a schematic diagram of main circuit mode 4 operation
FIG. 14 is a schematic diagram of main circuit mode 5 operation
FIG. 15 is a schematic diagram of the main circuit mode 6
FIG. 16 is a schematic diagram of the main circuit mode 7
FIG. 17 is a schematic diagram of the main circuit mode 8
FIG. 18 is a schematic diagram of the main circuit mode 9
Q in FIG. 11～Q4Is a high-frequency inverter MOS tube; s1～S4MOS tube of cycle converter; t is a high-frequency transformer; l is1Is a filter inductor; c1Is a filter capacitor; r1Is a load resistor; q in FIGS. 2 and 31、Q3、S1、S2、S3、S4Are MOS transistors Q respectively1、Q3、S1、S2、S3、S4Is applied to the gate-voltage signal of the gate-voltage signal,
Figure BDA0002563104750000041
is an MOS transistor Q2、Q4A non-signal of the applied gate voltage of (1); u. of(3)(4)The voltage between the third node 3 and the fourth node 4; u. of(10)(12)The voltage between the tenth node 10 and the twelfth node 12;
Figure BDA0002563104750000042
for flowing through the filter inductance L1The current of (a); q in FIGS. 7 and 121、Q2、Q3、Q4、S1、S2、S3、S4Are MOS transistors Q respectively1、Q2、Q3、Q4、S1、S2、S3、S4Applied gate voltage signal u(3)(4)The voltage between the third node 3 and the fourth node 4; i.e. iLkp、iLks1、iLks2Are respectively a current-through transformerT leakage inductance Lkp、Lks1、Lks2The current of (a); q in FIGS. 8 to 11 and 13 to 181～Q4Is a high-frequency inverter MOS tube; s1～S4MOS tube of cycle converter; t is a high-frequency transformer; l is1Is a filter inductor; c1Is a filter capacitor; r1Is a load resistance, Lkp、Lks1、Lks2The leakage inductance of the primary side of the transformer, the leakage inductance of the first winding of the secondary side of the transformer and the leakage inductance of the second winding of the secondary side of the transformer are respectively.
Detailed Description
The invention will now be further described with reference to the following examples and drawings:
as shown in FIG. 1, the inverter is a full-bridge full-wave high-frequency pulse AC link inverter, and the first switch tube Q1Drain electrode of (1), third switching tube Q3And a power supply anode connected to the first node 1, a first switch tube Q1Source electrode of and the second switch tube Q2The drain electrode of the first switching tube Q is connected with the same name end of the primary side of the transformer at a third node 32Source electrode and fourth switching tube Q4Is connected to the second node 2, and a third switching tube Q3Source electrode and fourth switching tube Q4And the drain of the transformer and the primary synonym terminal of the transformer are connected to a fourth node 4. Fifth switch tube S1The drain electrode of the first switch tube S is connected with the dotted terminal of the first winding on the secondary side of the transformer at a fifth node 5, and a sixth switch tube S2Source electrode of and fifth switching tube S1Is connected to the eighth node 8, and a seventh switching tube S3The drain electrode of the first switch tube S is connected with the synonym end of the secondary side second winding of the transformer at a seventh node 73Source electrode and eighth switching tube S4Is connected to the ninth node 9, the sixth switching tube S2Drain electrode of and the eighth switching tube S4And the first inductor L1One end of which is connected to the tenth node 10. First inductance L1And the other end of the first capacitor C1And a load R1One end of the capacitor is connected with the eleventh node 11, the other end of the capacitor is connected with the different name end of the first winding of the secondary side of the transformer, the same name end of the second winding of the secondary side of the transformer and the other end of the load to the sixth node6, first capacitance C1Another end of (1) and a load R1And the other end thereof is connected to a twelfth node 12, and the twelfth node 12 is grounded.
Referring to fig. 1, u is two points of an eleventh node 10 and a twelfth node 12(10)(12)Voltage and inductance L1The transfer function relationship of the current of (a) is:
Figure BDA0002563104750000061
the phase of the transfer function h(s) is:
Figure BDA0002563104750000062
when the phase phi is 0, the resistance R1The values of (A) are as follows:
Figure BDA0002563104750000063
where ω is the angular frequency of the fundamental wave of the output voltage, and the phase function ^ H(s) can be known from the formula (3) when L is1、C1When the value is constant, with respect to the resistance R1As an increasing function of resistance R1The larger the value of (A), the larger the phase value when R is1The smaller the value of (a) is, the smaller the phase value is.
Referring to the modulation strategy of the traditional high-frequency pulse alternating-current link inverter in FIG. 2, phi obtained through calculation can be obtained according to u within one modulation wave period(10)(12)And iL1The positive-negative relationship divides it into four parts: first region u(10)(12)>0,iL1<0, second region u(10)(12)>0,iL1>0, third region u(10)(12)<0,iL1>0, fourth region u(10)(12)<0,iL1<0. When the circuit works in the first region u(10)(12)>0,iL1<At 0, due to the inductor current iL1<0 and switch S2、S4Parasitic diodes in opposite directions, switch S2、S4Will turn off the leakage current of the transformer hard, at the tenth node 10 and the tenth nodeThe twelve nodes 12 generate voltage spikes; when the circuit operates in the second region u(10)(12)>0,iL1>At 0, due to the inductor current iL1>0 and switch S2、S4The body diodes are in the same direction, so the switch S2、S4When the transformer is turned off, the leakage inductance current of the transformer can pass through the switch S2、S4The body diode freewheels without voltage spikes at the tenth node 10 and the twelfth node 12. In the same way, the third area u(10)(12)<0,iL1>0 will generate voltage spikes at the tenth node 10 and the twelfth node 12, and the fourth region u(10)(12)<0,iL1<0 does not produce voltage spikes at the tenth node 10 and the twelfth node 12.
The conditions that do not generate voltage spikes at the tenth node 10 and the twelfth node 12 are:
Figure BDA0002563104750000064
referring to FIG. 3, the modulation strategy of the present invention includes SPWM and SPWPM modulation strategies, when
Figure BDA0002563104750000065
Then, an SPWM modulation mode is adopted, such as a white area in the image; when in use
Figure BDA0002563104750000071
When the method is used, the SPWPM modulation mode is adopted, as shown in a gray area in the figure. When the absolute value of the phase phi in the equation (2) is larger, the fundamental wave and the inductance L representing the voltages at the points of the tenth node 10 and the twelfth node 12 are larger1Current of
Figure BDA0002563104750000072
The larger the fundamental wave phase difference is, the larger the occupation ratio of the SPWM modulation strategy time needs to be increased; and conversely, the occupation ratio of the SPWM modulation strategy time is reduced. When the phase phi is 0 deg., the hybrid modulation strategy degenerates to a single SPWPM modulation strategy, and when the phase phi is 90 deg., the hybrid modulation strategy degenerates to a single SPWM modulation strategy. When 0 degree<φ<At 90 deg. two adjustmentsThe modulation strategies alternate and the two modulation strategies can be seamlessly joined.
The flux density of a transformer according to faraday's law has the following relationship:
Figure BDA0002563104750000073
where Δ B is the variation of the magnetic flux density of the transformer, V is the voltage of the primary side of the transformer, and TonFor the time of switching-on of the switching tube, AeThe effective magnetic core cross-sectional area, N is the number of transformer turns. This indicates when the parameter V, A is presenteAnd when N is unchanged, the longer the switching tube is switched on, the larger the magnetic flux density of the transformer is.
Referring to fig. 4, the primary side voltage u of the transformer is measured when the inverter is operating in a single SPWPM modulation strategy(3)(4)The voltage-second products of adjacent positive and negative pulses can be partially or completely offset, so that the magnetic flux density of the transformer is always kept at a relatively low value, and the transformer can be designed in a smaller volume, but because of the existence of the positive and negative pulses
Figure BDA0002563104750000074
This may cause relatively severe voltage spikes at the tenth node 10 and the twelfth node 12.
Referring to FIG. 5, when the inverter is operating in a single SPWM modulation strategy, the primary voltage u of the transformer(3)(4)Because the modulation wave is positive pulse in the positive half period, the magnetic flux density of the transformer is continuously increased, and the maximum magnetic flux density of the transformer in the mode is larger than that of the SPWPM modulation strategy, which also indicates that the transformer with larger volume needs to be designed under the same condition of the SPWM modulation strategy, but the switching tube state of the cycle converter is unchanged, so that the converter does not have voltage spikes at the tenth node 10 and the twelfth node 12.
Referring to fig. 6, when the inverter operates in the SPWM modulation and SPWPM hybrid modulation strategy, the inverter operates in the pwm modulation and SPWPM hybrid modulation strategy
Figure BDA0002563104750000075
The period adopts an SPWM (sinusoidal pulse Width modulation) strategy, so that the switch manufacture of a switching tube of a cycle converter is reducedVoltage spike of
Figure BDA0002563104750000081
The volume of the transformer is reduced by adopting the SPWPM modulation strategy, the power density of the converter is improved, the maximum magnetic flux density of the transformer is smaller by adopting the hybrid modulation strategy compared with that of a single SPWM modulation strategy, and the advantages of the two modulation strategies are combined.
Referring to FIG. 3, the phase φ is calculated by equation (2)0Phi is more than or equal to phi, and the ratio of:
Figure BDA0002563104750000082
wherein T is u(10)(12)The period of (c). The control strategy divides one duty cycle of the inverter into 5 phases as follows. When 0 is present<t<taWhen the method is adopted, an SPWM (sinusoidal pulse Width modulation) strategy is adopted, and when t isa<t<tbWhen the SPWPM modulation strategy is adopted, when t isb<t<tcThen, SPWM modulation is adopted, and when t isc<t<tdThen, the SPWPM modulation strategy is adopted, and finally when t isd<t<And T, selecting an SPWM modulation strategy.
Referring to fig. 3, the working mode of the high-frequency pulse ac link inverter can be divided into an SPWM modulation strategy and an SPWPM modulation strategy, and the following analysis is respectively made on the working processes of the SPWM and SPWPM modulation strategies:
referring to FIG. 7, at this time
Figure BDA0002563104750000083
Due to the fact that
Figure BDA0002563104750000084
The inverter works in an SPWM (sinusoidal pulse Width modulation) strategy, and the working process of the circuit is analyzed as follows:
mode 0[ t ]0,t1]
Referring to fig. 8, the switching tube Q2、Q4、S1、S2Primary side Q of open-circuit transformer2、Q4Turn on as leakage inductance LkpAnd an excitation inductor LmThe current flows continuously,at the secondary side S of the transformer1、S2Simultaneously switched on as leakage inductance Lks1And a filter inductance L1Follow current, at this time L1Current flows from the twelfth node 12 to the tenth node 10. Due to S3、S4Off, no current through Lks2。
Mode 1[ t ]1,t2]
Referring to fig. 9, the switching tube Q4、S1、S2On, Q2Body diode, Q4Turn on as leakage inductance LkpAnd an excitation inductor LmFollow current, on the secondary side S of the transformer1、S2Simultaneously switched on as leakage inductance Lks1And a filter inductance L1Follow current, at this time L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 2[ t ]2,t3]
Referring to fig. 10, the primary side Q of the transformer1、Q4Is switched on, and generates a positive pulse i at the third and fourth ends of the primary side of the transformerLkpIncrease of S1、S2Is opened, so that the positive pulse of the secondary side can reach the tenth node 10 point, iLks1Increase due to S3、S4And the switch is switched off, so that the negative pulse on the third side cannot reach C. Since no current passes through Lks2No voltage spike is generated at the twelfth node 12 to the tenth node 10.
Mode 3[ t ]3,t4]
Referring to fig. 11, the switching tube Q4、S1、S2On, Q2Body diode, Q4Turn on as leakage inductance LkpAnd an excitation inductor LmFollow current, the voltage of the third node 3 and the fourth node 4 gradually drops to 0, and the secondary side S of the transformer1、S2Simultaneously switched on as leakage inductance Lks1And a filter inductance L1Follow current, at this time L1Current flows from the twelfth node 12 to the tenth node 10.
Referring to FIG. 12, at this time
Figure BDA0002563104750000091
Due to the fact that
Figure BDA0002563104750000092
The inverter works in an SPWPM modulation strategy, and the working process of the circuit is analyzed as follows:
mode 4[ t ]5,t6]
Referring to fig. 13, the primary side switching tube Q of the transformer1、Q3Turn on as leakage inductance LkpAnd an excitation inductor LmFollow current, primary winding S on secondary side of transformer2Open, leakage inductance Lks1By S1Body diode and S2Follow current, secondary winding S on secondary side of transformer3、S4Turn-on, inductance L1And leakage inductance Lks2By S3And S4Follow current, at this time L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 5[ t ]6,t7]
Referring to fig. 14, a primary side switching tube Q of a transformer1、Q3Turn on as leakage inductance LkpAnd an excitation inductor LmFollow current iLkpReducing, at the secondary side of the transformer, the primary winding S2Turn-on, inductance L1And leakage inductance Lks1By S1Body diode and S2Follow current iLks1Increased size of secondary winding S on secondary side of transformer4Turn-on, inductance L1And leakage inductance Lks2By S3Body diode and S4Follow current iLks2Decrease in size when L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 6[ t ]7,t8]
Referring to fig. 15, the primary side switching tube Q of the transformer1、Q3Turn on as leakage inductance LkpAnd an excitation inductor LmFollow current iLkpContinuing to reduce, the primary winding S is arranged on the secondary side of the transformer1、S2Turn-on, inductance L1And leakage inductance Lks1By S1And S2Follow current iLks1Increasing in the reverse direction, in the secondary winding S of the transformer4Open, leakage inductance Lks2By S3Body diode and S4Follow current iLks2Decrease in size, L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 7[ t ]8,t9]
Referring to fig. 16, the primary side switching tube Q of the transformer3Open, leakage inductance LkpAnd an excitation inductor LmBy Q2Body diode and Q3Follow current, turning on Q for zero voltage2Preparation is made iLkpReducing, at the secondary side of the transformer, the primary winding S1、S2Turn-on, inductance L1And leakage inductance Lks1By S1And S2Follow current iLks1Increasing in the reverse direction, in the secondary winding S of the transformer4Open, leakage inductance Lks2By S3Body diode and S4Follow current iLks2Decrease in size, L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 8[ t ]9,t10]
Referring to fig. 17, the primary side switching tube Q of the transformer2、Q3On, a negative pulse i is generated on the primary side AB of the transformerLkpIncrease in negative direction, primary winding S on secondary side of transformer1、S2On so that a negative going pulse can reach the tenth node 10, iLks1Increasing in the reverse direction, in the secondary winding S of the transformer4Opening, S3Turn off, block positive pulse from reaching the tenth node 10, leakage inductance Lks2By S3Body diode and S4Follow current iLks2The size is reduced to 0, L1Current flows from the twelfth node 12 to the tenth node 10.
Mode 7[ t ]10,t11]
Referring to fig. 18, a primary side switching tube Q of a transformer2Open, leakage inductance LkpAnd an excitation inductor LmBy Q2And Q4Body diode freewheeling turning on Q for zero voltage4Preparation is made iLkpReduced size of primary winding S on secondary side of transformer1、S2Turn-on, inductance L1And leakage inductance Lks1By S1And S2Follow current iLks1Reduced size of secondary winding S on secondary side of transformer4Is on, iLks2Increasing in size, L1Current flows from the twelfth node 12 to the tenth node 10.
In conclusion, the mixed modulation strategy of the high-frequency pulse alternating-current link inverter can provide a leakage inductance and filter inductance follow current loop of the transformer, solve the problem that the leakage inductance current of the transformer is hard to be turned off by the cycle converter, and inhibit the secondary voltage peak of the transformer. The invention can improve the power density, reliability and service life of the full-bridge full-wave high-frequency pulse AC link inverter and reduce the cost of the circuit.

Claims (1)

1. A mixed modulation strategy of a high-frequency pulse alternating current link inverter comprises a first switching tube Q1A second switch tube Q2And a third switching tube Q3And a fourth switching tube Q4The fifth switch tube S1The sixth switching tube S2Seventh switching tube S3The eighth switching tube S4A first inductor L1A first capacitor C1Load R1And a transformer, a first switch tube Q1Drain electrode of (1), third switching tube Q3And the positive electrode of the power supply is connected with the first node (1), and the first switching tube Q1Source electrode of and the second switch tube Q2The drain electrode of the first switch tube Q is connected with the same name end of the primary side of the transformer at a third node (3)2Source electrode and fourth switching tube Q4The source electrode of the power supply and the negative electrode of the power supply are connected to a second node (2), and a third switching tube Q3Source electrode and fourth switching tube Q4The drain electrode of the first switching tube S is connected with the primary different name end of the transformer at a fourth node (4) and a fifth switching tube S1The drain electrode of the first switch tube is connected with the dotted end of the first winding on the secondary side of the transformer at a fifth node (5), and a sixth switch tube S2Source electrode of and fifth switching tube S1Is connected to an eighth node (8), a seventh switching tube S3The drain electrode of the first switch tube is connected with the synonym end of the secondary side second winding of the transformer to a seventh node (7), and a seventh switch tube S3Source electrode and eighth switching tube S4Is connected to the ninth node (9)) The sixth switching tube S2Drain electrode of and the eighth switching tube S4And the first inductor L1Is connected to the tenth node (10); first inductance L1And the other end of the first capacitor C1And a load R1Is connected to an eleventh node (11), a first capacitor C1Another end of (1) and a load R1The other end of the capacitor is connected with a twelfth node (12), the twelfth node (12) is grounded, and the other end of the capacitor is connected with the synonym end of the first secondary winding of the transformer, the homonymy end of the second secondary winding of the transformer and the other end of the load at a sixth node (6), and the transformer is characterized in that: when in use
Figure FDA0002563104740000011
Then, SPWM modulation is adopted, where u(10)(12)The voltage u of the tenth node (10) and the twelfth node (12)(10)(12)，
Figure FDA0002563104740000012
Is a first inductance L1The current of (a); when in use
Figure FDA0002563104740000013
Then, an SPWPM modulation mode is adopted; i.e. calculating the phase
Figure FDA0002563104740000014
Get phi0Phi is more than or equal to phi, and the ratio of:
Figure FDA0002563104740000015
wherein T is u(10)(12)The working period of (a), one working period T is divided into the following 5 phases: when 0 is present<t<taWhen the method is adopted, an SPWM (sinusoidal pulse Width modulation) strategy is adopted, and when t isa<t<tbWhen the SPWPM modulation strategy is adopted, when t isb<t<tcThen, SPWM modulation is adopted, and when t isc<t<tdThen, the SPWPM modulation strategy is adopted, and finally when t isd<t<And T, selecting an SPWM modulation strategy.
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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114123839A (en) * 2021-10-22 2022-03-01 深圳英飞源技术有限公司 Isolated bidirectional AC/DC conversion circuit and device

Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1489272A (en) * 2002-10-08 2004-04-14 中国科学院电工研究所 Phase-shift full-bridge high-frequency inverter based on DSP
CN101741272A (en) * 2009-12-09 2010-06-16 燕山大学 Regulating and controlling method and logic of self-adaption commutation integration of high frequency chain matrix inverter
CN101976965A (en) * 2010-10-08 2011-02-16 刘闯 Direct-current tracing control high-power photovoltaic grid-connected inverter
CN104065289A (en) * 2014-06-13 2014-09-24 南京理工大学 Flyback high-frequency isolating type three-level inverter
CN105703652A (en) * 2016-03-01 2016-06-22 北京交通大学 Control method of high-frequency isolation DC/AC inverter circuit and high-frequency isolation DC/AC inverter circuit
US20180091037A1 (en) * 2016-01-27 2018-03-29 Southeast University Four-port power electronic transformer based on hybrid modular multilevel converter
CN110707959A (en) * 2019-10-01 2020-01-17 西北工业大学 High-frequency transformer isolation voltage source inverter topology and control strategy based on pulse combination

Patent Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1489272A (en) * 2002-10-08 2004-04-14 中国科学院电工研究所 Phase-shift full-bridge high-frequency inverter based on DSP
CN101741272A (en) * 2009-12-09 2010-06-16 燕山大学 Regulating and controlling method and logic of self-adaption commutation integration of high frequency chain matrix inverter
CN101976965A (en) * 2010-10-08 2011-02-16 刘闯 Direct-current tracing control high-power photovoltaic grid-connected inverter
CN104065289A (en) * 2014-06-13 2014-09-24 南京理工大学 Flyback high-frequency isolating type three-level inverter
US20180091037A1 (en) * 2016-01-27 2018-03-29 Southeast University Four-port power electronic transformer based on hybrid modular multilevel converter
CN105703652A (en) * 2016-03-01 2016-06-22 北京交通大学 Control method of high-frequency isolation DC/AC inverter circuit and high-frequency isolation DC/AC inverter circuit
CN110707959A (en) * 2019-10-01 2020-01-17 西北工业大学 High-frequency transformer isolation voltage source inverter topology and control strategy based on pulse combination

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
HUI LI ET AL.: "Simulation of SPWPM high-frequency link inverter for EPS", 《2008 IEEE INTERNATIONAL CONFERENCE ON MECHATRONICS AND AUTOMATION》, pages 1106 - 1110 *
沙德尚 等: "高频链逆变器的全数字化混合 SPWPM 控制策略研究", 《电工技术学报》, vol. 20, no. 7, pages 57 - 62 *

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114123839A (en) * 2021-10-22 2022-03-01 深圳英飞源技术有限公司 Isolated bidirectional AC/DC conversion circuit and device

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