CN111796273A - Anti-radiation seeker signal processing method and system based on FPGA - Google Patents
Anti-radiation seeker signal processing method and system based on FPGA Download PDFInfo
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- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
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- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
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Abstract
The invention provides a method and a system for processing a signal of an anti-radiation seeker based on an FPGA (field programmable gate array), which comprises the steps of sampling serial data received by the anti-radiation seeker and converting the serial data into N paths of parallel data and units; obtaining effective channels and units according to the sampling data and the frequency threshold; calibrating and processing unit for signal amplitude output by effective channel; and detecting and measuring parameters and units for the calibrated signals through a gate.
Description
Technical Field
The invention relates to a signal processing technology, in particular to a method and a system for processing a back radiation seeker signal based on an FPGA.
Background
The performance of modern radar and electronic warfare has evolved to a very high level. The radar system is used as an electromagnetic sensor, plays a key role in modern war, and the advanced radar system can detect enemies first and respond quickly in battlefields. In order to counter the radar system, anti-radiation missiles are produced. The anti-radiation missile is the most effective weapon for electromagnetic radiation source equipment, and utilizes the electromagnetic wave guide of enemy equipment to destroy equipment such as enemy radar. In the process of intercepting and collecting enemy radar signals of the anti-radiation missile in a severe electromagnetic environment, an anti-radiation seeker system is of great importance.
The anti-radiation seeker system is a reconnaissance receiver which can adapt to complex modern electronic warfare. Since they tend to detect non-cooperative signals, they require wide input bandwidth, high sensitivity and resolution, large dynamic range, and the ability to process multiple signals simultaneously. In addition to this, it must be able to adapt to modern, highly intensive signal environments, and the large amounts of information received must be processed in real-time or near real-time. The main characteristic is wide frequency band, which almost covers the whole wireless frequency band.
Therefore, the anti-radiation seeker system for radar can be regarded as a detection and extraction of broadband radar signals. The digital channelized receiver in the software radio can meet the requirement of the frequency band coverage of the system. Meanwhile, by utilizing the characteristics of reconfigurable and parallelization of a Field Programmable Gate Array (FPGA), a digital channelized receiver and radar signal detection and processing after channelization are realized on the FPGA, and the real-time performance required by a counter radiation seeker system can be met. The anti-radiation seeker signal processing method and system based on the FPGA can achieve capture of large-bandwidth radar signals and real-time processing of high data rate, and provide technical support for accurate attack of enemy radars. However, the current phase characteristics of the channels of the FPGA chip and the high-speed AD chip are easily influenced by factors such as temperature and humidity, so that the system has inaccurate directional angle measurement results on the target radar.
Disclosure of Invention
The invention aims to solve the technical problem of eliminating the influence of the environment on the radar directional angle measurement result.
The problem is not solved, the method for processing the anti-radiation seeker signal based on the FPGA comprises the following steps: sampling serial data received by a back radiation seeker and converting the serial data into N paths of parallel data; obtaining an effective channel according to the sampling data and the frequency threshold; calibrating the signal amplitude output by the effective channel; and detecting and measuring parameters of the calibrated signals through a gate.
Furthermore, the amplitude average value of one channel data in a specified time is detected as the basis for adjusting the gain of the received signal of the anti-radiation seeker.
Further, deriving the effective channel based on the sampled data and the frequency threshold comprises: inputting N paths of parallel data into M paths of channels at the same time for delay processing in sequence; performing multi-phase filtering on the M paths of parallel data; carrying out FFT operation on the multiphase filtered signals to obtain channel frequency; a channel within a certain frequency range is selected as an active channel.
Further, calibrating the signal amplitude of the effective channel output comprises obtaining calibration values, the calibration values comprising an initial normal calibration value: the anti-radiation seeker transmits and receives a normal signal in a darkroom environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained; initial closed loop calibration value: the anti-radiation seeker transmits and receives a signal with a specific frequency under a laboratory environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained; real-time closed-loop calibration value: and in the working process of the anti-radiation guide head, the anti-radiation guide head transmits and receives a signal with a specific frequency, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained.
Further, calibrating the signal amplitude of the effective channel output further comprises compensating for channel-to-channel inconsistencies during operation of the anti-radiation seeker using the difference between the initial closed-loop calibration value and the real-time closed-loop calibration value.
The invention is characterized in that the system comprises a unit for sampling serial data received by the anti-radiation seeker and converting the serial data into N paths of parallel data; obtaining a unit of an effective channel according to the sampling data and the frequency threshold; a unit for calibrating the signal amplitude output by the effective channel; and a unit for detecting and measuring the parameter of the calibrated signal through the gate.
Furthermore, the system also comprises a unit for detecting the amplitude average value of one channel data in a specified time as the basis for adjusting the gain of the received signal of the anti-radiation seeker.
Further, the unit for obtaining the effective channel according to the sampling data and the frequency threshold value comprises: inputting N paths of parallel data into M paths of channels simultaneously to be sequentially used as a delay processing subunit; performing a multi-phase filtering subunit on the M paths of parallel data; performing FFT operation on the multiphase filtered signals to obtain a channel frequency subunit; channels within a certain frequency range are selected as valid channel subunits.
Further, the unit for calibrating the signal amplitude of the effective channel output comprises an obtaining calibration value subunit, wherein the calibration value comprises an initial normal calibration value: the anti-radiation seeker transmits and receives a normal signal in a darkroom environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained; initial closed loop calibration value: the anti-radiation seeker transmits and receives a signal with a specific frequency under a laboratory environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained; real-time closed-loop calibration value: and in the working process of the anti-radiation guide head, the anti-radiation guide head transmits and receives a signal with a specific frequency, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained.
Further, the unit for calibrating the signal amplitude of the effective channel output further comprises a subunit for compensating the inconsistency between the channels during the operation of the anti-radiation seeker by using the difference between the initial closed-loop calibration value and the real-time closed-loop calibration value.
The invention has the beneficial effects that: according to the method and the system for processing the anti-radiation seeker signal based on the FPGA, the radar signal in a 2-18 GHz frequency band can be captured, the signal in a 400MHz bandwidth can be processed in real time, and the channel phase error caused by working environment factors such as temperature and humidity is reduced through a real-time calibration system, so that the subsequent direction finding is more reliable compared with a traditional target radar direction finding system, and accurate striking on a target radar is facilitated.
The invention is further described below with reference to the accompanying drawings.
Drawings
FIG. 1 is a flow chart of the method of the present invention.
FIG. 2 is a flowchart of step S1 according to the present invention.
FIG. 3 is a flowchart of step S2 according to the present invention.
Fig. 4 is a flowchart of step S3 according to the present invention.
FIG. 5 is a flowchart of step S310 according to the present invention.
FIG. 6 is a flowchart of step S320 according to the present invention.
FIG. 7 is a schematic diagram of the system of the present invention.
FIG. 8 is a schematic diagram of a first unit of the system of the present invention.
FIG. 9 is a schematic diagram of a fifth unit of the system of the present invention.
FIG. 10 is a schematic diagram of a second unit of the system of the present invention.
FIG. 11 is a schematic diagram of a third unit of the system of the present invention.
FIG. 12 is a diagram of a fourth unit of the system of the present invention.
Fig. 13 is a schematic diagram of a rf front-end system of the system front-end of the present invention.
Detailed Description
Fig. 7 is a schematic diagram of an embodiment of a signal processing system according to the present invention, which includes a first unit 100 for sampling serial data received by a radiation protection seeker and converting the serial data into N parallel data; a second unit 200 for obtaining an effective channel according to the sampled data and the frequency threshold; a third unit 300 for calibrating the signal amplitude of the effective channel output; a fourth unit 400 for detecting and measuring parameters for the calibrated signal by passing through the gate.
Referring to fig. 8, the first unit 100 includes an AD sampling subunit 110, a serial-to-parallel conversion subunit 120, a system clock subunit 130, and a phase synchronization subunit 140. Further, the AD sampling sub-unit 110 performs AD sampling on the serial data received by the anti-radiation seeker; the serial-parallel conversion subunit 120 converts the serial data into N-way parallel data; the system clock subunit 130 realizes time synchronization of the N paths of parallel data; the phase synchronization subunit 140 implements phase synchronization of the N-way parallel data. In this embodiment, the AD sampling subunit 110 includes a selecteio Interface WizardIP core of Xilinx, and the IP core collects serial data. In this embodiment, the serial-parallel conversion subunit 120 includes N FIFO memories, and buffers N-way signals converted into parallel data. One of the channels where the N paths of data are located is a detection channel, and the rest are auxiliary channels. Arrows in fig. 8 to 13 indicate signal transmission directions.
Further, the system further includes a fifth unit 500 for detecting an amplitude average value of one channel data in a predetermined time as a basis for adjusting the gain of the received signal of the anti-radiation seeker. With reference to fig. 9, the fifth unit 500 includes a threshold sub-unit 510, a comparison sub-unit 520, and a gain adjustment feedback sub-unit 530. The threshold sub-unit 510 sets an upper threshold and a lower threshold according to specific data during system testing; the comparison subunit 520 compares the received signal amplitude with a threshold; the gain adjustment feedback sub-unit 530 feeds back a gain variation according to the compared information, and increases the gain if the amplitude average is smaller than the lower threshold, and decreases the gain if the amplitude average is larger than the upper threshold.
Referring to fig. 10, the second unit 200 includes a delay subunit 210 for inputting N paths of parallel data simultaneously into M paths of channels and sequentially performing delay processing; a polyphase filter subunit 220 for performing polyphase filtering on the M paths of parallel data; an FFT operation subunit 230 that performs FFT operation on the polyphase-filtered signal to obtain a channel frequency; a selection subunit 240 that selects a channel within a certain frequency range as an active channel. An anti-aliasing channelization model is set in the delay subunit 210, where the value of the anti-aliasing channelization model in this embodiment is M of 64, the input data is simultaneously input into 64 paths, and 64-fold extraction is performed after delay according to 0-63 clocks in sequence, so as to obtain 64 paths of parallel data. The polyphase filtering subunit 220 performs polyphase filtering on the M channels of data by using an FIR prototype low-pass filter, where parameters of the FIR prototype low-pass filter in this embodiment are as follows:
passband cutoff frequency: 10.5MHz
Starting frequency of stop band: 11.5MHz
Stop band suppression: 70dBc
Length: 512 steps.
In the FFT operation subunit 230 in this embodiment, N-point complex FFT operation is adopted to implement 2N-point real number FFT operation, and 64-point FFT is performed on the multiphase filtered signal to obtain 64-channel data. In this embodiment, the selection subunit 240 sets 400MHz as the threshold, and the channels with signal frequencies within 400MHz are effective channels.
Referring to fig. 11, the third unit 300 includes a calibration value subunit 310 and a calibration subunit 320. The calibration value subunit 310 obtains calibration values in three working states, where the three working states and the corresponding calibration values are respectively: initial normal calibration values in a darkroom environment, initial closed loop calibration values in a laboratory environment, real-time closed loop calibration values during operation of the anti-radiation seeker. The method for acquiring each calibration value comprises the following steps:
(1) initial normal calibration value: the anti-radiation seeker transmits and receives a normal signal in a darkroom environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
(2) initial closed loop calibration value: the anti-radiation seeker transmits and receives a signal with a specific frequency under a laboratory environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
(3) real-time closed-loop calibration value: and in the working process of the anti-radiation guide head, the anti-radiation guide head transmits and receives a signal with a specific frequency, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained.
The calibration subunit 320 calibrates the signal in the corresponding operating state by subtracting the calibration value from the amplitude phase of the actual signal. The consistency between channels is realized through calibration.
Further, considering that the working environment such as temperature may have an influence on the channel phase, a feedback sub-unit 330 for compensating the channel-to-channel inconsistency during the operation of the anti-radiation seeker by the difference between the initial closed-loop calibration value and the real-time closed-loop calibration value is further provided in the third unit 300. The difference value between the real-time closed-loop calibration value and the initial closed-loop calibration value represents the influence of the environment, and the influence of the environment can be reduced by compensating the value after the phase difference is calculated. This calibration is also implemented by the calibration subunit 320.
With reference to fig. 12, the fourth unit 400 includes a detection subunit 410 for selecting a detection channel and performing threshold detection on the signal amplitude in an effective channel within the detection channel; a parameter acquiring subunit 420 that acquires parameters such as arrival time, pulse width, amplitude, frequency, channel phase difference, and frequency modulation slope of the LFM signal of the detected pulse data; and an output subunit 430 for outputting the signal data of the effective channel according to the order of the arrival time and the channel label from small to large. The monitoring subunit 410 includes a signal detection threshold module 411 and a detection module 412. The signal detection threshold module 411 finds the signal amplitude of each effective channel for the detection channel, and the product of the minimum amplitude value and the preset threshold factor is the signal detection threshold. The detection module 412 compares the amplitude of the signal with a detection threshold, and obtains a pulse according to a comparison result of whether the current signal amplitude is greater than the threshold and whether a decision strategy of continuously passing the threshold by 8 points exists; if the current signal amplitude is larger than the threshold and 8 continuous points pass the threshold, the signal is detected. The method for acquiring each parameter in the parameter acquiring subunit 420 is as follows:
(1) signal arrival time: the system time corresponding to the pulse rising edge mark;
(2) pulse width: the corresponding system time difference between the rising and falling edges of the pulse;
(3) pulse amplitude: the method comprises the steps of performing amplitude accumulation on signals in a detection square wave and then taking an average value to obtain the detection square wave;
(4) pulse frequency: according to the digital phase discrimination method, the phase difference is obtained by measuring the delay phase difference between adjacent sampling points of a detection channel;
(5) channel phase difference: instantaneous phase difference values of each channel and the detection channel;
(6) the FM slope of LFM signal is calculated by using a sliding window method in pulse at 16-point sliding rate, taking the frequency value of each 16 points to calculate the FM slope, and then taking the average value.
Referring to fig. 13, the system is connected to a front-end rf front-end system 600 in a matching manner, where the rf front-end system 600 is configured to control a local oscillator frequency of a mixer of a rf front-end module, and may transmit and receive signals, and control a power amplifier to implement automatic gain control of signal power. Specifically, the rf front-end system 600 includes an rf signal receiving module 610, a down-conversion module 620, and a power amplification module 630. The radio frequency signal receiving module 610 is used for receiving a radio frequency signal of 2-18 GHz; the down-conversion module 620 is used for down-converting the radio-frequency signal of 2-18 GHz to the intermediate frequency of 1.5 GHz; the power amplification module 630 is configured to perform power amplification on the received signal to improve the signal-to-noise ratio.
Fig. 1 is a schematic diagram of an embodiment of a signal processing method based on the above system according to the present invention, and the FPGA-based anti-radiation seeker signal processing method is described with reference to fig. 1.
A method for processing a signal of a radiation-resistant seeker based on an FPGA (field programmable gate array) comprises the following steps of:
step S1, AD sampling is carried out on the signals input by each channel at the same time to obtain corresponding sampling data, and the amplitude of the sampling data is calculated and fed back to the front end of the receiver to be used as the basis of automatic gain control;
step S2, performing anti-aliasing channelization processing on the AD sampling data, and obtaining an effective channel containing an actual signal according to system indexes;
step S3, according to each channel IQ signal outputted by channelization, performing threshold-crossing detection and parameter measurement of the signal, where the measured signal parameters include arrival time, pulse width, pulse amplitude, pulse frequency, LFM signal frequency modulation slope, and phase difference between each channel and the detection channel.
Referring to fig. 2, step S1 includes the following steps:
step S110, AD sampling is carried out on serial data and the serial data are converted into N paths of parallel data;
step S120, realizing time synchronization and phase synchronization of N paths of parallel data;
step S130, selecting one path as a detection channel, calculating the amplitude mean value of the detection channel data in a fixed time, and comparing the amplitude mean value with a threshold value, so as to adjust the gain of the receiver and realize automatic gain control.
The amplitude of the signal is controlled within a suitable interval to facilitate the detection of the signal by the subsequent subunit, via step S1.
In this embodiment, in step S1, a selecteio Interface Wizard IP core of a vendor Xilinx is used to check 1.2Gsps serial data, but the technical solution of the present invention is not limited to the above product to sample data. The step S110 implemented based on the Xilinx product is specifically: and (3) acquiring 1.2Gsps serial data by adopting a selecteIO Interface Wizard IP core of Xilinx, and outputting parallel data of 150Msps after serial-to-parallel conversion. The IP core is realized by adopting special hardware resources such as ISERDESE2 and IDDR on FPGA hardware.
In step S120, N (8 in this embodiment) channels of AD sampled parallel data are buffered by N FIFO memories, and read by a system clock (in this embodiment, a 150MHz system clock is used in step S120 to match the above embodiment) to achieve time synchronization of N channels of signals.
In step S120, phase synchronization of the output data of the N FIFO buffers is realized through the test mode of the AD chip.
In step S130, an upper threshold and a lower threshold are set according to specific data during system testing, and if the amplitude mean value is smaller than the lower threshold, the gain is increased, and if the amplitude mean value is larger than the upper threshold, the gain is decreased.
Referring to fig. 3, step S2 includes the following steps:
step S210, inputting the data of step S1 simultaneously to obtain M paths of parallel data, and obtaining M paths of parallel data after delaying;
step S220, carrying out polyphase filtering on the M paths of parallel data;
step S230, carrying out FFT operation on the multiphase filtered signal according to a classic FFT butterfly graph algorithm to obtain channel data;
step S240, setting a channel with a signal frequency within a range as an effective channel according to the bandwidth of the rf front-end filter.
In this embodiment, in step S210, an anti-aliasing channelization model is constructed, that is, M takes a value of 64, the input data is simultaneously input into 64 channels, and 64-fold extraction is performed after delaying according to 0 to 63 clocks in sequence, so as to obtain 64 channels of parallel data.
In this embodiment, in step S220, a FIR prototype low-pass filter is used to perform multi-phase filtering on the M channels of data, where the parameters of the FIR prototype low-pass filter are as follows
Passband cutoff frequency: 10.5MHz
Starting frequency of stop band: 11.5MHz
Stop band suppression: 70dBc
Length: 512 steps.
In this embodiment, in step S230, an N-point complex FFT operation is adopted to implement a 2N-point real number FFT operation, and a 64-point FFT is performed on the multiphase-filtered signal to obtain 64-channel data.
In this embodiment, step S240 sets 400MHz as the threshold, and the signal with the frequency within 400MHz is the effective channel. In this embodiment, since the rf front end is 400MHz in bandwidth for filtering the rf signal, only 400MHz is effective for 1.2G sampling, and the others are aliased and meaningless for processing, it means processing the effective channel.
With reference to fig. 4, the specific process of step S3 is:
step S310, calibrating the signal processed in the step S2;
step S320, selecting a detection channel and carrying out threshold-crossing detection on the signal amplitude in the effective channel in the detection channel;
step S330, parameters such as the arrival time, the pulse width, the amplitude, the frequency and channel phase difference of the detected pulse data, the frequency modulation slope of the LFM signal and the like are obtained;
step S340, respectively outputting the signal data of the effective channel according to the order from the arrival time and the channel label from small to large.
Theoretically, the phase difference between the channels should be 0, but since the difference in performance of each channel under different operating conditions may cause amplitude-phase errors, the input signal needs to be calibrated. Referring to fig. 5, in step S310, the input signal is calibrated by loading a pre-stored calibration value. The calibration value comprises amplitude phase differences of the auxiliary channel and the reference channel under three working states, such as an implied environment, a laboratory environment, a seeker working process and the like, and the collected corresponding calibration value calibrates signals under the corresponding working state, namely the calibration value is subtracted to realize consistency among the channels. The method for obtaining each calibration value is as follows:
(1) pre-storing an initial normal calibration value in a darkroom environment: the radio frequency front end of the anti-radiation seeker transmits a normal signal, a receiving and transmitting channel of the anti-radiation seeker receives the signal, and the amplitude phase difference between the auxiliary channel and the detection channel is a calibration value;
(2) initial closed loop calibration values in a laboratory environment: controlling the radio-frequency front end of the anti-radiation seeker to transmit a signal with a specific frequency, receiving and transmitting the signal with the specific frequency by a receiving and transmitting channel of the anti-radiation seeker, and calibrating a value which is the amplitude phase difference between an auxiliary channel and a reference channel;
(3) real-time closed-loop calibration values in the working process of the seeker are as follows: before entering a tracking state, a radio frequency front end is controlled to transmit a signal with a specific frequency, a receiving and transmitting channel of the anti-radiation seeker receives the signal with the specific frequency, and the amplitude phase difference between the auxiliary channel and the reference channel is the calibration value.
The real-time closed-loop calibration value mainly considers that the working environment such as temperature has influence on the channel phase, the difference value of the real-time closed-loop calibration value and the initial closed-loop calibration value represents the influence of the environment, and the influence of the environment can be reduced by compensating the value after the phase difference is calculated.
In step S320, the effective channel may be the channel within the 400MHz bandwidth, and 64-point FFT is performed to obtain 64 channels of data, and then 6 to 28 channels are the effective channel. With reference to fig. 6, the specific process of step S320 includes:
step S321, calculating the signal amplitude of each effective channel for the detection channel, wherein the product of the minimum amplitude value and a preset threshold factor is a signal detection threshold;
step S322, comparing the signal amplitude with the detection threshold, and obtaining the pulse according to the comparison result of whether the current signal amplitude is larger than the threshold and whether a decision strategy that 8 continuous points pass the threshold exists; if the current signal amplitude is larger than the threshold and 8 continuous points pass the threshold, the signal is detected.
The method for acquiring each parameter in step S330 is:
(1) signal arrival time: the system time corresponding to the pulse rising edge mark;
(2) pulse width: the corresponding system time difference between the rising and falling edges of the pulse;
(3) pulse amplitude: the method comprises the steps of performing amplitude accumulation on signals in a detection square wave and then taking an average value to obtain the detection square wave;
(4) pulse frequency: according to the digital phase discrimination method, the phase difference is obtained by measuring the delay phase difference between adjacent sampling points of a detection channel;
(5) channel phase difference: instantaneous phase difference values of each channel and the detection channel;
(6) the FM slope of LFM signal is calculated by using a sliding window method in pulse at 16-point sliding rate, taking the frequency value of each 16 points to calculate the FM slope, and then taking the average value.
In step S330, the phase difference of the channels is the phase difference of the signals between the auxiliary channel and the detection channel. In this embodiment, the system has 7 channels, one channel of detection channel, and other 6 auxiliary channels, and the phase difference between the detection channel and the auxiliary channel needs to be calculated.
In step S340, different channel data is buffered by the FIFO buffer.
Claims (10)
1. A method for processing a signal of a radiation-resistant seeker based on an FPGA (field programmable gate array) is characterized by comprising the following steps of:
sampling serial data received by a back radiation seeker and converting the serial data into N paths of parallel data;
obtaining an effective channel according to the sampling data and the frequency threshold;
calibrating the signal amplitude output by the effective channel;
and detecting and measuring parameters of the calibrated signals through a gate.
2. The method of claim 1, wherein an average amplitude value of one channel data in a predetermined time is detected as a basis for adjusting a gain of a received signal of the anti-radiation seeker.
3. The method of claim 1 or 2, wherein deriving the effective channel based on the sampled data and the frequency threshold comprises:
inputting N paths of parallel data into M paths of channels at the same time for delay processing in sequence;
performing multi-phase filtering on the M paths of parallel data;
carrying out FFT operation on the multiphase filtered signals to obtain channel frequency;
a channel within a certain frequency range is selected as an active channel.
4. The method of claim 3, wherein calibrating the signal amplitude of the effective channel output comprises obtaining a calibration value, the calibration value comprising
Initial normal calibration value: the anti-radiation seeker transmits and receives a normal signal in a darkroom environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
initial closed loop calibration value: the anti-radiation seeker transmits and receives a signal with a specific frequency under a laboratory environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
real-time closed-loop calibration value: and in the working process of the anti-radiation guide head, the anti-radiation guide head transmits and receives a signal with a specific frequency, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained.
5. The method of claim 4, wherein calibrating the signal amplitude of the effective channel output further comprises compensating for channel-to-channel inconsistencies during operation of the backscatter seeker using the difference between the initial closed-loop calibration value and the real-time closed-loop calibration value.
6. An anti-radiation seeker signal processing system based on an FPGA, comprising:
a unit for sampling the serial data received by the anti-radiation seeker and converting the serial data into N paths of parallel data;
obtaining a unit of an effective channel according to the sampling data and the frequency threshold;
a unit for calibrating the signal amplitude output by the effective channel;
and a unit for detecting and measuring the parameter of the calibrated signal through the gate.
7. The system of claim 6, further comprising a unit for detecting an average amplitude value of one channel data in a predetermined time as a basis for adjusting the gain of the received signal of the anti-radiation seeker.
8. The system of claim 6 or 7, wherein the means for deriving the effective channel based on the sampled data and the frequency threshold comprises:
inputting N paths of parallel data into M paths of channels simultaneously to be sequentially used as a delay processing subunit;
performing a multi-phase filtering subunit on the M paths of parallel data;
performing FFT operation on the multiphase filtered signals to obtain a channel frequency subunit;
channels within a certain frequency range are selected as valid channel subunits.
9. The system of claim 8 wherein the means for calibrating the signal amplitude of the effective channel output comprises a get calibration value subunit, the calibration value comprising
Initial normal calibration value: the anti-radiation seeker transmits and receives a normal signal in a darkroom environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
initial closed loop calibration value: the anti-radiation seeker transmits and receives a signal with a specific frequency under a laboratory environment, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained;
real-time closed-loop calibration value: and in the working process of the anti-radiation guide head, the anti-radiation guide head transmits and receives a signal with a specific frequency, and then the amplitude phase difference between the auxiliary channel and the detection channel is obtained.
10. The system of claim 9 wherein the means for calibrating the signal amplitude of the effective channel output further comprises a means for compensating for channel-to-channel non-uniformity during operation of the backscatter seeker for differences between the initial closed-loop calibration value and the real-time closed-loop calibration value.
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