CN111585476B - Dual-motor drive system predicted torque control method without weight coefficient - Google Patents

Dual-motor drive system predicted torque control method without weight coefficient Download PDF

Info

Publication number
CN111585476B
CN111585476B CN202010498844.9A CN202010498844A CN111585476B CN 111585476 B CN111585476 B CN 111585476B CN 202010498844 A CN202010498844 A CN 202010498844A CN 111585476 B CN111585476 B CN 111585476B
Authority
CN
China
Prior art keywords
phase
pmsm
voltage
torque
formula
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
CN202010498844.9A
Other languages
Chinese (zh)
Other versions
CN111585476A (en
Inventor
周扬忠
潘斌
陈艳慧
钟天云
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Fuzhou University
Original Assignee
Fuzhou University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Fuzhou University filed Critical Fuzhou University
Priority to CN202010498844.9A priority Critical patent/CN111585476B/en
Publication of CN111585476A publication Critical patent/CN111585476A/en
Application granted granted Critical
Publication of CN111585476B publication Critical patent/CN111585476B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/04—Arrangements for controlling or regulating the speed or torque of more than one motor
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/12—Stator flux based control involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20—Estimation of torque
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters

Abstract

The invention relates to a control method for predicting torque of a dual-motor driving system without a weight coefficient. Providing a six-phase series three-phase double PMSM driving system, obtaining expected voltage and zero sequence voltage expected values of two PMSMs under a static coordinate system through a PI regulator according to information including six-phase current, direct current bus voltage and position angle obtained by system sampling, and further obtaining the expected voltage and zero sequence voltage expected values through the PI regulatorT 6And the inverse transformation of the matrix obtains the expected value of the six-phase voltage, and the 64 switching states are substituted into a cost function based on the six-phase voltage error, so that the switching state with the minimum cost function value acts on the next period of the system. The invention realizes the accurate control of the torque and the stator flux linkage amplitude of the two PMSM and the effective inhibition of the zero sequence current; the complicated weight coefficient setting work is saved; and greatly reduces the amount of calculation of the predicted torque control algorithm.

Description

Dual-motor drive system predicted torque control method without weight coefficient
Technical Field
The invention relates to a control method for predicting torque of a dual-motor driving system without a weight coefficient.
Background
A six-phase series three-phase dual Permanent Magnet Synchronous Motor (PMSM) driving system is a commonly used dual-Motor driving system. The six-phase series three-phase double-permanent-magnet synchronous motor driving system respectively connects U, V, W phases of a three-phase PMSM with AD, BE and CF of the six-phase PMSM, and realizes independent decoupling control of two PMSMs by a single inverter by utilizing the redundant freedom degree of the six-phase PMSM. Compared with the traditional single PMSM controlled by a single inverter, the six-phase series three-phase double PMSM driving system has the advantages of small driving system volume, low cost, easiness in realizing feedback braking and the like, and has wide application prospects in the industries of steel smelting, rewinders, electric automobiles, textile manufacturing and the like.
And (2) based on a motor system prediction model, performing traversal calculation on Torque and stator flux linkage amplitude values of the motor in the next Control period after different voltage vectors in the applied alternative voltage vector set act on the system, thereby obtaining different cost function values, and enabling the voltage vector with the minimum cost function to act on the next Control period of the system as an optimal voltage vector. The predicted torque control has the advantages of simple structure, intuitive realization, quick dynamic response, easy inclusion of constraint conditions and the like, and is a novel advanced control technology.
Compared with the traditional direct torque control method based on a switching vector table, the six-phase series three-phase dual PMSM driving system adopting the predicted torque control has more excellent steady-state characteristics and dynamic characteristics. However, the traditional cost function based on the torque and stator flux linkage amplitude errors needs to traverse and calculate the torque and stator flux linkage amplitude of the next control period of the two PMSM after different voltage vectors in the applied alternative voltage vector set act on the system, and the calculation amount of the predicted torque control algorithm is greatly increased while the control algorithm is complex. Meanwhile, because the torque and stator flux linkage amplitude dimensions and magnitudes are different, a weight coefficient needs to be introduced into the cost function, the setting work of the weight coefficient is complicated and lacks design basis, and the application of the predictive torque control is limited to a certain extent.
In a six-phase series three-phase dual PMSM driving system, the system has 5 degrees of freedom, and 4 degrees of freedom are needed for controlling the torque and the stator flux linkage amplitude of two PMSMs. In the system, if the last 1 degree of freedom is not effectively controlled, the zero sequence current of the system is overlarge. The large zero sequence current can cause the problems of phase current distortion, system loss increase and the like.
Aiming at the problems, the invention provides a predictive torque control method without a weight coefficient for a six-phase series three-phase double PMSM driving system, so that the accurate control of two PMSM torques, stator flux linkage amplitude and effective suppression of zero-sequence current are realized, the weight coefficient in a cost function is eliminated, and the calculated amount of a predictive torque control algorithm is greatly reduced.
Disclosure of Invention
The invention aims to provide a method for controlling the predicted torque of a dual-motor driving system without a weight coefficient, which realizes the accurate control of the torque and the stator flux linkage amplitude of two PMSMs and the effective inhibition of zero-sequence current; eliminating the weight coefficient of the cost function in the predictive torque control algorithm; and thirdly, reducing the calculation amount of the predicted torque control algorithm.
In order to achieve the purpose, the technical scheme of the invention is as follows: a double-motor driving system prediction torque control method without weight coefficients provides a six-phase series three-phase double-PMSM driving system, obtains expected voltage and zero sequence voltage expected values of two PMSMs under a static coordinate system through a PI regulator according to information including six-phase current, direct current bus voltage and position angle obtained by system sampling, and further obtains expected voltage and zero sequence voltage expected values through T6And the inverse transformation of the matrix obtains the expected value of the six-phase voltage, and the 64 switching states are substituted into a cost function based on the six-phase voltage error, so that the switching state with the minimum cost function value acts on the next period of the system.
In an embodiment of the present invention, the method of the present invention is specifically implemented as follows:
step S1, using constant power transformation matrix T6The six-phase current i obtained by samplingA～iFCurrent i transformed into α 1 β 1, α 2 β 2, o1o2 coordinate systemα1、iβ1、iα2、iβ2、io1、io2：
Figure BDA0002523862900000021
Wherein iα1、iβ1、iα2、iβ2、io1、io2Currents on the α 1, β 1, α 2, β 2, o1, o2 axes, respectively; i.e. io1、io2For two zero sequence currents, i is because the neutral point of the three-phase PMSM is not led outo1Is always 0;
step S2, obtaining stator flux psi of the two PMSMs on the static coordinate system according to the stator flux current model or the stator flux voltage modelsα1、ψsβ1、ψsα2、ψsβ2；ψsα1、ψsβ1、ψsα2、ψsβ2Stator flux linkages on the alpha 1, beta 1, alpha 2 and beta 2 axes respectively;
1) if a current model of the stator flux linkage is used, the stator flux linkage psisα1ψsβ1、ψsα2ψsβ2Comprises the following steps:
Figure BDA0002523862900000022
Figure BDA0002523862900000023
wherein psif1、ψf2Permanent magnet flux linkage for two PMSM; l issσ1Is the self leakage inductance, L, of a six-phase PMSM phase windingsm1＝(Ldm1+Lqm1)/2，Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1The permanent magnet synchronous motor is characterized by comprising six-phase PMSM phase windings, a main magnetic flux direct shaft inductor and a quadrature axis inductor respectively; l issσ2Is the self leakage inductance, L, of the three-phase PMSM phase windingsm2＝(Ldm2+Lqm2)/2，Lrs2＝(Ldm2-Lqm2)/2，Ldm2、Lqm2For main flux direct-axis and quadrature-axis inductances, theta, of three-phase PMSM phase windingsr1Is the angle between the d1 axis and the α 1 axis, θr2Is the included angle between the d2 axis and the alpha 2 axis;
2) if a stator is usedVoltage model of flux linkage, stator flux linkage psisα1ψsβ1、ψsα2ψsβ2Comprises the following steps:
Figure BDA0002523862900000031
Figure BDA0002523862900000032
step S3, obtaining two PMSM torques T according to the formula (6) and the formula (7)e1、Te2：
Te1＝p1(ψsα1iβ1-ψsβ1iα1) (6)
Te2＝p2(ψsα2iβ2-ψsβ2iα2) (7)
Wherein p is1Is six-phase PMSM pole pair number, p2The number of pole pairs of the three-phase PMSM is shown; t ise1For six-phase PMSM torque, Te2Three-phase PMSM torque;
step S4, obtaining two PMSM stator flux linkage amplitude psi according to the formula (8) and the formula (9)s1、ψs2：
Figure BDA0002523862900000033
Figure BDA0002523862900000034
Wherein psis1The six-phase PMSM stator flux linkage amplitude value; psis2The amplitude of the flux linkage of the three-phase PMSM stator is obtained;
step S5, according to the formula (10) and the formula (11), the given torque T of the two PMSMs is obtained by the PI regulators controlling the rotating speeds of the two PMSMs* e1、T* e2(ii) a Given value psi of six-phase PMSM stator flux linkage amplitude* s1Is composed of
Figure BDA0002523862900000035
Given value psi of three-phase PMSM stator flux linkage amplitude* s2Is composed of
Figure BDA0002523862900000036
Figure BDA0002523862900000037
Figure BDA0002523862900000038
Wherein, ω isr1、ωr1A given electrical angular velocity and an actual electrical angular velocity for the six-phase PMSM; omegar2、ωr2A given electrical angular velocity and an actual electrical angular velocity for the six-phase PMSM; kvp1、Kvi1Proportional coefficient and integral coefficient for controlling PI regulator of six-phase PMSM; kvp2、Kvi2Proportional coefficients and integral coefficients for controlling a PI regulator of the three-phase PMSM;
step S6, obtaining the torque and stator flux linkage amplitude error delta T of the two PMSMs according to the formulas (12) to (15)e1、Δψs1、ΔTe2、Δψs2：
Figure BDA0002523862900000039
Figure BDA00025238629000000310
Figure BDA0002523862900000041
Figure BDA0002523862900000042
Step S7, adopting fixed parameter PI controller or variable ratio coefficient PI controller to calculate the torque angle variation delta needed for accurately eliminating the two PMSM torque and stator flux linkage amplitude errors1、Δδ2；
1) If a fixed parameter PI controller is used, the amount of change in torque angle Δ δ is calculated according to equations (16) and (17)1、Δδ2Comprises the following steps:
Δδ1＝Kp1ΔTe1+Ki1∫ΔTe1dt (16)
Δδ2＝Kp2ΔTe2+Ki2∫ΔTe2dt (17)
wherein, Kp1、Ki1Proportional coefficient and integral coefficient of PI regulator for controlling six-phase PMSM torque; kp2、Ki2Proportional coefficient and integral coefficient of PI regulator for controlling three-phase PMSM torque;
2) if a variable scale coefficient PI controller is adopted, the calculation model is as follows:
2.1) obtaining the scaling factor coefficients of the two PMSMs according to the formula (18) to the formula (21)
Figure BDA0002523862900000043
Figure BDA0002523862900000044
Figure BDA0002523862900000045
Figure BDA0002523862900000046
Figure BDA0002523862900000047
Wherein, delta1Is a six-phase PMSM torque angle, δ2Is a three-phase PMSM torque angle; l isd1＝Lsσ1+3Lsm1+3Lrs1Is a six-phase PMSM planar d-axis inductor, Lq1＝Lsσ1+3Lsm1-3Lrs1A six-phase PMSM plane q-axis inductor; l isd2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2Is a three-phase PMSM planar d-axis inductor, Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2A three-phase PMSM plane q-axis inductor;
2.2) obtaining the torque angle variation delta needed for accurately eliminating the two PMSM torque and stator flux linkage amplitude errors according to the formula (22) and the formula (23)1、Δδ2：
Figure BDA0002523862900000048
Figure BDA0002523862900000049
2.3) simultaneously, an integral link of the torque error adopts integral separation, and the integral separation only acts when the torque error is small and is used for compensating the inaccuracy of the parameters of the two PMSM;
step S8, obtaining expected values of stator flux linkage variations in α 1 β 1 and α 2 β 2 coordinate systems according to the formulas (24) to (27)
Figure BDA0002523862900000051
Figure BDA0002523862900000052
Figure BDA0002523862900000053
Figure BDA0002523862900000054
Figure BDA0002523862900000055
Figure BDA0002523862900000056
Wherein the content of the first and second substances,
Figure BDA0002523862900000057
the expected value of the variation of the stator flux linkage in the alpha 1 beta 1 coordinate system is obtained;
Figure BDA0002523862900000058
the expected value of the variation of the stator flux linkage in the alpha 2 beta 2 coordinate system is obtained; t issIs a control period; omegar1、ωr2Electrical angular velocities of the six-phase PMSM and the three-phase PMSM, respectively;
step S9, obtaining expected values of stator voltage under the alpha 1 beta 1 and alpha 2 beta 2 coordinate system according to the formula (28) and the formula (29)
Figure BDA0002523862900000059
Figure BDA00025238629000000510
Figure BDA00025238629000000511
Figure BDA00025238629000000512
Wherein the content of the first and second substances,
Figure BDA00025238629000000513
the expected value of the stator voltage under an alpha 1 beta 1 coordinate system;
Figure BDA00025238629000000514
the expected value of the stator voltage under an alpha 2 beta 2 coordinate system;
step S10, obtaining a zero sequence voltage given value u according to a formula (30)* o2：
Figure BDA00025238629000000515
Wherein, Kip、KiiProportional coefficient and integral coefficient of PI regulator for controlling system zero sequence current;
step S11, obtaining the expected value of the system six-phase voltage according to the formula (31)
Figure BDA00025238629000000516
Figure BDA0002523862900000061
Wherein the content of the first and second substances,
Figure BDA0002523862900000062
a six-phase voltage desired value;
step S12 is to obtain six-phase voltage u corresponding to all 64 switching states that the six-phase inverter can output, from the formula (32)AO～uFO：
Figure BDA0002523862900000063
Wherein, UDCIs a DC bus voltage, Si1(i is a to f) represents that the upper tube of the ith phase bridge arm of the inverter is conducted and the lower tube is turned off; otherwise, SiWhen the upper tube of the ith phase bridge arm of the inverter is turned off, the lower tube of the ith phase bridge arm of the inverter is turned on, the upper tube of the ith phase bridge arm of the inverter is turned off;
step S13, obtaining cost function values corresponding to 64 switch states according to the formula (33), and taking the switch state with the minimum cost function value to act on the system in the next control period;
Figure BDA0002523862900000064
compared with the prior art, the invention has the following beneficial effects: 1) the accurate control of the torque and the flux linkage amplitude of the two PMSM and the effective inhibition of the zero-sequence current are realized; 2) the cost function is constructed by directly utilizing the expected value of the six-phase voltage and the error of the six-phase voltage output by the inverter, the weight coefficient is eliminated, and the complicated weight coefficient setting work is omitted; 3) the torque and stator flux linkage amplitude of two PMSMs in the next period after different voltage vectors are acted are not required to be calculated in a traversing mode, the optimal voltage vector is selected by directly utilizing a cost function based on six-phase voltage errors, and the calculation amount of a predictive torque control algorithm is greatly reduced.
Drawings
FIG. 1 is a block diagram of a control strategy according to the present invention.
FIG. 2 is a hardware configuration example of a driving system according to the present invention.
Fig. 3 shows a winding connection mode of a six-phase series three-phase dual PMSM driving system according to the invention.
Fig. 4 is a six-phase PMSM plane.
Fig. 5 is a three-phase PMSM plane.
Fig. 6 is a calculation of flux linkage variation in a six-phase PMSM plane.
Fig. 7 shows the flux linkage variation calculation of the stator in the plane of the three-phase PMSM.
FIG. 8 is a flow chart of the predicted torque control algorithm of the present invention.
Detailed Description
The technical scheme of the invention is specifically explained below with reference to the accompanying drawings.
The invention provides a control method for predicting torque of a dual-motor driving system without a weight coefficient. The purpose has three aspects: firstly, the accurate control of the torque and the stator flux linkage amplitude of two PMSM and the effective inhibition of zero sequence current are realized; eliminating the weight coefficient of the cost function in the predictive torque control algorithm; thirdly, the calculated amount of the predicted torque control algorithm is reduced. According to information such as six-phase current, direct current bus voltage, position angle and the like obtained by system sampling, expected voltage and zero sequence voltage expected values of two PMSMs under a static coordinate system are obtained through a PI (proportional-integral) regulator and then are subjected to T6The inverse transformation of the matrix results in the desired values for the six-phase voltages. And substituting 64 switching states into a cost function based on the six-phase voltage error, and enabling the switching state with the minimum cost function value to act on the next system period. The specific explanation is as follows. The structure block diagram of the system and the control strategy provided by the invention is shown in figure 1. Using T in equation 26Matrix, six-phase current i obtained by samplingA～iFConverted into currents i on the coordinate systems of alpha 1 beta 1, alpha 2 beta 2 and o1o2α1iβ1、iα2iβ2、 io1io2(ii) a Obtaining the stator flux linkage psi of the two PMSMs on the static coordinate system according to the stator flux linkage current models of the formula 3 and the formula 4 or the stator flux linkage voltage models of the formula 6 and the formula 7sα1ψsβ1、ψsα2ψsβ2. Obtaining two PMSM torques T according to the formula 9 and the formula 10e1、Te2(ii) a Obtaining two PMSM stator flux linkage amplitude psi according to formula 11 and formula 12s1、ψs2(ii) a According to the formula 41 and the formula 42, given torques T of the two PMSMs are obtained by a PI regulator for controlling the rotating speeds of the two PMSMs* e1、T* e2(ii) a Given value psi of six-phase PMSM stator flux linkage amplitude* s1Is composed of
Figure BDA0002523862900000071
Given value psi of three-phase PMSM stator flux linkage amplitude* s2Is composed of
Figure BDA0002523862900000072
Obtaining the torque and stator flux linkage amplitude error delta T of the two PMSMs according to the formulas 45 to 48e1、Δψs1、ΔTe2、Δψs2(ii) a According to the formula 49 and the formula 50, a PI controller with fixed parameters is adopted to obtain the torque angle variation delta needed for calculating and accurately eliminating the two PMSM torque and stator flux linkage amplitude errors1、Δδ2The delta can also be obtained by using a better proportional coefficient PI controller1、Δδ2: obtaining the scaling factor of the two PMSMs according to the formula 37 to the formula 40
Figure BDA0002523862900000076
Obtaining Δ δ according to equations 43 and 441、Δδ2Meanwhile, an integral link of the torque error adopts integral separation, and only acts when the torque error is small, so that the integral link is used for compensating the inaccuracy of the two PMSM parameters; obtaining expected values of stator flux linkage variation under the coordinate systems of alpha 1 beta 1 and alpha 2 beta 2 according to the formulas 51 to 54
Figure BDA0002523862900000073
Obtaining expected values of the stator voltage under the coordinate systems of alpha 1 beta 1 and alpha 2 beta 2 according to the formula 55 and the formula 56
Figure BDA0002523862900000074
Obtaining the given value u of the zero sequence voltage according to the formula 57* o2(ii) a The expected value of the system six-phase voltage is obtained according to equation 58
Figure BDA0002523862900000075
Obtaining six-phase voltage u corresponding to 64 switching states all of which can be output by the six-phase inverter according to formula 61AO～uFO(ii) a And obtaining cost function values corresponding to 64 switch states according to a formula 62, and taking the switch state with the minimum cost function value to act on the system in the next control period.
An example of a hardware configuration of a drive system embodying the present invention is shown in fig. 2. The method comprises the following steps: the device comprises a voltage regulator, a three-phase uncontrollable rectifying circuit, a large filtering capacitor, a direct current bus voltage detection circuit, a six-phase inverter, an isolation driving circuit, a six-phase winding current detection circuit, a six-phase PMSM, a three-phase PMSM, an encoder, a DSP, a CPLD, a human-computer interaction interface, a fault protection circuit, an AD conditioning circuit and the like. The power tube in the inverter adopts IGBT or MOSFET. The six-phase winding current detection circuit consists of a Hall current sensor and an operational amplifier circuit, and an output signal is input into the DSP through the AD conditioning circuit. The direct current bus voltage detection circuit consists of a Hall voltage sensor and an operational amplifier circuit, and an output signal also needs to be input into the DSP through the AD conditioning circuit. The detection signals of the six-phase winding current and the direct-current bus voltage also need to pass through the fault protection circuit, when the direct-current bus voltage and the winding current are abnormal, the fault protection circuit outputs signals to the CPLD, and the CPLD blocks the PWM output signals to turn off all the switch tubes. The rotor position angles of the two PMSM's are detected by two incremental photoelectric encoders. The six-phase winding current detection signal, the direct current bus voltage detection signal and the rotor position angle signals of the two PMSMs are input into the DSP, the DSP outputs a control signal of the switching tube according to the detected signal and the control strategy of the invention, then the CPLD detects whether the control signal can cause the switching tube to generate direct connection danger or not, and if not, the CPLD outputs the control signal to the isolation driving circuit to control the power switching tube in the inverter to act so as to control the two PMSMs.
The connection mode of the six-phase series three-phase dual-PMSM driving system windings is shown in figure 3, wherein the A-F phases are six-phase PMSM phase windings, the U-W phases are three-phase PMSM phase windings, and the windings of the two PMSM phases are symmetrically distributed in space. The three-phase PMSM adopts a star connection mode. U, V, W phases of the three-phase PMSM are connected to AD, BE, and CF of the six-phase PMSM, respectively. The power current components of the six-phase PMSM are mutually offset due to opposite phases at the connection part of the six-phase PMSM and the three-phase PMSM winding, so that the three-phase PMSM is not influenced; the power current component of the three-phase PMSM equally flows through the two-phase windings with opposite phases of the six-phase PMSM, and has no influence on the six-phase PMSM, so that the decoupling control of the two PMSMs is realized.
Fig. 4 and 5 are plane definitions of electromechanical energy conversion achieved by six-phase PMSM and three-phase PMSM. In fig. 4, α 1 β 1 and d1q1 coordinate systems are a stationary coordinate system and a synchronous rotating coordinate system of a six-phase PMSM plane, a to F are axes of windings of respective phases of the six-phase PMSM, respectively, and θr1Is the angle between the d1 axis and the α 1 axis, δ1Is the torque angle, ωr1、us1、is1、ψs1、ψf1The electrical angular velocity, the stator voltage vector, the stator current vector, the stator flux linkage vector and the rotor flux linkage vector of the six-phase PMSM are respectively. Three-phase PMSM plane variable definition in FIG. 5 and FIG. 4And (6) like. In addition, the system has 1 zero sequence which does not participate in electromechanical energy conversion, and the zero sequence is called a zero sequence plane.
According to the connection mode of the two PMSM windings in fig. 3, the expression of the output voltage of the six-phase inverter can be obtained as follows:
Figure BDA0002523862900000091
wherein u isAO～uFOThe voltages from the A-F output ends of the six-phase inverter, namely the A-F phase winding input ends of the six-phase PMSM, to the neutral point O of the three-phase PMSM are respectively referred to as A-F phase voltages; i.e. iA～iFPhase current for a six-phase PMSM; rs1Resistance, R, of each phase winding of a six-phase PMSMs2The resistance of each phase winding of the three-phase PMSM; psisA～ψsFEach phase of stator flux linkage is six-phase PMSM; psisU～ψsWThe magnetic flux linkage is a stator magnetic flux linkage of each phase of the three-phase PMSM.
Constant power transformation matrix T using equation 26And transforming the mathematical model of the dual PMSM driving system from an ABCDEF natural coordinate system to an alpha 1 beta 1 alpha 2 beta 2o1o2 static coordinate system.
Figure BDA0002523862900000092
The flux linkage model under the coordinate systems of alpha 1 beta 1, alpha 2 beta 2 and o1o2 is as follows:
Figure BDA0002523862900000093
Figure BDA0002523862900000094
Figure BDA0002523862900000095
wherein psisα1、ψsβ1、ψsα2、ψsβ2、ψso1、ψso2Stator flux linkages on the axes α 1, β 1, α 2, β 2, o1, o 2; i.e. iα1、 iβ1、iα2、iβ2、io1、io2The current on the axes of alpha 1, beta 1, alpha 2, beta 2, o1 and o 2; psif1Permanent magnet flux linkage, psi, for a six-phase PMSMf2A permanent magnet flux linkage of a three-phase PMSM; l issσ1Is the self leakage inductance, L, of a six-phase PMSM phase windingsm1＝(Ldm1+Lqm1)/2， Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1The permanent magnet synchronous motor is characterized by comprising six-phase PMSM phase windings, a main magnetic flux direct shaft inductor and a quadrature axis inductor respectively; l issσ2Is the self leakage inductance, L, of the three-phase PMSM phase windingsm2＝(Ldm2+Lqm2)/2，Lrs2＝(Ldm2-Lqm2)/2，Ldm2、Lqm2The three-phase PMSM phase winding is provided with a main magnetic flux direct-axis inductor and a quadrature-axis inductor.
The voltage model under the coordinate systems of alpha 1 beta 1, alpha 2 beta 2 and o1o2 is as follows:
Figure BDA0002523862900000101
Figure BDA0002523862900000102
Figure BDA0002523862900000103
wherein u isα1、uβ1、uα2、uβ2、uo1、uo2Voltages on the α 1, β 1, α 2, β 2, o1, o2 axes.
The torques of the two PMSM are:
Te1＝p1(ψsα1iβ1-ψsβ1iα1) (formula 9)
Te2＝p2(ψsα2iβ2-ψsβ2iα2) (formula 10)
Wherein, Te1For six-phase PMSM torque, Te2Three-phase PMSM torque; p is a radical of1Is six-phase PMSM pole pair number, p2Is the three-phase PMSM pole pair number.
The stator flux linkage amplitudes of the two PMSMs are respectively as follows:
Figure BDA0002523862900000104
Figure BDA0002523862900000105
wherein psis1The six-phase PMSM stator flux linkage amplitude value; psis2The three-phase PMSM stator flux linkage amplitude.
The quantities in the α 1 β 1 coordinate system are transformed to the d1q1 coordinate system using the six-phase PMSM plane rotation transformation matrix R (θ R1) of equation 13, and the quantities in the α 2 β 2 coordinate system are transformed to the d2q2 coordinate system using the three-phase PMSM plane rotation transformation matrix R (θ R2) of equation 14.
Figure BDA0002523862900000106
Figure BDA0002523862900000107
The flux linkage model under the d1q1 and d2q2 coordinate systems is as follows:
Figure BDA0002523862900000108
Figure BDA0002523862900000111
wherein psisd1、ψsq1、ψsd2、ψsq2Stator flux linkages on d1, q1, d2 and q2 axes; i.e. id1、iq1、id2、iq2Currents on d1, q1, d2 and q2 axes; l isd1＝Lsσ1+3Lsm1+3Lrs1Is a six-phase PMSM planar d-axis inductor, Lq1＝Lsσ1+3Lsm1-3Lrs1A six-phase PMSM plane q-axis inductor; l isd2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2Is a three-phase PMSM planar d-axis inductor, Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2Is a three-phase PMSM plane q-axis inductor.
The voltage model under the d1q1 and d2q2 coordinate systems is as follows:
Figure BDA0002523862900000112
Figure BDA0002523862900000113
wherein u isd1、uq1、ud2、uq2Voltages on d1, q1, d2 and q2 axes.
The torques of the two PMSM are:
Te1＝p1(ψsd1iq1-ψsq1id1) (formula 19)
Te2＝p2(ψsd2iq2-ψsq2id2) (formula 20)
The stator flux linkage amplitudes of the two PMSMs are respectively as follows:
Figure BDA0002523862900000114
Figure BDA0002523862900000115
as can be seen from fig. 4 and 5, the stator flux linkage in the d1q1 and d2q2 coordinate systems can be further represented as:
ψsd1＝ψs1cosδ1(formula 23)
ψsq1＝ψs1sinδ1(formula 24)
ψsd2＝ψs2cosδ2(equation 25)
ψsq2＝ψs2sinδ2(formula 26)
Wherein the content of the first and second substances,
Figure BDA0002523862900000116
Figure BDA0002523862900000117
wherein, thetas1The angle of the six-phase PMSM stator flux linkage vector is shown; thetas2Is the angle of the flux linkage vector of the three-phase PMSM stator.
Substituting equations 23 and 24 into 15 and equations 25 and 26 into 16 yields:
Figure BDA0002523862900000121
Figure BDA0002523862900000122
Figure BDA0002523862900000123
Figure BDA0002523862900000124
substituting equations 29 and 30 into equation 19 and equations 31 and 32 into equation 20, the torques of the two PMSM are:
Figure BDA0002523862900000125
Figure BDA0002523862900000126
the differential with respect to time is obtained on both sides of equation 33 and equation 34:
Figure BDA0002523862900000127
Figure BDA0002523862900000128
wherein the content of the first and second substances,
Figure BDA0002523862900000129
Figure BDA00025238629000001210
Figure BDA00025238629000001211
Figure BDA00025238629000001212
according to the formulas 37 to 40, the scaling coefficients of the two PMSMs are shown
Figure BDA00025238629000001213
Related to the operating state of the two PMSM's. When the amplitude of the stator flux linkage isAt the given value of the time,
Figure BDA00025238629000001214
the torque loop can adopt a variable proportion type PI regulator in order to accelerate the torque response of the system in the dynamic process.
According to the formula 41 and the formula 42, given torques T of the two PMSMs are obtained by a PI regulator for controlling the rotating speeds of the two PMSMs* e1、T* e2(ii) a Given value psi of six-phase PMSM stator flux linkage amplitude* s1Is composed of
Figure BDA0002523862900000136
Given value psi of three-phase PMSM stator flux linkage amplitude* s2Is composed of
Figure BDA0002523862900000137
Figure BDA0002523862900000131
Figure BDA0002523862900000132
Wherein, ω isr1、ωr1A given electrical angular velocity and an actual electrical angular velocity for the six-phase PMSM; omegar2、ωr2A given electrical angular velocity and an actual electrical angular velocity of the six-phase PMSM. Kvp1、Kvi1Proportional coefficient and integral coefficient for controlling PI regulator of six-phase PMSM; kvp2、Kvi2To control the proportionality and integral coefficients of a PI regulator of a three-phase PMSM.
Obtaining the torque angle variation quantity required when the two PMSM torques and the stator flux linkage amplitude errors can be accurately eliminated according to the formula 35 and the formula 36:
Figure BDA0002523862900000133
Figure BDA0002523862900000134
wherein the content of the first and second substances,
Figure BDA0002523862900000135
Figure BDA0002523862900000138
Figure BDA0002523862900000139
Figure BDA00025238629000001310
wherein, Delta Te1、Δψs1The torque error and the stator flux linkage amplitude error of the six-phase PMSM are obtained; delta Te2、Δψs2The torque error and the stator flux linkage amplitude error of the three-phase PMSM are shown.
Meanwhile, an integral link of the torque error adopts integral separation, and the integral separation only acts when the torque error is small and is used for compensating the inaccuracy of the parameters of the two PMSM.
In addition, Δ δ is simplified if necessary1、Δδ2The calculation process of (2) can also adopt a fixed parameter PI controller, the given value is a given torque value, the feedback value is an actual torque value, namely:
Δδ1＝Kp1ΔTe1+Ki1∫ΔTe1dt (equation 49)
Δδ2＝Kp2ΔTe2+Ki2∫ΔTe2dt (equation 50)
Wherein, Kp1、Ki1Proportional coefficient and integral coefficient of PI regulator for controlling six-phase PMSM torque; kp2、Ki2Proportional coefficient and integral coefficient of PI regulator for controlling six-phase PMSM torque;
schematic diagrams for calculating flux linkage variation of the two PMSM in-plane stators are shown in FIGS. 6 and 7. The expected values of the stator flux linkage variation under the coordinate systems of alpha 1 beta 1 and alpha 2 beta 2 are as follows:
Figure BDA0002523862900000141
Figure BDA0002523862900000142
Figure BDA0002523862900000143
Figure BDA0002523862900000144
wherein the content of the first and second substances,
Figure BDA0002523862900000145
the expected value of the variation of the stator flux linkage in the alpha 1 beta 1 coordinate system is obtained;
Figure BDA0002523862900000146
the expected value of the variation of the stator flux linkage in the alpha 2 beta 2 coordinate system is obtained; t issIs a control cycle.
In order to make the stator flux linkage variation amount track the expected value in the α 1 β 1 and α 2 β 2 coordinate systems, the expected value of the stator voltage in the α 1 β 1 and α 2 β 2 coordinate systems is obtained according to equations 6 and 7 as follows:
Figure BDA0002523862900000147
Figure BDA0002523862900000148
wherein the content of the first and second substances,
Figure BDA0002523862900000149
the expected value of the stator voltage under an alpha 1 beta 1 coordinate system;
Figure BDA00025238629000001410
the expected value of the stator voltage under the alpha 2 beta 2 coordinate system is obtained.
The leakage inductance of the six-phase PMSM is very small, very small zero-sequence voltage can generate very small zero-sequence current, the existence of the zero-sequence current can not only influence the THD of the six-phase current, but also cause the system loss to be increased, and the overall efficiency of the double-PMSM driving system is reduced. Introducing a zero-sequence current PI regulator, setting value as 0, feedback value as zero-sequence current value, and outputting as zero-sequence voltage setting value u* o2. When the zero sequence voltage output by the inverter is the value, the zero sequence current is controlled to be 0.
Figure BDA00025238629000001411
Wherein, Kip、KiiProportional coefficient and integral coefficient of PI regulator for controlling system zero sequence current.
In summary, the expected values of the six-phase voltages of the system are:
Figure BDA0002523862900000151
wherein the content of the first and second substances,
Figure BDA0002523862900000155
the six-phase voltage desired value.
As can be taken from fig. 3, the six-phase voltages can also be represented as:
Figure BDA0002523862900000152
wherein, UDCIs a DC bus voltage uNORepresenting the voltage from DC bus voltage ground N to neutral O, Si1(i is a to f) represents that the upper tube of the ith phase bridge arm of the inverter is conducted and the lower tube is turned off; otherwise, SiAnd 0 represents that the upper tube of the ith phase bridge arm of the inverter is turned off and the lower tube of the ith phase bridge arm of the inverter is turned on.
Since the neutral point of the system is not led out, u isAO+uBO+uCO+uDO+uEO+uFOWhen 0 is obtained, formula 59 is substituted to obtain:
Figure BDA0002523862900000153
substituting equation 60 into equation 59 yields:
Figure BDA0002523862900000154
according to a traditional cost function based on torque and stator flux linkage amplitude errors, two PMSM torque and stator flux linkage amplitude in the next period after different voltage vectors in an alternative voltage vector set are applied need to be calculated in a traversing mode according to prediction models of the two PMSM, the calculation amount is huge, and meanwhile, the weight coefficient setting work inevitably existing in the cost function is troublesome. Aiming at the problems, the cost function based on the six-phase voltage error is established according to the formula 62, the weight coefficient in the traditional cost function is eliminated by the cost function, and the complicated weight coefficient setting work is omitted; meanwhile, the cost does not need to predict the torque and the stator flux linkage amplitude of the next period of the two PMSMs, and the calculated amount of a predicted torque control algorithm is greatly reduced.
Figure BDA0002523862900000161
The six-phase inverter can output 64 different switching states, the different switching states correspond to different six-phase voltage values, and according to the formula 61, all the switching state pairs which can be output by the six-phase inverter can be obtainedCorresponding six-phase voltage uAO～uFO. And substituting the 64 groups of six-phase voltage values into a cost function for calculation, wherein the switching state with the minimum cost function value is the optimal switching state, and the optimal switching state acts on the next period of the system.
A flow chart of the proposed predicted torque control algorithm is shown in fig. 8.
The above are preferred embodiments of the present invention, and all changes made according to the technical scheme of the present invention that produce functional effects do not exceed the scope of the technical scheme of the present invention belong to the protection scope of the present invention.

Claims (1)

1. A double-motor driving system prediction torque control method without weight coefficients is characterized in that a six-phase series three-phase double-PMSM driving system is provided, according to information including six-phase current, direct-current bus voltage and position angle obtained by system sampling, expected voltage and zero-sequence voltage expected values of two PMSMs under a static coordinate system are obtained through a PI regulator, and then the expected voltage and the zero-sequence voltage expected values are subjected to T6Obtaining an expected value of the six-phase voltage through inverse transformation of the matrix, substituting 64 switching states into a cost function based on six-phase voltage errors, and enabling the switching state with the minimum cost function value to act on the next period of the system; the method is concretely realized as follows:
step S1, using constant power transformation matrix T6The six-phase current i obtained by samplingA～iFConverted into currents i on the coordinate systems of alpha 1 beta 1, alpha 2 beta 2 and o1o2α1、iβ1、iα2、iβ2、io1、io2：
Figure FDA0003176887220000011
Wherein iα1、iβ1、iα2、iβ2、io1、io2Currents on the α 1, β 1, α 2, β 2, o1, o2 axes, respectively; i.e. io1、io2Two zero sequence currents, because the neutral point of the three-phase PMSM is not introducedGo out of this pointo1Is always 0;
step S2, obtaining stator flux psi of the two PMSMs on the static coordinate system according to the stator flux current model or the stator flux voltage modelsα1、ψsβ1、ψsα2、ψsβ2；ψsα1、ψsβ1、ψsα2、ψsβ2Stator flux linkages on the alpha 1, beta 1, alpha 2 and beta 2 axes respectively;
1) if a current model of the stator flux linkage is used, the stator flux linkage psisα1ψsβ1、ψsα2ψsβ2Comprises the following steps:
Figure FDA0003176887220000012
Figure FDA0003176887220000013
wherein psif1、ψf2Permanent magnet flux linkage for two PMSM; l issσ1Is the self leakage inductance, L, of a six-phase PMSM phase windingsm1＝(Ldm1+Lqm1)/2，Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1The permanent magnet synchronous motor is characterized by comprising six-phase PMSM phase windings, a main magnetic flux direct shaft inductor and a quadrature axis inductor respectively; l issσ2Is the self leakage inductance, L, of the three-phase PMSM phase windingsm2＝(Ldm2+Lqm2)/2，Lrs2＝(Ldm2-Lqm2)/2，Ldm2、Lqm2For main flux direct-axis and quadrature-axis inductances, theta, of three-phase PMSM phase windingsr1Is the angle between the d1 axis and the α 1 axis, θr2Is the included angle between the d2 axis and the alpha 2 axis;
2) if a voltage model of the stator flux linkage is used, the stator flux linkage psisα1ψsβ1、ψsα2ψsβ2Comprises the following steps:
Figure FDA0003176887220000014
Figure FDA0003176887220000021
step S3, obtaining two PMSM torques T according to the formula (6) and the formula (7)e1、Te2：
Te1＝p1(ψsα1iβ1-ψsβ1iα1) (6)
Te2＝p2(ψsα2iβ2-ψsβ2iα2) (7)
Wherein p is1Is six-phase PMSM pole pair number, p2The number of pole pairs of the three-phase PMSM is shown; t ise1For six-phase PMSM torque, Te2Three-phase PMSM torque;
step S4, obtaining two PMSM stator flux linkage amplitude psi according to the formula (8) and the formula (9)s1、ψs2：
Figure FDA0003176887220000022
Figure FDA0003176887220000023
Wherein psis1The six-phase PMSM stator flux linkage amplitude value; psis2The amplitude of the flux linkage of the three-phase PMSM stator is obtained;
step S5, according to the formula (10) and the formula (11), the given torque T of the two PMSMs is obtained by the PI regulators controlling the rotating speeds of the two PMSMs* e1、T* e2(ii) a Given value psi of six-phase PMSM stator flux linkage amplitude* s1Is composed of
Figure FDA0003176887220000024
Three-phase PMSM decidesGiven value psi of amplitude of sub flux linkage* s2Is composed of
Figure FDA0003176887220000025
Figure FDA0003176887220000026
Figure FDA0003176887220000027
Wherein, ω isr1、ωr1A given electrical angular velocity and an actual electrical angular velocity for the six-phase PMSM; omegar2、ωr2A given electrical angular velocity and an actual electrical angular velocity for the six-phase PMSM; kvp1、Kvi1Proportional coefficient and integral coefficient for controlling PI regulator of six-phase PMSM; kvp2、Kvi2Proportional coefficients and integral coefficients for controlling a PI regulator of the three-phase PMSM;
step S6, obtaining the torque and stator flux linkage amplitude error delta T of the two PMSMs according to the formulas (12) to (15)e1、Δψs1、ΔTe2、Δψs2：
Figure FDA0003176887220000028
Figure FDA0003176887220000029
Figure FDA00031768872200000210
Figure FDA00031768872200000211
Step S7, adopting fixed parameter PI controller or variable ratio coefficient PI controller to calculate the torque angle variation delta needed for accurately eliminating the two PMSM torque and stator flux linkage amplitude errors1、Δδ2；
1) If a fixed parameter PI controller is used, the amount of change in torque angle Δ δ is calculated according to equations (16) and (17)1、Δδ2Comprises the following steps:
Δδ1＝Kp1ΔTe1+Ki1∫ΔTe1dt (16)
Δδ2＝Kp2ΔTe2+Ki2∫ΔTe2dt (17)
wherein, Kp1、Ki1Proportional coefficient and integral coefficient of PI regulator for controlling six-phase PMSM torque; kp2、Ki2Proportional coefficient and integral coefficient of PI regulator for controlling three-phase PMSM torque;
2) if a variable scale coefficient PI controller is adopted, the calculation model is as follows:
2.1) obtaining the scaling factor coefficients of the two PMSMs according to the formula (18) to the formula (21)
Figure FDA0003176887220000031
Figure FDA0003176887220000032
Figure FDA0003176887220000033
Figure FDA0003176887220000034
Figure FDA0003176887220000035
Wherein, delta1Is a six-phase PMSM torque angle, δ2Is a three-phase PMSM torque angle; l isd1＝Lsσ1+3Lsm1+3Lrs1Is a six-phase PMSM planar d-axis inductor, Lq1＝Lsσ1+3Lsm1-3Lrs1A six-phase PMSM plane q-axis inductor; l isd2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2Is a three-phase PMSM planar d-axis inductor, Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2A three-phase PMSM plane q-axis inductor;
2.2) obtaining the torque angle variation delta needed for accurately eliminating the two PMSM torque and stator flux linkage amplitude errors according to the formula (22) and the formula (23)1、Δδ2：
Figure FDA0003176887220000036
Figure FDA0003176887220000037
2.3) simultaneously, an integral link of the torque error adopts integral separation, and the integral separation only acts when the torque error is small and is used for compensating the inaccuracy of the parameters of the two PMSM;
step S8, obtaining expected values of stator flux linkage variations in α 1 β 1 and α 2 β 2 coordinate systems according to the formulas (24) to (27)
Figure FDA0003176887220000038
Figure FDA0003176887220000041
Figure FDA0003176887220000042
Figure FDA0003176887220000043
Figure FDA0003176887220000044
Figure FDA0003176887220000045
Wherein the content of the first and second substances,
Figure FDA0003176887220000046
the expected value of the variation of the stator flux linkage in the alpha 1 beta 1 coordinate system is obtained;
Figure FDA0003176887220000047
the expected value of the variation of the stator flux linkage in the alpha 2 beta 2 coordinate system is obtained; t issIs a control period;
step S9, obtaining expected values of stator voltage under the alpha 1 beta 1 and alpha 2 beta 2 coordinate system according to the formula (28) and the formula (29)
Figure FDA0003176887220000048
Figure FDA0003176887220000049
Figure FDA00031768872200000410
Figure FDA00031768872200000411
Wherein the content of the first and second substances,
Figure FDA00031768872200000412
the expected value of the stator voltage under an alpha 1 beta 1 coordinate system;
Figure FDA00031768872200000413
the expected value of the stator voltage under an alpha 2 beta 2 coordinate system;
step S10, obtaining a zero sequence voltage given value u according to a formula (30)* o2：
Figure FDA00031768872200000414
Wherein, Kip、KiiProportional coefficient and integral coefficient of PI regulator for controlling system zero sequence current;
step S11, obtaining the expected value of the system six-phase voltage according to the formula (31)
Figure FDA00031768872200000415
Figure FDA0003176887220000051
Wherein the content of the first and second substances,
Figure FDA0003176887220000052
a six-phase voltage desired value;
step S12 is to obtain six-phase voltage u corresponding to all 64 switching states that the six-phase inverter can output, from the formula (32)AO～uFO：
Figure FDA0003176887220000053
Wherein, UDCIs a DC bus voltage, Si1 (i-a-f) represents the upper tube of the ith bridge arm of the inverterConducting, and switching off the lower tube; otherwise, SiWhen the upper tube of the ith phase bridge arm of the inverter is turned off, the lower tube of the ith phase bridge arm of the inverter is turned on, the upper tube of the ith phase bridge arm of the inverter is turned off;
step S13, obtaining cost function values corresponding to 64 switch states according to the formula (33), and taking the switch state with the minimum cost function value to act on the system in the next control period;
Figure FDA0003176887220000054
CN202010498844.9A 2020-06-04 2020-06-04 Dual-motor drive system predicted torque control method without weight coefficient Active CN111585476B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202010498844.9A CN111585476B (en) 2020-06-04 2020-06-04 Dual-motor drive system predicted torque control method without weight coefficient

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202010498844.9A CN111585476B (en) 2020-06-04 2020-06-04 Dual-motor drive system predicted torque control method without weight coefficient

Publications (2)

Publication Number Publication Date
CN111585476A CN111585476A (en) 2020-08-25
CN111585476B true CN111585476B (en) 2021-09-28

Family

ID=72125909

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202010498844.9A Active CN111585476B (en) 2020-06-04 2020-06-04 Dual-motor drive system predicted torque control method without weight coefficient

Country Status (1)

Country Link
CN (1) CN111585476B (en)

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114142784B (en) * 2021-07-08 2023-11-21 福州大学 Low-calculation-amount dual-motor series system model prediction torque control method

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN105743404A (en) * 2016-03-23 2016-07-06 华中科技大学 Model prediction control method for four-quadrant induction motor driving system
CN106505927A (en) * 2016-12-26 2017-03-15 西南交通大学 A kind of five-phase PMSM finite aggregate model prediction current control method
CN107196571A (en) * 2017-07-13 2017-09-22 福州大学 A kind of bi-motor series connection forecasting type Direct Torque Control

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7965052B2 (en) * 2008-07-22 2011-06-21 Pratt & Whitney Canada Corp. Motor driving system and method for starting a motor
JP4915439B2 (en) * 2009-08-05 2012-04-11 株式会社デンソー Rotating machine control device
US9667232B2 (en) * 2015-05-13 2017-05-30 Raytheon Company System and method for parallel configuration of hybrid energy storage module
NO342168B1 (en) * 2016-11-04 2018-04-09 Diinef As A method for controlling torque equilibrium of a hydraulic motor

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN105743404A (en) * 2016-03-23 2016-07-06 华中科技大学 Model prediction control method for four-quadrant induction motor driving system
CN106505927A (en) * 2016-12-26 2017-03-15 西南交通大学 A kind of five-phase PMSM finite aggregate model prediction current control method
CN107196571A (en) * 2017-07-13 2017-09-22 福州大学 A kind of bi-motor series connection forecasting type Direct Torque Control

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
六相串联三相双永磁同步电机直接转矩控制的研究;黄志坡 等;《微电机》;20181231;第51卷(第12期);32-41 *
多相永磁电机容错型驱动控制研究;陈光团;《中国优秀硕士学位论文全文数据库 工程科技Ⅱ辑》;20200315(第03期);正文第33-59页 *

Also Published As

Publication number Publication date
CN111585476A (en) 2020-08-25

Similar Documents

Publication Publication Date Title
CN111585477B (en) Voltage vector preselection-based dual-motor drive system predicted torque control method
CN103441726B (en) Based on the double three-phase permanent-magnetic motor vector control method of ratio resonant regulator
CN107196571B (en) Double-motor series prediction type direct torque control method
CN107196543A (en) Common DC bus opens winding asynchronous electric system zero sequence circulation inhibition method
CN107231111B (en) Direct torque control method for high load capacity of five-phase permanent magnet synchronous motor
WO2008038338A1 (en) Permanent magnet synchronization motor vector control device
KR20050083958A (en) Motor and drive control device therefor
CN109347386B (en) SVPWM-based five-phase permanent magnet motor maximum torque current ratio fault-tolerant control method
CN110034719B (en) Control method for phase-lack fault of six-phase single-winding bearingless flux switching motor
CN112117941A (en) Fault-tolerant control method of open-winding permanent magnet synchronous motor based on model prediction current control
CN111431453A (en) Fault-tolerant control method for open-winding permanent magnet synchronous motor based on single-phase fault of inverter
Dyanamina et al. Adaptive neuro fuzzy inference system based decoupled control for neutral point clamped multi level inverter fed induction motor drive
CN108512473B (en) Direct torque control method for three-phase four-switch permanent magnet synchronous motor speed regulation system
CN111800050B (en) Permanent magnet synchronous motor three-vector model prediction torque control method based on voltage vector screening and optimization
CN106487308A (en) The input of series electric motor drive system lacks a phase error-tolerance type Direct Torque Control
CN105790666A (en) Brushless direct current motor direct torque control system and method based on Hall signals
CN111464099B (en) Control method for low torque and suspension force of single-winding bearingless flux switching motor
CN108429491B (en) Fault-tolerant control system and method for double permanent magnet synchronous motors
CN111585476B (en) Dual-motor drive system predicted torque control method without weight coefficient
CN111262491B (en) Incremental direct prediction speed control method suitable for permanent magnet motor system
JPH09149700A (en) Method and device for controlling ipm motor
JP2019115194A (en) Power converter control device
CN107276475B (en) Double-motor series open-phase fault-tolerant prediction type direct torque control method
CN109936320A (en) A kind of bi-motor series connection Direct Torque Control based on duty ratio modulation
CN107204727A (en) A kind of low capacity thin-film capacitor permagnetic synchronous motor direct-axis current is to locking control method

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant