CN111505600A - STPC-based FDA-MIMO radar signal processing method, device and medium - Google Patents

STPC-based FDA-MIMO radar signal processing method, device and medium Download PDF

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CN111505600A
CN111505600A CN202010424867.5A CN202010424867A CN111505600A CN 111505600 A CN111505600 A CN 111505600A CN 202010424867 A CN202010424867 A CN 202010424867A CN 111505600 A CN111505600 A CN 111505600A
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fda
clutter
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frequency
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CN111505600B (en
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文才
许世浩
彭进业
乐明楠
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Northwestern University
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G01S7/414—Discriminating targets with respect to background clutter

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  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
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  • Radar Systems Or Details Thereof (AREA)

Abstract

The embodiment of the invention discloses an STPC-based FDA-MIMO radar signal processing method, device and medium; the method can comprise the following steps: constructing a transmitting signal of the FDA-MIMO radar based on the STPC; receiving a received signal after the transmitted signal is affected by clutter; carrying out down-conversion processing on the signal of the received signal after the transmission freedom degree is extracted by utilizing the carrier frequency to obtain a baseband signal of the received signal; separating the baseband signal of the received signal according to fast Fourier transform to obtain a transmitting waveform signal; carrying out frequency correction on the transmitted waveform signals obtained by separation, and then carrying out matched filtering to obtain received echo signals; after the distance dependency of the emission space frequency is compensated for the echo signal, a self-adaptive weight vector is obtained through a Lagrange multiplier method; weighting the echo signal by using the self-adaptive weight vector to obtain a weighted signal to be estimated; and obtaining the estimated values of the distance gate where the target is located and the arrival direction based on the signal to be estimated.

Description

STPC-based FDA-MIMO radar signal processing method, device and medium
Technical Field
The embodiment of the invention relates to the technical field of radar signal processing, in particular to a frequency diversity Array Multiple-Input Multiple-Output (FDA-MIMO) radar signal processing method, device and computer storage medium based on Slow Time Phase-Coded waveforms (STPC).
Background
The detection and localization of moving objects is the most important task that needs to be accomplished by airborne radar systems. The main lobe and the side lobe clutter of the airborne or satellite-borne radar can be spread in a wide Doppler frequency range due to the motion of a platform, so that moving targets, particularly slow-speed moving targets, which are interested in the radar are submerged by scattered carriers. In order to improve the detection performance of a moving target, especially a slow moving target, a Space-time adaptive Processing (STAP) technique is usually adopted, and the technique combines a plurality of spatial channels and a plurality of coherent pulses to perform two-dimensional adaptive Processing, so that clutter and interference of Space-time coupling can be effectively suppressed, and the detection performance of the moving target is improved.
In recent years, Multiple Input Multiple Output (MIMO) radar has attracted considerable attention, compared to conventional STAP radar, MIMO radar has shown many advantages, mainly including increased spatial freedom, more accurate angle estimation, reduced minimum detectable speed (MDV) and lower probability of interception (L PI).
Currently, the new concept of electronically scanned arrays of Frequency Diversity Arrays (FDA) is of considerable interest. Unlike conventional phased arrays, frequency diversity arrays employ small frequency increments between array elements. It can provide far-field patterns that are distance and angle dependent and therefore can be widely used to improve target location performance and distance dependent interference rejection. It is noted, however, that the transmission pattern of the frequency diversity array is time-varying, which is detrimental to signal processing in practical radar systems. In order to utilize the characteristics of the FDA transmitting end, there are related technical solutions to combine the FDA and MIMO technologies to form an FDA-MIMO radar. This allows shifting the beamforming to the receiving end and a controllable degree of freedom (DOF) in the range of distance and angle can be obtained. FDA-MIMO radar technology shows great benefits of space-time adaptive radar processing in the aspects of time-varying directional diagram elimination, range ambiguity clutter suppression and target positioning. However, it is worth pointing out that in conventional FDA-MIMO radar technology, the sounding waveforms are only assumed to be perfectly orthogonal to each other. In practice, such perfect orthogonal waveforms do not exist in practice. Although Code Division Multiple Access (CDMA) waveforms have been used to recover the transmit freedom of ground-based FDA-MIMO radars, their adaptive clutter cancellation performance is very limited in airborne space-time adaptive processing radars.
Disclosure of Invention
In view of this, embodiments of the present invention are intended to provide an STPC-based FDA-MIMO radar signal processing method, apparatus, and medium; the clutter cancellation performance of the airborne space-time adaptive processing radar can be improved, and the problem of recovery of distance dimension emission freedom of the FDA-MIMO radar is solved.
The technical scheme of the embodiment of the invention is realized as follows:
in a first aspect, an embodiment of the present invention provides an STPC-based FDA-MIMO radar signal processing method, where the method includes:
constructing a transmitting signal of the FDA-MIMO radar based on the STPC;
receiving a received signal after the transmitted signal is affected by clutter;
carrying out down-conversion processing on the signal of the received signal after the transmission freedom degree is extracted by utilizing the carrier frequency to obtain a baseband signal of the received signal;
separating the baseband signal of the received signal according to fast Fourier transform to obtain a transmitting waveform signal;
carrying out frequency correction on the transmitted waveform signals obtained by separation, and then carrying out matched filtering to obtain received echo signals;
after the distance dependency of the emission space frequency is compensated for the echo signal, a self-adaptive weight vector is obtained through a Lagrange multiplier method;
weighting the echo signal by using the self-adaptive weight vector to obtain a weighted signal to be estimated;
and obtaining the estimated values of the distance gate where the target is located and the arrival direction based on the signal to be estimated.
In a second aspect, an embodiment of the present invention provides an STPC-based FDA-MIMO radar signal processing apparatus, where the apparatus includes: an FDA-MIMO radar antenna, a memory, and a processor; wherein the content of the first and second substances,
the FDA-MIMO radar antenna is used for receiving and transmitting FDA-MIMO radar signals;
the memory for storing a computer program operable on the processor;
the processor is configured to, when running the computer program, execute the steps of the STPC-based FDA-MIMO radar signal processing method according to the first aspect.
In a third aspect, an embodiment of the present invention provides a computer storage medium, where the computer storage medium stores an STPC-based FDA-MIMO radar signal processing program, and when the STPC-based FDA-MIMO radar signal processing program is executed by at least one processor, the method of processing an STPC-based FDA-MIMO radar signal according to the first aspect is implemented.
The embodiment of the invention provides an STPC-based FDA-MIMO radar signal processing method, device and medium; the slow time phase coding waveform has strong correlation, so that better cancellation performance can be obtained by taking the slow time phase coding waveform as a transmitting waveform, in addition, the problem of distance dimension transmitting freedom degree recovery of FDA-MIMO radar can be solved, and the problems of clutter suppression and target positioning of range ambiguity in the single pulse repetition frequency multi-input multi-output space-time self-adaptive processing radar can be effectively solved. .
Drawings
Fig. 1 is a schematic flow chart of an STPC-based FDA-MIMO radar signal processing method according to an embodiment of the present invention;
FIG. 2 is a schematic diagram of waveform auto-and cross-correlation at different windowing levels provided by an embodiment of the present invention;
FIG. 3 is a comparative schematic diagram of SCNR loss provided by an embodiment of the present invention;
FIG. 4(a) is a diagram illustrating DOA estimation performance as a function of signal-to-noise ratio provided by an embodiment of the present invention;
FIG. 4(b) is a diagram illustrating the performance of range gate estimation as a function of signal-to-noise ratio according to an embodiment of the present invention;
FIG. 4(c) is a diagram illustrating DOA estimation performance under different SNR conditions according to an embodiment of the present invention;
FIG. 4(d) is a schematic diagram of the performance of range gate estimation under different SNR conditions according to an embodiment of the present invention;
fig. 5 is a schematic diagram of a hardware structure of an STPC-based FDA-MIMO radar signal processing apparatus according to an embodiment of the present invention.
Detailed Description
The technical solution in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.
The technical scheme of the embodiment of the invention utilizes the strong correlation of the slow time phase coding waveform as the transmitting waveform of the FDA-MIMO radar, thereby expecting to improve the clutter cancellation performance of the airborne space-time adaptive processing radar, and provides a new signal processing mode by combining the transmitting waveform, expecting to successfully solve the problem of the distance dimension transmitting freedom degree recovery of the FDA-MIMO radar.
Based on the above explanation, referring to fig. 1, it shows a method for processing an STPC-based FDA-MIMO radar signal according to an embodiment of the present invention, where the method includes:
s11: constructing a transmitting signal of the FDA-MIMO radar based on the STPC;
s12: receiving a received signal after the transmitted signal is affected by clutter;
s13: carrying out down-conversion processing on the signal of the received signal after the transmission freedom degree is extracted by utilizing the carrier frequency to obtain a baseband signal of the received signal;
s14: separating the baseband signal of the received signal according to fast Fourier transform to obtain a transmitting waveform signal;
s15: carrying out frequency correction on the transmitted waveform signals obtained by separation, and then carrying out matched filtering to obtain received echo signals;
s16: after the distance dependency of the emission space frequency is compensated for the echo signal, a self-adaptive weight vector is obtained through a Lagrange multiplier method;
s17: weighting the echo signal by using the self-adaptive weight vector to obtain a weighted signal to be estimated;
s18: and obtaining the estimated values of the distance gate where the target is located and the arrival direction based on the signal to be estimated.
Through the technical scheme shown in fig. 1, it can be seen that the waveform based on slow time phase coding has strong correlation, so that better cancellation performance can be obtained by taking the waveform as a transmitting waveform, and in addition, the scheme can solve the problem of distance dimension transmission freedom recovery of the FDA-MIMO radar, and can also effectively solve the problems of clutter suppression and target positioning of range ambiguity in the radar processed by multiple input multiple output space-time adaptation with a single pulse repetition frequency.
For the technical solution shown in fig. 1, in some examples, the constructing an FDA-MIMO transmission signal based on an STPC includes:
according to the FDA-MIMO radar model, determining that the m array element transmitting signal at the k pulse is as follows:
Figure BDA0002498272670000051
wherein K is more than or equal to 1 and less than or equal to K, and K represents the total number of pulses; m is more than or equal to 1 and less than or equal to M, and M represents the total number of transmitting array elements of the transmitting array in the FDA-MIMO radar; u (t) represents a baseband waveform of a transmission signal; f. ofm＝f0+ (m-1) Δ f denotes the carrier frequency of the m-th array element, f0Denotes a reference carrier frequency, Δ f denotes a frequency increment;
Figure BDA0002498272670000052
indicating the phase encoding of the m-th element on the k-th pulse.
It should be noted that, based on the FDA-MIMO radar model, the transmit array is a frequency diversity array, and a slow time phase coding waveform with a slight frequency increment (or frequency offset) can be transmitted between array elements, so that the airborne space-time adaptive radar clutter cancellation performance can be improved by using the strong correlation of the slow time phase coding waveform.
For the solution shown in fig. 1, in some examples, the baseband waveform of the transmit signal
Figure BDA0002498272670000053
Wherein, Tpη is the frequency modulation rate, the rectangular function rect (x) is equal to 1 when | x ≦ 0.5, otherwise equal to 0;
the phase encoding
Figure BDA0002498272670000054
Linear function of m
Figure BDA0002498272670000055
Wherein, Δ fDmIs the Doppler shift, T, of the transmitted signal of the m-th array elementrRepresenting the pulse repetition interval.
In this embodiment, the L FM waveform is preferably selected as the baseband waveform to achieve a high compression ratio, a low Autocorrelation (AC) sidelobe level, and a low doppler loss, and the phase encoding is preferably selected as a linear function of m to achieve orthogonality of the waveform in the doppler domain.
For the solution shown in fig. 1, in some examples, the received signal after the transmit signal is affected by clutter includes a target-based signal portion ytn,k(t) and a clutter based signal part ycn,k(t); wherein the target signal received by the nth array element at the kth pulse is a superposition of delayed transmission signals, and thus the target-based signal portion is：
Figure BDA0002498272670000061
Where ρ istA reflection coefficient representing a target; tau istm,n,kShowing the round-trip propagation delay between the mth transmitting array element and the nth receiving array element on the kth pulse, in particular for a point-like target in the ith range gate
Figure BDA0002498272670000062
c is the speed of light, Rl,p＝Rl+(p-1)RuRepresenting the target distance, RlIs the maximum distance at which no range ambiguity occurs, Ru＝cT r2 denotes the maximum unambiguous distance, p denotes the p-th range gate in which range ambiguity occurs, dTAnd dRIndicating the spacing of the elements of the transmit and receive arrays, psitRepresenting the angle of incidence, v, of the objecttIs the target speed;
for the constructed clutter model, the clutter echo of each ambiguity distance can be modeled as a superposition of many independent clutter sources, so the clutter-based signal part received by the nth array element at the kth pulse is:
Figure BDA0002498272670000063
wherein N isaRepresenting the number of range ambiguities, NcRepresenting the number of independent clutter blocks, pcl,p,qRepresenting the complex reflection coefficient of the clutter block. Round-trip time delay tau of clutter between the mth transmit array element and the nth receive array element on the kth pulsecm,n,kExpressed as:
Figure BDA0002498272670000064
wherein R isl,pIs the distance of the clutter,. psil,p,qRepresents the clutter incident angle and satisfies
Figure BDA0002498272670000071
θqAnd
Figure BDA0002498272670000072
respectively, azimuth and pitch.
For the technical solution shown in fig. 1, in some examples, it is preferable to perform down-conversion using a carrier frequency for extracting a signal with a transmission degree of freedom from the STPC-based FDA-MIMO radar, and therefore, a baseband signal of the received signal is expressed as:
Figure BDA0002498272670000073
wherein, tau0＝2Rl,pThe/c represents the round-trip distance delay,
Figure BDA0002498272670000074
is a complex constant; f. ofD(vt)、fR(ψt) And fT(Rl,p,ψt) Respectively representing transmit spatial frequencies, receive spatial frequencies, and doppler frequencies.
For the solution shown in fig. 1, in some examples, the separating the baseband signal of the received signal according to the fast fourier transform to obtain the transmit waveform signal includes:
since doppler filtering can be performed directly in the fast time-pulse domain instead of the range-compressed-pulse domain, the baseband signal of the nth receive antenna can be added to a vector of K × 1:
Figure BDA0002498272670000075
using fmFor ytn(t) performing Doppler filtering to obtain an output corresponding to the mth transmitting antenna, wherein the signal of the nth receiving antenna is as follows:
Figure BDA0002498272670000076
wherein the content of the first and second substances,
Figure BDA0002498272670000077
based on a cut-off frequency of Δ fDThe low pass filter(s) will result in filtering out the other M-1 multiple input multiple output channels, ytm,n(t) the last sum term is set to approximately zero, and an approximation of the transmit waveform signal is obtained as:
Figure BDA0002498272670000081
wherein the content of the first and second substances,
Figure BDA0002498272670000082
and is represented as a complex amplitude with a doppler processing gain target.
With respect to the solution shown in fig. 1, in some examples, the signal after the filtering is completed according to the foregoing examples is a delayed baseband waveform frequency shift signal, and it is known that the frequency shift needs to be compensated before the matched filtering, so that the frequency correction needs to be performed first. Based on this, the obtaining of the received echo signal by performing the frequency correction on the transmission waveform signal obtained by the separation and then performing the matched filtering includes:
directly multiplying the transmitted waveform signal by a linear phase term to obtain a signal after the frequency correction is finished, wherein the signal is expressed as:
Figure BDA0002498272670000083
obtaining the time delay L FM signal of which the signal after the frequency correction is finished at t ═ τ based on the time delay L FM signal of which the signal after the frequency correction is finished is standard0The pulse compression output at (a) is:
Figure BDA0002498272670000084
wherein denotes the conjugate operator;
stacking and rearranging pulse compression outputs of all transmitting antennas and receiving antennas into a vector, and obtaining a transmitting-receiving snapshot vector of the ith range gate, wherein the vector is expressed as:
Figure BDA0002498272670000085
wherein the content of the first and second substances,
Figure BDA0002498272670000086
represents the Kronecker product (Kronecker product); a isT(fT(Rl,p,ψt) A and aR(fR(ψt) Expressed as a transmit steering vector and a receive steering vector, respectively;
obtaining the clutter snapshot vector of the ith range gate as:
Figure BDA0002498272670000087
wherein the content of the first and second substances,
Figure BDA0002498272670000091
and is represented as the complex amplitude of the clutter block with doppler processing gain;
the received echo in the ith range gate formed by the transmitting-receiving snapshot vector, the clutter snapshot vector and the noise is as follows:
xl＝sl+cl+nl
wherein n islRepresenting the noise component.
For the technical solution shown in fig. 1, in some examples, after compensating the echo signal for the distance dependency of the transmit spatial frequency, obtaining an adaptive weight vector by a lagrangian multiplier method includes:
to achieve adaptive clutter uniformity, a compensation vector d (R) may be used in the l-th range gatel) Compensating the distance dependence of the emission space frequency, and obtaining signals as follows:
Figure BDA0002498272670000092
wherein the content of the first and second substances,
Figure BDA0002498272670000093
conversion of a target steering vector in the transmit-receive spatial domain into an angle psi by distance-dependent compensation0And a distance gate p0The function of (d) is:
Figure BDA0002498272670000094
the adaptive weight vector obtained by the lagrange multiplier method is:
Figure BDA0002498272670000095
wherein the content of the first and second substances,
Figure BDA0002498272670000096
an estimate of the transmit-receive covariance matrix representing all clutter and noise for the ith range region.
For the technical solution shown in fig. 1, in some examples, the echo signal is weighted by using the adaptive weight vector to obtain a signal to be estimated after weighting, and in an implementation process, the signal to be estimated can be specifically obtained as
Figure BDA0002498272670000097
For the technical solution shown in fig. 1, in some examples, the estimated values of the range gate where the target is located and the direction of arrival (DOA) are obtained based on the signal to be estimated, and in a specific implementation, the range gate where the target is located and the direction of arrival (DOA) index may be preferably estimated by using a maximum likelihood (M L) method.
For the technical solution and the example shown in fig. 1, the embodiment of the present invention uses a specific simulation experiment to illustrate the advantages of the technical solution shown in fig. 1.
Firstly, an airborne radar is taken as an example in a simulation experiment, and experimental parameters and experimental conditions are specifically that a radar antenna adopts a planar array with 8 rows and × 4 columns, the spacing of receiving and transmitting array elements is 0.1m, the carrier frequency is 5GHz, 256 coherent accumulation pulses are transmitted in the same coherent pulse repetition interval, the pulse repetition frequency is 10kHz, the frequency increment is 500Hz, the height of a carrier is 4000m, the number of dividing range gates is 1000, the number of range gates generating range ambiguity is 3, horizontal uniform flight is realized, the speed is 40m/s, and the radius of the earth is 6378 km.
The cell level noise ratio CNR is 30dB and the baseband waveform is L FM waveform the transmit waveform uses a slow time phase encoded waveform, a 70dB Chebyshev window is used to localize the clutter in a narrow angular region by Doppler filtering, and the 500 th range gate is selected as the test cell the signal to noise ratio (SCNR) loss, defined as the ratio of the output of the adaptive process SCNR to the output of the matched filter with white noise only, is used as an indicator to evaluate the performance of suppressing clutter in each case.
Based on the above experimental parameters and experimental conditions, a simulation experiment is performed on the technical scheme and the example thereof shown in fig. 1, and the obtained experimental contents and results are as follows:
first, simulation experiments using slow time doppler filtering eliminated waveform cross-correlation and deeper windowing was used to ensure that clutter can be confined to a narrow angular region, as shown in fig. 2, the cross-correlation decreased with increasing windowing depth, with corresponding peak cross-correlations of-59.8 dB, -68.11dB and-76.91 dB, respectively. The result shows that the technical scheme shown in fig. 1 can successfully suppress the waveform cross-correlation. The distance profile also closely matches the ideal autocorrelation profile. This means that the solution shown in fig. 1 can effectively mitigate the effect of time-varying patterns.
Secondly, the SCNR penalty of the solution shown in fig. 1 is compared with PA radars and conventional slow-time MIMO radars. As can be seen from fig. 3, the performance of the slow-time FDA _ MIMO radar scheme shown in fig. 1, represented by the dashed box line, has similar performance to the conventional slow-time MIMO radar represented by the solid triangular line, and both have better performance than the Phased Array (PA) radar represented by the truncated solid line. Compared with PA radar and traditional slow-time mimo radar, the solution shown in fig. 1 has the capability of resolving range ambiguities at a single pulse repetition frequency, although there is a slight performance penalty in case of severe range ambiguities.
Finally, the experiment carries out simulation experiment on the parameter estimation performance of the technical scheme shown in the figure 1 under the conditions of different signal to noise ratios. The performance estimation characterizations using RMSE as the DOA estimation angle and PCR as the distance gate range region estimation, as shown in fig. 4(a) and 4(b), both DOA and distance gate estimation accuracy increase with increasing signal-to-noise ratio. In particular, the probability of correct estimation of the range gate reaches 100% when the signal-to-noise ratio SNR is greater than 0 dB. This means that in this case a radar system based on the solution shown in fig. 1 can produce a stable estimate of the range gate in which the target is located. As shown in fig. 4(c) and 4(d), in the case of a high snr, there is a 0.6 ° precision loss in the side lobe clutter region, and when the snr is 0dB, there is a 2 ° precision loss in the mainlobe clutter region. For the correct estimation probability of the range bin, the number of doppler channels that can stably resolve the range ambiguity is reduced from 27 to 25 at low snr. Therefore, the technical scheme shown in the figure 1 can provide satisfactory distance ambiguity resolution performance.
Based on the same technical concept of the foregoing technical solutions, if the above-mentioned solution can be implemented in the form of a software functional module and is not sold or used as an independent product, the solution may be stored in a computer readable storage medium, and based on such understanding, the technical solution of this embodiment essentially or a part contributing to the prior art, or all or part of the technical solution may be embodied in the form of a software product, which is stored in a storage medium and includes several instructions for enabling a computer device (which may be a personal computer, a server, a network device, or the like) or a processor (processor) to execute all or part of the steps of the method of this embodiment. And the aforementioned storage medium includes: various media capable of storing program codes, such as a usb disk, a removable hard disk, a Read Only Memory (ROM), a Random Access Memory (RAM), a magnetic disk, or an optical disk.
Therefore, the present embodiment provides a computer storage medium, which stores an STPC-based FDA-MIMO radar signal processing program, and when the STPC-based FDA-MIMO radar signal processing program is executed by at least one processor, the STPC-based FDA-MIMO radar signal processing program implements the steps of the STPC-based FDA-MIMO radar signal processing method in the technical solutions shown in the above embodiments.
Based on the same technical concept of the foregoing technical solution, referring to fig. 5, a specific hardware structure of an STPC-based FDA-MIMO radar signal processing apparatus 50 according to an embodiment of the present invention is shown, including: an FDA-MIMO radar antenna 501, a memory 502, and a processor 503; the various components are coupled together by a bus system 504. It is understood that the bus system 504 is used to enable communications among the components. The bus system 504 includes a power bus, a control bus, and a status signal bus in addition to a data bus. For clarity of illustration, however, the various buses are labeled as bus system 504 in fig. 5. The FDA-MIMO radar antenna 501 is used for receiving and transmitting FDA-MIMO radar signals;
a memory 502 for storing a computer program capable of running on the processor 503;
the processor 503 is configured to execute the steps of the STPC-based FDA-MIMO radar signal processing method in the foregoing technical solution when the computer program is executed.
It is to be understood that the Memory 502 in embodiments of the present invention may be either volatile Memory or non-volatile Memory, or may include both volatile and non-volatile Memory, wherein non-volatile Memory may be Read-Only Memory (ROM), Programmable Read-Only Memory (PROM), Erasable Programmable Read-Only Memory (EPROM), Electrically Erasable Programmable Read-Only Memory (EEPROM), or flash Memory volatile Memory may be Random Access Memory (RAM), which serves as external cache Memory, RAM, by way of example but not limitation, many forms of RAM are available, such as Static Random Access Memory (Static RAM, SRAM), Dynamic Random Access Memory (Dynamic RAM, DRAM), Synchronous Dynamic Random Access Memory (Synchronous DRAM, SDRAM), Double Data rate Synchronous Dynamic Random Access Memory (Double Data, ddrsted DRAM), Enhanced Synchronous Dynamic DRAM (Enhanced DRAM), or Synchronous DRAM (Synchronous DRAM), or any other type of RAM suitable for accessing a system including, but not limited to, SDRAM, and SDRAM, and SDRAM, and other suitable for use of the like, or SDRAM, or RAM, and SDRAM, and RAM, or RAM.
And the processor 503 may be an integrated circuit chip having signal processing capabilities. In implementation, the steps of the above method may be performed by integrated logic circuits of hardware or instructions in the form of software in the processor 503. The Processor 503 may be a general-purpose Processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), an off-the-shelf Programmable Gate Array (FPGA) or other Programmable logic device, discrete Gate or transistor logic device, or discrete hardware components. The various methods, steps and logic blocks disclosed in the embodiments of the present invention may be implemented or performed. A general purpose processor may be a microprocessor or the processor may be any conventional processor or the like. The steps of the method disclosed in connection with the embodiments of the present invention may be directly implemented by a hardware decoding processor, or implemented by a combination of hardware and software modules in the decoding processor. The software module may be located in ram, flash memory, rom, prom, or eprom, registers, etc. storage media as is well known in the art. The storage medium is located in the memory 502, and the processor 503 reads the information in the memory 502 and completes the steps of the above method in combination with the hardware thereof.
For a hardware implementation, the Processing units may be implemented within one or more Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable logic devices (P L D), Field-Programmable Gate arrays (FPGAs), general purpose processors, controllers, microcontrollers, microprocessors, other electronic units configured to perform the functions described herein, or a combination thereof.
For a software implementation, the techniques described herein may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory and executed by a processor. The memory may be implemented within the processor or external to the processor.
It should be noted that: the technical schemes described in the embodiments of the present invention can be combined arbitrarily without conflict.
The above description is only for the specific embodiments of the present invention, but the scope of the present invention is not limited thereto, and any person skilled in the art can easily conceive of the changes or substitutions within the technical scope of the present invention, and all the changes or substitutions should be covered within the scope of the present invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the appended claims.

Claims (10)

1. A frequency diversity array multiple-input multiple-output FDA-MIMO radar signal processing method based on a slow time phase coding waveform STPC is characterized by comprising the following steps:
constructing a transmitting signal of the FDA-MIMO radar based on the STPC;
receiving a received signal after the transmitted signal is affected by clutter;
carrying out down-conversion processing on the signal of the received signal after the transmission freedom degree is extracted by utilizing the carrier frequency to obtain a baseband signal of the received signal;
separating the baseband signal of the received signal according to fast Fourier transform to obtain a transmitting waveform signal;
carrying out frequency correction on the transmitted waveform signals obtained by separation, and then carrying out matched filtering to obtain received echo signals;
after the distance dependency of the emission space frequency is compensated for the echo signal, a self-adaptive weight vector is obtained through a Lagrange multiplier method;
weighting the echo signal by using the self-adaptive weight vector to obtain a weighted signal to be estimated;
and obtaining the estimated values of the distance gate where the target is located and the arrival direction based on the signal to be estimated.
2. The method of claim 1, wherein the STPC-based construction of FDA-MIMO transmission signals comprises:
according to the FDA-MIMO radar model, determining that the m array element transmitting signal at the k pulse is as follows:
Figure FDA0002498272660000011
wherein K is more than or equal to 1 and less than or equal to K, and K represents the total number of pulses; m is more than or equal to 1 and less than or equal to M, and M represents the total number of transmitting array elements of the transmitting array in the FDA-MIMO radar; u (t) represents a baseband waveform of a transmission signal; f. ofm＝f0+ (m-1) Δ f denotes the carrier frequency of the m-th array element, f0Denotes a reference carrier frequency, Δ f denotes a frequency increment;
Figure FDA0002498272660000012
indicating the phase encoding of the m-th element on the k-th pulse.
3. The method of claim 2, wherein the baseband waveform of the transmit signal
Figure FDA0002498272660000013
Wherein, TpRepresenting pulse duration, η is the frequency modulation rate;the rectangle function rect (x) is equal to 1 when | x | ≦ 0.5, otherwise is equal to 0;
the phase encoding
Figure FDA0002498272660000021
Linear function of m
Figure FDA0002498272660000022
Wherein, Δ fDmIs the Doppler shift, T, of the transmitted signal of the m-th array elementrRepresenting the pulse repetition interval.
4. The method of claim 3, wherein the received signal after the transmit signal is affected by clutter comprises a target-based signal portion ytn,k(t) and a clutter based signal part ycn,k(t); the target-based signal portion is:
Figure FDA0002498272660000023
where ρ istA reflection coefficient representing a target; tau istm,n,kShowing the round-trip propagation delay between the mth transmitting array element and the nth receiving array element on the kth pulse, in particular for a point-like target in the ith range gate
Figure FDA0002498272660000024
c is the speed of light, Rl,p＝Rl+(p-1)RuRepresenting the target distance, RlIs the maximum distance at which no range ambiguity occurs, Ru＝cTr2 denotes the maximum unambiguous distance, p denotes the p-th range gate in which range ambiguity occurs, dTAnd dRIndicating the spacing of the elements of the transmit and receive arrays, psitRepresenting the angle of incidence, v, of the objecttIs the target speed;
the clutter based signal part is:
Figure FDA0002498272660000025
wherein N isaRepresenting the number of range ambiguities, NcRepresenting the number of independent clutter blocks, pcl,p,qRepresenting the complex reflection coefficient of the clutter block. Round-trip time delay tau of clutter between the mth transmit array element and the nth receive array element on the kth pulsecm,n,kExpressed as:
Figure FDA0002498272660000026
wherein R isl,pIs the distance of the clutter,. psil,p,qRepresents the clutter incident angle and satisfies
Figure FDA0002498272660000027
θqAnd
Figure FDA0002498272660000028
respectively, azimuth and pitch.
5. The method of claim 4, wherein the baseband signal of the received signal is expressed as:
Figure FDA0002498272660000031
wherein, tau0＝2Rl,pThe/c represents the round-trip distance delay,
Figure FDA0002498272660000032
is a complex constant; f. ofD(vt)、fR(ψt) And fT(Rl,p,ψt) Respectively representing transmit spatial frequencies, receive spatial frequencies, and doppler frequencies.
6. The method of claim 5, wherein the separating the transmit waveform signal from the baseband signal of the received signal according to the fast Fourier transform comprises:
the baseband signal of the nth receiving antenna is added into a vector of K × 1:
Figure FDA0002498272660000033
using fmFor ytn(t) performing Doppler filtering to obtain an output corresponding to the mth transmitting antenna, wherein the signal of the nth receiving antenna is as follows:
Figure FDA0002498272660000034
wherein the content of the first and second substances,
Figure FDA0002498272660000035
based on a cut-off frequency of Δ fDThe low pass filter(s) will result in filtering out the other M-1 multiple input multiple output channels, ytm,n(t) the last sum term is set to approximately zero, and an approximation of the transmit waveform signal is obtained as:
Figure FDA0002498272660000036
wherein the content of the first and second substances,
Figure FDA0002498272660000037
and is represented as a complex amplitude with a doppler processing gain target.
7. The method of claim 6, wherein the frequency correcting the separated transmit waveform signals and performing matched filtering to obtain receive echo signals comprises:
directly multiplying the transmitted waveform signal by a linear phase term to obtain a signal after the frequency correction is finished, wherein the signal is expressed as:
Figure FDA0002498272660000041
obtaining the time delay L FM signal of which the signal after the frequency correction is finished at t ═ τ based on the time delay L FM signal of which the signal after the frequency correction is finished is standard0The pulse compression output at (a) is:
Figure FDA0002498272660000042
wherein denotes the conjugate operator;
stacking and rearranging pulse compression outputs of all transmitting antennas and receiving antennas into a vector, and obtaining a transmitting-receiving snapshot vector of the ith range gate, wherein the vector is expressed as:
Figure FDA0002498272660000043
wherein the content of the first and second substances,
Figure FDA0002498272660000044
represents the Kronecker product (Kronecker product); a isT(fT(Rl,p,ψt) A and aR(fR(ψt) Expressed as a transmit steering vector and a receive steering vector, respectively;
obtaining the clutter snapshot vector of the ith range gate as:
Figure FDA0002498272660000045
wherein the content of the first and second substances,
Figure FDA0002498272660000046
and is represented as the complex amplitude of the clutter block with doppler processing gain;
the received echo in the ith range gate formed by the transmitting-receiving snapshot vector, the clutter snapshot vector and the noise is as follows:
xl＝sl+cl+nl
wherein n islRepresenting the noise component.
8. The method of claim 7, wherein the compensating the echo signal for the distance dependency of the transmit spatial frequency is followed by obtaining an adaptive weight vector by a Lagrangian multiplier method, comprising:
using a compensation vector d (R) in the l-th range gatel) Compensating the distance dependence of the emission space frequency, and obtaining signals as follows:
Figure FDA0002498272660000051
wherein the content of the first and second substances,
Figure FDA0002498272660000052
conversion of a target steering vector in the transmit-receive spatial domain into an angle psi by distance-dependent compensation0And a distance gate p0The function of (d) is:
Figure FDA0002498272660000053
the adaptive weight vector obtained by the lagrange multiplier method is:
Figure FDA0002498272660000054
wherein the content of the first and second substances,
Figure FDA0002498272660000055
an estimate of the transmit-receive covariance matrix representing all clutter and noise for the ith range region.
9. An STPC-based FDA-MIMO radar signal processing apparatus, the apparatus comprising: an FDA-MIMO radar antenna, a memory, and a processor; wherein the content of the first and second substances,
the FDA-MIMO radar antenna is used for receiving and transmitting FDA-MIMO radar signals;
the memory for storing a computer program operable on the processor;
the processor, when executing the computer program, is configured to perform the steps of the STPC-based FDA-MIMO radar signal processing method according to any one of claims 1 to 8.
10. A computer storage medium storing an STPC-based FDA-MIMO radar signal processing program that, when executed by at least one processor, performs the steps of the STPC-based FDA-MIMO radar signal processing method of any one of claims 1 through 8.
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