CN111181407B - C-LLCT-LLT type resonance direct current converter - Google Patents
C-LLCT-LLT type resonance direct current converter Download PDFInfo
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- CN111181407B CN111181407B CN202010060719.XA CN202010060719A CN111181407B CN 111181407 B CN111181407 B CN 111181407B CN 202010060719 A CN202010060719 A CN 202010060719A CN 111181407 B CN111181407 B CN 111181407B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/32—Means for protecting converters other than automatic disconnection
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Abstract
The invention discloses a C-LLCT-LLT type resonance direct current converter, which comprises a voltage type half-bridge inverter circuit, a C-LLCT-LLT resonant cavity and a full-bridge rectifier circuit which are connected in sequence, wherein the C-LLCT-LLT resonant cavity comprises a first phase resonance loop and a second phase resonance loop, the first phase resonance loop comprises a first series branch inductor and a first parallel branch inductor, the second phase resonance circuit comprises a second series branch circuit inductor, a second parallel branch circuit inductor and a second high-frequency transformer, the input end of the first series branch circuit inductor and the input end of the second series branch circuit inductor are connected with one end of the series branch circuit capacitor, the other end of the series branch circuit capacitor is connected with the output port of the half-bridge inverter circuit, and secondary windings of the first high-frequency transformer and the second high-frequency transformer are connected with the input port of the full-bridge rectifier circuit in series. The invention has wider voltage gain adjustment range and lower voltage stress of the resonant capacitor.
Description
Technical Field
The invention belongs to the technical field of direct current converters, and particularly relates to a C-LLCT-LLT type resonant direct current converter.
Background
In recent years, the related art in the field of server data power supply systems has been vigorously developed due to the increasing demand. In these power supply systems, since the ac voltage generated by the ac power grid is rectified and actively corrected to output a dc voltage significantly higher than the voltage level at which the server data power supply operates, a dc-dc converter with a high step-down ratio is often required to efficiently convert the voltage. However, to prevent data loss due to system outage, it is necessary to ensure that the dc voltage ultimately provided to the server data power supply is maximally immune to bus voltage sag. Therefore, a large-capacity voltage stabilizing capacitor is often required to be connected in parallel to the voltage input side of the server data power supply to inhibit voltage drop, so that an Uninterruptible Power Supply (UPS) can provide backup guarantee for the system in time. Since the capacitance value of the voltage-stabilizing capacitor has a positive correlation with the volume, a large increase in the capacitance value can suppress the possibility of a failure of the server data power supply to a certain extent, but also brings about a large increase in the overall volume of the system, which leads to a cost increase. Therefore, in recent years, most researchers turn the voltage stabilizing capacitor in the system to the dc-dc converter providing voltage conversion, and hope to maintain the output voltage of the dc-dc converter from being affected by the bus voltage drop by widening the voltage gain range of the dc-dc converter, thereby improving the reliability of the operation of the server data power supply system.
Among the dc-dc converters used for this scenario, the LLC type resonant converter is widely used due to excellent overall performance. The LLC type converter can realize zero voltage switching-on (ZVS) of an inverter switching tube and zero current switching-off (ZCS) of a rectifier diode in a wide load range, and further enables a direct current-direct current voltage conversion process in a system to obtain high conversion efficiency. However, when the bus voltage drops, the inductance of the parallel branch inductor (magnetizing inductor) needs to be as small as possible to ensure that the LLC converter achieves higher voltage gain. Therefore, when the converter operates in a normal bus voltage state without high voltage gain, the reduction of the resonant cavity impedance caused by the parallel branch inductance with a low inductance value can cause the current in the resonant cavity to increase, and further the efficiency of the converter is reduced. In view of this, in recent years, some researchers have proposed an LCLC resonant converter. The only difference with respect to LLC converters is that the parallel-branch inductance (magnetizing inductance) is replaced by a pair of series-connected inductance and capacitance (i.e. parallel-branch inductance and parallel-branch capacitance). When the switching frequency is higher than the resonance frequency between the inductor and the capacitor, the inductor and the capacitor will form a "variable excitation inductor". Since the capacitor can be regarded as a negative inductor, and the absolute value of the inductance value of the capacitor is opposite to the changing direction of the switching frequency, the inductance value of the "variable excitation inductance" is naturally large when the switching frequency is high, and the inductance value of the "variable excitation inductance" is naturally small when the switching frequency is low. Therefore, when the bus voltage is in a normal state, the switching frequency is higher due to the lower gain requirement, and the switching frequency can be set to be equal to the resonant frequency of the converter for higher efficiency, so that the gain is independent of the inductance value of the excitation inductor. Because the inductance value of the variable excitation inductance is larger at the moment, the converter can further improve the conversion efficiency due to larger resonant cavity impedance and lower resonant current; when the bus voltage drops, the switching frequency is reduced, and high voltage gain can be obtained due to the reduction of the inductance value of the variable excitation inductance (due to the action of an Uninterruptible Power Supply (UPS), the total duration of the high gain state of the converter caused by the drop of the bus voltage is short, so the efficiency of the converter in the bus voltage drop state can be disregarded). For this reason, it can be seen that the LCLC converter can achieve dual advantages in terms of efficiency and voltage gain.
When the bus voltage drops and the switching frequency is reduced, so that the inductance value of the variable excitation inductance is reduced, for the series branch capacitor in the LCLC converter, even if the input impedance of the resonant cavity is reduced, the current flowing through the series branch capacitor cannot be obviously increased due to the simultaneous drop of the input voltage, and therefore, when the bus voltage drops, the voltage stress of the series branch capacitor cannot be obviously increased. However, to maintain the output voltage constant, the decrease in the inductance of the "variable magnetizing inductance" directly results in a significant increase in the current flowing through the parallel branch capacitance. In addition, in order to ensure that the inductance value of the variable excitation inductance changes obviously in the process of reducing the switching frequency so as to obviously improve the gain, the capacitance value of the parallel branch capacitor is as small as possible. Both factors will therefore lead to a drastic increase in the voltage stress of the parallel branch capacitance. The voltage stress of the parallel branch capacitors becomes a major factor that limits the performance of the conventional LCLC converter, since the voltage stress of the capacitors is too high to reduce the reliability of the converter operation.
Disclosure of Invention
The invention aims to: aiming at the defects of the prior art, the C-LLCT-LLT type resonant DC converter is provided, the wide voltage gain characteristic consistent with that of an LCLC converter can be obtained, the problem of overhigh voltage stress of a parallel branch capacitor under the condition of bus voltage drop can be avoided, meanwhile, the winding loss of a magnetic element when the converter works in a normal bus voltage state is further reduced through a parallel shunt mode, and the power conversion efficiency is improved.
In order to achieve the purpose, the invention adopts the following technical scheme:
a C-LLCT-LLT type resonance direct current converter comprises a voltage type half-bridge inverter circuit, a C-LLCT-LLT resonant cavity and a full-bridge rectifier circuit which are sequentially connected, wherein the C-LLCT-LLT resonant cavity comprises a first phase resonance loop and a second phase resonance loop, and the first phase resonance loop comprises a first series branch circuit inductor L r1 First parallel branch inductor L p1 A first parallel branch capacitor C p1 And a first high-frequency transformer T 1 The second phase resonant circuit comprises a second series branch inductor L r2 And a second parallel branch inductor L m2 And a second high-frequency transformer T 2 The first series branch inductor L r1 And the second series branch inductor L r2 The input ends of the two capacitors are connected with a series branch capacitor C r One end of said series branch capacitor C r The other end of the first high-frequency transformer T is connected with an output port of the half-bridge inverter circuit 1 And a second high-frequency transformer T 2 The secondary winding of the rectifier is connected with the input port of the full-bridge rectifier circuit in series.
As an improvement of the C-LLCT-LLT type resonant DC converter, the half-bridge inverter circuit comprises an input voltage stabilizing capacitor C i A first high-frequency power switch device S 1 And a second high frequency power switching device S 2 Said input voltage stabilizing capacitor C i The first high-frequency power switching device S 1 And the second high-frequency power switch device S 2 A loop is formed.
As an improvement of the C-LLCT-LLT resonant dc converter according to the present invention, the output port of the half-bridge inverter circuit is disposed on the first high-frequency power switch device S 1 And said second high frequency power switching device S 2 In the meantime.
As an improvement of the C-LLCT-LLT type resonant DC converter, the full-bridge rectification circuit comprises a first diode D 1 A second diode D 2 A third diode D 3 A fourth diode D 4 And an output voltage-stabilizing filter capacitor C o The first diode D 1 The second diodeD 2 The third diode D 3 And the fourth diode D 4 Form a loop, and output voltage-stabilizing filter capacitor C o And the third diode D 3 And the fourth diode D 4 And (4) connecting in parallel.
As an improvement of the C-LLCT-LLT type resonant DC converter of the present invention, the first diode D 1 And the second diode D 2 A first input port of the full-bridge rectification circuit is arranged between the first input port and the second input port, and the first input port is connected with the first high-frequency transformer T 1 The secondary winding of (2).
As an improvement of the C-LLCT-LLT type resonant DC converter of the present invention, the first diode D 1 And the second diode D 2 Is connected to the cathode.
As an improvement of the C-LLCT-LLT type resonant DC converter of the present invention, the third diode D 3 And the fourth diode D 4 A second input port of the full-bridge rectification circuit is arranged between the first input port and the second high-frequency transformer T 2 The secondary winding of (2).
As an improvement of the C-LLCT-LLT type resonant DC converter of the present invention, the third diode D 3 And the fourth diode D 4 Is connected to the cathode.
As an improvement of the C-LLCT-LLT type resonant DC converter of the present invention, the first diode D 1 The second diode D 2 The third diode D 3 And the fourth diode D 4 Are all unidirectional diodes.
As an improvement of the C-LLCT-LLT type resonant DC converter, the first parallel branch circuit capacitor C p1 And the second parallel branch inductor L m2 Is connected at one end.
The invention has the beneficial effects that:
(1) when the switching frequency of the converter is higher than the inductance L p1 And a capacitor C p1 At the resonant frequency of (1), inductance L p1 And a capacitor C p1 It can be equivalent to a "variable magnetizing inductance" whose inductance value decreases as the switching frequency decreases. Therefore, when the bus voltage drops, the inductance value of the variable excitation inductance can be reduced only by reducing the switching frequency, so that the voltage gain of the converter is greatly improved, the output voltage is finally stabilized, and the output voltage is prevented from being influenced by the bus voltage drop; when the bus voltage is in a normal state, the switching frequency needs to be increased for reducing the voltage gain, and if the switching frequency at the moment is set to be equal to the resonant frequency of the converter, the gain at the moment is irrelevant to the inductance value of the excitation inductor, so that the resonant cavity impedance caused by the variable excitation inductor with a higher inductance value can be increased to reduce the magnitude of the resonant current, and further higher conversion efficiency can be obtained.
(2) The C-LLCT-LLT type resonant DC converter can reduce the voltage at two ends of the variable excitation inductor simultaneously by means of a special resonant cavity structure of the C-LLCT-LLT type resonant DC converter when the inductance value of the variable excitation inductor is reduced, so that the current flowing through the variable excitation inductor is prevented from being excessively increased under the bus voltage drop state, and compared with the traditional LCLC type resonant DC converter, the capacitor (namely the parallel branch capacitor) in the variable excitation inductor can be prevented from excessively high voltage stress, so that the manufacturing cost of the capacitor is reduced, and the system reliability is improved.
(3) Compared with a single-phase resonant circuit formed by 5 elements in the traditional LCLC type converter, the C-LLCT-LLT type converter can expand the resonant circuit into two phases under the condition that only 2 elements are added, and can further reduce the winding loss of the magnetic elements when the converter works in a normal bus voltage state by means of the action of two-phase parallel shunt under the condition that the number of the elements is increased in a small proportion, so that the power conversion efficiency is improved.
Drawings
FIG. 1 is a circuit diagram of the present invention.
FIG. 2 is a fundamental equivalent circuit diagram of the C-LLT-LLT resonant structure of the present invention.
FIG. 3 is a voltage gain of the C-LLT-LLT resonant structure of the present inventionWith L m1 And (5) a change trend graph of the change of the sensitivity value.
FIG. 4 is a graph of voltage gain versus switching frequency according to the present invention.
FIG. 5 is a diagram showing an output voltage E of the first phase resonant tank of the C-LLT-LLT resonant structure according to the present invention o1 And the output voltage E of the second phase resonant tank o2 With L m1 And (5) a change trend graph of the change of the sensitivity value.
FIG. 6 shows L in the present invention p1 、C p1 Voltage sum of both ends L m2 Voltage waveforms across the terminals are compared.
Detailed Description
As used in the specification and in the claims, certain terms are used to refer to particular components. As one skilled in the art will appreciate, manufacturers may refer to a component by different names. This specification and claims do not intend to distinguish between components that differ in name but not function. In the following description and in the claims, the terms "include" and "comprise" are used in an open-ended fashion, and thus should be interpreted to mean "include, but not limited to. "substantially" means within an acceptable error range, and a person skilled in the art can solve the technical problem within a certain error range to substantially achieve the technical effect.
In the description of the present invention, it is to be understood that the terms "upper", "lower", "front", "rear", "left", "right", horizontal ", and the like indicate orientations or positional relationships based on those shown in the drawings, and are only for convenience in describing the present invention and simplifying the description, but do not indicate or imply that the referred device or element must have a specific orientation, be constructed in a specific orientation, and be operated, and thus, should not be construed as limiting the present invention.
In the present invention, unless otherwise expressly specified or limited, the terms "mounted," "connected," "secured," and the like are to be construed broadly and can, for example, be fixedly connected, detachably connected, or integrally connected; can be mechanically or electrically connected; they may be connected directly or indirectly through intervening media, or they may be interconnected between two elements. The specific meanings of the above terms in the present invention can be understood by those skilled in the art according to specific situations.
The present invention will be described in further detail with reference to fig. 1 to 6, but the present invention is not limited thereto.
Example 1
A C-LLCT-LLT type resonance DC converter is shown in figure 1, and comprises a voltage type half-bridge inverter circuit, a C-LLCT-LLT resonant cavity and a full-bridge rectifier circuit which are connected in sequence, wherein the C-LLCT-LLT resonant cavity comprises a first phase resonance loop and a second phase resonance loop, the first phase resonance loop comprises a first series branch circuit inductance L r1 First parallel branch inductor L p1 A first parallel branch capacitor C p1 And a first high-frequency transformer T 1 The second phase resonant circuit comprises a second series branch inductor L r2 And a second parallel branch inductor L m2 And a second high-frequency transformer T 2 First series branch inductance L r1 With the second series branch inductance L r2 The input ends of the two capacitors are connected with a series branch capacitor C r One end of (1), series branch circuit capacitance C r The other end of the first high-frequency transformer T is connected with an output port of the half-bridge inverter circuit 1 And a second high-frequency transformer T 2 The secondary winding of the transformer is connected with the input port of the full-bridge rectifying circuit in series.
Preferably, the half-bridge inverter circuit comprises an input voltage stabilizing capacitor C i A first high-frequency power switch device S 1 And a second high-frequency power switch device S 2 Input voltage stabilizing capacitor C i A first high-frequency power switch device S 1 And a second high-frequency power switch device S 2 Forming a loop.
Preferably, the output port of the half-bridge inverter circuit is arranged on the first high-frequency power switch device S 1 And a second high frequency power switch device S 2 In the meantime.
Preferably, the full-bridge rectification circuit includes a first diode D 1 A second diode D 2 A third diode D 3 The fourth diodeD 4 And an output voltage-stabilizing filter capacitor C o First diode D 1 A second diode D 2 A third diode D 3 And a fourth diode D 4 Form a loop and output a voltage-stabilizing filter capacitor C o And a third diode D 3 And a fourth diode D 4 And (4) connecting in parallel.
Preferably, the first diode D 1 And a second diode D 2 A first input port of the full-bridge rectification circuit is arranged between the first input port and the second input port, and the first input port is connected with a first high-frequency transformer T 1 The secondary winding of (2).
Preferably, the first diode D 1 And a second diode D 2 Is connected to the cathode.
Preferably, a third diode D 3 And a fourth diode D 4 A second input port of the full-bridge rectification circuit is arranged between the first input port and the second input port, and the second input port is connected with a second high-frequency transformer T 2 The secondary winding of (2).
Preferably, a third diode D 3 And a fourth diode D 4 Is connected to the cathode.
Preferably, the first diode D 1 A second diode D 2 A third diode D 3 And a fourth diode D 4 Are all unidirectional diodes.
Preferably, the first parallel branch capacitor C p1 And the second parallel branch inductor L m2 Is connected at one end.
The working principle of the invention is as follows:
similar to LCLC converters, if the switching frequency is higher than L p1 And C p1 Resonant frequency of (2), then L p1 And C p1 Can form a variable excitation inductance L m1 . When L is p1 And C p1 Can be regarded as a variable excitation inductance L m1 In this case, the C-LLCT-LLT resonant cavity can be regarded as a capacitance-inductance-transformer-inductance-transformer (C-LLT) resonant structure, and its fundamental equivalent circuit can be shown in fig. 2. The two branches in the resonator can be viewed as two LLT cells (LLT-1 and LLT-2). E o1 And E o2 Respectively generation by generationThe output voltages of tables LLT-1 and LLT-2. E i 、E o And R eq Respectively the input voltage, the output voltage and the equivalent load resistance of the whole C-LLT-LLT resonant structure. I.C. A 1 、I 2 And I o Respectively flows through L r1 、L r2 And R eq The current of (2). s is the Laplace operator. From fig. 2, the following system of equations can be derived:
the voltage gain M of the converter can be derived from equation (1):
wherein:
according to the formula (2), the imaginary part in the expression M is zero, and the resonant frequency f of the converter can be obtained r Is composed of
When the formula (2) and the formula (3) are combined, L can be drawn m1 The voltage gain curves of the C-LLT-LLT type resonant structure are shown in FIG. 3 when different values are taken. It can be seen that in the C-LLT-LLT structure, L m1 The reduction of the inductance value can also enable the converter to obtain higher voltage gain in the process of reducing the switching frequency, namely, the C-LLCT-LLT converter can also obtain the same excellent characteristics as the LCLC converter, namely high gain characteristics when the bus voltage drops and high efficiency characteristics when the bus voltage is normal.
For "variable excitation inductance" L m1 Of which with L p1 And C p1 Can be expressed as：
By combining the equations (2), (3) and (5), the complete voltage gain curve of the C-LLCT-LLT resonant dc converter can be drawn, as shown in fig. 4. It can be seen that L is due to p1 And C p1 There is also a resonant zero in the converter. Frequency f of resonance zero 0 Can be found by making M equal to 0, f 0 As shown in equation (6).
Parallel branch capacitance voltage stress:
defining an inductance ratio:
by combining the formula (1) and the formula (7), the output voltage E of the first phase LLT (LLT-1) can be obtained o1 The expression of (a) is:
as can be seen from the formula (8), if L r2 n 1 -L r1 n 2 When the value is 0, then E o1 Then with s and a load R eq Are irrelevant. For this purpose, L may be set taking into account the current sharing between the two branches r1 ＝L r2 ，n 1 ＝n 2 . Thus, E o1 And E o2 The expression of (c) can be simplified to:
due to L in FIG. 2 m1 Is a variable excitation inductor, the inductance value of which is to beDecreasing with decreasing switching frequency. Thus, k 1 Will increase with decreasing switching frequency. For this reason, when k is plotted according to equation (9) 2 When not changed at 0.1, E o1 And E o2 Following k 1 As shown in fig. 5.
As can be seen from FIG. 5, only when k is 1 ＝k 2 When E is greater o1 And E o2 Is equal to E o /2. Once k is completed 1 Becomes ratio k 2 Higher and higher (L) m1 Becomes smaller than L m2 Lower and lower) E o1 Will become more than E o A/2 is lower and lower, and E o2 Will become more than E o The/2 is higher and higher. Since the first phase LLT (LLT-1) and the second phase LLT (LLT-2) are connected in series at the output, it can be concluded that: whenever L is present m1 Becomes smaller than L m2 The lower, the lower the factor L m1 The higher voltage gain obtained is increasingly reflected in the total output voltage by the second phase LLT (LLT-2). Thus, by this "exchange of functions", the C-LLT can not only obtain a high voltage gain by virtue of the presence of the "variable excitation inductance", but also cause the voltage across the "variable excitation inductance" to continuously decrease as the gain increases. Thus when L is m1 Quilt L p1 And C p1 After the replacement, compared with the situation that the voltage at two ends of the variable excitation inductance in the LCLC converter is always kept unchanged, the C-LLCT-LLT converter can prevent the current flowing through the variable excitation inductance from increasing sharply due to the reduction of the terminal voltage while the bus voltage drops and the inductance value of the variable excitation inductance is reduced. Therefore, the voltage stress of the capacitor (parallel branch capacitor) in the "variable excitation inductor" can be relieved to a certain extent. Therefore, the biggest defects of the LCLC converter can be solved to a certain extent.
To verify that the terminal voltage of the "variable excitation inductance" of the C-LLCT-LLT converter decreases with the decrease of the inductance value of the "variable excitation inductance", FIG. 6 shows that the C-LLCT-LLT converter operates at normal bus voltage (FIG. 6(a), converter input voltage 400V, low gain state, high switching frequency 150kHz) and bus voltage drop (FIG. 6(b), converter input voltage 170V, high switching frequency 150kHz)Gain state, low switching frequency: 90kHz) in both cases. When the bus voltage is normal, according to the transformer T 1 、T 2 The secondary side current can show that the converter works in a resonant frequency state due to the inductance value and the L of the variable excitation inductance m2 The same "variable excitation inductance" and inductance L m2 The voltages at the two ends are the same, the converter can obtain smaller resonant current due to larger resonant cavity impedance, and the converter works in a resonant state to finally obtain higher power conversion efficiency; when the bus voltage drops, according to the transformer T 1 、T 2 The secondary side current can be seen that the switching frequency of the converter is smaller than the resonant frequency, so that the inductance value of the variable excitation inductance is reduced, the gain is increased to keep the output voltage unchanged (50V), meanwhile, the voltage at two ends of the variable excitation inductance is reduced compared with the voltage at the two ends of the bus, and more output voltage is added to the inductance L m2 By means of the special function exchange effect, the parallel branch capacitor in the C-LLCT-LLT converter can be prevented from being subjected to overhigh voltage stress.
Combined advantages in terms of volume and efficiency:
it is known that to reduce the winding loss of the magnetic elements in the resonant cavity of the converter, the current can be divided in a multi-phase parallel connection mode, and the larger the number of phases divided in parallel, the smaller the total winding loss of the converter. However, as the number of phases connected in parallel increases, the size of the converter also increases due to the increase in the number of components. Therefore, the volume and the efficiency are often two factors that are mutually restricted. For an LCLC converter, the parallel branch inductance cannot be realized by the excitation inductance of the transformer due to the presence of the parallel branch capacitance. Similarly, for the C-LLCT-LLT converter in FIG. 1, the capacitance C is used p1 Presence of an inductance L p1 Cannot rely on transformer T 1 However, L in the C-LLCT-LLT converter m2 But can be made of a transformer T 2 Is achieved by the excitation inductance of (1). Thus, there are 5 elements in the LCLC converter cavity and 7 elements in the C-LLCT-LLT converter cavity. Thus, it can be seen that C-LLCTThe LLT converter can increase the number of parallel phases in the resonant cavity by 100% (1 phase to 2 phases) under the condition that the number of resonant elements is increased by only 40% (5 to 7), namely, the C-LLCT-LLT converter can further reduce the winding loss of magnetic elements when the converter works in a normal bus voltage state through the action of two-phase parallel shunt under the condition that the number of elements is increased by a small proportion, and the power conversion efficiency is improved.
Variations and modifications to the above-described embodiments may also occur to those skilled in the art, which fall within the scope of the invention as disclosed and taught herein. Therefore, the present invention is not limited to the above-mentioned embodiments, and any obvious improvement, replacement or modification made by those skilled in the art based on the present invention is within the protection scope of the present invention. Furthermore, although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation.
Claims (9)
1. A C-LLCT-LLT type resonance direct current converter is characterized in that: the resonant circuit comprises a voltage type half-bridge inverter circuit, a C-LLCT-LLT resonant cavity and a full-bridge rectifier circuit which are sequentially connected, wherein the C-LLCT-LLT resonant cavity comprises a first phase resonant circuit and a second phase resonant circuit, and the first phase resonant circuit comprises a first series branch inductor L r1 First parallel branch inductor L p1 A first parallel branch capacitor C p1 And a first high-frequency transformer T 1 The second phase resonant circuit comprises a second series branch inductor L r2 And a second parallel branch inductor L m2 And a second high-frequency transformer T 2 The first series branch inductor L r1 And the second series branch inductor L r2 The input ends of the two capacitors are connected with a series branch capacitor C r One end of said series branch capacitor C r The other end of the first high-frequency transformer T is connected with an output port of the half-bridge inverter circuit 1 And a second high-frequency transformer T 2 The secondary winding of the transformer is connected with the input port of the full-bridge rectification circuit in series, and the first parallel branch circuit capacitor C p1 Is connected in parallel with the secondBranch inductor L m2 When the switching frequency of the converter is higher than L p1 And C p1 At the resonant frequency of (1), then L p1 And C p1 Can form a variable excitation inductance L m1 ，
Wherein, f s Is the switching frequency.
2. A C-LLCT-LLT resonant dc converter as claimed in claim 1, characterized in that: the half-bridge inverter circuit comprises an input voltage-stabilizing capacitor C i A first high-frequency power switch device S 1 And a second high-frequency power switch device S 2 Said input voltage stabilizing capacitor C i The first high-frequency power switching device S 1 And the second high-frequency power switch device S 2 A loop is formed.
3. The C-LLCT-LLT resonant dc-converter according to claim 2, characterized in that: an output port of the half-bridge inverter circuit is arranged in the first high-frequency power switch device S 1 And said second high frequency power switching device S 2 In the meantime.
4. A C-LLCT-LLT resonant dc converter as claimed in claim 1, characterized in that: the full-bridge rectification circuit comprises a first diode D 1 A second diode D 2 A third diode D 3 A fourth diode D 4 And an output voltage-stabilizing filter capacitor C o The first diode D 1 The second diode D 2 The third diode D 3 And the fourth diode D 4 Form a loop, and output voltage-stabilizing filter capacitor C o And the third diode D 3 And the fourth diode D 4 Are connected in parallel.
5. The C-LLCT-LLT resonant DC converter as set forth in claim 4, whereinCharacterized in that: the first diode D 1 And the second diode D 2 A first input port of the full-bridge rectification circuit is arranged between the first input port and the second input port, and the first input port is connected with the first high-frequency transformer T 1 The secondary winding of (2).
6. The resonant dc-to-dc converter of the C-LLCT-LLT type as set forth in claim 5, wherein: the first diode D 1 And the second diode D 2 Is connected to the cathode.
7. The resonant dc-to-dc converter of the C-LLCT-LLT type according to claim 4, wherein: the third diode D 3 And the fourth diode D 4 A second input port of the full-bridge rectification circuit is arranged between the first input port and the second high-frequency transformer T 2 The secondary winding of (2).
8. A C-LLCT-LLT resonant dc converter according to claim 7, characterized in that: the third diode D 3 And the fourth diode D 4 Is connected to the cathode.
9. The resonant dc-to-dc converter of the C-LLCT-LLT type according to claim 4, wherein: the first diode D 1 The second diode D 2 The third diode D 3 And the fourth diode D 4 Are all unidirectional diodes.
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| TWI822511B (en) * | 2022-12-09 | 2023-11-11 | 群光電能科技股份有限公司 | Voltage converter circuit |
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| Publication number | Priority date | Publication date | Assignee | Title |
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| CN112787517A (en) * | 2021-01-28 | 2021-05-11 | 国网宁夏电力有限公司电力科学研究院 | LLC resonant converter with double voltage gain curves |
| CN115296543A (en) * | 2022-07-11 | 2022-11-04 | 南京航空航天大学 | Multiphase parallel resonant converter with automatic current equalization of secondary side discrete windings |
Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR20140144013A (en) * | 2013-06-10 | 2014-12-18 | 청주대학교 산학협력단 | High voltage switching power supply |
| CN106533181A (en) * | 2016-06-29 | 2017-03-22 | 江苏镇安电力设备有限公司 | Double transformer parallel series LLC resonant DC-DC converter and control method of the same |
| CN107968471A (en) * | 2017-11-20 | 2018-04-27 | 山东鲁能智能技术有限公司 | LCLC resonance circuits, wide scope constant power output direct current charger and control method |
| CN108964469A (en) * | 2018-07-16 | 2018-12-07 | 江南大学 | The double LLC resonant converters of a kind of and cascaded structure full-bridge |
| CN109861543A (en) * | 2019-01-28 | 2019-06-07 | 浙江大学 | A kind of wide crisscross parallel type LCLC controlled resonant converter for loading wide gain |
| CN209787049U (en) * | 2019-04-08 | 2019-12-13 | 深圳市航嘉驰源电气股份有限公司 | Resonant converter |
Family Cites Families (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US10491123B2 (en) * | 2015-09-18 | 2019-11-26 | Murata Manufacturing Co., Ltd. | Modular parallel technique for resonant converter |
| CN106787757B (en) * | 2016-12-29 | 2019-07-09 | 天津大学 | A kind of CLTCL resonance DC converter |
| CN107809178B (en) * | 2017-11-03 | 2021-06-11 | 上海联影医疗科技股份有限公司 | High-voltage generator and control method thereof |
-
2020
- 2020-01-19 CN CN202010060719.XA patent/CN111181407B/en not_active Expired - Fee Related
Patent Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR20140144013A (en) * | 2013-06-10 | 2014-12-18 | 청주대학교 산학협력단 | High voltage switching power supply |
| CN106533181A (en) * | 2016-06-29 | 2017-03-22 | 江苏镇安电力设备有限公司 | Double transformer parallel series LLC resonant DC-DC converter and control method of the same |
| CN107968471A (en) * | 2017-11-20 | 2018-04-27 | 山东鲁能智能技术有限公司 | LCLC resonance circuits, wide scope constant power output direct current charger and control method |
| CN108964469A (en) * | 2018-07-16 | 2018-12-07 | 江南大学 | The double LLC resonant converters of a kind of and cascaded structure full-bridge |
| CN109861543A (en) * | 2019-01-28 | 2019-06-07 | 浙江大学 | A kind of wide crisscross parallel type LCLC controlled resonant converter for loading wide gain |
| CN209787049U (en) * | 2019-04-08 | 2019-12-13 | 深圳市航嘉驰源电气股份有限公司 | Resonant converter |
Non-Patent Citations (1)
| Title |
|---|
| "Analysis on Two Structures of Interleaved LLC Converter";Xiangjun Zhang,等;《2018 IEEE International Power Electronics and Applications Conference and Exposition(PEAC)》;20181227;全文 * |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| TWI822511B (en) * | 2022-12-09 | 2023-11-11 | 群光電能科技股份有限公司 | Voltage converter circuit |
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