CN111130333B - Control method, control device, PFC circuit, motor driving device and air conditioner - Google Patents
Control method, control device, PFC circuit, motor driving device and air conditioner Download PDFInfo
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- CN111130333B CN111130333B CN201911343012.3A CN201911343012A CN111130333B CN 111130333 B CN111130333 B CN 111130333B CN 201911343012 A CN201911343012 A CN 201911343012A CN 111130333 B CN111130333 B CN 111130333B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
- H02M1/4208—Arrangements for improving power factor of AC input
- H02M1/4233—Arrangements for improving power factor of AC input using a bridge converter comprising active switches
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Abstract
The invention relates to the field of PFC control, and discloses a control method, a control device, a PFC circuit, motor driving equipment and an air conditioner. The method comprises the steps of detecting phase current of each phase of branch circuit and direct-current bus voltage output by a three-phase PFC circuit, determining a three-phase duty ratio signal conducted by a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage, determining a bidirectional switch driving signal of the three-phase bidirectional switch tube according to the three-phase duty ratio signal, determining a forward switch driving signal of each phase of forward switch tube and a reverse switch driving signal of a reverse switch tube according to the phase current and the bidirectional switch driving signal, and finally respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch tube. The problem of switch tube like IGBT generate heat greatly and have reverse conduction risk among the full conduction scheme among the prior art to and complementary conduction mode current distortion among the prior art is solved, thereby promoted the operational reliability of whole three-phase PFC circuit.
Description
Technical Field
The invention relates to the field of PFC control, in particular to a control method, a control device, a PFC circuit, motor driving equipment and an air conditioner.
Background
When the three-phase PFC circuit based on the VIENNA rectifier is applied, the adopted control mode is generally a full conduction mode or a complementary conduction mode, and the full conduction mode is that three-phase bidirectional switching tubes in the circuit are all conducted. When the control mode is adopted for working, the switch tube has large heat productivity and risks of reverse conduction, so that the switch tube is damaged. And when complementary conduction is carried out, each switching tube of each phase is respectively conducted independently when the voltage of the phase is in a forward direction or a reverse direction, and during control, because zero-crossing points adopt factors such as delay, filtering processing and direct current offset, control errors caused by misjudgment of voltage polarity are easily caused, and current is distorted.
Disclosure of Invention
The invention aims to solve the problem that a switch tube is damaged or current distortion is caused by control errors when a three-phase PFC circuit based on a VIENNA rectifier is controlled, and provides a control method, a control device, a PFC circuit, motor driving equipment and an air conditioner.
In order to achieve the above object, in a first aspect of the present invention, there is provided a control method for a three-phase PFC circuit for a VIENNA rectifier, the control method including:
the method comprises the steps of obtaining phase current of each phase of branch in three-phase branches and direct-current bus voltage output by a three-phase PFC circuit;
determining a three-phase duty ratio signal conducted by a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage;
determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal;
determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal;
and respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal.
Optionally, the determining the forward switch driving signal of the forward switch tube and the reverse switch driving signal of the reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal comprises:
under the condition that the phase current of the positive phase of one phase branch is larger than the preset positive current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the positive switch tube of the phase and the reverse switch tube of the phase is closed;
under the condition that the phase current of the negative phase of one phase of branch circuit is greater than negative preset current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the reverse switch tube of the phase and the positive switch tube of the phase is closed;
and under the condition that the phase current of the positive phase of one phase is less than or equal to a positive preset current and the phase current of the negative phase of the one phase is less than or equal to a negative preset current, determining the bidirectional switch driving signal of the phase as the duty ratio of the positive switch tube of the phase and the duty ratio of the reverse switch tube of the phase.
Optionally, obtaining the phase current of each of the three-phase branches includes:
obtaining phase currents of two phases of the current signals;
phase currents of a third phase are determined based on the phase currents of the two phases.
In a second aspect of the present invention, there is provided a control device for a three-phase PFC circuit of a VIENNA rectifier, the control device comprising:
phase current detection equipment for detecting the phase current of each phase branch in the three-phase branches;
the bus voltage detection equipment is used for detecting the direct-current bus voltage output by the three-phase PFC circuit;
a PFC processor configured to:
acquiring a phase current from a phase current detecting device;
acquiring direct-current bus voltage from bus voltage detection equipment;
determining a three-phase duty ratio signal conducted by a three-phase bidirectional switching tube according to the phase current and the direct-current bus voltage;
determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal;
determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal;
and respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal.
Optionally, the PFC processor is further configured to:
under the condition that the positive phase electric current of one phase branch is larger than the positive preset current, determining the duty ratio of the phase as the duty ratio of the positive switch tube of the phase;
under the condition that the negative phase current of one phase branch is larger than the negative preset current, determining the duty ratio of the phase as the duty ratio of the opposite-direction switching tube;
and under the condition that the positive phase current of one phase is less than or equal to the positive preset current and the negative phase current of one phase is less than or equal to the negative preset current, controlling the conduction of the positive switch tube and the reverse switch tube of the phase.
Optionally, the PFC processor is further configured to:
obtaining phase currents of two phases of the current signals;
phase currents of a third phase are determined based on the phase currents of the two phases.
In a third aspect of the invention, a three-phase PFC circuit for a VIENNA rectifier is provided, which includes the control device for the three-phase PFC circuit for the VIENNA rectifier.
In a fourth aspect of the present invention, there is provided a motor driving device including: a three-phase PFC circuit;
the direct current output end of the three-phase PFC circuit is connected with the power input end of the intelligent power module to provide working high-voltage direct current for the intelligent power module, and the output end of the intelligent power module outputs a three-phase alternating current signal to drive the motor to operate;
the direct current bus current sampling device is used for sampling direct current bus current of the three-phase PFC circuit for supplying power to the intelligent power module; and
a motor processor configured to:
acquiring direct current bus current and direct current bus voltage;
and determining six switching signals for controlling the intelligent power module according to the direct-current bus voltage and the direct-current bus current so as to control the intelligent power module to drive the motor to operate.
Optionally, the motor processor is further configured to:
estimating the rotor position of the motor to obtain a rotor angle estimation value and a motor speed estimation value of the motor;
calculating a Q-axis given current value according to the motor target rotating speed value and the motor speed estimated value;
calculating a D-axis given current value according to the maximum output voltage of the inverter and the output voltage amplitude of the inverter;
and calculating according to the Q-axis given current value, the D-axis given current value, the motor speed estimation value, the direct-current bus voltage value and the phase current value to generate a pulse width signal, and generating a PWM control signal to the intelligent power module according to the triangular carrier signal and the pulse width signal to drive the motor to operate.
In a fifth aspect of the present invention, there is provided an air conditioner including the motor driving apparatus described above.
According to the control method for the three-phase PFC circuit of the VIENNA rectifier, the phase current of each phase branch and the direct-current bus voltage output by the three-phase PFC circuit are detected, the three-phase duty ratio signal of the three-phase bidirectional switch tube is determined according to the phase current and the direct-current bus voltage, the bidirectional switch driving signal of the three-phase bidirectional switch tube is determined according to the three-phase duty ratio signal, the forward switch driving signal of each phase of forward switch tube and the reverse switch driving signal of each phase of reverse switch tube are determined according to the phase current and the bidirectional switch driving signal, and finally the forward switch tube and the reverse switch tube are respectively controlled to work according to the forward switch driving signal and the reverse switch tube. The invention solves the problems of high heating and reverse conduction risk of a switching tube such as an Insulated Gate Bipolar Transistor (IGBT) in a full conduction scheme in the prior art and the problem of current distortion of a complementary conduction mode in the prior art. Therefore, the working reliability of the whole three-phase PFC circuit is improved.
Drawings
Fig. 1 schematically illustrates an application circuit block diagram of a three-phase PFC circuit for a VIENNA rectifier according to an embodiment of the present invention;
fig. 2 schematically illustrates a flow chart of a control method for a three-phase PFC circuit for a VIENNA rectifier in accordance with an embodiment of the present invention;
fig. 3 schematically shows waveform diagrams of the bidirectional switch drive signal a _ SW of the a-phase, the duty signal DutyA of the a-phase, and the carrier signal Ca;
fig. 4 schematically shows waveform diagrams of the a-phase actual voltage Ua, the corrected a-phase voltage VA _ adj, the detected a-phase voltage VA _ samp, and the a-phase forward-switch drive signal T1 and the reverse-switch drive signal T2;
FIG. 5 schematically illustrates a control logic block diagram for determining the forward and reverse switch drive signals for each phase;
FIG. 6 schematically illustrates a block diagram internal to the motor processor 50;
fig. 7 schematically shows a corresponding relationship diagram of a PWM signal for controlling the inverter and an isosceles triangle carrier signal.
Detailed Description
The following detailed description of embodiments of the invention refers to the accompanying drawings. It should be understood that the detailed description and specific examples, while indicating the present invention, are given by way of illustration and explanation only, not limitation.
It should be noted that if the present invention relates to directional indications (such as up, down, left, right, front, and back \8230;), the directional indications are only used to explain the relative position relationship between the components, the motion situation, etc. in a specific posture (as shown in the attached drawings), and if the specific posture is changed, the directional indications are correspondingly changed.
In addition, if there is a description of "first", "second", etc. in the embodiments of the present invention, the description of "first", "second", etc. is for descriptive purposes only and is not to be construed as indicating or implying relative importance or implicitly indicating the number of technical features indicated. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one such feature. In addition, technical solutions between the various embodiments can be combined with each other, but must be realized by a person skilled in the art, and when the technical solutions are contradictory or cannot be realized, the combination of the technical solutions should be considered to be absent and not be within the protection scope of the present invention.
The embodiment of the invention provides a control method of a three-phase PFC circuit for a VIENNA rectifier.
Fig. 1 schematically shows an application circuit block diagram of a three-phase PFC circuit for a VIENNA rectifier according to an embodiment of the present invention. Referring to fig. 1, the vienna rectifier mainly comprises three-phase boost inductors La, lb and Lc, three-phase diode rectifier bridges D1 to D6 and three-phase bidirectional switches T1 to T6, wherein each phase of bidirectional switch has two IGBT (insulated gate bipolar transistors) to form a common emitter back-to-back type, and bidirectional conduction is realized by using intrinsic freewheeling diodes therein. The processor outputs six switching tube signals to control the three-phase bidirectional switches T1-T6 to work, so that the VIENNA rectifier works to convert input alternating current into high-voltage direct current, for example, three-phase alternating current such as power frequency 220V is processed by a three-phase PFC circuit of the VIENNA rectifier, a high-voltage direct current circuit of about 650V is output to supply power to a following load such as a three-phase inverter in the figure 1, and meanwhile, the three-phase inverter drives a motor such as a variable frequency compressor or a direct current motor to run through the control of the processor, so that a complete control process is realized.
Fig. 2 schematically shows a flowchart of a control method for a three-phase PFC circuit of a VIENNA rectifier according to an embodiment of the present invention. Referring to fig. 1, the control method includes:
step S100: the method comprises the steps of obtaining phase current of each phase of branch in three-phase branches and direct-current bus voltage output by a three-phase PFC circuit;
step S200: determining a three-phase duty ratio signal conducted by a three-phase bidirectional switching tube according to the phase current and the direct-current bus voltage;
step S300: determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal;
step S400: determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal;
step S500: and respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal.
In step S100, phase current of each phase may be sampled by a current sampling circuit in an ac branch of the three-phase input, where the current sampling circuit may adopt an existing current sampling circuit based on a sampling resistor, and the phase current of the phase may be determined by detecting a voltage across the sampling resistor. To obtain phase currents Ia, ib and Ic for each phase, respectively. The output end of the three-phase PFC circuit can sample the output direct-current bus voltage Udc through a simple resistance voltage division circuit.
In step S200, reference may be made to a scheme of determining a three-phase duty ratio signal in the prior art. Such as the one-cycle integral reset PFC control methods referred to in the prior art.
In the method for controlling the single-cycle integral reset PFC, taking the determination of the duty ratio signal of the A phase as an example, the A phase duty ratio signal D can be determined by detecting the A phase current Ia and the direct current bus voltage Udc based on the following formula:
in the formula, vm = Vo Rs/Re, where Vo is the dc bus voltage, rs is the input current equivalent detection resistor, re is the a-phase converter equivalent resistor, rs and Re can be determined according to empirical values, lin is the a-phase current, T is the switching period of the bidirectional switching tube, and D is the duty ratio signal.
The duty cycle signals of the other two phases are similar to the above-described determination process of the duty cycle signal of the a phase.
In step S300, after obtaining the three-phase duty ratio signals Da, db, and Dc, the carrier signal Ca may be further combined to determine the bidirectional switch driving signals a _ SW, B _ SW, and C _ SW of the three-phase bidirectional switch tube. It can be specifically determined according to the following formulas (1) to (3):
the bidirectional switching device comprises a bidirectional switching tube, a DutyA, dutyB and DutyC, wherein A _ SW, B _ SW and C _ SW are respectively a bidirectional switching tube corresponding to A, B and C three phases, dutyA, dutyB and DutyC are respectively duty ratio signals for conducting the bidirectional switching tube of the A, B and C three phases, and Ca is a carrier signal.
Based on the above formula (1), fig. 3 schematically shows waveform diagrams of the bidirectional switch drive signal a _ SW of the a phase, the duty ratio signal DutyA of the a phase, and the carrier signal Ca.
In steps S400 and S500, the duty ratios of the two switching tubes of each phase, i.e., the forward switching tube and the reverse switching tube, are determined according to the three-phase currents Ia, ib and Ic and the bidirectional switch driving signals a _ SW and B _ SW of the three-phase bidirectional switching tubes, so as to control the forward switching tube and the reverse switching tube to be respectively conducted when the voltage of each phase is the positive phase and the negative phase.
In the prior art, as mentioned in the background, there are two control modes for controlling the forward switch tube and the reverse switch tube of each phase, i.e. a full conduction mode or a complementary conduction mode. Wherein, the full conduction is that the forward switch tube and the reverse switch tube of each phase are controlled to be conducted simultaneously according to the duty ratio signal of each phase; the complementary conduction is to control the conduction of the forward switch tube and the reverse switch tube respectively according to the phase voltage polarity of each phase, namely, when the phase voltage is a positive phase voltage, the conduction of the forward switch tube is controlled, and when the phase voltage is a negative phase voltage, the conduction of the reverse switch tube is controlled.
In the embodiment of the invention, the difference from strictly controlling the conduction of the forward switching tube and the reverse switching tube according to the phase voltage polarity of each phase is that the embodiment of the invention controls the forward switching tube and the reverse switching tube to work according to the sampled positive phase current and negative phase preset current, and in a preset interval of transition of the positive phase current and the negative phase preset current, the forward switching tube and the reverse switching tube are both conducted simultaneously.
The specific control mode is as follows:
taking phase a as an example, the forward switch drive signal and the reverse switch drive signal of phase a can be determined by equation (4) according to phase a current Ia and the bidirectional switch drive signal of phase a:
wherein Ia is phase A current, and T1 and T2 are respectively a forward switch driving signal and a reverse switch driving signal of phase A. Ia >0 when the A-phase current is positive, ia <0, izero _upis a positive preset current and Izero _ up >0, izero _dwis a negative preset current and Izero _ dw <0 when the A-phase current is negative. According to the formula (4), when the A-phase current Ia is greater than the forward preset current Izero _ up, the A-phase current is in a positive phase, the forward switch tube is controlled to be switched on by the bidirectional switch driving signal A _ SW, and the reverse switch tube is controlled to be switched off; when the A-phase current Ia is smaller than the negative preset current Izero _ dw, the A-phase current is in a negative phase, and the negative switch tube is controlled to be switched on by the bidirectional switch driving signal A _ SW and the positive switch tube is controlled to be switched off; when the A-phase current Ia is larger than the negative preset current Izero _ dw and smaller than the positive preset current Izero _ up, the A-phase current is in a section of transition of the positive phase current and the negative phase current, and the positive switch tube and the reverse switch tube are controlled to be simultaneously switched on by the bidirectional switch driving signal A _ SW. Wherein Izero _ up and Izero _ dw can be determined by equation (5) and equation (6), respectively:
izero _ up = Iaup _ peak X% formula (5)
Izero _ dw = Iadw _ peak X% formula (6)
Wherein Iaup _ peak is a positive phase current peak value, iadw _ peak is a negative phase current peak value, and X% is a preset percentage, which can be determined according to previous experiments, and generally can be taken as 4% -11%, for example, 5%.
As can be seen from the above formula (4), formula (5) and formula (6), when Izero _ dw is not less than Ia and not more than Izero _ up, the A-phase current is in a narrow range around the zero crossing point of the current, and at this time, the forward switch tube and the reverse switch tube are both controlled to be simultaneously conducted by the bidirectional switch driving signal A _ SW.
Based on the above equation (4), fig. 4 schematically shows waveform diagrams of the a-phase actual current Ia, the a-phase forward switch drive signal T1, and the reverse switch drive signal T2.
Similarly, the forward switching drive signal and the reverse switching drive signal corresponding to the two phases B and C can be determined by the following equations (7) and (8):
wherein, T3 and T4 are respectively a forward switch driving signal and a reverse switch driving signal of the B phase.
Wherein T5 and T6 are the forward switch driving signal and the reverse switch driving signal of the C phase, respectively.
Fig. 5 schematically shows a control logic block diagram for determining the forward switch drive signal and the reverse switch drive signal for each phase according to the above equations (1) to (4), equation (7), and equation (8).
Taking an a phase as an example, a 40KHz triangular wave carrier signal Ca and a duty ratio signal DutyA of an a-phase bidirectional switch tube are compared by a seventh comparator to output a bidirectional switch driving signal a _ SW, namely, a determination method of formula (1), the a-phase current is filtered by a first low-pass filter and then respectively enters a non-inverting input terminal of a first comparator and an inverting input terminal of a second comparator, a positive preset current Izero _ up is simultaneously input to the non-inverting input terminal of the second comparator, a negative preset current Izero _ dw is input to the inverting input terminal of the first comparator to be compared respectively, the output terminal of the first comparator enters an input terminal of a first and gate, the output terminal of the second comparator enters an input terminal of a second and gate, the bidirectional switch driving signal a _ SW simultaneously enters the other input terminals of the first and gate and the second and gate to realize the determination method of formula (4), and finally, the a positive switch driving signal T1 and the reverse switch driving signal T2 of the a phase are respectively output from the first and the second and gate.
And finally, controlling the forward switching tube and the reverse switching tube of each phase to work according to the switch driving signals, thereby realizing the work of the three-phase PFC circuit of the VIENNA rectifier.
The control method for the three-phase PFC circuit of the VIENNA rectifier comprises the steps of detecting phase current of each phase branch and direct-current bus voltage output by the three-phase PFC circuit, determining a three-phase duty ratio signal of conduction of a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage, determining a bidirectional switch driving signal of the three-phase bidirectional switch tube according to the three-phase duty ratio signal, determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal, and finally respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal. The invention solves the problems of large heating and reverse conduction risk of a switching tube such as an IGBT in a full conduction scheme in the prior art and the problem of current distortion of a complementary conduction mode in the prior art, and controls the forward switching tube and the reverse switching tube to be simultaneously conducted in a narrow range width (realized by X percent) above and below a phase current zero crossing point position according to the detected phase current, thereby solving the problems of current distortion and heating. Therefore, the working reliability of the whole three-phase PFC circuit is improved. Meanwhile, the control method of the embodiment of the invention does not need to detect the phase voltage of the three-phase branch, thereby reducing phase voltage sampling circuits and simplifying the whole circuit structure and processing process.
In a preferred embodiment of the present invention, obtaining the phase current of each of the three phase legs comprises:
step S110: obtaining phase currents of two phases of the current signals;
step S120: phase currents of a third phase are determined based on the phase currents of the two phases.
Taking the phase currents of the a phase and the B phase as an example, after the a phase current Ia and the B phase current Ib are detected, the C phase current Ic may be calculated by the company, and may be determined in formula (9):
ic = -Ia-Ib equation (9)
Compared with the independent sampling of the phase current of each phase, only two phase currents need to be sampled, so that the processor can sample only through two ports, and because the sampling voltage value requires that the port of the processor is an AD (analog-to-digital conversion) port, one path of AD port can be saved, thereby saving the port resource of the processor and reducing the resource requirement of the processor.
The invention also provides a control device of the three-phase PFC circuit for the VIENNA rectifier. The specific circuit block diagram of the control device can refer to fig. 1. The control device includes:
phase current detection equipment for detecting the phase current of each phase branch in the three-phase branches;
a bus voltage detection device 34 for detecting a dc bus voltage output from the three-phase PFC circuit;
a PFC processor 20 configured to: obtaining phase current from phase current detection equipment, obtaining direct-current bus voltage from bus voltage detection equipment, and determining a three-phase duty ratio signal of the conduction of a three-phase bidirectional switch tube according to the phase voltage and the direct-current bus voltage; determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal; determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal; and respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal.
The phase current of each phase can be sampled by a current sampling circuit in an alternating current branch of the three-phase input, wherein the current sampling circuit can adopt an existing current sampling circuit based on a sampling resistor, and the phase current of the phase can be determined by detecting the voltage on the sampling resistor so as to respectively obtain the phase currents Ia, ib and Ic of each phase.
The bus voltage detection device 34 may specifically sample the dc bus voltage Udc output by the three-phase PFC circuit through a simple resistance voltage division circuit at the output terminal of the three-phase PFC circuit.
Or as in the scheme shown in fig. 1, the detection of the two-phase currents is realized by detecting the phase currents of two phases, namely, the phase current detection device 31 of the a-phase and the phase current detection device 32 of the B-phase, when the PFC processor 20 is further configured to: acquiring a middle stream; phase currents of a third phase are determined based on the phase currents of the two phases.
Taking the phase currents of the a phase and the B phase as an example, after the a phase current Ia and the B phase current Ib are detected, the C phase current Ic may be calculated by the company, and may be determined in formula (9):
ic = -Ia-Ib equation (9)
When a three-phase duty ratio signal of the conduction of a three-phase bidirectional switching tube is determined according to phase voltage and direct-current bus voltage, a single-period integral reset PFC control method in the prior art is referred.
In the method for controlling the one-cycle integral reset PFC, taking the determination of the duty ratio signal of the A phase as an example, the A phase duty ratio signal D can be determined by detecting the A phase current Ia and the direct current bus voltage Udc based on the following formula:
in the formula, vm = Vo Rs/Re, where Vo is the dc bus voltage, rs is the input current equivalent detection resistor, re is the a-phase converter equivalent resistor, rs and Re can be determined according to empirical values, lin is the a-phase current, T is the switching period of the bidirectional switching tube, and D is the duty ratio signal.
The duty cycle signals of the other two phases are similar to the above-described determination process of the duty cycle signal of the a phase.
In fig. 1, the duty ratio calculation module 21 in the PFC processor 20 is used to realize the calculation function of the three-phase duty ratio signal.
After the three-phase duty ratio signals Da, db and Dc are obtained, the carrier signal Ca can be further combined to determine bidirectional switch driving signals A _ SW, B _ SW and C _ SW of the three-phase bidirectional switch tube. It can be specifically determined according to the following formulas (1) to (3):
the bidirectional switching device comprises a bidirectional switching tube, a DutyA, dutyB and DutyC, wherein A _ SW, B _ SW and C _ SW are respectively a bidirectional switching tube corresponding to A, B and C three phases, dutyA, dutyB and DutyC are respectively duty ratio signals for conducting the bidirectional switching tube of the A, B and C three phases, and Ca is a carrier signal.
Based on the above formula (1), fig. 3 schematically shows waveform diagrams of the bidirectional switch drive signal a _ SW of the a phase, the duty ratio signal DutyA of the a phase, and the carrier signal Ca.
And determining the duty ratios of two switching tubes of each phase, namely a forward switching tube and a reverse switching tube according to the three-phase current Ia, ib and Ic and the bidirectional switch driving signals A _ SW and B _ SW of the three-phase bidirectional switching tubes so as to control the forward switching tube and the reverse switching tube to be respectively conducted when the voltage of each phase is a positive phase and a negative phase.
In the prior art, as mentioned in the background, there are two control modes for controlling the forward switch tube and the reverse switch tube of each phase, i.e. a full conduction mode or a complementary conduction mode. Wherein, the full conduction is that the forward switch tube and the reverse switch tube of each phase are controlled to be conducted simultaneously according to the duty ratio signal of each phase; the complementary conduction is to control the conduction of the forward switch tube and the reverse switch tube respectively according to the phase voltage polarity of each phase, namely, when the phase voltage is a positive phase voltage, the conduction of the forward switch tube is controlled, and when the phase voltage is a negative phase voltage, the conduction of the reverse switch tube is controlled.
In the embodiment of the invention, the difference from strictly controlling the conduction of the forward switching tube and the reverse switching tube according to the phase voltage polarity of each phase is that the embodiment of the invention controls the forward switching tube and the reverse switching tube to work according to the sampled positive phase current and negative phase preset current, and in a preset interval of transition of the positive phase current and the negative phase preset current, the forward switching tube and the reverse switching tube are both conducted simultaneously.
The specific control mode is as follows:
taking phase a as an example, the forward switch drive signal and the reverse switch drive signal of phase a can be determined by equation (4) according to phase a current Ia and the bidirectional switch drive signal of phase a:
wherein Ia is phase A current, and T1 and T2 are respectively a forward switch driving signal and a reverse switch driving signal of phase A. Ia >0 when the A-phase current is positive, ia <0, izero _upis a positive preset current and Izero _ up >0, izero _dwis a negative preset current and Izero _ dw <0 when the A-phase current is negative. According to the formula (4), when the A-phase current Ia is greater than the forward preset current Izero _ up, the A-phase current is in a positive phase, the forward switch tube is controlled to be switched on by the bidirectional switch driving signal A _ SW, and the reverse switch tube is controlled to be switched off; when the A-phase current Ia is smaller than the negative preset current Izero _ dw, the A-phase current is in a negative phase, the negative switch tube is controlled to be switched on by a bidirectional switch driving signal A _ SW, and the positive switch tube is controlled to be switched off; when the A-phase current Ia is larger than the negative preset current Izero _ dw and smaller than the positive preset current Izero _ up, the A-phase current is in a section of transition of the positive phase current and the negative phase current, and the positive switch tube and the reverse switch tube are controlled to be simultaneously switched on by the bidirectional switch driving signal A _ SW. Wherein Izero _ up and Izero _ dw can be determined by equation (5) and equation (6), respectively:
izero _ up = Iaup _ peak X% formula (5)
Izero _ dw = Iadw _ peak X% formula (6)
Wherein Iaup _ peak is a positive phase current peak value, iadw _ peak is a negative phase current peak value, and X% is a preset percentage, which can be determined according to previous experiments, and generally can be taken as 4% -11%, for example, 5%.
As can be seen from the above formula (4), formula (5) and formula (6), when Izero _ dw is not less than Ia and not more than Izero _ up, the A-phase current is in a narrow range around the zero crossing point of the current, and at this time, the forward switch tube and the reverse switch tube are both controlled to be simultaneously conducted by the bidirectional switch driving signal A _ SW.
Based on the above equation (4), fig. 4 schematically shows waveform diagrams of the a-phase actual current Ia, the a-phase forward switch drive signal T1, and the reverse switch drive signal T2.
Similarly, the forward switching drive signal and the reverse switching drive signal corresponding to the two phases B and C can be determined by the following equations (7) and (8):
wherein, T3 and T4 are respectively a forward switch driving signal and a reverse switch driving signal of the B phase.
Wherein T5 and T6 are the forward switch driving signal and the reverse switch driving signal of the C phase, respectively.
Fig. 5 schematically shows a control logic block diagram for determining the forward switch drive signal and the reverse switch drive signal for each phase according to the above equations (1) to (4), equation (7), and equation (8).
Taking an a phase as an example, a 40KHz triangular wave carrier signal Ca and a duty ratio signal DutyA of an a-phase bidirectional switch tube are compared by a seventh comparator to output a bidirectional switch driving signal a _ SW, namely, a determination method of formula (1), the a-phase current is filtered by a first low pass filter and then respectively enters a non-inverting input terminal of a first comparator and an inverting input terminal of a second comparator, a positive preset current Izero _ up is simultaneously input to a non-inverting input terminal of the second comparator, a negative preset current Izero _ dw is input to an inverting input terminal of the first comparator to be compared respectively, the positive preset current Izero _ up enters an input terminal of the first and gate from an output terminal of the first comparator, the negative preset current Izero _ dw enters an inverting input terminal of the second and gate from an output terminal of the second comparator, the bidirectional switch driving signal a _ SW simultaneously enters the other input terminals of the first and gate and the second and gate, the determination method of formula (4) is implemented, and finally, the a positive switch driving signal T1 and the reverse switch driving signal T2 of the a phase are respectively output from the first and the second and gate.
And finally, controlling the forward switch tube and the reverse switch tube of each phase to work according to the switch driving signals, so that the work of the three-phase PFC circuit of the VIENNA rectifier is realized.
The control device for the three-phase PFC circuit of the VIENNA rectifier detects the phase current of each phase branch through the phase current detection equipment, detects the direct-current bus voltage output by the three-phase PFC circuit through the bus voltage detection equipment, determines a three-phase duty ratio signal of conduction of a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage, determines a bidirectional switch driving signal of the three-phase bidirectional switch tube according to the three-phase duty ratio signal, determines a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal, and finally controls the forward switch tube and the reverse switch tube to work respectively according to the forward switch driving signal and the reverse switch driving signal. The invention solves the problems of large heating and reverse conduction risk of a switching tube such as an IGBT in a full conduction scheme in the prior art and the problem of current distortion of a complementary conduction mode in the prior art, and controls the forward switching tube and the reverse switching tube to be simultaneously conducted in a narrow range width (realized by X percent) above and below a phase current zero crossing point position according to the detected phase current, thereby solving the problems of current distortion and heating. Therefore, the working reliability of the whole three-phase PFC circuit is improved. Meanwhile, the control device of the embodiment of the invention does not need to detect the phase voltage of the three-phase branch, thereby reducing phase voltage sampling circuits and simplifying the whole circuit structure and processing process.
The invention also provides a three-phase PFC circuit for a VIENNA rectifier, which can refer to the circuit simplified diagram of FIG. 1 and comprises the control device for the three-phase PFC circuit of the VIENNA rectifier. Referring to fig. 1, the three-phase rectifier circuit further includes inductors La, lb and Lc and VIENNA rectifiers 10 respectively connected to the three-phase input terminals. The switching driving signals for controlling the three-phase bidirectional switches T1 to T6 are output by the control device and are driven by the first driving signal driving device 33, so as to realize amplification and signal level conversion of the six switching driving signals, because the signal level output from the PFC processor 20 of the control device is relatively low, for example, between 3V and 5V, and the level of 12V or more is required for driving the three-phase bidirectional switches T1 to T6 to operate, and therefore, the level conversion of the six switching driving signals needs to be performed by the first driving signal driving device 33, so that the six three-phase bidirectional switches T1 to T6 can be normally driven to operate. The high-voltage direct-current bus voltage is output through the three-phase PFC circuit, for example, for the input 220V power frequency three-phase alternating current, the high-voltage direct-current voltage output by the three-phase PFC circuit can reach about 650V, and power is supplied to subsequent loads.
The invention also provides motor driving equipment. The motor driving apparatus may refer to a circuit block diagram shown in fig. 1, wherein the three-phase PFC circuit for the VIENNA rectifier includes:
the direct current output end of the three-phase PFC circuit is connected with the power input end of the inversion module 40 to provide working high-voltage direct current for the inversion module 40, and the output end of the inversion module 40 outputs a three-phase alternating current signal to drive the motor 60 to operate;
the dc bus current sampling device 35 is configured to sample a dc bus current for supplying power to the inverter power module by the three-phase PFC circuit;
a motor processor 50 configured to:
acquiring direct current bus current and direct current bus voltage;
and determining six switching signals for controlling the inverter module 30 according to the direct-current bus voltage and the direct-current bus current so as to control the inverter module 30 to drive the motor to operate.
The following steps may be specifically performed by the motor processor 50 determining and controlling the six switching signals of the inverter module 30 according to the dc bus voltage and the dc bus current:
estimating the rotor position of the motor to obtain a rotor angle estimated value and a motor speed estimated value of the motor;
calculating a Q-axis given current value according to the motor target rotating speed value and the motor speed estimated value;
calculating a D-axis given current value according to the maximum output voltage of the inverter and the output voltage amplitude of the inverter;
and calculating according to the Q-axis given current value, the D-axis given current value, the motor speed estimation value, the direct current bus voltage value and the phase current value to generate a pulse width signal, generating a PWM control signal according to the triangular carrier signal and the pulse width signal, and performing level conversion and amplification on the signal to the inverter module 30 through the second driving module 36 so as to drive the motor 60 to operate.
Fig. 6 schematically shows a block diagram of the inside of the motor processor 50. Referring to fig. 1, in order to implement the above-mentioned step of determining the six switching signals for controlling the inverter module 30 according to the dc bus voltage and the dc bus current, the motor processor 50 may specifically be implemented by the following processing modules:
a position/speed estimation module 51 for estimating a rotor position of the motor to obtain a rotor angle estimation value θ est and a motor speed estimation value ω est of the motor 60;
a Q-axis given current value Iqref calculation module 52, configured to calculate a Q-axis given current value Iqref according to the motor target rotation speed value ω ref and the motor speed estimation value ω est;
a D-axis given current value Idref calculation module 53 for calculating a D-axis given current value Idref from the maximum output voltage Vmax of the inverter and the output voltage amplitude V1 of the inverter;
and the current control module 54 is configured to calculate phase current values Iu, iv, and Iw sampled by the motor 60 according to the Q-axis given current value Iqref, the D-axis given current value Idref, the motor speed estimation value ω est, the dc bus voltage value Udc, and generate a PWM control signal to the inverter 8 according to the triangular carrier signal and the pulse width signal, so as to drive the motor 60 to operate.
Specifically, the motor 60 in the embodiment of the present invention may be a motor without a position sensor, and when the position/speed estimation module 51 determines the rotor angle estimation value θ est and the motor speed estimation value ω est of the motor 60, the above-mentioned functions may be implemented by flux linkage observation, specifically, first, the voltage V on the two-phase stationary coordinate system may be used as the reference α 、V β And current I α 、I β And calculating the estimated values of the effective magnetic fluxes of the compressor motor in the directions of the alpha axis and the beta axis of the two-phase static coordinate system according to the following formula (21):
wherein the content of the first and second substances,
and
estimated values of the effective flux of the motor in the alpha and beta directions, V α And V β Voltages in the alpha and beta axis directions, I α And I β Current in the alpha and beta axis directions, R is stator resistance, L q Is the q-axis inductance parameter of the motor.
Then, a rotor angle estimation value θ est of the compressor motor and an actual rotation speed value ω est of the motor are calculated according to the following formula (22):
wherein, K p_pll And K i_pll Respectively, a proportional integral parameter, theta err As an estimate of the deviation angle, ω f The bandwidth of the velocity low pass filter.
Specifically, the Q-axis given current value calculation block 522 includes a superposition unit and a PI regulator. The PI regulator is used for carrying out PI regulation according to the difference between the motor target rotating speed value omega ref and the motor speed estimation value omega est output by the superposition unit so as to output a Q-axis given current value Iqref.
Specifically, the D-axis given current value calculation module 523 includes a weak magnetic controller and a limiting unit, wherein the weak magnetic controller is configured to calculate a maximum output voltage Vmax of the inverter 8 and an output voltage amplitude V1 of the inverter 8 to obtain a D-axis given current value initial value Id0, and the limiting unit is configured to perform limiting processing on the D-axis given current value initial value Id0 to obtain a D-axis given current value Idref.
In an embodiment of the present invention, the field weakening controller may calculate the D-axis given current value initial value Id0 according to the following equation (23):
wherein, I d0 Setting the initial value of current for D axis, K i In order to integrate the control coefficients of the motor,
V 1 is the output voltage amplitude, v, of the inverter d Is D-axis voltage, v q Is Q-axis voltage, V max Is the maximum output voltage, V, of the inverter 8 dc Which is the dc bus voltage output by the rectifier 4.
In an embodiment of the present invention, the clipping unit obtains the D-axis given current value according to the following formula (24):
where Idref is the D-axis given current value, I demag Is the demagnetization current limit value of the motor.
Specifically, the current control module 54 calculates as follows:
the current I of the motor in the directions of the alpha axis and the beta axis of the two-phase static coordinate system is obtained by sampling the motor 60 to obtain the U, V and W three-phase current values Iu, iv and Iw, performing Clark conversion through a three-phase static-two-phase static coordinate conversion unit and obtaining the current I of the motor in the directions of the alpha axis and the beta axis of the two-phase static coordinate system based on the following formula (25) α And I β
I α ＝I u
Then according to the rotor angle estimated value theta est The real current values Iq and Id of the D axis and the Q axis in the two-phase rotating coordinate system are calculated by the following formula (26) through Park conversion performed by the two-phase stationary-two-phase rotating coordinate conversion unit.
I d ＝I α cosθ est +I β sinθ est
I q ＝-I α sinθ est +I β cosθ est (26)
The calculation of the actual current values Iq, id of the D axis and Q axis by the Q axis current value and D axis current value calculating unit in the current control module 54 is realized by the above-described formula (25) and formula (26).
Further, the current control module 54 may calculate the Q-axis given voltage value and the D-axis given voltage value according to the following equation (27):
V d ＝V d0 －ωL q I q
V q ＝V q0 +ωL d I d +ωK e (27)
vq is a Q-axis given voltage value, vd is a D-axis given voltage value, iqref is a Q-axis given current value, idref is a D-axis given current value, iq is Q-axis current, id is D-axis current, kpd and Kid are D-axis current control proportional gain and integral gain respectively, kpq and Kiq are Q-axis current control proportional gain and integral gain respectively, omega is motor rotation speed, ke is motor 60 back electromotive force coefficient, ld and Lq are D-axis and Q-axis inductors respectively, the two parameters can be provided by a motor manufacturer, and particularly can be provided according to the D-axis and Q-axis follow-up current of the motor provided by the motor manufacturerThe nominal value is taken from the flow profile,
denotes the integral of x (τ) over time.
Further, in order to further accurately obtain the D-axis inductor Ld and the Q-axis inductor Lq, the current control module 54 is further configured to: the method comprises the steps of obtaining phase current values of motor operation, calling a first Q-axis inductance, a second Q-axis inductance, a first D-axis inductance and a second D-axis inductance which correspond to a prestored first phase current value and a prestored second phase current value respectively, and calculating the Q-axis inductance and the D-axis inductance according to the phase current values, the first phase current value, the second phase current value, the first Q-axis inductance, the second Q-axis inductance, the first D-axis inductance and the second D-axis inductance. Specifically, the phase current signals Iu, iv, and Iw of the motor 60 acquired by the current sampling unit 9 are obtained, and the three phase currents have the same magnitude and only need to be one of the three phase currents. A D-axis inductance and Q-axis inductance variation curve chart of the motor provided by a motor manufacturer, wherein i is a winding current of the motor, i.e. a phase current value, at this time, a first Q-axis inductance Lq1, a second Q-axis inductance Lq2, a first D-axis inductance Ld1, and a second D-axis inductance Ld2, which correspond to a first phase current value i1 and a second phase current value i2, respectively, are prestored through the curve chart, and a D-axis inductance Ld and a Q-axis inductance Lq, which correspond to a currently detected phase current i, can be calculated according to the following difference calculation formula (28):
Ld＝Ld1+(Ld2-Ld1)*(i-i1)/(i2-i1)
Lq＝Lq1+(Lq2-Lq1)*(i-i1)/(i2-i1) (28)
the D-axis inductance Ld and the Q-axis inductance Lq corresponding to the current phase current of the motor 60 can be relatively accurately determined through the formula (28).
After the Q-axis given voltage value Vq and the D-axis given voltage value Vd are obtained, the angle estimation value theta of the motor rotor can be obtained est And carrying out Park inverse transformation on Vq and Vd through a two-phase rotation-two-phase static coordinate conversion unit to obtain voltage values V alpha and V beta on a fixed coordinate system, wherein a specific transformation formula (29) is as follows:
where θ is the rotor angle of the motor 60, the rotor angle estimate θ est may be used.
Further, clark inverse transformation can be performed through a two-phase static-three-phase static coordinate conversion unit according to the voltage values V α and V β on the fixed coordinate system to obtain three-phase voltages Vu, vv and Vw, and the specific transformation formula (30) is as follows:
V u ＝V α
then, the duty ratio calculation unit can perform duty ratio calculation according to the direct-current bus voltage Udc and the three-phase voltages Vu, vv and Vw to obtain duty ratio control signals, namely three-phase duty ratios Du, dv and Dw, and a specific calculation formula (31) is as follows:
D u ＝(V u +0.5V dc )/V dc
D v ＝(V v +0.5V dc )/V dc
D w ＝(V w +0.5V dc )/V dc (31)
wherein Udc is a dc bus voltage.
The three-phase duty ratio signal includes three-phase pulse width signals, such as the duty ratio signals Du1, du2, and Du3 corresponding to the duty ratio Du of one phase at different times in fig. 7, and finally generates a corresponding three-phase PWM control signal to the three-way switching tube of the upper bridge arm of the inverter 30 by using a triangular carrier signal generated by a timer in the operation control unit, and the three-phase control signal of the lower bridge arm and a corresponding complementary three-phase PWM control signal, so that the three-phase duty ratio signal actually includes six-way PWM control signals, and finally controls the six-way switching tube of the inverter 30 according to the six-way PWM control signals corresponding to the three-phase duty ratios Du, dv, and Dw, so as to implement the driving operation of the motor 60.
The invention also provides an air conditioner which comprises the motor driving device. The air conditioner is preferably a variable frequency air conditioner which comprises an indoor unit part and an outdoor unit part, wherein an outdoor unit controller and/or an indoor unit controller can comprise the motor driving device in the embodiment of the invention so as to control an indoor fan or an outdoor compressor to operate, and the reliability of the whole variable frequency air conditioner can be effectively improved.
Embodiments of the present invention also provide a machine-readable storage medium having stored thereon instructions, which when executed by a processor, enable the processor to execute the control method for the three-phase PFC circuit of the VIENNA rectifier described in any of the above embodiments.
Those skilled in the art can understand that all or part of the steps in the method for implementing the above embodiments may be implemented by a program instructing related hardware, where the program is stored in a storage medium and includes several instructions to enable a (which may be a single chip, a chip, etc.) or a processor (processor) to execute all or part of the steps in the method for implementing each embodiment of the present invention. And the aforementioned storage medium includes: a U-disk, a removable hard disk, a Read-Only Memory (ROM), a Random Access Memory (RAM), a magnetic disk or an optical disk, and other various media capable of storing program codes.
In addition, various different embodiments of the present invention may be arbitrarily combined with each other, and the embodiments of the present invention should be considered as disclosed in the disclosure of the embodiments of the present invention as long as the embodiments do not depart from the spirit of the embodiments of the present invention.
Claims (8)
1. A method of controlling a three-phase PFC circuit for a VIENNA rectifier, the method comprising:
obtaining phase current of each phase of branch in three-phase branches and direct-current bus voltage output by the three-phase PFC circuit;
determining a three-phase duty ratio signal of the conduction of a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage;
determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal;
determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal;
respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal;
the determining the forward switch driving signal of each phase of forward switch tube and the reverse switch driving signal of each phase of reverse switch tube according to the phase current and the bidirectional switch driving signal comprises:
under the condition that the phase current of the positive phase of one phase branch is larger than the preset positive current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the positive switch tube of the phase and the reverse switch tube of the phase is closed;
under the condition that the phase current of the negative phase of the branch circuit of one phase is larger than negative preset current, determining the bidirectional switch driving signal of the phase as the duty ratio of the reverse switch tube of the phase and the closing of the positive switch tube of the phase;
and under the condition that the positive phase current of one phase is less than or equal to the positive preset current, and the negative phase current of the one phase is less than or equal to the negative preset current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the positive switch tube of the phase and the duty ratio of the negative switch tube of the phase, wherein the positive preset current is 4% -11% of the peak value of the positive phase current, and the negative preset current is 4% -11% of the peak value of the negative phase current.
2. The control method according to claim 1, wherein the obtaining phase currents of each of the three-phase legs comprises:
obtaining phase currents of two phases of the current signals;
phase currents of a third phase are determined based on the phase currents of the two phases.
3. A control device for a three-phase PFC circuit for a VIENNA rectifier, the control device comprising:
phase current detection equipment for detecting the phase current of each phase branch in the three-phase branches;
the bus voltage detection device is used for detecting the direct-current bus voltage output by the three-phase PFC circuit;
a PFC processor configured to:
acquiring the phase current from the phase current detection apparatus;
acquiring the direct-current bus voltage from the bus voltage detection equipment;
determining a three-phase duty ratio signal of the conduction of a three-phase bidirectional switch tube according to the phase current and the direct-current bus voltage;
determining a bidirectional switch driving signal of a three-phase bidirectional switch tube according to the three-phase duty ratio signal;
determining a forward switch driving signal of a forward switch tube and a reverse switch driving signal of a reverse switch tube of each phase according to the phase current and the bidirectional switch driving signal;
respectively controlling the forward switch tube and the reverse switch tube to work according to the forward switch driving signal and the reverse switch driving signal;
the PFC processor is further configured to:
under the condition that the phase current of the positive phase of one phase branch is larger than the preset positive current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the positive switch tube of the phase and the reverse switch tube of the phase is closed;
under the condition that the phase current of the negative phase of the branch circuit of one phase is larger than negative preset current, determining the bidirectional switch driving signal of the phase as the duty ratio of the reverse switch tube of the phase and the closing of the positive switch tube of the phase;
and under the condition that the positive phase current of one phase is less than or equal to the positive preset current, and the negative phase current of the one phase is less than or equal to the negative preset current, determining that the bidirectional switch driving signal of the phase is the duty ratio of the positive switch tube of the phase and the duty ratio of the negative switch tube of the phase, wherein the positive preset current is 4% -11% of the peak value of the positive phase current, and the negative preset current is 4% -11% of the peak value of the negative phase current.
4. The control device of claim 3, wherein the PFC processor is further configured to:
obtaining phase currents of two phases of the current signals;
phase currents of a third phase are determined based on the phase currents of the two phases.
5. A three-phase PFC circuit for a VIENNA rectifier, characterized in that the three-phase PFC circuit comprises a control device for the three-phase PFC circuit of the VIENNA rectifier according to any one of claims 3 or 4.
6. A motor drive apparatus characterized by comprising: the three-phase PFC circuit of claim 5;
the direct current output end of the three-phase PFC circuit is connected with the power input end of the intelligent power module to provide working high-voltage direct current for the intelligent power module, and the output end of the intelligent power module outputs a three-phase alternating current signal to drive a motor to operate;
the direct current bus current sampling equipment is used for sampling the direct current bus current of the three-phase PFC circuit for supplying power to the intelligent power module; and
a motor processor configured to:
acquiring the direct current bus current and the direct current bus voltage;
and determining six switching signals for controlling the intelligent power module according to the direct-current bus voltage and the direct-current bus current so as to control the intelligent power module to drive the motor to run.
7. The motor drive apparatus of claim 6 wherein the motor processor is further configured to:
estimating the rotor position of the motor to obtain a rotor angle estimation value and a motor speed estimation value of the motor;
calculating a Q-axis given current value according to the motor target rotating speed value and the motor speed estimated value;
calculating a D-axis given current value according to the maximum output voltage of the inverter and the output voltage amplitude of the inverter;
and calculating according to the Q-axis given current value, the D-axis given current value, the motor speed estimation value, the direct current bus voltage value and the phase current value to generate a pulse width signal, and generating a PWM control signal to the intelligent power module according to a triangular carrier signal and the pulse width signal to drive the motor to operate.
8. An air conditioner characterized by comprising the motor driving device according to claim 6 or 7.
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| CN107994798A (en) * | 2018-01-11 | 2018-05-04 | 福州大学 | A kind of two-way double buck inverters and its method of work containing on-line fault diagnosis |
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| CN111130333A (en) | 2020-05-08 |
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