CN109600061B - Novel fixed-frequency model prediction current control method based on dynamic weight - Google Patents

Novel fixed-frequency model prediction current control method based on dynamic weight Download PDF

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CN109600061B
CN109600061B CN201910022417.0A CN201910022417A CN109600061B CN 109600061 B CN109600061 B CN 109600061B CN 201910022417 A CN201910022417 A CN 201910022417A CN 109600061 B CN109600061 B CN 109600061B
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王志强
左志文
谷鑫
张国政
李新旻
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Tianjin Polytechnic University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade

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Abstract

The invention belongs to the field of power converter control, and relates to a dynamic variable weight novel fixed frequency model prediction current control method. The traditional fixed frequency model prediction current control accurately calculates a voltage vector required at the next moment according to a system discretization mathematical model, so that the system has the characteristic of quick response in a dynamic process, the system does not have good control performance in a steady state process due to the influence of direct current bus voltage double frequency fluctuation in a triple converter system, and a Proportional Resonance (PR) control algorithm can realize the non-static tracking of an alternating current signal in the steady state process, but has the defect of control regulation lag in the dynamic process. Therefore, the method combines the advantages of two control algorithms, takes the PR control algorithm as a compensation link of model prediction current control, constructs dynamic variable weight novel fixed frequency model prediction current control, takes a d-axis current error value as a judgment basis, and designs a time-varying weight function m according to a Gaussian distribution function in a bell curve. According to the change rule of the m function, smooth conversion of two algorithms can be realized, so that the influence of double-frequency fluctuation in the direct-current side bus voltage on the performance of the control system is weakened, and the control system is ensured to have good performance in the dynamic and steady processes.

Description

Novel fixed-frequency model prediction current control method based on dynamic weight
Technical Field
The invention belongs to the field of power converter control, and relates to a novel fixed-frequency model prediction current control method based on dynamic weight, which can be applied to the fields of motor speed regulation, renewable energy power generation and the like.
Background
With the improvement of the power grade of a permanent magnet synchronous motor system, a topological structure formed by traditional power devices cannot meet the requirements of high-voltage and high-power application occasions. In order to meet the actual engineering requirements, scientific researchers at home and abroad adopt a multiple current transformation technology to combine the traditional two-level six-switch topological structure to form a multiple current transformer topological structure. Due to the characteristics of the topological structure, when the system runs, the bus voltage of the multiple converter structure has double-frequency fluctuation. Therefore, when a system is controlled, if a traditional fixed frequency model is adopted to predict current control, when a system mathematical model is accurate, the system has the capability of quick response, but the influence of bus voltage fluctuation on control performance in a steady state cannot be weakened, and Proportional Resonance (PR) control can realize ac signal non-static tracking, so that a PR control algorithm can weaken the influence of bus voltage fluctuation on control performance, but the PR control algorithm has the defect of regulation lag in a dynamic process.
Disclosure of Invention
The invention aims to overcome the defects in the prior art, provides a novel fixed frequency model prediction current control method based on dynamic weight, can realize that a control system has good control performance in both dynamic and steady processes, and has the advantages of simple realization and high reliability. The invention discloses a novel fixed frequency model prediction current control method based on dynamic variable weight, which comprises the following steps:
step one, sampling signals required by a control system at a moment k, and specifically comprising the following steps: three groups of DC bus voltage Udc1，Udc2，Udc3Three-phase voltage e input from power grid sideA、eB、eCAnd three-phase current iA、iB、iC；
Step two, making the q-axis given current zero, and obtaining a d-axis current given value i through a proportional integral controller according to the difference value between the voltage outer ring given value and the feedback valued ref；id refAnd id refAre respectively:
Figure GSB0000179787990000011
in the formula id ref、iq refGiven current values, K, for d-q axes, respectivelypIs the proportional coefficient, K, of a voltage outer loop PI controlleriFor voltage outer loop PI controller integral coefficient, Ueq refIs given value of voltage, UeqIs twice the average value of the feedback voltage.
Step three, sampling the voltage e obtained by kA、eB、eCObtaining position information theta through phase-locked loop operation, and converting an alternating current input voltage signal (obtained by sampling) and a current signal (obtained by sampling and calculating) into α - β shafting lower voltage and current components through a coordinate transformation theory, wherein the expression of the step is as follows:
Figure GSB0000179787990000012
in the formula eα(k)、eβ(k) The input voltage at the AC side at the time k is the voltage component under the shaft system of α - β, eA(k)、eB(k)、eC(k) Inputting three-phase voltage for the AC side at the time k;
Figure GSB0000179787990000021
in the formula iα(k)、iβ(k) The current component i of the input current at the AC side at the time k under the shaft system of α - β isA(k)、iB(k)、iC(k) Inputting three-phase current for the AC side at the time k;
Figure GSB0000179787990000022
in the formula iα ref、iβ refRespectively, i is the current component of a given current in the α - β shaftingd ref、iq refRespectively setting current values for d-q axes, and enabling theta to be position information;
calculating the model prediction control delay compensation current value by using the current and voltage sampling values at the moment k according to the system discretization mathematical model; the expression is as follows:
Figure GSB0000179787990000023
in the formula iα(k+1)、iβ(k +1) is the current component of the alternating current side input current at the moment of k +1 under the shaft system of α - β, namely the time delay compensation current value TsFor the system control period, LgxIs equivalent AC side inductance uα(k-1)、uβ(k-1) are voltage components of the output voltage vector at the moment of k-1 under the shaft system of α - β respectively;
step five, obtaining according to step fourThe delay compensation value is up to, the delay compensation value iα，β(k +1) instead of the current value i at time kα，β(k) Substituting the system discretization mathematical model into a model predictive control output voltage vector under a α - β shafting, wherein the expression is as follows:
Figure GSB0000179787990000024
in the formula uα p(k)、uβ p(k) Outputting voltage components of model prediction control under α - β shafting;
solving the output voltage vector of the PR controller under an alpha-beta shaft system according to the discretization mathematical model of the system; the expression is as follows:
Figure GSB0000179787990000025
in the formula uα pr(k)、uβ pr(k) Voltage component, G, at α - β axis for PR controller outputPR(s) is the transfer function of the PR controller, as shown in the following equation:
Figure GSB0000179787990000031
in the formula KpAnd KrProportional coefficient and resonance coefficient, omega, of the PR controllerc、ω0Cut-off frequency and resonance frequency respectively;
and step seven, the PR control is used as a compensation link of the model prediction current control, and the novel dynamic variable-weight fixed-frequency model prediction current control is constructed. Taking the d-axis current error value as a judgment basis, designing a time-varying weight function m according to a Gaussian distribution function in a bell curve, and carrying out amplitude limiting on the maximum value of m as 1; the expression is as follows:
Figure GSB0000179787990000032
in the formula uα end(k)、uβ end(k) For the final reference voltage vector, m is a time-varying weighting function, and the expression is:
Figure GSB0000179787990000033
in the formula,. DELTA.idD-axis current error value, sigma is standard deviation, and is constantly larger than 0; assuming that the d-axis current difference value fluctuates between +/-0.2A when the control system is in a steady state, the following rules are required when the m function is subjected to parameter setting:
1) when the system is in a steady state process and has no sampling error, the d-axis current difference value fluctuates between +/-0.2A, PR control is expected, therefore 0.2A is taken as a turning point, and when the absolute value of the error is less than or equal to 0.2A, m is constantly 1.
2) When the system is in a steady state process and sampling errors occur, the absolute value of the current difference is larger than 0.2A, at the moment, m is expected to change slowly from 0 to 1, and the two control algorithms act simultaneously, so that the purposes of quick response and weakening the influence of bus voltage fluctuation on the control performance are achieved.
3) When the system is in a dynamic process and there is no sampling error, assuming that the dynamic process current difference becomes 1A, then it is expected that the model predictive current control dominates, and m should decrease rapidly to 0.
Step eight, according to the position information theta obtained at the moment k and the three groups of bus voltages, obtaining a finally obtained reference voltage vector uα end(k)、uβ end(k) Dividing by 2, respectively carrying out duty ratio calculation by three SVPWM calculating units, wherein the carrier phases of the three SVPWM calculating units are different when the SVPWM calculating units calculate, the carrier phases of the second SVPWM calculating unit and the third SVPWM calculating unit lag behind the first 1/3 control periods of the calculating unit at the same time, so as to carry out PWM duty ratio calculation, recombine obtained PWM signals, carry out duty ratio updating at the moment of k +1, and simultaneously repeat the steps from one step to seven at the moment of k +1, thereby carrying out circulation.
Compared with the prior art, the invention has the following effects:
(1) the method adopts the model to predict the current control in the dynamic process, and has the advantage of quick dynamic response compared with PR control.
(2) The method adopts PR control to compensate the model prediction current control, can weaken the influence of the double frequency fluctuation of the bus voltage on the system control performance, and has good steady-state performance compared with the traditional model prediction current control.
Drawings
FIG. 1: a line voltage cascade type triple converter topological diagram;
FIG. 2: the line voltage cascade type triple converter equivalent circuit;
FIG. 3: system control structure diagram
Detailed Description
When the triplex converter is used as a grid-side rectifier circuit, the topology is as shown in fig. 1. Neglecting the influence of the input line resistance at the ac side on the system, the loop formed by the unit 1 and the unit 2 in fig. 1 is generalized, and the following relationship can be obtained:
Figure GSB0000179787990000041
in the formula of UAB、UBC、UCAFor tripling the LVC-VSC AC side line voltage, EAB、EBC、ECAFor mains input line voltage, Iai、Ibi、IciPhase currents, U, of the phases of the constituent unit iaibi、Ubici、UciaiThe fundamental wave components of line voltage between the ac sides ab, bc and ca in the constituent unit i are respectively, wherein i is 1, 2 and 3. U shapeb1a2、Uc2b3、Ua1c3Respectively, the voltage of the current-limiting inductor connected between the units.
From the above analysis, it can be seen that the triple-commutated converter can be equivalent to a conventional two-level power converter, as shown in fig. 2. In the figure, LgxThe equivalent alternating-current side inductance is expressed as follows:
Figure GSB0000179787990000042
in the formula, kLIs LgAnd LxThe ratio between, namely:
Figure GSB0000179787990000043
because the voltage of the alternating-current side wire of the equivalent switch circuit is equal to the sum of the voltages of two adjacent cascaded VSC unit wires, the equivalent direct-current bus voltage UeqEqual to twice Uav，UavThe average value of 3 groups of direct current bus voltages is shown, according to the mathematical model expression of the triple converter equivalent switch circuit under the shaft line α - β can be obtained according to the diagram of FIG. 2, and the mathematical model expression is as follows:
Figure GSB0000179787990000044
in the formula eα、eβ、iα、iβGrid voltage and grid current under the axis α - β respectively, and the obtaining mode thereof is as described in the step three, and can be respectively expressed as:
Figure GSB0000179787990000045
in the formula eα(k)、eβ(k) The input voltage at the AC side at the time k is the voltage component under the shaft system of α - β, eA(k)、eB(k)、eC(k) And inputting three-phase voltage to the AC side at the moment k.
Figure GSB0000179787990000051
In the formula iα(k)、iβ(k) The current component i of the input current at the AC side at the time k under the shaft system of α - β isA(k)、iB(k)、iC(k) And inputting three-phase current to the AC side at the moment k.
In the second step, the d-axis given current value of the control system is calculated by a voltage outer ring PI controller, and can be represented as:
Figure GSB0000179787990000052
in the formula id ref、iq refRespectively giving current values, K, to d-q axes on the power grid sidepIs the proportional coefficient, K, of a voltage outer loop PI controlleriFor voltage outer loop PI controller integral coefficient, Ueq refIs given value of voltage, UeqIs twice the average value of the feedback voltage. Coordinate transformation is performed on equation (7) as described in step three, and the following can be obtained:
Figure GSB0000179787990000053
in the formula iα ref、iβ refRespectively, i is the current component of a given current in the α - β shaftingd ref、iq refThe current values are given for the d-q axes, respectively, and θ is position information.
In the fourth step, model prediction control delay compensation is calculated, i.e. i is solved according to the formula (4)α，β(k +1), the formula (4) is discretized and solved. It is possible to obtain:
Figure GSB0000179787990000054
in the formula iα(k+1)、iβ(k +1) is the current component of the alternating current side input current at the moment of k +1 under the shaft system of α - β, namely the time delay compensation current value TsFor the system control period, uα(k-1)、uβAnd (k-1) is the voltage component of the given voltage vector at the moment k-1 in the axis α - β respectively.
In the fifth step, according to the delay compensation current value obtained in the fourth step, the output voltage vector u is subjected to model prediction controlα p、uβ pThe calculation can be expressed as:
Figure GSB0000179787990000055
in the sixth step, the PR control compensation link is controlled to output a voltage vector uα pr(k)，uβ pr(k) The calculation can be expressed as:
Figure GSB0000179787990000056
in the formula GPR(s) is the transfer function for PR control, as shown below:
Figure GSB0000179787990000061
in the formula KpAnd KrProportional coefficient and resonance coefficient, omega, of the PR controllerc、ω0Respectively, a cut-off frequency and a resonance frequency.
In the seventh step, a dynamic variable weight novel fixed frequency model is constructed to predict current control, which can be expressed as:
Figure GSB0000179787990000062
in the formula uα end(k)、uβ end(k) For the final reference voltage vector, m is a time-varying weighting function, and the expression is:
Figure GSB0000179787990000063
in the formula,. DELTA.idAnd the d-axis current error value is sigma which is a standard deviation and is constantly larger than 0. Meanwhile, according to the requirements of actual operating conditions, a proper sigma can be selected to construct the function m, and because different sigma can enable m to be larger than 1, amplitude limitation needs to be carried out on the maximum value of m to be 1.
Step eight, according to the structural characteristics of the triple converter and the existing carrier phase-shift modulation strategy, three SVPWM (space vector pulse width modulation) calculation units are adopted when the system is controlledAnd each SVPWM computing unit adopts 2Uav/3 as the module length of the coordinate axis in the space vector coordinate system, where UavThe average value of the voltages of three groups of capacitors on the direct current side is shown. Thus, the reference voltage vector u obtained at time kα end(k)、uβ end(k) 1/2 is needed to be multiplied as a final reference voltage vector of each SVPWM calculating unit, when the SVPWM calculating units calculate, the carrier phases of the three SVPWM calculating units are different, wherein the carrier phases of the second SVPWM calculating unit and the third SVPWM calculating unit lag behind the calculating unit for 1/3 control periods at the same time, so that PWM duty ratio calculation is carried out, the obtained PWM signals are recombined, duty ratio updating is carried out at the moment of k +1, and the steps from one step to the seventh step are repeated, so that the cycle is carried out. The control block diagram of the system at this time is shown in fig. 3.

Claims (7)

1. A dynamic variable weight novel fixed frequency model prediction current control method is characterized by comprising the following steps:
step one, at the time k, a control system samples a required signal, and the method specifically comprises the following steps: three groups of direct current bus voltages and three-phase voltage e input by power grid sideA，B，CAnd three-phase current iA，B，C；
Step two, enabling the q-axis given current to be zero, obtaining a d-axis current given value i through a proportional-integral controller according to a difference value between a voltage outer ring given value and a feedback value, wherein the feedback value is two times of an average value of three groups of direct-current bus voltage sampling valuesd ref；
Step three, the voltage e obtained by samplingA，B，CObtaining position information theta through phase-locked loop operation, and converting the alternating current input voltage signal and the current signal obtained by sampling in the step one into α - β shafting lower voltage and current components through a coordinate transformation theory;
step four, current and voltage sampling values at the moment k are utilized, wherein the current sampling value at the moment k is the current i of the three-phase power gridA，B，CThe voltage sampling value at the moment k is the three-phase power grid voltage eA，B，C(ii) a Calculating the delay compensation value of model prediction current control according to the discretization mathematical model of the system,i.e. the current value i at the moment k +1α，β(k+1)；
Step five, delaying the compensation value iα，β(k +1) instead of the current value i at time kα，β(k) Substituting the voltage vector into a system discretization mathematical model, solving α - β shafting when adopting model prediction current control, and outputting a given voltage vector u by the model prediction current controlα p、uβ p；
Step six, according to the discretization mathematical model of the system, the output given voltage vector u is obtained by the PR controller when the PR control is adoptedα pr、uβ pr；
Step seven, constructing a time-varying function m according to a Gaussian distribution function in a bell-shaped curve so as to realize smooth conversion between model prediction current control and PR control, wherein the value of m is between 0 and 1;
step eight, according to the position information theta obtained at the moment k and the three groups of bus voltages, obtaining a finally obtained reference voltage vector uα end(k)、uβ end(k) Dividing by 2, performing PWM wave duty ratio calculation by three SVPWM calculation units respectively, updating the duty ratio at the moment of k +1, and repeating the steps from one step to seven at the moment of k +1, thereby performing circulation.
2. The method for controlling the predictive current of the dynamic variable-weight novel fixed-frequency model according to claim 1, wherein the calculation formula in the second step is as follows:
Figure FSB0000188007460000011
in the formula id ref、iq refRespectively giving current values, K, to d-q axes on the power grid sidepIs the proportional coefficient, K, of a voltage outer loop PI controlleriFor voltage outer loop PI controller integral coefficient, Ueq refIs given value of voltage, UeqIs twice the average value of the feedback voltage.
3. The method for controlling the predictive current of the dynamic variable weight novel fixed frequency model according to claim 1, wherein the calculation formulas of the coordinate transformation of each signal in the three steps are respectively as follows:
Figure FSB0000188007460000012
in the formula eα、eβIs the voltage component of the grid voltage in the α - β axis system, eA、eB、eCThe three-phase voltages of the power grid are respectively;
Figure FSB0000188007460000021
in the formula iα、iβInputting current components i of three-phase current in α - β shafting for the side of the power gridA、iB、iCInputting three-phase current for the power grid respectively;
Figure FSB0000188007460000022
in the formula iα ref、iβ refRespectively, the current components of the given current in the α - β shafting.
4. The method for controlling the predicted current of the novel dynamic variable weight constant frequency model according to claim 1, wherein the delay compensation value in the fourth step is calculated according to the following formula:
Figure FSB0000188007460000023
in the formula iα，β(k +1) is the current value of the current at the moment of k +1 in the axis system of α - β, namely the delay compensation value uα，β(k-1) is the component of a given voltage vector at the time of k-1 in the axis line of α - β, LgxIs equivalent AC side inductance, TsA system control cycle.
5. The method for predicting the current control by the dynamic variable weight novel fixed frequency model as claimed in claim 1, wherein the model prediction current control output voltage vector in the fifth step can be calculated according to the following formula:
Figure FSB0000188007460000024
in the formula uα p(k)，uβ p(k) Predicting the current control output for the model, TsA system control cycle.
6. The method as claimed in claim 1, wherein the output of the PR controller in step six is calculated according to the following formula:
Figure FSB0000188007460000025
in the formula uα pr(k)，uβ pr(k) Is the output of the PR controller, GPR(s) is a PR controller.
7. The method according to claim 1, wherein the novel fixed-frequency model predictive current control in step seven is calculated according to the following formula:
Figure FSB0000188007460000026
in the formula uα end(k)、uβ end(k) For the final reference voltage vector, m is a time-varying duty ratio function, and the expression is as follows:
Figure FSB0000188007460000031
in the formula,. DELTA.idIs d-axis current error value, sigma is standard deviation, constantGreater than 0; meanwhile, according to the requirements of actual operating conditions, a proper sigma can be selected to construct the function m, and because different sigma can enable m to be larger than 1, amplitude limitation needs to be carried out on the maximum value of m to be 1.
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