CN109256968B - Sliding mode power offset direct power control method of three-phase voltage type PWM converter - Google Patents

Sliding mode power offset direct power control method of three-phase voltage type PWM converter Download PDF

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CN109256968B
CN109256968B CN201811095634.4A CN201811095634A CN109256968B CN 109256968 B CN109256968 B CN 109256968B CN 201811095634 A CN201811095634 A CN 201811095634A CN 109256968 B CN109256968 B CN 109256968B
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reactive power
active power
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CN109256968A (en
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王延敏
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Jiaxing Juteng Information Technology Co ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output

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Abstract

The invention discloses a sliding mode power offset direct power control method of a three-phase voltage type PWM converter, which is characterized in that when an actual value of active power is reduced, an actual value of reactive power is increased and an actual value of reactive power of the active power is reduced, another non-zero voltage vector is searched to obtain an improved vector table, then three optimal voltage vectors capable of simultaneously reducing errors between the actual values of the active power and the reactive power and reference values of the active power and the reactive power are selected from the improved vector table according to a sector where a grid voltage position angle is located, the actual values of the active power and the reactive power and the reference values of the active power and the reactive power, and the action time of each vector is calculated according to the reference values of the active power and the reactive power. The method has simple control algorithm, is easy to realize, can improve the approaching speed of the arrival stage, inhibit the output buffeting of the system, simultaneously inhibit the steady-state ripples of active power and reactive power, and has low current distortion rate so that the system has excellent dynamic and static performances.

Description

Sliding mode power offset direct power control method of three-phase voltage type PWM converter
Technical Field
The invention relates to the field of control of power electronic power conversion devices, in particular to a sliding mode power offset direct power control method of a three-phase voltage type PWM converter.
Background
With the continuous development of power electronic technology and control technology, the performance of the power converter is continuously improved, and the power converter is widely applied to the fields of rectification, alternating current speed regulation, active filtering, reactive compensation, new energy grid-connected power generation and the like. Through rapid development in recent years, more and more control methods are proposed and applied to the control of the PWM power converter. The control method of the power converter mainly includes grid Voltage Oriented Control (VOC), Virtual Flux Oriented Control (VFOC), Sliding Mode Control (SMC), Model Predictive Control (MPC), and Direct Power Control (DPC). The sliding mode control is a special nonlinear control, and the control strategy is different from other control strategies in that the structure of the system is not fixed, but can be purposefully and continuously changed according to the current state of the system in a dynamic process to force the system to move according to the state track of a preset sliding mode. The sliding mode control system has the advantages of simple algorithm, high response speed and strong robustness to external interference and parameter perturbation. The DPC does not need a current loop, and the power of the system can be directly adjusted by only selecting a proper voltage vector through looking up a switch table, so that the deep research of domestic and foreign scholars is obtained. However, the control performance of the DPC is affected by the quality of the switch table, and at present, there are a plurality of ways to establish the switch table: modeling and analyzing the converter from the power conservation angle; analyzing the relation between the current and space voltage vector and power; selecting a proper voltage vector to realize the control of active power and reactive power through an optimal vector general criterion based on DPC; and analyzing the relation between the voltage vector and the input active power and reactive power by combining an instantaneous power theory on the basis of a converter mathematical model.
By analyzing the change rate of the voltage vector on the input side of the converter to the active power and the reactive power by predicting the dead beat direct power control, two non-zero vectors are provided, wherein the active power is increased and the reactive power is increased, and the active power is increased and the reactive power is reduced. In order to suppress steady-state ripples of active power and reactive power at the same time and further improve the dynamic response of the system, patent No. CN201510097432.3 discloses a three-phase voltage type PWM converter three-vector direct power control method, which selects another non-zero voltage vector with an approximate goal (small increase in active power but large increase in reactive power and small increase in active power but large decrease in reactive power) when active power is decreased and reactive power is increased and active power is decreased and reactive power is decreased, and the ripples of active power and reactive power obtained by this method are unstable and the dynamic response of the system is poor.
Disclosure of Invention
The invention aims to provide a sliding mode power offset direct power control method of a three-phase voltage type PWM converter, which can obviously reduce the fluctuation of active power and reactive power and has low current distortion rate so that a system has excellent dynamic and static performances.
In order to achieve the purpose, the invention adopts the following technical scheme:
a sliding mode power offset direct power control method of a three-phase voltage type PWM converter sequentially comprises the following steps:
(1) three-phase network voltage u for collecting three-phase voltage type PWM converterga、ugb、ugcThree-phase input current iga、igb、igcAnd the DC bus voltage Udc；
(2) The collected three-phase power grid voltage and three-phase input current are processed by an 3/2 coordinate transformation module to obtain the power grid voltage u under a two-phase static coordinate systemgα、ugβAnd the input current sample value igα、igβAnd calculating instantaneous active power p and instantaneous reactive power q, wherein subscripts g alpha and g beta are both distinguishing functions and are not variables;
(3) the three-phase power grid voltage signal is subjected to a software phase-locked loop to obtain a power grid voltage position angle theta, and the power grid voltage position angle theta is divided into six sectors of 0-60 degrees, 60-120 degrees, 120-180 degrees, 180-240 degrees, 240-300 degrees and 300-360 degrees;
(4) reference value of DC bus voltage
Figure GDA0002439275830000021
Comparing with the actual value U of the DC bus voltage obtained in the step (1)dcMaking difference, and obtaining a current reference value through a sliding mode controller constructed by a self-adaptive function
Figure GDA0002439275830000022
Reference value of current
Figure GDA0002439275830000023
And the actual value U of the DC bus voltagedcMultiplying to obtain an active power reference value prefAnd a reference value q of reactive power is setref0, where dc, d and ref are all discriminatory functions and are not variables;
(5) obtaining the power grid voltage u under the two-phase static coordinate system through the step (2)gα、ugβThe instantaneous active power p, the instantaneous reactive power q and 8 different voltage vectors are calculated, and the active power change rate and the reactive power change rate under the action of the corresponding voltage vectors are calculated;
8 different voltage vectors represent the input voltage of the down converter of the two-phase static alpha and beta coordinate system;
(6) respectively obtaining two nonzero vectors which can simultaneously increase active power and increase reactive power, increase active power and reduce reactive power and a nonzero vector which can reduce active power, increase reactive power and reduce reactive power in 8 different voltage vectors according to the active power change rate and the reactive power change rate, and searching another nonzero voltage vector for counteracting the influence of the zero vector on the active power, namely, two nonzero vectors which increase the active power and reduce the reactive power and consider the addition of the zero vector, namely, each sector can obtain 4 groups of vector combinations which respectively comprise two nonzero vectors and one zero vector, and the voltage of the power grid is in an interval of 0-360 degrees, thereby obtaining 24 groups of vector combinations;
(7) the actual values of the active power and the reactive power obtained in the step (2) and the reference value p of the active power obtained in the step (4) are comparedrefReference value q of reactive powerrefComparing to obtain the variation trend of the actual value of the active power and the actual value of the reactive power, wherein the upper mark ref is a distinguishing function and is not a variable;
(8) establishing an improved vector table according to the change trends of the active power and the reactive power obtained in the step (7) when the active power and the reactive power are acted by different voltage vectors in different sectors;
(9) according to the sector of the grid voltage position angle and the variation trend of the actual values of the active power and the reactive power, selecting a corresponding voltage vector group from the improved vector table obtained in the step (8), and calculating the variation rate of the active power and the reactive power and the difference value between the reference value and the actual value of the active power and the reactive power according to each vector of the vector group to obtain the time t of the corresponding vector action0、t1、t2；
(10) The time t of the control vector action obtained in the step (9)0、t1、t2Converted into a switching signal that controls the power device.
Preferably, in the step (2), according to the instantaneous power theory, the obtained instantaneous active power p and the instantaneous reactive power q are respectively
p＝1.5(ugαigα+ugβigβ)
q＝1.5(ugβigα-ugαigβ) (1)
Here, ugαAnd ugβGrid voltage i in two-phase stationary coordinate systemgα、igβThe sampling values of the input current under the two-phase static coordinate system are respectively.
Preferably, the current reference value in the step (4)
Figure GDA0002439275830000031
The calculation process of (2) is as follows: firstly, selecting the state variable of the system as
Figure GDA0002439275830000032
In the formula,
Figure GDA0002439275830000033
is a reference value of DC bus voltage, UdcIs the actual value of the DC bus voltage, x1The intermediate variable is self-defined;
designing the switching function of the switch plane in the adaptive continuous variable structure as s ═ x1Reconstructing an approximation law of variable structure control by using an adaptive continuous function, wherein the approximation law is constructed as follows:
Figure GDA0002439275830000034
wherein,
Figure GDA0002439275830000035
k1、k2are all adjustment coefficients, k1>0，k2>k1>0, m is (1+ e)-s) N is the number of switching functions s, 1<m<3，1<n<3, s is the switching function, s' is the derivative of the switching function s,
Figure GDA0002439275830000036
sgn(s) is a sign function for the adaptive continuous function;
the derivation is carried out for the formula (2) because
Figure GDA00024392758300000310
Is a given value, so the derivative result is:
Figure GDA0002439275830000038
according to the current relation of the three-phase PWM rectifier on the direct current side under the d and q coordinate system
Figure GDA0002439275830000039
Wherein C is a DC side capacitor.
According to the formulae (4) and (5):
Figure GDA0002439275830000041
wherein idAnd iqRespectively, the network side current, s, in a synchronous rotating coordinate systemdAnd sqAre respectively d-axis and q-axis switching functions i under a synchronous rotating coordinate systemRSubscripts d, q, dc, and R are all discriminative, non-variable, for load current;
by substituting formula (4) for formula (6), a compound of formula (4) can be obtained
Figure GDA0002439275830000042
Assuming that the input voltage is three-phase symmetrical voltage, in steady state, there are
Figure GDA0002439275830000043
eq＝0,
Figure GDA0002439275830000044
iq＝0，
Figure GDA0002439275830000045
Figure GDA0002439275830000046
Wherein e isdAnd eqRespectively the grid voltage, U, of the three-phase PWM rectifier in a two-phase rotating coordinate systemRMSThe subscript RMS is a distinguishing function and is not a variable;
mathematical model of three-phase PWM rectifier under synchronous rotation coordinate system
Figure GDA0002439275830000047
It can be deduced that:
Figure GDA0002439275830000048
where ω is the angular frequency of the grid voltage, and R and L represent the resistance and inductance values of the PWM converter, respectively;
when formula (9) is substituted into formula (7), it can be obtained:
Figure GDA0002439275830000049
at steady state, there are
Figure GDA00024392758300000410
Namely, it is
Figure GDA00024392758300000411
Then the formula (9) can be rewritten as
Figure GDA00024392758300000412
Namely the d-axis current reference value under the synchronous rotating coordinate system
Figure GDA00024392758300000413
Preferably, the equation (1) is differentiated, and assuming that the three-phase grid voltage is balanced and sinusoidal, the obtained instantaneous active power p and the instantaneous reactive power q have the following differential values:
Figure GDA00024392758300000414
Figure GDA00024392758300000415
where ω is the angular frequency of the grid voltage, R and L represent the resistance and inductance values of the PWM converter, respectively, p and q represent the instantaneous active and reactive power, respectively, vgαAnd vgβThe input voltages of the alpha axis and the beta axis of the converter are respectively expressed, and the active power change rate and the reactive power change rate under the action of different voltage vectors can be obtained according to the formula (12).
Preferably, in the step (8), the active power change rates corresponding to the selected non-zero vector and the selected zero vector at the time k are respectively assumed to be fp1、fp2And fp0The rate of change of the reactive power is fq1、fq2And fq0The active power and the reactive power at the k moment are respectively pk、qkThen the active power p at time k +1k+1Reactive power qk+1Are respectively as
pk+1＝pk+fp1t1+fp2t2+fp0(tsc-t1-t2)
qk+1＝qk+fq1t1+fq2t2+fq0(tsc-t1-t2) (13)
Wherein, t1And t2Representing the action time, t, of two non-zero vectorsscSubscripts p1, p2, p0, q1, q2, q0 and sc are all distinguishing functions, and are not variables, and a subscript k represents time;
suppose that the active and reactive powers at time k +1 are equal to the reference value p, respectivelyref、qrefThe action time t of the non-zero vector can be obtained from the formula (13)1、t2Time of action t of sum zero vector0Respectively as follows:
Figure GDA0002439275830000051
Figure GDA0002439275830000052
t0＝tsc-t1-t2 (14)
preferably, the control of the switch state is based on the principle of minimum switching times, and the following improvement is made to the formula (14)
Figure GDA0002439275830000053
Figure GDA0002439275830000054
Figure GDA0002439275830000055
When in a certain switching period, the action time t of two non-zero vectors1+t2>tscThen the zero vector does not act in the period, and the action time of two non-zero vectors is adjusted to
Figure GDA0002439275830000056
Figure GDA0002439275830000057
The invention searches for another non-zero voltage vector for counteracting the influence of the zero vector on the active power by analyzing and predicting the reason of larger steady-state ripple of the dead-beat direct power control reactive power when the actual value of the active power is reduced, the actual value of the reactive power is increased and the actual value of the reactive power of the active power is reduced, obtains an improved vector table, then selects three optimal voltage vectors which can simultaneously reduce the errors between the actual values of the active power and the reactive power and the reference values of the active power and the reactive power from the improved vector table according to the sector where the position angle of the grid voltage is located, the actual values of the active power and the reactive power and the reference values of the active power and the reactive power, and calculates the action time of each vector according to the reference values of the active power and the reactive power. The method has simple control algorithm, is easy to realize, can improve the approaching speed of the arrival stage, inhibit the output buffeting of the system, simultaneously inhibit the steady-state ripples of active power and reactive power, and has low current distortion rate so that the system has excellent dynamic and static performances.
Drawings
FIG. 1 is a main circuit topology diagram of a three-phase voltage type PWM converter according to the present invention;
FIG. 2 is a schematic diagram of the active power change rate and the reactive power change rate under the input side voltage vector of the converter when the active power is 1000W and the reactive power is 0 Var;
FIG. 3 is a table of 8 voltage vector values in a two-phase stationary α, β coordinate system;
FIG. 4 is a table showing the influence of V1-V6 on the actual values of active power and reactive power in different sectors;
FIG. 5 is a table of predicted dead-beat direct power control switch vectors;
FIG. 6 is a table of the improved vectors employed in the present invention;
FIG. 7 is a schematic diagram of a method for generating a switching pattern with three vectors;
FIG. 8 is a diagram illustrating the relationship between the approach process S and time of the conventional exponential approach law;
FIG. 9 is a schematic output diagram of a conventional slip film control system;
FIG. 10 is a diagram illustrating the relationship between the approach process S and the time of the approach law according to the present invention;
FIG. 11 is a schematic output diagram of the synovial control system of the invention;
FIG. 12 is a control block diagram of the present invention;
FIG. 13 is a diagram of the AC side a-phase current waveform and spectrum using a predicted dead-beat direct power control vector table;
FIG. 14 is a waveform diagram and a frequency spectrum diagram of the phase current at the a side of the AC using the modified vector table;
FIG. 15 is a waveform diagram of active and reactive power experiments using a predicted dead-beat direct power control vector table;
fig. 16 is a waveform diagram of active and reactive power experiments using an improved vector table.
Detailed Description
The technical solutions of the present invention will be described clearly and completely with reference to the accompanying drawings, and it is obvious that the described embodiments are only some embodiments of the present invention, not all embodiments. All other embodiments that can be obtained by a person skilled in the art based on the embodiments of the present invention without any creative effort belong to the protection scope of the present invention.
The main circuit topology structure of the three-phase voltage type PWM converter used in the present invention is shown in FIG. 1, in which uga、ugb、ugcRespectively, three-phase voltage of AC side, iga、igb、igcRespectively, three-phase current, v, on the AC sidega、vgb、vgcAre respectively three-phase voltage, U, at the input side of the converterdcThe direct current bus voltage is represented by L and R respectively representing an incoming line inductor and an equivalent resistor thereof, C representing a direct current filter capacitor, and a direct current side load is represented by a resistor RL in an equivalent manner.
The mathematical model of the three-phase voltage type PWM converter can be converted into a two-phase static coordinate system through coordinate conversion, and is represented as
Figure GDA0002439275830000071
Wherein u isgα、ugβGrid voltages of the alpha and beta axes, igα、igβ、vgα、vgβThe input current and the input voltage of the alpha axis and the beta axis of the converter are respectively. As shown in fig. 3, the two-phase stationary alpha, beta coordinate system down-converter input voltages can be represented by 8 voltage vectors, respectively, wherein6 are non-zero vectors (V1-V6), and 2 are zero vectors (V0, V7).
According to the PWM converter, the invention discloses a sliding mode power offset direct power control method of a three-phase voltage type PWM converter, which sequentially comprises the following steps:
(1) three-phase power grid voltage u of three-phase voltage type PWM converter is acquired by respectively utilizing voltage transformer and current transformerga、ugb、ugcThree-phase input current iga、igb、igcAnd the DC bus voltage Udc；
(2) The collected three-phase power grid voltage and three-phase input current are processed by an 3/2 coordinate transformation module to obtain the power grid voltage u under a two-phase static coordinate systemgα、ugβAnd the input current sample value igα、igβAnd calculating instantaneous active power p and instantaneous reactive power q, wherein subscripts g alpha and g beta are both distinguishing functions and are not variables;
according to the instantaneous power theory, the obtained instantaneous active power p and the instantaneous reactive power q are respectively
p＝1.5(ugαigα+ugβigβ)
q＝1.5(ugβigα-ugαigβ) (1)
Here, ugαAnd ugβGrid voltage i in two-phase stationary coordinate systemgα、igβRespectively obtaining input current sampling values under a two-phase static coordinate system, wherein the four values are known;
(3) the three-phase power grid voltage signal is subjected to a software phase-locked loop to obtain a power grid voltage position angle theta, and the power grid voltage position angle theta is divided into six sectors of 0-60 degrees, 60-120 degrees, 120-180 degrees, 180-240 degrees, 240-300 degrees and 300-360 degrees;
(4) reference value of DC bus voltage
Figure GDA0002439275830000072
Comparing with the actual value U of the DC bus voltage obtained in the step (1)dcMake a difference, subject to adaptationObtaining a current reference value by a sliding mode controller with a function structure
Figure GDA0002439275830000073
Reference value of current
Figure GDA0002439275830000074
And the actual value U of the DC bus voltagedcMultiplying to obtain an active power reference value prefAnd a reference value q of reactive power is setrefIs 0; reference value of DC bus voltage
Figure GDA0002439275830000075
The method is set according to experience, the set value is a fixed value, the upper mark ref is a distinguishing function, and the set value is not a variable;
reference value of current
Figure GDA0002439275830000081
The calculation process of (2) is as follows: firstly, selecting the state variable of the system as
Figure GDA0002439275830000082
In the formula,
Figure GDA0002439275830000083
is a reference value of DC bus voltage, UdcIs the actual value of the DC bus voltage, x1The intermediate variable is self-defined;
designing the switching function of the switch plane in the adaptive continuous variable structure as s ═ x1Reconstructing an approximation law of variable structure control by using an adaptive continuous function, wherein the approximation law is constructed as follows:
Figure GDA0002439275830000084
wherein,
Figure GDA0002439275830000085
k1、k2are all adjustment coefficients, k1>0，k2>k1>0, m is (1+ e)-s) N is the number of switching functions s, 1<m<3，1<n<3, s is the switching function, s' is the derivative of the switching function s,
Figure GDA0002439275830000086
sgn(s) is a sign function for the adaptive continuous function;
the term k of the exponential function when the system is far from the switch plane, i.e. s is large2s|s|nPlays a major role, since n is a constant greater than 1, | snThe introduction of the method enables the system to have a larger approaching speed in a stage away from the switch surface, obviously accelerates the approaching movement in the reaching stage, further shortens the approaching time compared with the traditional exponential approaching law, and enables the speed of the movement point reaching the switch surface to be small along with the reduction of s. When the approach point of the system approaches the switch surface, i.e. s is close to zero, the adaptive continuous function
Figure GDA0002439275830000087
The approach speed is adaptive and reduced along with the reduction of the s value, the smooth transition of the approach switch surface in the arrival stage is realized, the approach of the low speed of the system is finally ensured when the system is close to a steady state, and the buffeting of the system output is greatly reduced compared with the traditional constant speed approach. Increasing n and decreasing m properly can increase approaching speed of the reaching stage and reduce output buffeting of the system.
The derivation is carried out for the formula (2) because
Figure GDA0002439275830000088
Is a given value, so the derivative result is:
Figure GDA0002439275830000089
according to the current relation of the three-phase PWM rectifier on the direct current side under the d and q coordinate system
Figure GDA00024392758300000810
Wherein C is a DC side capacitor.
According to the formulae (4) and (5):
Figure GDA00024392758300000811
wherein idAnd iqRespectively, the network side current, s, in a synchronous rotating coordinate systemdAnd sqAre respectively d-axis and q-axis switching functions i under a synchronous rotating coordinate systemRFor load current, subscripts d, q, R, and dc are all discriminatory and are not variables.
By substituting formula (4) for formula (6), a compound of formula (4) can be obtained
Figure GDA00024392758300000812
Assuming that the input voltage is three-phase symmetrical voltage, in steady state, there are
Figure GDA0002439275830000091
eq＝0,
Figure GDA0002439275830000092
iq＝0，
Figure GDA0002439275830000093
Figure GDA0002439275830000094
Wherein e isdAnd eqRespectively the grid voltage, U, of the three-phase PWM rectifier in a two-phase rotating coordinate systemRMSThe subscript RMS is a distinguishing function and is not a variable;
mathematical model of three-phase PWM rectifier under synchronous rotation coordinate system
Figure GDA0002439275830000095
It can be deduced that:
Figure GDA0002439275830000096
where ω is the angular frequency of the grid voltage, and R and L represent the resistance and inductance values of the PWM converter, respectively;
when formula (9) is substituted into formula (7), it can be obtained:
Figure GDA0002439275830000097
at steady state, there are
Figure GDA0002439275830000098
Namely, it is
Figure GDA0002439275830000099
Then the formula (9) can be rewritten as
Figure GDA00024392758300000910
Namely the d-axis current reference value under the synchronous rotating coordinate system
Figure GDA00024392758300000911
(5) Obtaining the power grid voltage u under the two-phase static coordinate system through the step (2)gα、ugβThe instantaneous active power p and the instantaneous reactive power q and 8 different voltage vectors in the figure 3 calculate the active power change rate and the reactive power change rate under the action of the corresponding voltage vectors;
differentiating the formula (1), and assuming that the three-phase power grid is balanced and sinusoidal, the obtained instantaneous active power p and the instantaneous reactive power q have the following differential values:
Figure GDA00024392758300000912
Figure GDA00024392758300000913
where ω is the angular frequency of the grid voltage, R and L represent the resistance and inductance values of the PWM converter, respectively, p and q represent the instantaneous active and reactive power, respectively, vgαAnd vgβThe input voltages of the alpha axis and the beta axis of the converter are respectively shown, the active power change rate and the reactive power change rate under the action of different voltage vectors can be obtained according to the formula (12), as shown in fig. 2, when the active power is 1000W and the reactive power is 0Var, the active power change rate under the voltage vector of the input side of the converter is shown in fig. 2(a), and when the active power is 1000W and the reactive power is 0Var, the reactive power change rate under the voltage vector of the input side of the converter is shown in fig. 2 (b).
(6) Respectively obtaining two nonzero vectors which can simultaneously increase active power and increase reactive power, increase active power and reduce reactive power and a nonzero vector which can reduce active power, increase reactive power and reduce reactive power in 8 different voltage vectors according to the active power change rate and the reactive power change rate, and searching another nonzero voltage vector for counteracting the influence of the zero vector on the active power, namely, two nonzero vectors which increase the active power and reduce the reactive power and consider the addition of the zero vector, namely, each sector can obtain 4 groups of vector combinations which respectively comprise two nonzero vectors and one zero vector, and the voltage of the power grid is in an interval of 0-360 degrees, thereby obtaining 24 groups of vector combinations;
as shown in fig. 5, in the predicted dead-beat direct power control vector table, a non-zero vector and a zero vector selected by the vector table are introduced simultaneously in one control period, but the method can only calculate the action time of each vector according to the active power reference value, thereby improving the steady-state ripple of the active power. Therefore, the present inventionThe method aims to calculate the action time of each vector according to an active power reference value and a reactive power reference value at the same time, and introduce two non-zero vectors and a zero vector in a control period at the same time, so that the steady-state ripples of the active power and the reactive power can be obviously improved at the same time. As can be seen from fig. 2, there are two non-zero vectors in each sector that can both increase active power and increase reactive power and that can both increase active power and decrease reactive power, whereas there is only one non-zero vector that can both decrease active power and increase reactive power and that can both decrease active power and decrease reactive power, for which one more non-zero vector has to be selected. As can be seen from FIG. 2, in sector IV, to reduce the active power and increase the reactive power, the selection V for the predicted deadbeat direct power control is made5And V0Function but V5Active power change rate fPv5Varies sinusoidally with theta, and V0Active power change rate fPv0Is a constant value, wherein fPv5<0，fPv0>0 and fPv0Larger, it can be seen that in one control period tscIn, if P is to be decreased, V5Time of action t1And V0Time of action t0It is inevitable to satisfy: t is t1>t0. At this time, V5Rate of change of reactive power fQv5Varies sinusoidally with theta, fQv0Is a constant value, wherein fQv5>0 and fQv0>0, then at tscInner, V5And V0The reactive power is continuously increased and exceeds the reference value, so that the fluctuation of the reactive power is large, namely, the coupling characteristic exists between the active power and the reactive power. To solve this problem, a non-zero vector is found in the IV sector and its influence on the active power is suppressed to cancel the zero vector V0The influence on the active power is only V as shown in FIG. 24And (6) conforming to the standard. In this case, two non-zero vectors V are selected to decrease the real value of active power and increase the real value of reactive power in the IV sector4、V5And a zero vector V0According to P, Q, the reference value p of active powerrefReference value q of reactive powerrefTo calculate the action time of the vectorThus, the fluctuation of the reactive power can be suppressed. Similarly, another non-zero vector can be found in other sectors, i.e. the optimized switching vector table, as shown in fig. 6. And comparing the actual values of the active power and the reactive power with the reference values to obtain the variation trend of the actual values, and selecting a corresponding voltage vector group from the improved vector table according to the sector where the grid voltage position angle is located. The subscripts Pv5, Pv0, Qv0, and Qv5 are all discriminatory and are not variables.
(7) The actual values of the active power and the reactive power obtained in the step (2) and the reference value p of the active power obtained in the step (4) are comparedrefReference value q of reactive powerrefComparing to obtain the variation trend of the active power actual value and the reactive power actual value;
(8) establishing an improved vector table according to the change trends of the active power and the reactive power obtained in the step (7) when the active power and the reactive power are acted by different voltage vectors in different sectors;
(9) according to the sector of the grid voltage position angle and the variation trend of the actual values of the active power and the reactive power, selecting a corresponding voltage vector group from the improved vector table obtained in the step (8), and calculating the variation rate of the active power and the reactive power and the difference value between the reference value and the actual value of the active power and the reactive power according to each vector of the vector group to obtain the time t of the corresponding vector action0、t1、t2；
Assuming that the active power change rates corresponding to the selected non-zero vector and the selected zero vector at the moment k are respectively fp1、fp2And fp0The rate of change of the reactive power is fq1、fq2And fq0The active power and the reactive power at the k moment are respectively pk、qkThen the active power p at time k +1k+1Reactive power qk+1Are respectively as
pk+1＝pk+fp1t1+fp2t2+fp0(tsc-t1-t2)
qk+1＝qk+fq1t1+fq2t2+fq0(tsc-t1-t2) (13)
Wherein, t1And t2Representing the action time, t, of two non-zero vectorsscSubscripts p1, p2, p0, q1, q2, q0 and sc are all distinguishing functions, and are not variables, and a subscript k represents time;
suppose that the active and reactive powers at time k +1 are equal to the reference value p, respectivelyref、qrefThe action time t of the non-zero vector can be obtained from the formula (13)1、t2Time of action t of sum zero vector0Respectively as follows:
Figure GDA0002439275830000111
Figure GDA0002439275830000112
t0＝tsc-t1-t2 (14)
the control of the switch state should follow the principle of minimum switching times, therefore, the following improvement is made to the equation (14)
Figure GDA0002439275830000121
Figure GDA0002439275830000122
Figure GDA0002439275830000123
When in a certain switching period, the action time t of two non-zero vectors1+t2>tscIn time, the zero vector does not act in the period, and the action time of the two non-zero vectors is adjusted as follows:
Figure GDA0002439275830000124
Figure GDA0002439275830000125
the switching pattern generation method is shown in fig. 7.
(10) The time t of the control vector action obtained in the step (9)0、t1、t2Converted into a switching signal that controls the power device.
Fig. 8 and 9 are performance diagrams of the conventional exponential approach law, and fig. 10 and 11 are performance diagrams of the approach law of the present invention. As can be seen from fig. 8, 9, 10, and 11, the approach velocity in the arrival phase using the conventional exponential approach law is slow, and the output chattering is large, whereas the approach velocity in the arrival phase using the approach law of the present invention is significantly increased, and the output chattering of the system is suppressed.
Fig. 13(a) is an ac side a-phase current waveform diagram of a predicted dead beat direct power control vector table, fig. 13(b) is an ac side a-phase frequency spectrum diagram of a predicted dead beat direct power control vector table, fig. 14(a) is an ac side a-phase current waveform diagram of an improved vector table, and fig. 14(b) is an ac side a-phase frequency spectrum diagram of an improved vector table, as is apparent from a comparison between fig. 13 and fig. 14, the current distortion can be reduced by using the improved vector table, higher power quality can be easily obtained, and the total harmonic distortion of the current is reduced from 7.64% to 0.9%.
Fig. 15 is an active power and reactive power experimental waveform of a prediction dead beat direct power control vector table, fig. 16 is an active power and reactive power experimental waveform of an improved vector table for canceling direct power control by adopting sliding mode power, and as is apparent from comparison between fig. 15 and fig. 16, active power and fluctuation of reactive power can be simultaneously and obviously reduced by adopting the improved vector table.
In conclusion, the method disclosed by the invention is simple in control algorithm and easy to implement, can improve the approaching speed of the arrival stage, inhibit the output buffeting of the system, inhibit the steady-state ripples of active power and reactive power, and is low in current distortion rate, so that the system has excellent dynamic and static performances.

Claims (5)

1. A sliding mode power offset direct power control method of a three-phase voltage type PWM converter is characterized by sequentially comprising the following steps:
(1) three-phase network voltage u for collecting three-phase voltage type PWM converterga、ugb、ugcThree-phase input current iga、igb、igcAnd the DC bus voltage Udc；
(2) The collected three-phase power grid voltage and three-phase input current are processed by an 3/2 coordinate transformation module to obtain the power grid voltage u under a two-phase static coordinate systemgα、ugβAnd the input current sample value igα、igβAnd calculating instantaneous active power p and instantaneous reactive power q, wherein subscripts g alpha and g beta are both distinguishing functions and are not variables;
(3) the three-phase power grid voltage signal is subjected to a software phase-locked loop to obtain a power grid voltage position angle theta, and the power grid voltage position angle theta is divided into six sectors of 0-60 degrees, 60-120 degrees, 120-180 degrees, 180-240 degrees, 240-300 degrees and 300-360 degrees and respectively corresponds to sectors I-VI;
(4) reference value of DC bus voltage
Figure FDA0002439275820000011
Comparing with the actual value U of the DC bus voltage obtained in the step (1)dcMaking difference, and obtaining a current reference value through a sliding mode controller constructed by a self-adaptive function
Figure FDA0002439275820000012
Reference value of current
Figure FDA0002439275820000013
And the actual value U of the DC bus voltagedcMultiplying to obtain an active power reference value prefAnd a reference value q of reactive power is setref0, where dc, d and ref are all discriminatory functions and are not variables, the current reference value in step (4)
Figure FDA0002439275820000014
The calculation process of (2) is as follows: firstly, selecting the state variable of the system as
Figure FDA0002439275820000015
In the formula,
Figure FDA0002439275820000016
is a reference value of DC bus voltage, UdcIs the actual value of the DC bus voltage, x1The intermediate variable is self-defined;
designing the switching function of the switch plane in the adaptive continuous variable structure as s ═ x1Reconstructing an approximation law of variable structure control by using an adaptive continuous function, wherein the approximation law is constructed as follows:
Figure FDA0002439275820000017
wherein,
Figure FDA0002439275820000018
k1、k2are all adjustment coefficients, k1＞0，k2＞k1Is greater than 0, m is (1+ e)-s) N is the number of switching functions s, m < 1 > m < 3, n < 1 > n < 3, s is the switching function, s' is the derivative of the switching function s,
Figure FDA0002439275820000019
sgn(s) is a sign function for the adaptive continuous function;
the derivation is carried out for the formula (2) because
Figure FDA00024392758200000110
Is a given value, so the derivative result is:
Figure FDA00024392758200000111
according to the current relation of the three-phase PWM rectifier on the direct current side under the d and q coordinate system
Figure FDA00024392758200000112
In the formula, C is a direct current side capacitor;
according to the formulae (4) and (5):
Figure FDA0002439275820000021
wherein idAnd iqRespectively, the network side current, s, in a synchronous rotating coordinate systemdAnd sqAre respectively d-axis and q-axis switching functions i under a synchronous rotating coordinate systemRSubscripts d, q, dc, and R are all discriminative, non-variable, for load current;
by substituting formula (4) for formula (6), a compound of formula (4) can be obtained
Figure FDA0002439275820000022
Assuming that the input voltage is three-phase symmetrical voltage, in steady state, there are
Figure FDA0002439275820000023
eq＝0，
Figure FDA0002439275820000024
iq＝0，
Figure FDA0002439275820000025
Figure FDA0002439275820000026
Wherein e isdAnd eqRespectively the grid voltage, U, of the three-phase PWM rectifier in a two-phase rotating coordinate systemRMSThe subscript RMS is a distinguishing function and is not a variable;
mathematical model of three-phase PWM rectifier under synchronous rotation coordinate system
Figure FDA0002439275820000027
It can be deduced that:
Figure FDA0002439275820000028
where ω is the angular frequency of the grid voltage, and R and L represent the resistance and inductance values of the PWM converter, respectively;
when formula (9) is substituted into formula (7), it can be obtained:
Figure FDA0002439275820000029
at steady state, there are
Figure FDA00024392758200000210
Namely, it is
Figure FDA00024392758200000211
Then the formula (9) can be rewritten as
Figure FDA00024392758200000212
Namely the d-axis current reference value under the synchronous rotating coordinate system
Figure FDA00024392758200000213
(5) The two phases obtained by step (2) are stationaryGrid voltage u under coordinate systemgα、ugβThe instantaneous active power p, the instantaneous reactive power q and 8 different voltage vectors are calculated, and the active power change rate and the reactive power change rate under the action of the corresponding voltage vectors are calculated;
8 different voltage vectors represent the input voltage of the down converter of the two-phase static alpha and beta coordinate system;
(6) respectively obtaining two nonzero vectors which can simultaneously increase active power and increase reactive power, increase active power and reduce reactive power and a nonzero vector which can reduce active power, increase reactive power and reduce reactive power in 8 different voltage vectors according to the active power change rate and the reactive power change rate, and searching another nonzero voltage vector for counteracting the influence of the zero vector on the active power, namely, two nonzero vectors which increase the active power and reduce the reactive power and consider the addition of the zero vector, namely, each sector can obtain 4 groups of vector combinations which respectively comprise two nonzero vectors and one zero vector, and the voltage of the power grid is in an interval of 0-360 degrees, thereby obtaining 24 groups of vector combinations;
(7) the actual values of the active power and the reactive power obtained in the step (2) and the reference value p of the active power obtained in the step (4) are comparedrefReference value q of reactive powerrefComparing to obtain the variation trend of the actual value of the active power and the actual value of the reactive power, wherein the upper mark ref is a distinguishing function and is not a variable;
(8) establishing an improved vector table according to the change trends of the active power and the reactive power obtained in the step (7) when the active power and the reactive power are acted by different voltage vectors in different sectors;
(9) according to the sector of the grid voltage position angle and the variation trend of the actual values of the active power and the reactive power, selecting a corresponding voltage vector group from the improved vector table obtained in the step (8), and calculating the variation rate of the active power and the reactive power and the difference value between the reference value and the actual value of the active power and the reactive power according to each vector of the vector group to obtain the time t of the corresponding vector action0、t1、t2；
(10) Step (9)) The time t of the resulting control vector action0、t1、t2Converted into a switching signal that controls the power device.
2. The sliding-mode power-canceling direct power control method of a three-phase voltage-type PWM converter according to claim 1, characterized by: in the step (2), according to the instantaneous power theory, the obtained instantaneous active power p and the instantaneous reactive power q are respectively
p＝1.5(ugαigα+ugβigβ)
q＝1.5(ugβigα-ugαigβ) (1)
Here, ugαAnd ugβGrid voltage i in two-phase stationary coordinate systemgα、igβThe sampling values of the input current under the two-phase static coordinate system are respectively.
3. The sliding-mode power-canceling direct power control method of a three-phase voltage-type PWM converter according to claim 2, characterized by: differentiating the formula (1), and assuming that the three-phase power grid is balanced and sinusoidal, the obtained instantaneous active power p and the instantaneous reactive power q have the following differential values:
Figure FDA0002439275820000031
Figure FDA0002439275820000032
where ω is the angular frequency of the grid voltage, R and L represent the resistance and inductance values of the PWM converter, respectively, p and q represent the instantaneous active and reactive power, respectively, vgαAnd vgβThe input voltages of the alpha axis and the beta axis of the converter are respectively expressed, and the active power change rate and the reactive power change rate under the action of different voltage vectors can be obtained according to the formula (12).
4. The sliding-mode power-canceling direct power control method of a three-phase voltage-type PWM converter according to claim 1, characterized by: in the step (8), it is assumed that the active power change rates of the selected non-zero vector and the selected zero vector at the time k are respectively fp1、fp2And fp0The rate of change of the reactive power is fq1、fq2And fq0The active power and the reactive power at the k moment are respectively pk、qkThen the active power p at time k +1k+1Reactive power qk+1Are respectively as
pk+1＝pk+fp1t1+fp2t2+fp0(tsc-t1-t2)
qk+1＝qk+fq1t1+fq2t2+fq0(tsc-t1-t2) (13)
Wherein, t1And t2Representing the action time, t, of two non-zero vectorsscSubscripts p1, p2, p0, q1, q2, q0 and sc are all distinguishing functions, and are not variables, and a subscript k represents time;
suppose that the active and reactive powers at time k +1 are equal to the reference value p, respectivelyref、qrefThe action time t of the non-zero vector can be obtained from the formula (13)1、t2Time of action t of sum zero vector0Respectively as follows:
Figure FDA0002439275820000041
Figure FDA0002439275820000042
t0＝tsc-t1-t2 (14)。
5. the sliding-mode power-canceling direct power control method of a three-phase voltage-type PWM converter according to claim 4, characterized by: the control of the switch state should follow the principle of minimum switching times, and the following improvement is made to the formula (14)
Figure FDA0002439275820000043
Figure FDA0002439275820000044
Figure FDA0002439275820000045
When in a certain switching period, the action time t of two non-zero vectors1+t2＞tscThen the zero vector does not act in the period, and the action time of two non-zero vectors is adjusted to
Figure FDA0002439275820000051
Figure FDA0002439275820000052
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Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008242

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: WENLING JUFENG MACHINERY Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008237

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Hengtai Forging Tools Manufacturing Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008235

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Fengling Auto Parts Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008232

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Yifeng Electromechanical Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008171

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Taiping Gangfeng Plastic Mould Factory

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008170

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Southeast Instrument Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008165

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Zhejiang Jingfan Saddle Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008164

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Wenling Nuobo Shoes Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008163

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

Application publication date: 20190122

Assignee: Taizhou Dingyue Shoes Co.,Ltd.

Assignor: Jiaxing Juteng Information Technology Co.,Ltd.

Contract record no.: X2024980008160

Denomination of invention: A sliding mode power cancellation direct power control method for three-phase voltage source PWM converters

Granted publication date: 20201120

License type: Common License

Record date: 20240628

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