CN102946110A - Fixed frequency model prediction control method for voltage type PWM (Pulse Width Modulation) rectifier in process of voltage unbalance drop - Google Patents
Fixed frequency model prediction control method for voltage type PWM (Pulse Width Modulation) rectifier in process of voltage unbalance drop Download PDFInfo
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- CN102946110A CN102946110A CN2012104160034A CN201210416003A CN102946110A CN 102946110 A CN102946110 A CN 102946110A CN 2012104160034 A CN2012104160034 A CN 2012104160034A CN 201210416003 A CN201210416003 A CN 201210416003A CN 102946110 A CN102946110 A CN 102946110A
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Abstract
The invention discloses a fixed frequency model prediction control method for a voltage type PWM (Pulse Width Modulation) rectifier in the process of voltage unbalance drop. The key point of the technical scheme is as follows: the prediction control method comprises the steps as follows: carrying out a subtraction operation on second harmonic frequency given values of active power and reactive power and input active power and reactive power obtained by calculation; passing through a proportional resonant controller; calculating through a compensation voltage to obtain a second harmonic frequency compensation voltage in a two-phase static coordinate system and compensating into a predication model; obtaining three vector function time by a fixed frequency prediction control principle and sending to a modulator; and obtaining a switch signal for controlling a power converter by modulating. According to the fixed frequency model prediction control method, the second harmonic frequency fluctuation of the input power of the PWM rectifier can be effectively inhibited, so that the system is guaranteed to run in unit power factor, and the running quality of the rectifier is effectively improved.
Description
Technical field
The control method of voltage type PWM rectifier when the present invention relates to a kind of unbalanced source voltage and falling, belong to electric and electronic power converting means control field, the fixed model predictive control method frequently of voltage type PWM rectifier when particularly a kind of unbalanced source voltage falls.
Background technology
Power inverter control method based on Model Predictive Control is rapidly developed in recent years.Model Predictive Control is a kind of control algolithm of coming the following response of PREDICTIVE CONTROL object based on Mathematical Modeling, comprise a cost function that defines according to the control target in the algorithm, by minimizing cost function, algorithm obtains the effect vector of optimum voltage vector as next cycle in each sampling period prediction.Fixed frequently formula Model Predictive Control is a branch of Model Predictive Control, and the method adds modulation technique in Model Predictive Control, has a switching frequency lower, is easy to the advantages such as electromagnetic interference design.
When unbalanced source voltage falls, will produce the secondary pulsation based on the PWM rectifier AC power of deciding frequency formula model predictive controller, have a strong impact on the riding quality of rectifier.In the situation that unbalanced source voltage is when falling, traditional PWM rectifier surely frequency formula model predictive control method is to adopt the symmetrical component theory that system is carried out positive and negative order to decompose, calculate positive and negative order given value of current value, adopt again that controller aligns, negative-sequence current is regulated.Positive and negative phase sequence decomposable process when system moves will exert an influence to response speed, control precision and the stable operation of control system.In addition, active power and reactive power fluctuation that the method control structure is complicated, operand is large, can not suppress system simultaneously.
Therefore, be necessary to design that a kind of unbalanced source voltage falls voltage type PWM rectifier under the condition decides frequency formula model predictive control method, when control system is moved, the input active power, the reactive power that do not need complicated positive and negative order decomposition can suppress simultaneously again the PWM rectifier are pulsed, and guarantee the control quality of system.
Summary of the invention
The technical problem that the present invention solves has provided a kind of Voltage unbalance when falling voltage type PWM rectifier is model predictive control method frequently surely, the method can be in the situation that unbalanced source voltage when falling, the input active power of PWM rectifier, reactive power pulsation are suppressed simultaneously, improve fixed frequently formula Model Predictive Control and fall under the condition control effect to Three-Phase PWM Rectifier at Voltage unbalance.
Technical scheme of the present invention is: the fixed model predictive control method frequently of voltage type PWM rectifier when a kind of Voltage unbalance falls is characterized in that may further comprise the steps: (1), detect Three-phase PWM Voltage Rectifier system three phase network voltage, three-phase input current and DC bus-bar voltage; (2), the three phase network voltage that detects and three-phase input current are obtained line voltage and input current under the two-phase rest frame through 3/2 conversion module, with the process of the mains voltage signal under two-phase rest frame software phase-lock loop, obtain line voltage position angle and line voltage angular velocity of rotation, take the voltage location angle as angle of transformation the line voltage under the two-phase rest frame and input current value are carried out the Park conversion, obtain line voltage and input current under the two-phase rotating coordinate system; (3), according to the line voltage under two cordic phase rotators and input current computing system active power and reactive power actual value, with system's active power, reactive power two frequency multiplication set-point and active power, the reactive power actual value is poor, pass through respectively ratio resonant controller (1), ratio resonant controller (2) also calculates active power by bucking voltage, reactive power two frequency multiplication bucking voltages, it is poor that the DC bus-bar voltage actual value that DC bus-bar voltage reference value and step (1) are obtained is done, obtain d shaft current reference value under the synchronous rotating frame through the PI controller, establishing q shaft current reference value is 0; (4), the α β plane is divided into six sectors, according to the voltage location angle, determine sector, line voltage vector place, select adjacent with sector, line voltage vector place two voltage vectors and zero vector as the effect vector, obtain two voltage vectors and magnitude of voltage corresponding to zero vector under the two-phase rest frame of above selection according to switch list and DC bus-bar voltage, this magnitude of voltage through the Park conversion, is obtained magnitude of voltage corresponding under the two-phase rotating coordinate system; (5), two frequency multiplication bucking voltages that two voltage vectors of obtaining in the step (4) and zero vector corresponding voltage and step (3) under the two-phase rotating coordinate system are obtained are poor, the converter input voltage after being compensated; (6), the converter input voltage after the compensation that obtains of the line voltage under the two-phase rotating coordinate system that step (2) is obtained and input current value, step (5) is as the input of current forecasting model, obtains the rate of change of d, q shaft current; (7), adopt the rate of change of input current value under the two-phase rotating coordinate system that step (2) obtains, d that step (3) obtains, q shaft current reference value, d that step (6) obtains, q shaft current as the input of vector duration computing module, obtain the action time of each vector; (8), with in the step (7) to each vector be input to modulator action time, with the switch position signal of the modulator output switching signal as the power ratio control device.
As further execution mode, the expression formula of vector duration computing module is described in the step (7):
In the formula:
,
e Dm,
e Dn,
e DlBe respectively d shaft current rate of change under the selected vector effect,
e Qm,
e Qn,
e QlBe respectively q shaft current rate of change under the selected vector effect, each vector satisfies action time
t 0+
t 1+
t 2=
T s
The fixed model predictive control method frequently of voltage type PWM rectifier when unbalanced source voltage of the present invention falls, by sector, line voltage vector place adjacent voltage vector and zero vector are carried out real-time ripple compensation, eliminate unbalanced source voltage and fall the impact that control system is caused, establishment the fluctuation of input power, strengthened and decided the runnability of frequency formula Model Predictive Control system when unbalanced source voltage falls.
Description of drawings
Fig. 1 is Three-phase PWM Voltage Rectifier main circuit structure figure; Fig. 2 is that sector definition figure and each voltage vector are to the influence of peak current schematic diagram; Fig. 3 is control structure schematic diagram of the present invention; Fig. 4 adopts the input power oscillogram before and after the control algolithm of the present invention.
Specific implementation method
The present invention will be further described below in conjunction with accompanying drawing.Among Fig. 1,
u Ga,
u Gb,
u GcBe the AC three-phase voltage source,
i Ga,
i Gb,
i GcBe three-phase alternating current side electric current,
u Ca,
u Cb,
u CcBe power bridge input side three-phase voltage,
u DcBe dc voltage,
L gWith
R gBe respectively inlet wire inductance and equivalent resistance thereof,
CBe dc filter capacitor, O is the electrical network mid point,
i LBe load current, the DC side load is by resistance
R LEquivalently represented.Definition unipolarity two-valued function switch function
S k ,When
S k =1(
k=a, b, c) the expression converter
kGo up mutually brachium pontis open-minded, lower brachium pontis is closed;
S k The upper brachium pontis of=0 expression is closed, and lower brachium pontis is open-minded.
Under electrical network three-phase voltage balance and stable case, can get the Mathematical Modeling of PWM rectifier under synchronous rotating frame:
In the formula:
u Gd,
u GqBe respectively d, the q axle component of line voltage;
i Gd , i GqBe respectively d, the q axle component of ac-side current;
u Cd , u CqBe respectively d, the q axle component of rectifier bridge AC input voltage;
w gBe the line voltage angular velocity of rotation;
Each electric weight all contains two frequency multiplication of acs in the forward rotating coordinate system when unbalanced source voltage falls, so line voltage, rectifier bridge AC voltage and input current can be expressed as
In the formula: subscript 0,2 represents respectively DC component and two harmonics.
By formula (2) as can be known, unbalanced source voltage falls that each electric weight shows as DC component and two harmonic sums under the condition in the forward synchronous rotating frame.For two harmonics are controlled separately, bring formula (2) into formula (1) and DC quantity in the formula is separated with two harmonics, can get system's two frequency multiplication voltage equations
Fall at unbalanced source voltage that system's active power and reactive power equation are under the condition
In the formula:
P G0,
Q G0Be respectively active power, reactive power DC component;
P C2,
Q C2Be respectively active power, reactive power cosine two harmonic amplitudes;
P S2,
Q S2Be respectively active power, the sinusoidal two harmonic amplitudes of reactive power;
Power two harmonics in the formula (4) are
Wherein
In the formula: subscript n, p represent respectively the forward and backward synchronous rotating frame; Subscript n, p represents respectively positive and negative sequence component.
Each electric weight in the reverse sync rotating coordinate system in the formula (5) is forwarded in the forward synchronous rotating frame, draw the system power that is rotated in the forward when unbalanced source voltage falls in the coordinate system and be
By formula (6) as can be known, positive sequence component is DC component, and negative sequence component is two harmonics.Make d axle and line voltage vector in the same way, at this moment
=0.Because two harmonics of line voltage q axle fluctuate near 0, in order to reduce the control system complexity, ignore this wave component, can get system power two frequency multiplication equations and be
In order to embody the control to power two harmonics, to formula (7) differentiate, and the DC component when considering stable situation can think constant,
The governing equation that can get active power and reactive power two harmonics according to formula (3) and formula (8) is
In the formula:
,
Be respectively PWM rectifier two frequency multiplication bucking voltages in the synchronous rotating frame, its in proportion resonance control principle be designed to
Wherein
(11)
(12)
In the formula:
k Pr,
k IrBe respectively ratio, the resonance coefficient of ratio resonant controller;
ω gBe the line voltage angular velocity of rotation;
,
Be respectively active power, reactive power two frequency multiplication command value.In the formula (10)
v Gd2,
v Gq2Coefficient can compensate with proportionality coefficient, formula (11) can be by compensation of resonators for Front Feed Compensation.
The rate of change that can be got grid side converter d, q shaft current by formula (1) is
By formula (13) as can be known, d, q shaft current rate of change are subjected to system parameters, line voltage, converter input voltage and the influence of peak current.Fig. 2 has provided space vector of voltage and sector dividing condition, and as can be seen from Figure, the grid side converter input voltage can represent by eight voltage vectors respectively, wherein six be effective vector (
V 1~
V 6), two be zero vector (
V 0,
V 7), each vector is static in two-phase
α,
βSize in the coordinate system and corresponding on off state thereof are as shown in the table.
| Voltage vector ( S c S b S a） | u c α | u cβ |
| V 1 （001） | 2 u dc /3 | 0 |
| V 2 （010） | - u dc /3 |  |
| V 3 （011） | u dc /3 | u dc /3 |
| V 4 （100） | u dc /3 |  |
| V 5 （101） | -2 u dc /3 | 0 |
| V 6 （110） | - u dc /3 |  |
| V 0 （000）, V 7 （111） | 0 | 0 |
Take sector III as example, Fig. 2 has provided each voltage vector to the influence of peak current, because inlet wire reactance equivalent resistance is generally less, does not consider resistance in the analysis
R gImpact.As can be seen from Figure, in whole sector, when selecting voltage vector
V 6,
V 4 , V 0Or
V 7The time d
i d/ d
tFor just; Voltage vector from
θ=0o rotates to
θIn=60o the process, vector
V 3Effect make d
i d/ d
t θFor just, then become negative in the time of near=the 0o.And vector
V 1Effect make d
i d/ d
t θFor just, interval for negative at other in the time of near=the 60o.Therefore when selecting vector
V 3,
V 1The time, d in the overwhelming majority of sector III is interval
i d/ d
tFor negative.Vector
V 1,
V 5,
V 4Keep d
i q/ d
tFor just; And vector
V 3,
V 2,
V 6Make d
i q/ d
tFor negative.In brief, voltage vector
V 1In most intervals of sector III the d shaft current is reduced, the q shaft current is increased; And voltage vector
V 3In the most intervals at sector III the d shaft current is reduced, in whole sector the q shaft current is reduced.Other sector voltage vector action effect and sector III are similar.
The present invention is a sampling period
T sThree voltage vectors of interior selection are respectively two effective vectors and a zero vector.In each cycle, two adjacent vectors of sector, chosen distance line voltage vector place are effective vector, and zero vector can adopt vector
V 0Or
V 7, then in three vectors the vector that makes electric current increase and reduce must be arranged simultaneously.
Corresponding switch list be can make up by above voltage vector selection rule, zero vector and vector selected at sector I
V 2,
V 3Select zero vector and vector at sector II
V 1,
V 5Select zero vector and vector at sector III
V 1,
V 3Select zero vector and vector at sector IV
V 4,
V 6Select zero vector and vector at sector V
V 2,
V 6Select zero vector and vector at sector VI
V 4,
V 5
To be transformed under the synchronous rotating frame according to the effective voltage vector that the line voltage angle is determined, obtain magnitude of voltage corresponding to effective voltage vector under the synchronous rotating frame, the magnitude of voltage under the corresponding synchronous rotating frame of effective voltage vector that the two frequency multiplication bucking voltages compensation of employing formula (10) is chosen
In the formula, m, n, l represent respectively three voltage vectors choosing, m, n, l ∈ [0,7].
With the pwm converter input voltage after the compensation
u Cdc
χ With
u Cqc
χ Substitution formula (13) can obtain the rate of change of d, q shaft current
Can be obtained respectively under the different voltage vector effects in each sector the rate of change of d, q shaft current by formula (15).Therefore in the current effect vector duration
t n In, d, q shaft current variable quantity can be expressed as
In the formula:
i Gd(
k),
i Gq(
k) be respectively current vector and begin effect constantly d, q shaft current value;
i Gd(
k+ 1),
i Gq(
k+ 1) is respectively the current vector effect d finish time, q shaft current value.
If
t 0,
t 1,
t 2Three voltage vectors that represent respectively selection in each switch periods
V m,
V n,
V lAction time.Among the figure,
kWhen individual switch periods finished, the current tracking error can be expressed as
In the formula:
e Dm,
e Dn,
e DlBe respectively d shaft current rate of change under the selected vector effect;
e Qm,
e Qn,
e QlBe respectively q shaft current rate of change under the selected vector effect; Each vector satisfies action time
t 0+
t 1+
t 2=
T s
The control target of MPC is in each switch periods finish time, makes actual current and given current error minimum.In order in each control cycle, to reduce to greatest extent d, q shaft current error, adopt least square optimized algorithm definition target function
With target function
WMinimum is constraints, can obtain each control cycle
T sInterior three vectors
V m,
V n , V lThe best use of time.The calculating of action time should be satisfied following condition
Simultaneous formula (17), formula (18) and formula (19) can get each vector action time
t 0,
t 1With
t 2For
After three voltage vectors were determined by the line voltage vector position, it was at next control cycle
T sAction time can be calculated by formula (20).But in certain control cycle, when the sum action time of two effective voltage vectors
t 1+
t 2 T sThe time, zero vector no longer acts on, and be adjusted into the action time of two effective voltage vectors respectively
(21)
Sent to modulator the action time of zero vector and two effective voltage vectors, get final product the switching signal of controlled power inverter by modulation.
Fig. 3 is control structure schematic diagram of the present invention, and its control method specifically comprises the steps:
(1), adopt voltage, current sensor to detect Three-phase PWM Voltage Rectifier system three phase network voltage
u Ga,
u Gb,
u Gc, three-phase input current
i Ga,
i Gb,
i GcAnd DC bus-bar voltage
u Dc
(2), with the three phase network voltage that detects
u Ga,
u Gb,
u GcAnd three-phase input current
i Ga,
i Gb,
i GcObtain line voltage under the two-phase rest frame through 3/2 conversion module
u G α,
u G βAnd input current
i G α,
i G β, the mains voltage signal under the two-phase rest frame through software phase-lock loop, is obtained the line voltage position angle
θWith the line voltage angular velocity of rotation
ω g, with the voltage location angle
θFor angle of transformation to the line voltage under the two-phase rest frame
u G α,
u G βAnd input current
i G α,
i G βCarry out the Park conversion, obtain the line voltage under the two-phase rotating coordinate system
u Gd,
u GqAnd input current
i Gd,
i Gq
(3), according to the line voltage under two cordic phase rotators
u Gd,
u GqAnd input current
i Gd,
i GqComputing system active power
P gAnd reactive power
Q g, with system's active power, reactive power two frequency multiplication set-points
P g *,
Q g *With active power, reactive power actual value
P g,
Q gPoor, pass through respectively ratio resonant controller (1), ratio resonant controller (2) and calculate active power, reactive power two frequency multiplication bucking voltages by bucking voltage
u Cd2 *,
u Cq2 *With the DC bus-bar voltage reference value
u Dc *The DC bus-bar voltage actual value that obtains with step (1)
u DcIt is poor to do, and obtains d shaft current reference value under the synchronous rotating frame through the PI controller
i Gd * , establish q shaft current reference value
i Gq * Be 0;
(4), the α β plane is divided into six sectors, according to the voltage location angle
θ, determine sector, line voltage vector place, select two voltage vectors adjacent with sector, line voltage vector place
V m,
V nAnd zero vector
V lAs the effect vector, obtain two voltage vectors and magnitude of voltage corresponding to zero vector under the two-phase rest frame of above selection according to switch list and DC bus-bar voltage, this magnitude of voltage through the Park conversion, is obtained magnitude of voltage corresponding under the two-phase rotating coordinate system
u Cd
χ ,
u Cq
χ ,
(5), with two voltage vectors of obtaining in the step (4) and zero vector corresponding voltage under the two-phase rotating coordinate system
u Cd
χ ,
u Cq
χ The two frequency multiplication bucking voltages that obtain with step (3)
u Cd2 *,
u Cq2 *Differ from the converter input voltage after being compensated
u Cdc
χ ,
u Cqc
χ
(6), the line voltage under the two-phase rotating coordinate system that step (2) is obtained
u Gd,
u GqAnd input current value
i Gd,
i Gq, the converter input voltage after the compensation that obtains of step (5)
u Cdc
χ ,
u Cqc
χ As the input of current forecasting model, obtain the rate of change of d, q shaft current
e d
χ ,
e q
χ
(7), adopt input current value under the two-phase rotating coordinate system that step (2) obtains
i Gd,
i Gq, step (3) d, the q shaft current reference value that obtain
i Gd * ,
i Gq * , the d that obtains of step (6), the rate of change of q shaft current
e d
χ ,
e q
χ As the input of vector duration computing module, obtain the action time of each vector
t 0,
t 1,
t 2
(8), with each vector action time of arriving in the step (7)
t 0,
t 1,
t 2Be input to modulator, with the switch position signal of the modulator output switching signal as the power ratio control device.
As further execution mode, the expression formula of vector duration computing module is described in the step (7):
In the formula:
,
e Dm,
e Dn,
e DlBe respectively d shaft current rate of change under the selected vector effect,
e Qm,
e Qn,
e QlBe respectively q shaft current rate of change under the selected vector effect, each vector satisfies action time
t 0+
t 1+
t 2=
T s
Fig. 4 is for adopting the surely input power oscillogram of the PWM rectifier of frequency model predictive control method front and back of the present invention, as seen from Figure 4, before adopting control algolithm of the present invention, the input active power of PWM rectifier, the fluctuation of two frequencys multiplication occurs in reactive power, in addition, because controller can not accurately be followed the tracks of q shaft current set-point, therefore, reactive power is non-vanishing, can not realize unity power factor control, when 0.15s, adopt control algolithm of the present invention, as can be seen from Figure, rectifier input active power, the two frequencys multiplication fluctuation of reactive power is effectively suppressed rapidly, simultaneously, the input reactive power is zero, has realized the accurate tracking to given electric current.
In sum, fixed frequently model predictive control method of the present invention unbalanced source voltage fall under the condition can establishment PWM rectifier input power the fluctuation of two frequencys multiplication, guarantee the operation of system unit power factor, the riding quality of Effective Raise rectifier, to compare amount of calculation little with conventional method, control structure is simple, and system delay is less, the secondary pulsation of simultaneously elimination system input active power, reactive power.
Claims (2)
1. the fixed model predictive control method frequently of voltage type PWM rectifier when a Voltage unbalance falls is characterized in that may further comprise the steps: (1), detect Three-phase PWM Voltage Rectifier system three phase network voltage, three-phase input current and DC bus-bar voltage; (2), the three phase network voltage that detects and three-phase input current are obtained line voltage and input current under the two-phase rest frame through 3/2 conversion module, with the process of the mains voltage signal under two-phase rest frame software phase-lock loop, obtain line voltage position angle and line voltage angular speed, take the voltage location angle as angle of transformation the line voltage under the two-phase rest frame and input current value are carried out the Park conversion, obtain line voltage and input current under the two-phase rotating coordinate system; (3), according to the line voltage under two cordic phase rotators and power network current computing system active power and reactive power actual value, with system's active power, reactive power two frequency multiplication set-point and active power, the reactive power actual value is poor, pass through respectively ratio resonant controller (1), ratio resonant controller (2) also calculates active power by bucking voltage, reactive power two frequency multiplication bucking voltages, it is poor that the DC bus-bar voltage actual value that DC bus-bar voltage reference value and step (1) are obtained is done, obtain d shaft current reference value under the synchronous rotating frame through the PI controller, establishing q shaft current reference value is 0; (4), the α β plane is divided into six sectors, according to the voltage location angle, determine sector, line voltage vector place, select adjacent with sector, line voltage vector place two voltage vectors and zero vector as the effect vector, obtain two voltage vectors and magnitude of voltage corresponding to zero vector under the two-phase rest frame of above selection according to switch list and DC bus-bar voltage, this magnitude of voltage through the Park conversion, is obtained magnitude of voltage corresponding under the two-phase rotating coordinate system; (5), two frequency multiplication bucking voltages that two voltage vectors of obtaining in the step (4) and zero vector corresponding voltage and step (3) under the two-phase rotating coordinate system are obtained are poor, the converter input voltage after being compensated; (6), the converter input voltage after the compensation that obtains of the line voltage under the two-phase rotating coordinate system that step (2) is obtained and input current value, step (5) is as the input of current forecasting model, obtains the rate of change of d, q shaft current; (7), adopt the rate of change of input current value under the two-phase rotating coordinate system that step (2) obtains, d that step (3) obtains, q shaft current reference value, d that step (6) obtains, q shaft current as the input of vector duration computing module, obtain the action time of each vector; (8), with in the step (7) to each vector be input to modulator action time, with the switch position signal of the modulator output switching signal as the power ratio control device.
2. the fixed model predictive control method frequently of voltage type PWM rectifier when Voltage unbalance according to claim 1 falls, it is characterized in that: the expression formula of vector duration computing module is in the described step (7):
In the formula:
,
e Dm,
e Dn,
e DlBe respectively d shaft current rate of change under the selected vector effect,
e Qm,
e Qn,
e QlBe respectively q shaft current rate of change under the selected vector effect, each vector satisfies action time
t 0+
t 1+
t 2=
T s
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| CN107769219A (en) * | 2017-11-24 | 2018-03-06 | 华南理工大学 | A kind of substation capacitors operation/cutting method for taking into account non-equilibrium among three phase voltages improvement |
| CN111525591A (en) * | 2020-04-30 | 2020-08-11 | 陕西科技大学 | VSC control method under three-phase unbalanced state |
| CN111969594A (en) * | 2020-05-20 | 2020-11-20 | 国网山东省电力公司青岛供电公司 | Control method and system of back-to-back converter |
| CN112713835A (en) * | 2020-12-21 | 2021-04-27 | 南通斯迈尔精密设备有限公司 | Permanent magnet synchronous linear motor model prediction current control method considering prediction model decomposition |
Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO1999014836A1 (en) * | 1997-09-15 | 1999-03-25 | Siemens Westinghouse Power Corporation | Power flow controller with dc-to-dc converter linking shunt and series connected inverters |
| US20030007368A1 (en) * | 2001-06-29 | 2003-01-09 | Hironari Kawazoe | Power generation apparatus using permanent-magnet generator |
| JP4204846B2 (en) * | 2002-10-31 | 2009-01-07 | 三菱電機株式会社 | Control unit for AC excitation generator motor |
| CN101478283A (en) * | 2009-01-22 | 2009-07-08 | 北京清能华福风电技术有限公司 | Dual feedback asynchronous wind power generator rotor side inverter control method under unbalanced electric grid voltage |
| CN101951174A (en) * | 2010-09-11 | 2011-01-19 | 天津大学 | Constant-frequency direct power control method for PWM converter under condition of power grid voltage imbalance |
| CN102655332A (en) * | 2012-04-28 | 2012-09-05 | 北方工业大学 | Method for controlling network side rectifier of brushless double-feed wind-driven generator under unbalanced electric network |
-
2012
- 2012-10-26 CN CN201210416003.4A patent/CN102946110B/en not_active Expired - Fee Related
Patent Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO1999014836A1 (en) * | 1997-09-15 | 1999-03-25 | Siemens Westinghouse Power Corporation | Power flow controller with dc-to-dc converter linking shunt and series connected inverters |
| US20030007368A1 (en) * | 2001-06-29 | 2003-01-09 | Hironari Kawazoe | Power generation apparatus using permanent-magnet generator |
| JP4204846B2 (en) * | 2002-10-31 | 2009-01-07 | 三菱電機株式会社 | Control unit for AC excitation generator motor |
| CN101478283A (en) * | 2009-01-22 | 2009-07-08 | 北京清能华福风电技术有限公司 | Dual feedback asynchronous wind power generator rotor side inverter control method under unbalanced electric grid voltage |
| CN101951174A (en) * | 2010-09-11 | 2011-01-19 | 天津大学 | Constant-frequency direct power control method for PWM converter under condition of power grid voltage imbalance |
| CN102655332A (en) * | 2012-04-28 | 2012-09-05 | 北方工业大学 | Method for controlling network side rectifier of brushless double-feed wind-driven generator under unbalanced electric network |
Non-Patent Citations (3)
| Title |
|---|
| CHANGLIANG XIA: "Robust Model Predictive Current Control of Three-Phase Voltage Source PWM Rectifier With Online Disturbance Observation", 《IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS》, 31 August 2012 (2012-08-31), pages 459 - 471, XP011454357, DOI: doi:10.1109/TII.2012.2187912 * |
| YAN YAN: "Proportional-Resonant Control of Doubly-Fed Induction Generator Wind Turbines for Low-Voltage Ride-Through Enhancement", 《ENERGIES》, 31 May 2012 (2012-05-31) * |
| 王萌: "不平衡电网电压条件下PWM整流器功率谐振补偿控制策略", 《中国电机工程学报》, vol. 32, no. 21, 25 July 2012 (2012-07-25) * |
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| CN107769219A (en) * | 2017-11-24 | 2018-03-06 | 华南理工大学 | A kind of substation capacitors operation/cutting method for taking into account non-equilibrium among three phase voltages improvement |
| CN107769219B (en) * | 2017-11-24 | 2020-12-22 | 华南理工大学 | Transformer substation capacitor switching method considering improvement of three-phase voltage unbalance degree |
| CN111525591A (en) * | 2020-04-30 | 2020-08-11 | 陕西科技大学 | VSC control method under three-phase unbalanced state |
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