CN109239744B - Rapid anti-bit-reversal rapid capturing method based on complex signal phase - Google Patents

Rapid anti-bit-reversal rapid capturing method based on complex signal phase Download PDF

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CN109239744B
CN109239744B CN201811221034.8A CN201811221034A CN109239744B CN 109239744 B CN109239744 B CN 109239744B CN 201811221034 A CN201811221034 A CN 201811221034A CN 109239744 B CN109239744 B CN 109239744B
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code phase
code
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phase
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CN109239744A (en
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吴超
孙闽红
刘二小
滕旭阳
简志华
钟华
苏明坤
乔磊
仇兆炀
宋慧娜
汪立新
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Hangzhou Dianzi University
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
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Abstract

The invention mainly discloses a fast anti-bit reversal fast capturing method based on complex signal phase, which comprises the steps of carrying out down-conversion processing on a received signal, partitioning a local related signal and the received signal, carrying out code phase searching processing on an array of an obtained matrix to obtain a variable to be detected, carrying out coherent integration detection to obtain a coherent integral value of the code phase searching processing, carrying out code phase processing on the received signal to obtain a variable to be detected by the code phase processing, obtaining an estimated bit data reversal position, carrying out coherent integration detection, finally obtaining a parameter code phase to be solved, and finally obtaining an estimated captured parameter code phase and frequency, thereby realizing fast capturing of GNSS signals. The method can quickly and accurately estimate the code phase, thereby realizing the estimation of the capture parameter under the condition of bit symbol inversion.

Description

Rapid anti-bit-reversal rapid capturing method based on complex signal phase
Technical Field
The invention belongs to the technical field of communication, and particularly relates to a fast anti-bit reversal fast capturing method based on complex signal phase.
Background
With the progress of the state of technology, wireless communication technology and Global Positioning System (GPS) technology are increasingly applied to aspects of daily life. In a Global Navigation Satellite System (GNSS), positioning is performed by using observations such as pseudo ranges, ephemeris, and satellite emission time of a set of satellites, and a user clock error must be known. The satellite navigation receiver acquires and tracks signals of a plurality of GNSS satellites and then demodulates navigation data modulated therein. The satellite navigation receiver calculates the relative distance between the GNSS satellite and the user by using the ranging code, and calculates the satellite position and the time model by using ephemeris data in the navigation data, thereby calculating the position of the user. The satellite navigation positioning technology has basically replaced the ground-based radio navigation, the traditional geodetic survey and the astronomical survey navigation positioning technology at present, and promotes the brand new development of the field of geodetic survey and navigation positioning. Therefore, GNSS systems are the infrastructure for national security and socioeconomic development.
In the process of detecting a received GNSS signal (i.e., capturing a satellite signal), the frequency and code phase of a parameter to be estimated may be affected by data inversion (for example, a GPS L1C/a code received signal, where L1 is a carrier signal, and the C/a code is a pseudo random code emitted by a GPS satellite and used for coarse ranging and capturing the GPS satellite), resulting in a reduction in the detected peak value, so that the inaccurate estimated parameter affects the detection speed.
In order to improve the detection efficiency, the current main capture method adopts a fast Fourier transform-based method to estimate capture parameters. However, the capture speed is yet to be further improved, and a more effective capture strategy needs to be formulated for the randomness of bit inversion of the signal to be detected. In order to further increase the capture speed and reduce the complexity of the capture algorithm, the problem of fast capture of the capture parameters under the condition of bit data sign inversion needs to be further researched.
Disclosure of Invention
The invention aims to solve the problem of improving the speed of a capturing method under the condition of data bit inversion, and provides a quick and accurate code phase and frequency estimation method for GNSS signals, so that a quick anti-bit-inversion quick capturing method based on complex signal phases is realized.
In order to achieve the purpose, the invention adopts the following specific technical scheme:
a fast anti-bit reversal fast acquisition method based on complex signal phase comprises the following steps:
s1, performing down-conversion processing on a received signal R (n), and partitioning a local related signal and the received signal to obtain a partitioned matrix
Figure GDA0002620616840000021
And
Figure GDA0002620616840000022
wherein i represents the ith integration time, D represents the local pseudo random code when influenced by data bits, ND represents the local pseudo random code when not influenced by data bits, and M, N represents the partition matrix parameter;
s2, carrying out code phase search processing on the matrix obtained in the step S1 to obtain the variable to be detected
Figure GDA0002620616840000023
And
Figure GDA0002620616840000024
then obtaining coherent integration value of code phase search processing by coherent integration detection
Figure GDA0002620616840000025
Wherein i0For the estimated bit data inversion position, fd,k0To estimate the frequency, drEstimating a value of a code phase parameter for a code phase search process, i representing an i-th integration time, D representing a local pseudo-random code when affected by a data bit, ND representing a local pseudo-random code when not affected by a data bit, fd,kIndicating the search frequency, dcPresentation searchA chip offset of the index;
s3, carrying out code phase processing on the received signal to obtain a code phase processing variable to be detected
Figure GDA0002620616840000027
Estimated bit data inversion position i obtained from S10To, for
Figure GDA0002620616840000028
And performing coherent integration detection to finally obtain a phase of the parameter code to be solved and finally obtain the estimated phase and frequency of the captured parameter code, thereby realizing the rapid capture of the GNSS signal.
Further, the received signal r (n) is:
Figure GDA0002620616840000026
where q is the code phase of the received signal, fdIs Doppler frequency, C () is pseudo-random code, D () is modulated data, fsFor the sampling frequency, p (n) denotes that both the real and imaginary parts obey a mean of 0 and a variance of σ2N denotes a sample point, n is 0,1, …,
the received signal R (n) is processed by down-conversion, and the down-conversion signal is obtained as follows:
Figure GDA0002620616840000031
wherein, the down-conversion signal f (n) is:
Figure GDA0002620616840000032
wherein f isd,kTo search for frequency, fsFor the sampling frequency, n represents a sampling point.
The two local correlation signals are:
LD(n)＝(-1)k-1C(n)；
LND(n)＝C(n)；
wherein n ═ iT + k0T is the number of sampling points of PN code period, k0T-1, C () is a pseudo random code, i represents the i-th integration time, ld (n) and lnd (n) represent the local pseudo random code when there is a data bit influence and the local pseudo random code when there is no data bit influence, respectively;
the locally correlated signal and the down-converted signal are partitioned (all alphabetic symbols, etc. referred to in the following equations are to be interpreted, and are to be re-interpreted if present):
Figure GDA0002620616840000033
Figure GDA0002620616840000041
Figure GDA0002620616840000042
wherein N isL＝2(i-1)MN+dc，i＝1,...,Nc，NcFor coherent integration time, NRThe M and N denote partition matrix parameters, and i denotes the i-th integration time.
Further, the code phase search processing is performed on the matrix obtained in S1, and the specific method is as follows:
Figure GDA0002620616840000043
Figure GDA0002620616840000044
wherein, mod ((c)2+n),2N2) Is represented by (c)2+ N) divided by 2N2I denotes the ith integration time, D denotes the local pseudo-random code when there is a data bit influence, ND denotes the local pseudo-random code when there is no data bit influence, fd,kIndicating the search frequency, dcIndicates the chip offset of the search when dc-1<0，dc-1＝dc-1+ M; if d isc+1＞M，dc+1＝dc+1-M, detected by coherent integration:
Figure GDA0002620616840000045
Figure GDA0002620616840000046
Figure GDA0002620616840000051
wherein i1Representing possible flip positions of the symbols of the data bits, D representing the local pseudo-random code when affected by a data bit, ND representing the local pseudo-random code when unaffected by a data bit, fd,kIndicating the search frequency, dcChip offset indicating search, drCode offset, i, representing the code phase search process prediction0Indicating the predicted data bit flip position, fd,k0Is the estimated signal frequency.
Further, the code phase estimation processing for performing code phase processing on the received signal specifically includes:
Figure GDA0002620616840000052
Figure GDA0002620616840000053
wherein D represents the local pseudo random code when affected by the data bits, ND represents the local pseudo random code when unaffected by the data bits,
Figure GDA0002620616840000054
indicating the ith without bit flipping in code phase estimation for code phase processingNSub-integration time chip offset d2The time of reception of the signal integral value,
Figure GDA0002620616840000055
indicating the ith bit flip in the code phase estimation of the code phase processingNSub-integration time chip offset d2Integral value of time-of-reception signal, iNIndicating ith without bit flippingNTime of sub-integration, iD＝i0+1 denotes the ith bit flipDIntegration time, M, N denotes the blocking matrix parameter, i0Indicating the predicted data bit flip position, N ═ M2N2，M2、N2Representing the code phase prediction parameters, and,
and then carrying out coherent integration detection:
Figure GDA0002620616840000056
Figure GDA0002620616840000057
Figure GDA0002620616840000061
wherein d isr2A code phase prediction parameter representing a code phase process,
Figure GDA0002620616840000062
detection amount after coherent integration in code phase prediction of code phase processing, D represents local pseudo-random code when there is influence of data bit, ND represents local pseudo-random code when there is no influence of data bit, i0Indicating the predicted data bit flip position, d2The chip offset at the time of code phase prediction representing code phase processing can be obtained as d according to the PN code correlation property2＝dr2The maximum value is obtained:
Figure GDA0002620616840000063
wherein, cr2、dr2Code phase prediction parameter representing code phase processing, M represents a scoreBlock matrix parameter, M2、N2Representing the code phase prediction parameter, M2、N2A code phase prediction blocking matrix parameter representing a code phase process. Let X2(dr2) Phase angle of
Figure GDA0002620616840000064
Then
Figure GDA0002620616840000065
Wherein, cr2Code phase prediction parameter, N, representing code phase processing2Representing the code phase prediction parameters. Finally, the received estimated code phase may be expressed as
q＝(cr2M2+dr2)M+dr；
Wherein, cr2、dr2Code phase prediction parameters representing the code phase processing, drCode phase prediction parameters representing a code phase search process, M represents code phase prediction blocking matrix parameters of a code phase search process, M2A code phase prediction blocking matrix parameter representing a code phase process.
By adopting the technical scheme, the invention has the beneficial effects that: the bit symbol inversion of the invention can adopt a two-step signal code phase fast estimation method based on the transformation signal complex phase estimation in an integral interval, and the code phase is divided into different intervals to estimate the code phase. The code phase can be estimated quickly and accurately based on the two-step code phase estimation method, so that the acquisition parameter estimation under the condition of bit symbol inversion is realized.
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Fig. 1 is a flowchart of a fast anti-bit-reversal fast acquisition method based on complex signal phase according to the present invention.
Fig. 2 is a flowchart comparing the prior art method with the fast capture method based on complex signal phase for fast anti-bit reversal.
FIG. 3 is a diagram of the comparison of the complex addition computation amount between the prior art method and the fast anti-bit-reversal fast acquisition method based on the complex signal phase according to the present invention.
FIG. 4 is a comparison graph of complex multiplication computation of the prior art method and the fast anti-bit-reversal fast acquisition method based on complex signal phase according to the present invention.
Detailed Description
As shown in fig. 1, a fast anti-bit-reversal fast acquisition method based on complex signal phase includes the following steps:
s1, performing down-conversion processing on a received signal R (n), and partitioning a local related signal and the received signal to obtain a partitioned matrix
Figure GDA0002620616840000071
And
Figure GDA0002620616840000072
wherein i represents the ith integration time, D represents the local pseudo random code when affected by a data bit, ND represents the local pseudo random code when not affected by a data bit, and M, N represents the partition matrix parameter.
The received signal R (n) is:
Figure GDA0002620616840000073
where q is the code phase of the received signal, fdIs Doppler frequency, C () is pseudo-random code, D () is modulated data, fsFor the sampling frequency, p (n) denotes that both the real and imaginary parts obey a mean of 0 and a variance of σ2N denotes a sample point, n is 0,1, …,
the received signal R (n) is processed by down-conversion, and the down-conversion signal is obtained as follows:
Figure GDA0002620616840000074
wherein, the down-conversion signal f (n) is:
Figure GDA0002620616840000075
wherein f isd,kTo search for frequency, fsFor the sampling frequency, n represents a sampling point.
The two local correlation signals are:
LD(n)＝(-1)k-1C(n)；
LND(n)＝C(n)；
wherein n ═ iT + k0T is the number of sampling points of PN code period, k0T-1, C () is a pseudo random code, i represents the ith integration time, LD(n) and LND(n) respectively representing a local pseudo-random code when influenced by data bits and a local pseudo-random code when not influenced by the data bits;
partitioning the local correlation signal and the down-converted signal:
Figure GDA0002620616840000081
Figure GDA0002620616840000082
Figure GDA0002620616840000083
wherein N isL＝2(i-1)MN+dc，i＝1,...,Nc，NcFor coherent integration time, NRThe M and N denote partition matrix parameters, and i denotes the i-th integration time.
S2, carrying out code phase search processing on the matrix obtained in the step S1 to obtain the variable to be detected
Figure GDA0002620616840000091
And
Figure GDA0002620616840000092
then obtaining coherent integration value of code phase search processing by coherent integration detection
Figure GDA0002620616840000093
Wherein i0In order to estimate the bit data inversion position,
Figure GDA0002620616840000094
to estimate the frequency, drEstimating a value of a code phase parameter for a code phase search process, i representing an i-th integration time, D representing a local pseudo-random code when affected by a data bit, ND representing a local pseudo-random code when not affected by a data bit, fd,kIndicating the search frequency, dcIndicating the chip offset of the search.
The code phase search processing is performed on the matrix obtained in S1, and the specific method is as follows:
Figure GDA0002620616840000095
Figure GDA0002620616840000096
wherein M, N represents the partition matrix parameter, i represents the ith integration time, D represents the local pseudo random code when there is data bit influence, ND represents the local pseudo random code when there is no data bit influence, fd,kIndicating the search frequency, dcIndicates the chip offset of the search when dc-1<0，dc-1＝dc-1+ M; if d isc+1＞M，dc+1＝dc+1-M, detected by coherent integration:
Figure GDA0002620616840000097
Figure GDA0002620616840000098
Figure GDA0002620616840000099
wherein i1Indicating possible flip positions of data bit symbols, D indicating a local pseudo-random code when affected by a data bit, ND indicating a local pseudo-random code when not affected by a data bitRandom code, fd,kIndicating the search frequency, dcChip offset indicating search, drCode offset, i, representing the code phase search process prediction0Indicating the predicted data bit flip position, fd,k0Is the estimated signal frequency.
S3, carrying out code phase processing on the received signal to obtain a code phase processing variable to be detected
Figure GDA0002620616840000101
Estimated bit data inversion position i obtained from S10To, for
Figure GDA0002620616840000102
And performing coherent integration detection to finally obtain a phase of the parameter code to be solved and finally obtain the estimated phase and frequency of the captured parameter code, thereby realizing the rapid capture of the GNSS signal.
The code phase estimation processing for performing code phase processing on the received signal specifically includes:
Figure GDA0002620616840000103
Figure GDA0002620616840000104
wherein D represents the local pseudo random code when affected by the data bits, ND represents the local pseudo random code when unaffected by the data bits,
Figure GDA0002620616840000105
indicating the ith without bit flipping in code phase estimation for code phase processingNSub-integration time chip offset d2The time of reception of the signal integral value,
Figure GDA0002620616840000106
indicating the ith bit flip in the code phase estimation of the code phase processingNSub-integration time chip offset d2Integral value of time-of-reception signal, iNMeans for indicating nothingIth bit flipNTime of sub-integration, iD＝i0+1 denotes the ith bit flipDIntegration time, M, N denotes the blocking matrix parameter, i0Indicating the predicted data bit flip position, N ═ M2N2，M2、N2Representing the code phase prediction parameters, and,
and then carrying out coherent integration detection:
Figure GDA0002620616840000107
Figure GDA0002620616840000108
Figure GDA0002620616840000111
wherein d isr2A code phase prediction parameter representing a code phase process,
Figure GDA0002620616840000112
detection amount after coherent integration in code phase prediction of code phase processing, D represents local pseudo-random code when there is influence of data bit, ND represents local pseudo-random code when there is no influence of data bit, i0Indicating the predicted data bit flip position, d2The chip offset at the time of code phase prediction representing code phase processing can be obtained as d according to the PN code correlation property2＝dr2The maximum value is obtained:
Figure GDA0002620616840000113
wherein, cr2、dr2Code phase prediction parameters representing code phase processing, M representing block matrix parameters, M2、N2Representing the code phase prediction parameter, M2、N2A code phase prediction blocking matrix parameter representing a code phase process. Let X2(dr2) Phase angle of
Figure GDA0002620616840000114
Then
Figure GDA0002620616840000115
Wherein, cr2Code phase prediction parameter, N, representing code phase processing2Representing the code phase prediction parameters. Finally, the received estimated code phase may be expressed as
q＝(cr2M2+dr2)M+dr；
Wherein, cr2、dr2Code phase prediction parameters representing the code phase processing, drCode phase prediction parameters representing a code phase search process, M represents code phase prediction blocking matrix parameters of a code phase search process, M2A code phase prediction blocking matrix parameter representing a code phase process.
As shown in fig. 2, where c (n) is a local signal and NF is the number of sampling points in a unit integration time. R0(m) integral value, R, obtained without data modulation for a pseudo code period1And (m) the integral value is obtained by modulating data for one pseudo code period, and m is the chip offset. The integral value is assumed to be R when no data modulation is carried out in the ith pseudo code period0，i(m) assuming that the integral value is R when no data modulation is performed in the ith pseudo code period1，i(m), the cumulative result of the upper graph is:
Figure GDA0002620616840000121
Figure GDA0002620616840000122
where K is the accumulation time, K is 1, …, K. And obtaining a maximum value according to the accumulated peak value, wherein the chip offset corresponding to the maximum value is the code phase estimation value of the received signal.
Table 1 shows the comparison between the complex addition calculation and the complex multiplication calculation of the prior art method and the fast anti-bit-reversal fast acquisition method based on complex signal phase according to the prior art and the proposed method:
Figure GDA0002620616840000123
TABLE 1
Wherein N iscRepresenting integration time, NFThe number of sampling points per unit integration time. M, N, M therein2And N2The parameters represented by the formula steps S2 and S3 represent the first and second fractional block parameters.
Fig. 3 is a comparison graph of complex addition computation of the prior art method and the fast anti-bit-reversal fast acquisition method based on complex signal phase according to the present invention.
Fig. 4 is a comparison graph of complex multiplication computation quantities of a prior art method and a fast anti-bit-reversal fast acquisition method based on complex signal phase.
From fig. 3 and 4, we can find that the method of the present invention employs the complex multiplication calculation amount and the complex addition calculation amount which are less consumed by converting the estimated code phase to the estimated complex signal phase than the existing method based on fast fourier transform, so that the acquisition parameters can be estimated more quickly, and the purpose of fast acquisition is achieved.
Wherein Nc is 1-11 ms, NF＝1023，M＝N＝32，M2＝4，N2＝8，P1＝P 28. From the above figure, it can be seen that the proposed method can greatly reduce the amount of computation compared to the current generation method.
Wherein, FFT is FFT (fast Fourier transform) is fast algorithm of Discrete Fourier Transform (DFT).
Herein, a Pseudo-random code refers to a code sequence consisting of 0 and 1 (PN code) having autocorrelation properties similar to white Noise.
It will be appreciated by those skilled in the art that the specific embodiments of the invention are merely illustrative of the principles of the invention and are not limiting of the invention. All equivalent changes or modifications made according to the design spirit of the present invention shall fall into the protection scope of the present invention.

Claims (4)

1. A fast anti-bit reversal fast acquisition method based on complex signal phase is characterized by comprising the following steps:
s1, performing down-conversion processing on a received signal R (n), and partitioning a local related signal and the received signal to obtain a partitioned matrix
Figure FDA0002620616830000011
And
Figure FDA0002620616830000012
wherein i represents the ith integration time, D represents the local pseudo random code when influenced by data bits, ND represents the local pseudo random code when not influenced by data bits, and M, N represents the partition matrix parameter;
s2, carrying out code phase search processing on the matrix obtained in the step S1 to obtain the variable to be detected
Figure FDA0002620616830000013
And
Figure FDA0002620616830000014
then obtaining coherent integration value of code phase search processing by coherent integration detection
Figure FDA0002620616830000015
Wherein i0In order to estimate the bit data inversion position,
Figure FDA0002620616830000016
to estimate the frequency, drEstimating a value of a code phase parameter for a code phase search process, i representing an i-th integration time, D representing a local pseudo-random code when affected by a data bit, ND representing a local pseudo-random code when not affected by a data bit, fd,kIndicating the search frequency, dcA chip offset representing the search;
s3, carrying out code phase processing on the received signal to obtain a code phase processing variable to be detected
Figure FDA0002620616830000017
Estimated bit data inversion position i obtained from S20To, for
Figure FDA0002620616830000018
And performing coherent integration detection to finally obtain a phase of the parameter code to be solved and finally obtain the estimated phase and frequency of the captured parameter code, thereby realizing the rapid capture of the GNSS signal.
2. The method as claimed in claim 1, wherein the received signal r (n) is:
Figure FDA0002620616830000019
where q is the code phase of the received signal, fdIs Doppler frequency, C () is pseudo-random code, D () is modulated data, fsFor the sampling frequency, p (n) denotes that both the real and imaginary parts obey a mean of 0 and a variance of σ2N denotes a sample point, n is 0,1, …,
the received signal R (n) is processed by down-conversion, and the down-conversion signal is obtained as follows:
Figure FDA0002620616830000021
wherein, the down-conversion signal f (n) is:
Figure FDA0002620616830000022
wherein f isd,kTo search for frequency, fsFor the sampling frequency, n represents the sampling point,
the two local correlation signals are:
LD(n)＝(-1)iC(n)；
LND(n)＝C(n)；
where n represents the number of sample points, n can be written as iT + k0T is the number of sampling points of PN code period, k0T-1, C () is a pseudo random code, i represents the ith integration time, LD(n) and LND(n) respectively representing a local pseudo-random code when influenced by data bits and a local pseudo-random code when not influenced by the data bits; partitioning the local correlation signal and the down-converted signal:
Figure FDA0002620616830000023
Figure FDA0002620616830000024
Figure FDA0002620616830000031
wherein N isL＝2(i-1)MN+dc，i＝1,...,Nc，NcFor coherent integration time, NRThe M and N denote partition matrix parameters, and i denotes the i-th integration time.
3. A fast anti-bit-reversal fast acquisition method based on complex signal phase as claimed in claim 1, characterized in that:
the code phase search processing is performed on the matrix obtained in S1, and the specific method is as follows:
Figure FDA0002620616830000032
wherein M, N represents the partition matrix parameter, i represents the ith integration time, D represents the local pseudo random code when there is data bit influence, ND represents the local pseudo random code when there is no data bit influence, fd,kIndicating the search frequency, dcIndicates the chip offset of the search when dc-1<0，dc-1＝dc-1+ M; if d isc+1＞M，dc+1＝dc+1-M, detected by coherent integration:
Figure FDA0002620616830000033
Figure FDA0002620616830000034
Figure FDA0002620616830000035
wherein i1Representing possible flip positions of the symbols of the data bits, D representing the local pseudo-random code when affected by a data bit, ND representing the local pseudo-random code when unaffected by a data bit, fd,kIndicating the search frequency, dcChip offset indicating search, drCode offset, i, representing the code phase search process prediction0Indicating the predicted data bit flip position, fd,k0Is the estimated signal frequency.
4. A fast anti-bit-reversal fast acquisition method based on complex signal phase as claimed in claim 1, characterized in that:
the code phase estimation processing for performing code phase processing on the received signal specifically includes:
Figure FDA0002620616830000041
Figure FDA0002620616830000042
wherein, mod ((c)2+n),2N2) Is represented by (c)2+ N) divided by 2N2The remainder of (1); d denotes the local pseudo-random code when affected by a data bit, ND denotes the local pseudo-random code when not affected by a data bit,
Figure FDA0002620616830000043
indicating the ith without bit flipping in code phase estimation for code phase processingNSub-integration time chip offset d2The time of reception of the signal integral value,
Figure FDA0002620616830000044
indicating the ith bit flip in the code phase estimation of the code phase processingNSub-integration time chip offset d2Integral value of time-of-reception signal, iNIndicating ith without bit flippingNTime of sub-integration, iD＝i0+1 denotes the ith bit flipDIntegration time, M, N denotes the blocking matrix parameter, i0Indicating the predicted data bit flip position, N ═ M2N2，M2、N2Representing the code phase prediction parameters, and,
and then carrying out coherent integration detection:
Figure FDA0002620616830000045
Figure FDA0002620616830000051
Figure FDA0002620616830000052
wherein d isr2A code phase prediction parameter representing a code phase process,
Figure FDA0002620616830000053
detection amount after coherent integration in code phase prediction of code phase processing, D represents local pseudo-random code when there is influence of data bit, ND represents local pseudo-random code when there is no influence of data bit, i0Indicating the predicted data bit flip position, d2The chip offset at the time of code phase prediction representing code phase processing can be obtained from the PN code correlation propertyWhen d is2＝dr2The maximum value is obtained:
Figure FDA0002620616830000054
wherein, cr2、dr2Code phase prediction parameters representing code phase processing, M representing block matrix parameters, M2、N2Representing code phase prediction parameters, let X2(dr2) Phase angle of
Figure FDA0002620616830000056
Then
Figure FDA0002620616830000055
Wherein, cr2Code phase prediction parameter, N, representing code phase processing2Representing the code phase prediction parameters. Finally, the received estimated code phase may be expressed as
q＝(cr2M2+dr2)M+dr；
Wherein, cr2、dr2Code phase prediction parameters representing the code phase processing, drCode phase prediction parameters representing a code phase search process, M represents code phase prediction blocking matrix parameters of a code phase search process, M2A code phase prediction blocking matrix parameter representing a code phase process.
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