CN109067193B - Cascade power electronic transformer and unbalance compensation control method thereof - Google Patents

Cascade power electronic transformer and unbalance compensation control method thereof Download PDF

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CN109067193B
CN109067193B CN201810939314.6A CN201810939314A CN109067193B CN 109067193 B CN109067193 B CN 109067193B CN 201810939314 A CN201810939314 A CN 201810939314A CN 109067193 B CN109067193 B CN 109067193B
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CN109067193A (en
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曲正伟
姚云枭
王云静
郭蕊
孟令楠
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Nanjing Yanzhan Technology Co.,Ltd.
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Yanshan University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
    • H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
    • H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M5/4585—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only having a rectifier with controlled elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/26—Arrangements for eliminating or reducing asymmetry in polyphase networks
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2203/00—Indexing scheme relating to details of circuit arrangements for AC mains or AC distribution networks
    • H02J2203/20—Simulating, e g planning, reliability check, modelling or computer assisted design [CAD]
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/50—Arrangements for eliminating or reducing asymmetry in polyphase networks

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Abstract

The invention provides an unbalance compensation control method of a cascade power electronic transformer, which comprises a control component and a transformer, wherein the transformer consists of an input rectification stage, an isolation stage and an output inversion stage. The input stage is a three-phase star-connected cascade H-bridge rectifier, the isolation stage is a plurality of independent double-active-bridge converters, and the output stage is a single-phase PWM inverter. The control method comprises an input rectification level hierarchical control unit, an isolation level voltage closed-loop control unit and an output inverter level constant voltage and current sharing control unit, wherein the input level hierarchical control unit is divided into an upper layer control unit and a lower layer control unit, the upper layer control unit consists of coordinate transformation, positive sequence decoupling control and negative sequence current suppression, and the lower layer control unit is in-phase voltage sharing control. By adopting the control method provided by the invention, the problem of negative sequence current compensation of the PET when the grid side voltage and the three-phase load are unbalanced can be solved, so that the application of the power electronic transformer in the engineering practice is promoted.

Description

Cascade power electronic transformer and unbalance compensation control method thereof
Technical Field
The invention belongs to the technical field of power electronic transformers, and particularly relates to an unbalance compensation control method of a cascade power electronic transformer.
Background
A Power Electronic Transformer (PET), also called Solid State Transformer (SST), is a Power transformation device that replaces a traditional Power frequency Transformer in a Power Electronic Power system, and has functions of reactive compensation, harmonic suppression, Power bidirectional transmission, and multi-output Power supply in addition to realizing traditional voltage transformation and isolation.
In recent years, with the continuous development of PET technology, the PET technology is more and more focused on application to medium and high voltage power distribution networks. In high voltage applications, because the voltage withstanding level of the power switching device is limited, a modular or cascade structure is often adopted to increase the input voltage that the power electronic transformer can withstand. Two topologies which are widely applied are a single-phase PET topology based on a cascaded H-bridge rectifier adopted in Future Renewable Electric Energy transmission and Management project (fresh) in the united states and a three-phase PET topology based on a Modular Multilevel Converter (MMC) developed by the chinese academy. The two topologies can well realize various functions of the PET under the working conditions of an ideal power grid and a three-phase symmetric load, but in an actual power distribution network, due to the input of system faults or large-scale single-phase loads, the three-phase asymmetric working conditions often occur to the grid side voltage and the output side load, if the unbalance cannot be compensated, the PET can inject negative sequence current into the power grid, so that the large power grid is polluted, and meanwhile, the voltage of each phase of high-voltage direct-current capacitor caused by the negative sequence current is unbalanced, so that a switching device of an input stage is damaged due to overvoltage. Therefore, the research on the compensation control strategy under the three-phase imbalance has very important significance for the popularization of PET in the engineering practice.
The compensation of the PET to the unbalanced load can be realized from two aspects of a topology and a control link, but a topology improvement method usually increases the complexity of a circuit, a matched control strategy is usually required, the cost is increased, and in comparison, the method for compensating from the control link is more practical. The conventional compensation method adopts a symmetric component method to realize the separation of positive and negative sequence components, and then respectively controls the positive and negative sequence components under respective rotating coordinate systems. The method needs to calculate the amplitude and the phase angle of the compensation voltage in real time according to the voltage fluctuation of the direct current side, so that the sine performance of the compensated zero-sequence voltage is poor, harmonic pollution is easy to inject to the network side, and the compensation range is limited.
Disclosure of Invention
In order to overcome the defects of the prior art, the invention provides an unbalance compensation control system and method of a cascade power electronic transformer, which can simultaneously solve the negative sequence current compensation problem of PET when the grid side voltage and the three-phase load are unbalanced, thereby promoting the application of the power electronic transformer in engineering practice.
In particular, the invention provides a cascaded power electronic transformer comprising a transformer and a control assembly, the transformer comprising an input rectification stage, an isolation stage and an output inverter stage,
the input rectification stage consists of three single-phase cascade H-bridge rectifiers which are connected in a star shape; the isolation stage is a plurality of independent double-active-bridge converters, and the output ends of the double-active-bridge converters are connected in parallel to form a low-voltage direct-current bus; the output inverter stage is a single-phase PWM inverter;
the input rectifier stage, the isolation stage and the output inverter stage are all controlled by a single-phase H-bridge converter, and switching devices in the single-phase H-bridge converter are all high-power IGBTs;
the control assembly comprises an input rectification level hierarchical control unit, an isolation level control unit and an output inversion level control unit, wherein the hierarchical control unit of the input level is divided into an upper layer control unit and a lower layer control unit, a reduced-order generalized integrator is embedded in the upper layer control unit to inhibit negative sequence current on the network side, and positive sequence decoupling control is adopted to realize decoupling of network side power control and constant total direct current voltage; the lower layer control unit of the input rectification stage is used for overcoming influences caused by circuit parameters and switching delay and maintaining voltage balance among all sub-modules of each phase, the isolation stage adopts a closed-loop PI controller to realize constant voltage of a low-voltage direct-current bus of each sub-module, and the output inversion stage adopts a voltage-current double closed-loop control method to maintain constant output voltage.
Preferably, the isolation-level dual-active-bridge converter includes a primary-side H-bridge converter, a high-frequency transformer, and a secondary-side H-bridge converter, and a Single Phase Shift Pulse Width Modulation unit is configured between the primary-side H-bridge converter and the secondary-side H-bridge converter, and the Single Phase Shift Pulse Width Modulation unit implements bidirectional flow of power in the isolation-level converter by using Single Phase Shift-Pulse Width Modulation (SPS-PWM).
Preferably, the controller of the output inverter stage is a quasi-PR controller.
Preferably, the three single-phase cascaded H-bridge rectifiers of the input rectification stage are star-connected and then connected to an ac grid or an ac load via inductive filtering.
Preferably, the output end of the input stage H-bridge rectifier is connected with the input end of the DAB primary side H-bridge converter; the output end of the DAB primary side H-bridge converter is connected with the primary side of the high-frequency transformer, and the secondary side of the high-frequency transformer is connected with the input end of the DAB secondary side H-bridge converter; the output end of the DAB secondary side H-bridge converter is connected with the input end of an output-stage single-phase PWM inverter; the output end of the output-stage single-phase PWM inverter is connected with a low-voltage alternating-current power grid or an alternating-current load through a filter.
Preferably, the present invention further provides an unbalance compensation control method for a cascaded power electronic transformer, which includes:
s1, dividing a hierarchical control unit of the input stage into an upper-layer control unit and a lower-layer control unit, wherein the upper-layer control unit consists of three parts, namely coordinate transformation, positive sequence decoupling control and negative sequence current suppression.
And (3) coordinate transformation: three-phase voltage e on network sideabcObtaining an active voltage component e after dq coordinate transformationdAnd a reactive voltage component eq(ii) a Three-phase current i on network sideabcObtaining an active current measurement value i after dq coordinate transformationsdAnd a reactive current measurement isqActive current measurement value isdAnd a reactive current measurement isqRespectively pass through a wave trap to obtain respective positive sequence component, active current isdHas a positive sequence component of
Figure BDA0001768650900000031
Reactive current isqHas a positive sequence component of
Figure BDA0001768650900000032
Positive sequence decoupling: high-voltage direct-current capacitor voltage average value U of all modules is collecteddc_aveAnd is compared with a reference value
Figure BDA0001768650900000033
Making difference, and obtaining reference value of active current through a PI regulator
Figure BDA0001768650900000037
While simultaneously converting the reference value of the reactive current
Figure BDA0001768650900000038
Setting the value to 0 for realizing the unit power factor correction of the network side and obtaining the active current reference value
Figure BDA0001768650900000039
And a reactive current reference value
Figure BDA00017686509000000310
Respectively obtaining an active voltage reference signal u through a decoupling controller after being differenced with respective positive sequence component measured valuessdAnd a reactive voltage reference signal usq；
Negative sequence current suppression: reference value of active current isd *With active current measurement isdObtaining a compensation signal u of active voltage through a reduced-order generalized integrator after difference makingdcom(ii) a Reference value of reactive current isq *With a reactive current measurement isqAfter difference is made, a compensation signal u of reactive voltage is obtained through a reduced-order generalized integratorqcom；
Compensating signal u of active voltagedcomSubtracting the reference signal usdAnd with the feed-forward voltage e of the network sidedCompensation signal u of additive reactive voltageqcomSubtracting a reference signal u derived from the output of the positive sequence decoupling controlsqAnd with the feed-forward voltage e of the network sideqAdding, and finally respectively carrying out dq inverse transformation to obtain modulation voltage signals of the input stage
Figure BDA0001768650900000034
S2, the lower layer control unit of the input rectification stage is in-phase voltage-sharing control, which specifically comprises the following steps: collecting direct current capacitance voltage U of each submodule in the x-th phase (x epsilon { a, b, c })dc_x_1,...,Udc_x_NAnd is connected with the mean voltage U of the x-th phase DC capacitordc_x_aveMaking difference, multiplying the difference by the sign function of active current after passing through a proportional controller, and finally obtaining a modulation voltage reference signal by upper layer control
Figure BDA0001768650900000035
Multiplying and adding to finally obtain the modulation voltage u of each submodulexm1,uxm1,...uxmN；
S3, the isolation stage adjusts the voltage of the low-voltage direct-current bus of each sub-module to be constant by adopting closed-loop control, bidirectional flow of power in the isolation stage converter is realized by adopting an SPS-PWM modulation technology, the control methods of N sub-modules are the same, and the voltage measured value U of the low-voltage direct-current capacitor of the xth sub-module (x is 1,2,., N) isdc_LxAnd a low voltage DC bus voltage reference value Udc_L *After difference making, obtaining the phase-shifting control angle of each DAB converter through a PI regulator
Figure BDA0001768650900000036
Respectively controlling the power flow of each DAB converter;
s4, output stage closed-loop control: in output stage control, the inverter stage outputs a measured value u of the voltagexFirst and the reference value u of the output voltagex *Making a difference, and dividing the difference by N to obtain an output current reference signal i of each submodule through a quasi-PR controllerx *The output current is used as the input of the current inner loop and is respectively compared with the output current measured value i of each submodulex_1,ix_2,...,ix_NMaking difference, and finally obtaining modulation voltage signal u of each submodule through a quasi-PR controllerx_1,ux_2,...,ux_N。
Preferably, the modulated voltage reference signal of the input stage rectifier under the αβ axis can be expressed as:
Figure BDA0001768650900000041
in the formula of UαSum of UβDenotes the reference signal of the modulated voltage in the axis αβ, Ed,EqThe active and reactive components of the grid electromotive force,
Figure BDA0001768650900000042
and
Figure BDA0001768650900000043
representing active and reactive voltages, omega, of the PI controller output1Representing the angular frequency of the grid, t representing time, I2d,I2qThe amplitudes of the negative sequence double frequency active and reactive components,
Figure BDA0001768650900000044
is the initial phase angle of the negative sequence component;
Uαsum of UβThe converter only contains fundamental positive sequence components and negative sequence components, wherein the positive sequence components are used for controlling the power transmitted by the system, and the negative sequence components are used for suppressing the negative sequence current on the grid side.
Preferably, the trap frequency in step S1 is 100 HZ.
Compared with the prior art, the invention has the following advantages:
1. the auxiliary control strategy based on the reduced-order generalized integrator disclosed by the invention can improve the capability of the cascade PET to cope with unbalanced working conditions, so that the PET can still realize high-quality power transmission when three phases are unbalanced on both the network side and the load side.
2. The compensation method based on the ROGI regulator can effectively restrain the network side negative sequence current under the unbalanced working condition, and has obvious advantages in response speed and compensation effect compared with the traditional sequence division compensation method based on the double dq shafting.
3. The unbalance compensation control method disclosed by the invention only plays a role when the three phases are unbalanced, and the normal operation of the PET under the condition of voltage and load balance cannot be influenced.
4. The unbalanced control scheme disclosed by the invention can ensure that the PET still maintains the high power quality of AC/DC output when the grid side voltage is seriously asymmetric, so that the PET has certain fault ride-through capability.
5. The unbalance compensation control strategy disclosed by the invention is suitable for application occasions of any voltage class and cascade PET with any module number, and can be expanded to three-phase four-wire system PET.
Drawings
Fig. 1 is a schematic diagram of a main circuit topology of a cascaded power electronic transformer adopted by the invention.
FIG. 2 is a schematic diagram of a conventional control scheme for imbalance compensation in a split sequence based on a double dq shafting system.
Fig. 3 is a schematic diagram of the upper-level control of the input stage based on the auxiliary control of the ROGI regulator.
Fig. 4 is a schematic diagram of the voltage-sharing control in the lower phase of the input rectification stage adopted by the invention.
Fig. 5 is a voltage closed loop control schematic of the isolation stage employed in the present invention.
Fig. 6 is a voltage-current double closed-loop control schematic diagram of an output-stage single-phase PWM inverter based on a quasi-PR controller adopted by the invention.
Fig. 7 is a simulated waveform diagram when three-phase asymmetry occurs in the grid-side voltage.
Fig. 8 is a waveform diagram of simulation when unbalance of three-phase load power occurs.
FIG. 9 is a net side current simulation waveform diagram when using a dual dq shafting sequential compensation controller.
Fig. 10 is a graph of net side current simulation waveforms when compensated using the imbalance control scheme of the present invention.
Fig. 11 is a graph of net side a-phase voltage and current waveforms compensated using the control scheme of the present invention.
Fig. 12 is a simulated waveform diagram of the grid side voltage with severe three-phase asymmetry.
Fig. 13 is a simulated waveform diagram of the net side current that is compensated for when the net side voltage is severely three-phase asymmetric using the imbalance control scheme of the present invention.
Detailed Description
Exemplary embodiments, features and aspects of the present invention will be described in detail below with reference to the accompanying drawings. In the drawings, like reference numbers can indicate functionally identical or similar elements. While the various aspects of the embodiments are presented in drawings, the drawings are not necessarily drawn to scale unless specifically indicated.
The invention discloses a cascade PET unbalance compensation strategy based on reduced-order generalized integrator auxiliary control, which is characterized in that an ROGI regulator is embedded in a decoupling control link of a forward dq shafting to uniformly control positive and negative sequence currents, and the control of an input stage is obviously simplified because instruction current calculation and positive and negative sequence separation links are not required.
A typical PET topology is shown in fig. 1 and includes three parts, rectification, isolation and inversion. The rectification stage consists of three star-connected single-phase cascade H-bridge rectifiers, and the switching control is carried out by adopting a carrier phase shift modulation (CPS-PWM) technology. The isolation level is a plurality of independent Double Active Bridge (DAB) converters; the output stage is a single-phase PWM inverter. The three single-phase rectifier stages are connected in star connection and then connected with a public power grid through inductive filtering. The output ends of DABs can be connected in parallel to form a low-voltage direct-current bus for connecting a distributed power supply, an energy storage device and a direct-current load. The output end of the inverter can be connected with a low-voltage alternating current power grid or an alternating current load.
When the three-phase load at the alternating current output end is unbalanced, the currents of the transmission power of each phase are different, so that the three-phase current at the network side has negative sequence components, and the pollution to a power grid is caused. Some documents adopt a double dq shafting lower imbalance control strategy based on positive and negative sequence separation as shown in fig. 2, and although negative sequence current on a network side can be effectively suppressed, a control system is complex, compensation delay caused by positive and negative sequence separation cannot be avoided, and dynamic performance of a controller is poor.
The control strategy provided by the patent is to suppress the negative sequence current of the network side by embedding an ROGI regulator in a decoupling control link, so that the control strategy not only can adapt to the working condition that the voltage and the load of the network side are asymmetric, but also can obtain the AC/DC output with high electric energy quality, and is a simple and practical compensation method. The imbalance compensation strategy herein is analyzed below in conjunction with a mathematical model of the PET input stage.
Referring to fig. 1, a typical cascaded PET input stage topology, each phase is composed of N cascaded H-bridge rectifier modules, three-phase inputs are connected in a star shape, and no neutral line exists between a network side power supply and an output end load, so that a zero sequence current is not included in the system.
When an imbalance occurs between the grid side and the load side, the supply voltage on the grid side can be expressed as:
Figure BDA0001768650900000061
in the formula, ea,eb,ecInstantaneous electromotive force of the net side; ep,En,EzThe amplitudes of the positive, negative and zero sequence components of the network side voltage are obtained;
Figure BDA0001768650900000062
the phase angle of the positive and negative zero sequence components.
If the system imbalance is not controlled, the net side input current will have a negative sequence component, which can be expressed as:
Figure BDA0001768650900000063
in the formula, LacA filter inductor on the network side; ra,Rb,RcThe equivalent series resistance of the rectifying output end is used for representing the switching loss of the system, and the power consumption of the system is usually negligible compared with the power consumed by the alternating current load of the output stage.
Dq conversion is carried out on the formula (3), and an expression of the current at the input end of the three-phase bridge arm in a dq shaft system can be obtained
Figure BDA0001768650900000064
Through the decoupling control, the active and reactive current controllers can be represented as:
Figure BDA0001768650900000065
Figure BDA0001768650900000066
in the formula ia,ib,icThree-phase instantaneous current at the side of the grid; i isp,InThe amplitude of the positive and negative sequence current at the network side; thetap,θnIs the initial phase of positive and negative sequence current.
The input current of the cascaded H-bridge rectifier can be expressed as:
when the system is unbalanced, the negative sequence component of the network side current has 2 frequency multiplication components under dq shafting, and the active and reactive current at this time can be expressed as:
Figure BDA0001768650900000067
in the formula i1d,i1qIs the direct component of the active and reactive currents, I2d,I2qThe amplitude of the negative sequence double frequency component,
Figure BDA0001768650900000068
is the initial phase angle of the negative sequence component.
The PI controller can not be used for carrying out no-difference regulation on the 2-time-multiplied negative sequence component, so that an ROGI regulator is embedded on the basis of positive sequence decoupling control and is used for suppressing the negative sequence current on the network side. The hierarchical control of the input stage comprises an upper-layer control and a lower-layer control, and the upper-layer control block diagram is shown in FIG. 3 and comprises three parts of coordinate transformation, positive sequence decoupling and negative sequence current suppression.
The purpose of coordinate transformation is to convert voltage and current to dq axis system, and facilitate the respective real and reactive current on network sideAnd (5) controlling. As shown in FIG. 3, eabcAnd iabcRespectively representing three-phase voltage and current at the network side, and obtaining respective active and reactive components (e) after dq coordinate transformationsd,esq),(isd,isq)。isdAnd isqAnd also respectively pass through a 100Hz wave trap to obtain respective positive sequence components
Figure BDA0001768650900000071
And the positive sequence decoupling link is used for realizing the decoupling of the active and reactive current controllers. As shown in figure 3 of the drawings,
Figure BDA0001768650900000072
a reference value representing the voltage of the high voltage dc capacitor,
Figure BDA0001768650900000073
the average value of the high-voltage direct-current capacitor voltage of all the modules is represented, and the difference between the average value and the average value is passed through a reference value of active current of a PI (proportional-integral) controller
Figure BDA0001768650900000075
Reference value of reactive current
Figure BDA0001768650900000076
Set to 0 for net-side unity power factor correction. i.e. isd *And isq *Respectively and the respective positive sequence component measurements
Figure BDA0001768650900000074
Obtaining active and reactive voltage reference signals u through decoupling control after difference makingsdAnd usq. Here, s is an abbreviation of source, which refers to a power source, namely a power grid, so that the current with the upper s and the lower mark refers to the current on the power grid side, and similarly, L is an abbreviation of load, and the current with the upper L and the lower mark refers to the current on the load side.
The negative sequence current suppression link comprises an active part and a reactive part, the active part and the reactive part are controlled completely, taking the active current as an example, and a reference value id *Measured value idObtaining a compensation signal u of active voltage through an ROGI regulator after difference makingdcom。
Compensation term u for active and reactive voltagedcomAnd uqcomSubtracting the reference signal u separatelysdAnd usqRe-summing the network-side voltage edAnd eqAnd adding, and finally carrying out dq inverse transformation to obtain a modulation voltage signal of the input stage. Compared with the double dq shafting-based sequence compensation method shown in the figure 2, the method has the advantage that the complexity of the control unit is greatly simplified because the positive and negative sequence separation links are not needed.
The lower-layer control of the input stage is used for overcoming the influence caused by circuit parameters and switching delay and maintaining the voltage balance among the sub-modules of each phase, and a control block diagram is shown in fig. 4. Collecting direct current capacitance voltage U of each submodule in the x-th phase (x epsilon { a, b, c })dc_x_1,...,Udc_x_NAnd is connected with the mean voltage U of the x-th phase DC capacitordc_x_aveMaking difference, multiplying the difference by the sign function of active current after passing through a proportional controller, and finally obtaining a modulation voltage reference signal by master control
Figure BDA0001768650900000077
Multiplying and adding to finally obtain the modulation voltage u of each submodulexm1,uxm1,...uxmN。
The isolation-level DAB converter is used for power exchange of the input stage and the output stage. Fig. 5 is a schematic diagram of the closed-loop control of the voltage of the isolation stage according to the present invention. And the voltage of a low-voltage direct-current bus of each submodule is regulated to be constant by adopting closed-loop PI control, and the power bidirectional flow in the isolation-stage converter is realized by adopting an SPS-PWM modulation technology. In the figure, Udc_L *Is a reference signal, U, of a low voltage DC busdc_LxIs the low voltage dc capacitor voltage measurement for the xth sub-module, where x is 1, 2. Measured value Udc_LxAnd a reference value Udc_L *Obtaining the phase-shift control angle of each DAB converter after PI control
Figure BDA0001768650900000081
The power flow of the respective DAB is controlled separately.
FIG. 6 is a schematic diagram of voltage-current dual closed-loop control of an output-stage single-phase PWM inverter based on a quasi-PR controller according to the present invention, in which ux *And uxReference and measured values, i, respectively, of the output voltage of the inverter stagex *Reference value, i, representing the output current of each submodulex_1,ix_2,...,ix_NRespectively representing the output current measurements, u, of N submodules of each phasex_1,ux_2,...,ux_NIs the modulated voltage signal of the N parallel inverters. The control methods of the three phases are completely the same, taking a U phase as an example, in the control of an output stage, firstly, a measured value of output voltage of an inverter stage is differenced with a reference value, then, the measured value is divided by N after passing through a quasi-PR controller to obtain an output current reference signal of each sub-module, the output current reference signal is used as the input of a current inner ring, then, the output current reference signal is differenced with the measured value of the output current of each sub-module respectively, and finally, a modulation voltage signal of each sub-module is. The quasi-PR controller is adopted instead of the PI controller, the quasi-PR controller has better effect of tracking an alternating current signal without static error, and the voltage and current double-closed-loop control method can ensure that the output voltage is constant and realize the current sharing of each parallel sub-module.
The principle of the cascade-type PET imbalance compensation control method disclosed in the present invention is analyzed below in conjunction with the transfer function of the ROGI modulator.
The transfer functions of the ROGI modulators are respectively shown in the formula (7)
Figure BDA0001768650900000082
The transfer function of the ROGI is only-j omegarOne pole, therefore when ωrAt 2 pi x 100rad/s, the controller only has the maximum gain at-100 Hz and almost 0 at other frequencies, so the ROGI can adjust the negative sequence component without affecting the positive sequence current component.
The output of the negative-sequence current controller based on the ROGI can be expressed as
Figure BDA0001768650900000083
In the formula
Figure BDA0001768650900000084
And
Figure BDA0001768650900000085
respectively representing the active and reactive voltage compensation quantities output by the controller,
Figure BDA0001768650900000086
representing an inverse Laplace transform and a convolution operation.
As time t increases, the exponential term of the second term in equation (8) increases rapidly, with values far exceeding the first term. Equation (8) can therefore be approximately equivalent to:
Figure BDA0001768650900000087
through positive sequence decoupling control and negative sequence current control, the reference voltage signal of the input stage rectifier under the dq shafting can be expressed as:
Figure BDA0001768650900000091
by performing dq inverse transformation on the formula (10), an expression of the reference voltage signal in the axis αβ is obtained as follows:
Figure BDA0001768650900000092
in the formula of UαSum of UβDenotes the modulated voltage reference signal at axis αβ.
From equation (11), it can be seen that the modulated signal of the input stage cascade H-bridge rectifier only contains fundamental positive sequence and negative sequence components, where the positive sequence component is used for controlling the power transmitted by the system, and the negative sequence component is used for suppressing the negative sequence current on the network side.
In actual operation, the allowable fluctuation range of the power grid frequency is-2.5- +1.5Hz, if the ROGI regulator is directly adopted, the robustness of the controller is poor when the power grid frequency fluctuates, and therefore the cut-off frequency omega needs to be introducedcTo improve the gain bandwidth range of the controller. The ROGI regulator at this time becomes a Reduced order quasi-resonant regulator (ROQR), and the transfer function is as shown in equation (12).
Figure BDA0001768650900000093
In order to verify the invention, a three-phase cascade PET simulation model with the same topology as that of the graph 1 is built on the basis of Matlab/Simulink. The main simulation parameters are as follows:
rated capacity: 2MVA
Grid side line voltage: 10kV (effective value)
Number of cascade modules: 3
Three-phase rated load power: 360kW
The simulation process is as follows: before 0.45s, the voltage on the network side is three-phase symmetrical, and the three-phase load is 360 kW. After 0.45s, the voltage on the network side is slightly asymmetric, the load of the U phase is kept unchanged, the load power of the V phase is reduced by 20%, and the load power of the W phase is changed to 50% of the original load power. And the sequence division compensation controller under the double dq shafting and the unbalance compensation scheme provided by the invention are respectively put into the system at 0.45s, so that the negative sequence current compensation effect is compared. Fig. 7-13 show simulation results of the present invention.
Fig. 7 is a simulation waveform diagram of the grid-side voltage variation of the present invention, wherein the grid-side voltage is three-phase symmetric before 0.45s, and slight asymmetry appears after 0.45 s.
Fig. 8 is a waveform diagram of the three-phase load power variation simulation of the present invention. Before 0.45s, the three-phase load power is 1:1:1, and after 0.45s, the load ratio is changed to 1:0.8: 0.5.
Fig. 9 and 10 show the compensation effect of the double dq shafting based sequential compensation control and the imbalance control scheme of the present invention, respectively. It can be seen from the waveform comparison that when the compensation scheme of the present invention is adopted, the current on the grid side is restored to balance after about 0.04 seconds, whereas when the sequential compensation controller with the dual dq shafting is adopted, the response time is long after 0.15 seconds, and the current on the grid side after compensation still has slight asymmetry.
The reasons for the above results are mainly that when three-phase imbalance is compensated by adopting sequence division, detection delay of a quarter period caused by positive and negative sequence separation cannot be avoided, and excessive coordinate transformation and complex instruction calculation in the control scheme influence compensation precision. By adopting the unbalance control scheme, the links of instruction current calculation and positive-negative sequence separation are avoided, and the response speed of the unbalance compensation controller is greatly improved.
Referring to fig. 11, it can be seen that the compensated PET operates in a unit power factor state and the output power quality is high.
As shown in fig. 12, which is a simulation waveform diagram when the grid-side voltage is severely asymmetric, the grid-side voltage is three-phase symmetric before 0.45s, and the a-phase voltage drops to 70% of the original voltage after 0.45 s.
Fig. 13 is a simulated waveform diagram of the grid-side current for compensation when the grid-side voltage is severely asymmetric with three phases, according to the unbalanced control scheme of the present invention. Simulation results show that when the voltage of a power grid drops seriously, the control method can still enable the current at the grid side to keep three-phase symmetry, which shows that the three-phase PET which is controlled by the ROGI regulator in an auxiliary mode not only can compensate unbalanced load, but also has certain fault ride-through capability.
Finally, it should be noted that: the above-mentioned embodiments are only used for illustrating the technical solution of the present invention, and not for limiting the same; although the present invention has been described in detail with reference to the foregoing embodiments, it will be understood by those of ordinary skill in the art that: the technical solutions described in the foregoing embodiments may still be modified, or some or all of the technical features may be equivalently replaced; and the modifications or the substitutions do not make the essence of the corresponding technical solutions depart from the scope of the technical solutions of the embodiments of the present invention.

Claims (7)

1. An unbalance compensation control method of a cascade power electronic transformer is characterized by comprising a transformer and a control component, wherein the transformer comprises an input rectifying stage, an isolation stage and an output inversion stage,
the input rectification stage consists of three single-phase cascade H-bridge rectifiers which are connected in a star shape; the isolation stage is a plurality of independent double-active-bridge converters, and the output ends of the double-active-bridge converters are connected in parallel to form a low-voltage direct-current bus; the output inverter stage is a single-phase PWM inverter;
the input rectifier stage, the isolation stage and the output inverter stage are all controlled by a single-phase H-bridge converter, and switching devices in the single-phase H-bridge converter are all high-power IGBTs;
the control assembly comprises an input rectification level hierarchical control unit, an isolation level control unit and an output inversion level control unit, wherein the hierarchical control unit of the input level is divided into an upper layer control unit and a lower layer control unit, a reduced-order generalized integrator is embedded in the upper layer control unit to inhibit negative sequence current on the network side, and positive sequence decoupling control is adopted to realize decoupling of network side power control and constant total direct current voltage; the lower layer control unit of the input rectification stage is used for overcoming the influence caused by circuit parameters and switching delay and maintaining the voltage balance among all the submodules of each phase, the isolation stage adopts a closed-loop PI controller to realize the constant voltage of the low-voltage direct-current bus of each submodule, and the output inversion stage adopts a voltage-current double closed-loop control method to maintain the constant output voltage;
the unbalance compensation control method comprises the following steps:
s1, dividing a hierarchical control unit of the input stage into an upper-layer control unit and a lower-layer control unit, wherein the upper-layer control unit consists of three parts, namely coordinate transformation, positive sequence decoupling control and negative sequence current suppression;
and (3) coordinate transformation: three-phase voltage e on network sideabcObtaining an active voltage component e after dq coordinate transformationdAnd a reactive voltage component eq(ii) a Three-phase current i on network sideabcObtaining an active current measurement value i after dq coordinate transformationsdAnd a reactive current measurement isqActive current measurement value isdAnd a reactive current measurement isqRespectively pass through a wave trap to obtain respective positive sequence component, active current isdHas a positive sequence component of
Figure FDA0002321182800000011
Reactive current isqHas a positive sequence component of
Figure FDA0002321182800000012
Positive sequence decoupling: high-voltage direct-current capacitor voltage average value U of all modules is collecteddc_aveAnd is compared with a reference value
Figure FDA0002321182800000013
Making difference, and obtaining reference value i of active current through a PI regulatorsd p*While simultaneously applying a reference value i of the reactive currentsq p*Setting the value to 0 for realizing the unit power factor correction of the network side and obtaining the active current reference value isd p*And a reactive current reference value isq p*Respectively obtaining an active voltage reference signal u through a decoupling controller after being differenced with respective positive sequence component measured valuessdAnd a reactive voltage reference signal usq；
Negative sequence current suppression: reference value of active current isd *With active current measurement isdObtaining a compensation signal u of active voltage through a reduced-order generalized integrator after difference makingdcom(ii) a Reference value of reactive current isq *With a reactive current measurement isqAfter difference is made, a compensation signal u of reactive voltage is obtained through a reduced-order generalized integratorqcom；
Compensating signal u of active voltagedcomSubtracting the reference signal usdAnd with the feed-forward voltage e of the network sidedCompensation signal u of additive reactive voltageqcomSubtracting a reference signal u derived from the output of the positive sequence decoupling controlsqAnd the feed forward voltage of the network sideeqAdding, and finally respectively carrying out dq inverse transformation to obtain modulation voltage signals of the input stage
Figure FDA0002321182800000021
S2, the lower layer control unit of the input rectification stage is in-phase voltage-sharing control, which specifically comprises the following steps: collecting direct current capacitance voltage U of each submodule in the x-th phase and x epsilon { a, b, c }dc_x_1，...，Udc_x_NAnd is connected with the mean voltage U of the x-th phase DC capacitordc_x_aveMaking difference, multiplying the difference by the sign function of active current after passing through a proportional controller, and finally obtaining a modulation voltage reference signal by upper layer control
Figure FDA0002321182800000022
Multiplying and adding to finally obtain the modulation voltage u of each submodulexm1，uxm1，...uxmN；
S3, the isolation stage adjusts the voltage of the low-voltage direct-current bus of each sub-module to be constant by adopting closed-loop control, and realizes the bidirectional flow of power in the isolation stage converter by adopting an SPS-PWM modulation technology, the control methods of N sub-modules are the same, and the x sub-module is equal to 1,2dc_LxAnd a low voltage DC bus voltage reference value Udc_L *After difference making, obtaining the phase-shifting control angle of each DAB converter through a PI regulator
Figure FDA0002321182800000023
Respectively controlling the power flow of each DAB converter;
s4, output stage closed-loop control: in output stage control, the inverter stage outputs a measured value u of the voltagexFirst and the reference value u of the output voltagex *Making a difference, and dividing the difference by N to obtain an output current reference signal i of each submodule through a quasi-PR controllerx *The output current is used as the input of the current inner loop and is respectively compared with the output current measured value i of each submodulex_1，ix_2，...，ix_NMake a difference, finally pass throughObtaining the modulation voltage signal u of each sub-module by a quasi-PR controllerx_1，ux_2，...，ux_N。
2. The unbalance compensation control method for a cascade-type power electronic transformer according to claim 1, characterized in that: the isolation-level double-active-bridge converter comprises a primary side H-bridge converter, a high-frequency transformer and a secondary side H-bridge converter, wherein a single phase-shifting pulse width modulation unit is arranged between the primary side H-bridge converter and the secondary side H-bridge converter and used for realizing the bidirectional flow of power in the isolation-level converter.
3. The unbalance compensation control method for a cascade-type power electronic transformer according to claim 1, characterized in that: and the controller of the output inverter stage is a quasi-PR controller.
4. The unbalance compensation control method for a cascade-type power electronic transformer according to claim 1, characterized in that: and the three single-phase cascaded H-bridge rectifiers of the input rectifier stage are connected in a star shape and then connected with an alternating current power grid or an alternating current load through inductive filtering.
5. The unbalance compensation control method for a cascade-type power electronic transformer according to claim 1, characterized in that: the output end of the input-stage H-bridge rectifier is connected with the input end of the DAB primary-side H-bridge converter; the output end of the DAB primary side H-bridge converter is connected with the primary side of the high-frequency transformer, and the secondary side of the high-frequency transformer is connected with the input end of the DAB secondary side H-bridge converter; the output end of the DAB secondary side H-bridge converter is connected with the input end of an output-stage single-phase PWM inverter; the output end of the output-stage single-phase PWM inverter is connected with a low-voltage alternating-current power grid or an alternating-current load through a filter.
6. The unbalance compensation control method for the cascade power electronic transformer according to claim 1, wherein the reference signal of the modulated voltage of the input stage rectifier under the αβ axis is represented as:
Figure FDA0002321182800000031
in the formula of U* αAnd U* βIndicating the modulated voltage reference signal in the axis αβ, Ed，EqThe active and reactive components of the grid electromotive force,
Figure FDA0002321182800000032
and
Figure FDA0002321182800000033
representing active and reactive voltages, omega, of the PI controller output1Representing the angular frequency of the grid, t representing time, I2d，I2qThe amplitudes of the negative sequence double frequency active and reactive components,
Figure FDA0002321182800000034
is the initial phase angle of the negative sequence component;
U* αand U* βOnly contains a positive sequence component and a negative sequence component of a fundamental wave, wherein the positive sequence component is used for controlling the power transmitted by the system, and the negative sequence component is used for suppressing the negative sequence current on the network side.
7. The unbalance compensation control method for a cascaded power electronic transformer according to claim 6, wherein: the trap frequency in step S1 is 100 HZ.
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