CN109004836B - Frequency conversion optimization control method suitable for modular multilevel direct current transformer - Google Patents

Frequency conversion optimization control method suitable for modular multilevel direct current transformer Download PDF

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CN109004836B
CN109004836B CN201810744869.5A CN201810744869A CN109004836B CN 109004836 B CN109004836 B CN 109004836B CN 201810744869 A CN201810744869 A CN 201810744869A CN 109004836 B CN109004836 B CN 109004836B
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current transformer
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CN109004836A (en
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梅军
管州
丁然
葛锐
范光耀
何梦雪
吴夕纯
王冰冰
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Southeast University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer

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Abstract

The invention discloses a frequency conversion optimization control method suitable for a modular multilevel direct current transformer, which comprises the steps of setting an MMC side as a nearest level modulation strategy and setting an H bridge side as square wave modulation for an MMC-H bridge type direct current transformer consisting of an MMC bridge arm and an H bridge, and analyzing the running characteristic of the direct current transformer; deducing a transmission power and reflux power expression of the direct-current transformer based on a nearest level approximation modulation strategy according to the topology and the operating characteristics of the direct-current transformer; for the deduced power expression, analyzing the distribution condition of the power changing along with the parameters, summarizing factors influencing the backflow power, discussing the optimal configuration of the voltage regulation ratio, and condensing the boundary condition of the zero backflow power so as to obtain a zero backflow power area; according to the derived power expression, the working frequency is inversely proportional to the transmission power, the phase-shifting ratio is adjusted by the direct-current transformer through the change of the working frequency to fall into a zero reflux power area, and the inhibition of the reflux power of the direct-current transformer is realized. This method can reduce the reflow power and current stress.

Description

Frequency conversion optimization control method suitable for modular multilevel direct current transformer
Technical Field
The invention belongs to the field of a modular multilevel converter and direct current conversion, in particular relates to a modular multilevel MMC direct current transformer, and belongs to the field of modular multilevel converters and direct current conversion.
Background
Renewable energy power generation has the characteristic of random fluctuation, and the power supply reliability and stability of the renewable energy power generation are seriously affected when a large number of renewable energy power generation is connected into an alternating current power grid, and the renewable energy power generation has the problems of frequency fluctuation, reactive power compensation and the like. The direct-current power distribution network does not have the problems, and meanwhile, with the development of power electronic technology, the advantages of the direct-current power distribution network in the aspects of new energy power generation access, electric energy quality, transmission loss and the like are increasingly highlighted.
Unlike the ac power grid, the dc power grid must rely on power electronics to achieve voltage matching and energy interaction between the different power grids. At present, researchers at home and abroad have conducted many researches on direct-current transformer transformers for direct-current distribution networks.
The Dual Active Bridge (DAB) DC-DC converter topology reduces the size, reduces the number of components and improves the efficiency of the converter compared to a conventional unidirectional DC-DC converter, and bidirectional energy flow can be realized without changing the polarity of the primary side and the secondary side. However, the presence of the return power will significantly reduce the efficiency of the converter in case the voltage amplitudes on both sides do not match. Therefore, in addition to the conventional single phase shift, extended phase shift, double phase shift, triple phase shift and hybrid phase shift have been proposed to suppress the backflow power.
A plurality of DC-DC converters are connected in series and parallel to form a combined converter structure, so that electric isolation and voltage matching among different power grids are realized. The medium-voltage side adopts a series connection structure to solve the contradiction between low voltage resistance and high voltage of a power grid of the switching semiconductor device. Each DC-DC converter adopts a phase-shifting bridge structure or a resonant bridge structure to realize high-efficiency energy conversion and correct energy transfer. However, the combined converter uses a plurality of high frequency transformers, each of which has low transmission power and high isolation voltage requirement (usually much higher than the medium voltage side grid voltage), resulting in low core utilization and difficult insulation design.
The Modular Multilevel Converter (MMC) topology overcomes the problem of low voltage stress in the semiconductor device, and has been widely applied in the field of flexible high voltage direct current transmission. In view of this, some scholars propose MMC-DAB type DC-DC transformer topologies combining DAB and MMC. H bridges in DAB are replaced by MMC bridge arms, and original secondary sides are still connected by a transformer to form an MMC-DAB topology. And when only one H bridge is replaced in the DAB, namely the primary side adopts an MMC bridge arm, and the secondary side still adopts an H bridge topology to form an MMC-H bridge topology. The use of the MMC bridge arm can improve the voltage level and increase the transmission power. The MMC type DAB can also realize the stabilization of output voltage through phase shift control or direct voltage transformation control. However, the existing control strategy of the topology adopts simultaneous switching-on of a plurality of or all sub-modules, and the working frequency is high, so that large dv/dt is easily generated, and the stability of the circuit is influenced.
Traditional DAB adopts fixed working frequency control, and a frequency conversion control strategy is correspondingly proposed in consideration of the characteristics of transmission power and frequency.
Disclosure of Invention
The invention aims to provide a variable frequency optimization control method suitable for a modular multilevel direct current transformer, which can reduce backflow power and current stress.
In order to achieve the above purpose, the solution of the invention is:
a frequency conversion optimization control method suitable for a modular multilevel DC transformer comprises the following steps:
step 1, setting an MMC side as a nearest level modulation strategy and an H bridge side as square wave modulation for an MMC-H bridge type direct current transformer consisting of an MMC bridge arm and an H bridge, and analyzing the running characteristic of the direct current transformer;
step 2, deducing a transmission power and reflux power expression of the direct current transformer based on a recent level approximation modulation strategy according to the topological and operating characteristics of the MMC-H bridge type direct current transformer;
step 3, analyzing the distribution condition of the power along with parameter change for the deduced power expression, summarizing factors influencing the backflow power, discussing the optimal configuration of the voltage regulation ratio and condensing the boundary condition of the zero backflow power so as to obtain a zero backflow power area;
and 4, according to the deduced power expression, the working frequency is inversely proportional to the transmission power, and the direct-current transformer adjusts the phase-shifting ratio to fall into a zero reflux power area through the change of the working frequency, so that the inhibition of the reflux power of the direct-current transformer is realized.
After the scheme is adopted, the MMC side of the MMC-H bridge type direct current transformer formed by the MMC bridge arm and the H bridge is set as the latest level modulation strategy, the H bridge side is set as square wave modulation, the voltage grade can be improved by using the MMC bridge arm, and the transmission power can be increased. The MMC-H bridge type direct current transformer can realize the stabilization of output voltage through phase shift control or direct voltage transformation control.
In order to restrain the backflow power, the invention provides a frequency conversion control strategy, the phase shift ratio is realized to fall into a zero backflow power area through the change of the working frequency, and the backflow power and the current stress are reduced. And verifying the feasibility of a transmission power formula and a variable frequency optimization control strategy by using a simulation result. The invention has strong practicability, simple control and high reliability.
Drawings
FIG. 1 is a topological diagram of an MMC-H bridge type direct current transformer, wherein an MMC bridge arm is adopted on a primary side, a common H bridge is adopted on a secondary side, and the middles are connected by a medium-frequency voltage device;
FIG. 2 is a simplified schematic diagram of a DC transformer, including a simplified schematic diagram of an MMC bridge arm and a simplified power transmission model of the DC transformer;
the system comprises a direct-current transformer, a phase MMC bridge arm;
FIG. 3 is a waveform of the AC chain operation of the DC transformer;
FIG. 4 is a backflow power area distribution, dividing the backflow power into three areas;
wherein (a) is a return power region I, II distribution, and (b) is a return power region III distribution;
FIG. 5 is a graph of voltage regulation ratio p versus phase shift ratio D;
wherein, (a) is a curve of iteration results of p ═ f (D, p), and (b) is D and thetayA curve;
FIG. 6 is a power transmission curve of a DC transformer, including curves of transmission power and return power as a function of 1/p and D;
wherein, (a) is a change curve of transmission power along with 1/p and D, and (b) is a change curve of reflux power along with 1/p and D;
FIG. 7 shows the operating frequency fs、fsA/2 and fsA power transfer curve of 2;
FIG. 8 is a frequency conversion optimization control strategy, including a control strategy diagram and a control flow diagram;
wherein, (a) is a frequency conversion optimization strategy flow chart, and (b) is a control strategy chart;
FIG. 9 is a simulated waveform with a 500Hz operating frequency;
wherein, (a) is voltage current waveform when 400Hz working frequency, and (b) is reflux power and phase shift ratio when 400Hz working frequency;
FIG. 10 is a simulated waveform with a working frequency of 1 kHz;
wherein, (a) is voltage current waveform when the working frequency is 1kHz, and (b) is reflux power and phase shift ratio when the working frequency is 1 kHz.
Detailed Description
The technical solution and the advantages of the present invention will be described in detail with reference to the accompanying drawings.
The average power transmitted in a half cycle of the dc transformer is defined as transmission power P, and the transmission power is negative and defined as return power Q.
As shown in fig. 1, the primary side of the dc transformer constructed by the present invention uses an MMC bridge arm, and the secondary side still uses an H-bridge topology, thereby forming an MMC-H bridge topology. Considering that the MMC bridge arms in the direct-current transformer have the same structure and two phases/three phases are symmetrical, taking a phase a as an example for analysis, the p1 point and the p2 point can be regarded as equipotential, so that the two points are short-circuited in the phase a, and the analysis is simplified.
Similarly, when the secondary side is an H-bridge, the simplified power transfer model in fig. 2(a) can be obtained. The phase difference between the primary and secondary may enable power transfer.
According to the simplified circuit shown in fig. 2(a), neglecting the loss of the switching tube, the MMC-H bridge type dc transformer is converted to the primary side, which is equivalent to the primary and secondary sides and their connecting inductances, and herein, considering only the two-phase bridge arm condition, the primary and secondary output ac voltages act on the inductance L (L ═ L)p+LT) In the above, the transmission of power is realized, as shown in FIG. 2(b)A simplified power transfer model is shown.
The primary side adopts Nearest level approximation Modulation (N L M), and the inverse output voltage is set as up. The secondary side H bridge adopts square wave modulation to output square wave u with the same frequency as the primary sides. By controlling the phase difference between the primary and secondary, the magnitude and direction of the power can be controlled. The analysis is here carried out taking as an example the transmission of power from the primary side to the secondary side, i.e. upPhase lead usAs shown in fig. 3.
In FIG. 3, U1、U2Is the primary and secondary voltage peak value, wherein U1＝Udc1Defined as the primary side modulation ratio, U2＝Udc2Will U is2Converted to primary side, nU2(ii) a n is transformer transformation ratio, and defines voltage regulation ratio p ═ U1/(nU2)＝Udc1/(nUdc2) (ii) a An inductive current of iL(ii) a T is half of the working period and the switching frequency f s1/(2T); d is defined as the phase shift ratio when DT is the phase shift time in a half period, and D is more than or equal to 0 and less than or equal to 1; assuming that the primary side inverted output voltage is 2N +1 level (then N is the number of steps in 1/4 cycles), according to the definition of the nearest level approximation modulation, we can obtain:
Figure BDA0001724092980000041
for the sake of convenience of discussion, it is assumed that 1 is the same as
Figure BDA0001724092980000042
Wherein: thetaN+k＝θN+1-kk＝1,…,N
Then there are:
Figure BDA0001724092980000043
the MMC side inverter voltage can be represented as according to FIG. 3
Figure BDA0001724092980000044
Let t be the steady state of the DC transformerx≤DT≤tx+1(x is 0, …, N), in which case up＝(x-1)U1and/N, according to the figure 3, the working mode of the direct current transformer can be divided into 2N +1 states, and the inductance current equation can be listed under different states:
Figure BDA0001724092980000045
from the symmetry, i is knownL(t2N+1)＝-iL(t0) The current value of each state is solved by combining the current expression.
At tx<tDWhen, the inductive current is:
Figure BDA0001724092980000051
tdthe inductor current is:
Figure BDA0001724092980000052
at tx>tDWhen, the inductive current is:
Figure BDA0001724092980000053
according to the definition of the transmission power P, where tx≤DT≤tx+1(x ═ 0, …, N), from which:
Figure BDA0001724092980000054
namely, the transmission power expression is:
Figure BDA0001724092980000055
the above derivation is based on tx≤DT≤tx+1(x ═ 0, …, N) when t isN+x≤DT≤tN+x+1(x is 0, …, N), the same can be said:
Figure BDA0001724092980000056
for convenient analysis, per-unit the transmission power is taken and per-unit power P is takenN＝nUdc1Udc2/(8fsL), a per-unit expression of the transmission power P can be obtained:
Figure BDA0001724092980000057
region 1 for analyzing existence of reflow power, when U1>nU2A return power region I may be generated while assuming that the inductor current of this region is at tyThe moment reaches the minimum value iL(ty). 2. If the voltage is still greater than zero before the inductor current rises to zero, there may be a power return region II. 3. There may be a reflow power region III. The distribution of the regions is shown in fig. 4.
If the backflow power of the area I is restrained, the following requirements are met:
Figure BDA0001724092980000058
and (y-1)/N is less than 1/p and less than or equal to y/N
Order to
Figure BDA0001724092980000059
The voltage regulation ratio p threshold is then found according to the equation p ═ f (D, p). And f (D, p) is a nonlinear function, and an iterative method is directly adopted for solving. Different results are iterated for different values D, and taking N as an example 10, a curve of the phase shift ratio D and the voltage regulation ratio p shown in fig. 5(a) can be obtained.
In fact, analyzing the condition of the reflow region I, the reflow region I is determined to be D ≦ θyThere are, therefore, on the basis of fig. 5(a), a curve D ═ g of D with respect to 1/p is drawn1(1/p) according to thetayMaking theta in relation to 1/py＝g2(1/p) curve, as shown in FIG. 5 (b).
Combining the curves (a) and (b), it can be seen that when N is 10, D is not less than 0.18537, D is not less than θyThe constant true, the backflow zone I does not exist; when D is present<0.18537, if p is less than f (D, p), i.e. p is less than curve (a), D is more than or equal to thetayIn this case, the return power range I is zero.
From the curve (a), the minimum value p of the voltage regulation ratio p can be obtainedminTherefore, if p ≦ 1 is set in the design of the dc transformer, the backflow power region I is directly suppressed.
In region II, it is assumed that the inductor current is present for a time Tx’The moment rises to 0 for the first time and t is satisfiedd1≤tx’(＝θx'T/π)≤td1+1. In region III, it is assumed that the inductor current is present for a time Tx’The time drops to 0 for the first time and t is satisfiedd2≤tx’(＝θx'T/π)≤td2+1D1 and d2 are positive integers respectively, d1 is less than N, d2<N。
The solving process of the reflux power is omitted, the reflux power is subjected to per unit, and per unit power Q is obtainedN＝nUdc1Udc2/(8fsL), obtaining a return power Q per unit expression:
Figure BDA0001724092980000061
wherein:
Figure BDA0001724092980000062
Figure BDA0001724092980000063
Figure BDA0001724092980000064
Figure BDA0001724092980000065
Figure BDA0001724092980000066
Figure BDA0001724092980000067
Figure BDA0001724092980000068
region of zero reflow power
Figure BDA0001724092980000069
P*、Q*A graph of the variation of the step ratio D and the voltage regulation ratio p is shown in fig. 6. Wherein D is more than or equal to 0 and less than or equal to 1, and 1/p is more than or equal to 1 and less than or equal to 2.
The rated working frequency of the known direct current transformer is fsThe above analysis is all established at a fixed frequency fsUnder control, its power reference value PN、QNAlso fixed, in fact transmitting power and operating frequency fsIn inverse proportion, f is reducedsThe transmission power can be increased.
N is 10, the voltage regulation ratio p is 1, and the working frequency is fs、fsA/2 and fsTransmission power curve of 2, as shown in figure 7. As shown in the figure at transmission power PaAt an operating frequency of fs、fsA/2 and fsThe corresponding phase shift ratios are D1、D2And D3And satisfy D3<D1<D2. Lowering the operating frequency can reduce the phase shift ratio, which in turn increases. As can be seen from the analysis of the zero backflow power region Z, since the change in frequency does not affect the region Z, the phase shift ratio D can be changed by changing the operating frequency, so that D falls in the zero backflow power region Z again, and thus the state where Q is equal to 0 is achieved.
Setting the operating frequency to fwWhile a transformer is arrangedFrequency range of operation fmin，fmax]And satisfies the following conditions:
fw＝kfs
consider a curve where P is symmetric about D0.5, and D>The reflux power is deteriorated at 0.5, so that D.ltoreq.0.5 is defined. As mentioned above, for the nominal operating frequency fsThe specific MMC-H bridge type DC transformer is shown in formula (13)
Figure BDA0001724092980000071
The frequency conversion optimization control strategy is expressed as
Figure BDA0001724092980000072
A variable-frequency optimized controlled (VFOC) control block diagram and a control flow diagram of the dc transformer optimization control strategy are available according to the above formula, as shown in fig. 8.
The DC transformer shown in FIG. 8 is based on VFOC control block diagram, using phase shift ratio D and frequency fwTo achieve regulation of output voltage and power. The control model comprises an outer ring voltage, an inner ring current and a VFOC controller, wherein the output value of an outer ring voltage control ring is the reference value of an inner ring current ring, and the output value of the current ring is the phase shift ratio D. The phase shift ratio D and the rated control frequency f obtained by the control loopsInputting the power into a VFOC controller, and calculating zero backflow power Dmin、DmaxAnd adjusting the shift ratio and the working frequency by comparing the size of the upper limit and the lower limit of the shift ratio D and the Z, so as to realize that the shift ratio D falls into the region Z.
TABLE 1 simulation parameters
Figure BDA0001724092980000073
From table 1, it can be seen that the dc voltage regulation ratio p ═ U of the dc transformer1/(nU2) 1, and simultaneously calculating D according to an upper and lower boundary formula of a zero backflow region of the formulamin＝0.0918，Dmax0.2556, so zero reflux powerA region can be represented as Z ═ 0.0918,0.2556]。
FIGS. 9 and 10 are simulated waveforms at an operating frequency of 500Hz and 1 kHz. FIG. 9 shows a simulated waveform at an operating frequency of 500Hz, where (a) shows a voltage-current waveform, where the phase shift ratio D is 0.045 and D is<DminThat is, this state has a reflux power of about 11.3W, as shown in fig. (b). FIG. 10 is a simulation waveform at an operating frequency of 1kHz, and (a) is a waveform when a frequency conversion optimization control strategy is adopted, wherein the operating frequency is fw1kHz voltage and current waveform, and a phase shift ratio D of 0.091 and DminAnd therefore the return power drops to zero as shown in graph (b).
The above embodiments are only for illustrating the technical idea of the present invention, and the protection scope of the present invention is not limited thereby, and any modifications made on the basis of the technical scheme according to the technical idea of the present invention fall within the protection scope of the present invention.

Claims (5)

1. A frequency conversion optimization control method suitable for a modular multilevel DC transformer is characterized by comprising the following steps:
step 1, setting an MMC side of an MMC-H bridge type direct current transformer consisting of an MMC bridge arm and an H bridge as a nearest level approximation modulation strategy, setting an H bridge side as square wave modulation, and analyzing the running characteristic of the direct current transformer;
step 2, deducing a transmission power and reflux power expression of the direct current transformer based on a recent level approximation modulation strategy according to the topological and operating characteristics of the MMC-H bridge type direct current transformer;
step 3, analyzing the distribution condition of the transmission power and the reflux power changing along with the parameters for the deduced power expression, summarizing factors influencing the reflux power, discussing the optimized configuration of the voltage regulation ratio, summarizing the boundary condition of the zero reflux power and obtaining a zero reflux power area;
and 4, according to the deduced power expression, the working frequency is inversely proportional to the transmission power, and the direct-current transformer adjusts the phase-shifting ratio to fall into a zero reflux power area through the change of the working frequency, so that the inhibition of the reflux power of the direct-current transformer is realized.
2. The frequency conversion optimization control method suitable for the modular multilevel dc transformer according to claim 1, wherein: in step 2, the derived transmission power per unit expression is:
Figure FDA0002431126360000011
wherein the reference power PN＝nU1U2/(8fsL)＝nUdc1Udc2/(8fsL)，U1、U2Is the primary and secondary voltage peak value, wherein U1＝Udc1，U2＝Udc2Will U is2Converted to primary side, nU2(ii) a n is the transformer transformation ratio; DT is the phase-shifting time in a half period, D is the phase-shifting ratio, and D is more than or equal to 0 and less than or equal to 1; t is half of the working period and the switching frequency fs1/(2T); n is the number of steps in 1/4 cycles; x is a positive integer and x<N，
Figure FDA0002431126360000012
3. The frequency conversion optimization control method suitable for the modular multilevel dc transformer according to claim 1, wherein: in step 2, the derived reflow power per unit expression is as follows:
Figure FDA0002431126360000013
wherein the reference power QN＝nU1U2/(8fsL)＝nUdc1Udc2/(8fsL)，U1、U2Is the primary and secondary voltage peak value, wherein U1＝Udc1，U2＝Udc2Will U is2Converted to primary side, nU2(ii) a n is the transformer transformation ratio; p is U1/(nU2)＝Udc1/(nUdc2) A voltage regulation ratio;
Figure FDA0002431126360000021
Figure FDA0002431126360000022
d is a phase shift ratio, and D is more than or equal to 0 and less than or equal to 1; t is half of the working period and the switching frequency fs＝1/(2T)；a1、a2、b1、b2、c1、c2Are coefficients.
4. The frequency conversion optimization control method suitable for the modular multilevel dc transformer according to claim 1, wherein: in step 3, the expression of the zero reflow power region is:
Figure FDA0002431126360000023
wherein p is a voltage regulation ratio; n is the number of steps in 1/4 cycles;
Figure FDA0002431126360000024
Figure FDA0002431126360000025
5. the frequency conversion optimization control method suitable for the modular multilevel dc transformer according to claim 1, wherein: in said step 4, the phase shift ratio D and the frequency f are usedwThe output voltage and power adjustment is realized, wherein the control model comprises an outer ring voltage, an inner ring current and a VFOC controller, the output value of an outer ring voltage control ring is the reference value of an inner ring current ring, and the current ring output is a shift ratio D; the phase shift ratio D and the rated control frequency f obtained by the control loopsInputting the power into a VFOC controller, and calculating zero backflow power Dmin、DmaxAdjusting the phase shift ratio by comparing the magnitudes of the upper and lower limits of the phase shift ratio D and ZAnd the implementation phase shift ratio D falls within the zero backflow power region Z compared to the operating frequency.
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