CN108848047B - Method for realizing filter bank multicarrier transmitter - Google Patents

Method for realizing filter bank multicarrier transmitter Download PDF

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CN108848047B
CN108848047B CN201810705720.6A CN201810705720A CN108848047B CN 108848047 B CN108848047 B CN 108848047B CN 201810705720 A CN201810705720 A CN 201810705720A CN 108848047 B CN108848047 B CN 108848047B
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龙恳
程志安
余翔
段思睿
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Chongqing University of Post and Telecommunications
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    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
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Abstract

The invention relates to a method for realizing a filter bank multicarrier transmitter, belonging to the technical field of wireless communication. The method comprises the following steps: s1: carrying out OQAM modulation on the transmitted data stream, then grouping data blocks of the data stream of the modulation number, and carrying out IFFT respectively; s2: designing a PAPR suppression module to realize optimal phase search, and multiplying the optimal phase factors by data blocks after IFFT respectively; s3: designing a PPN module, inputting each optimized data block into a multiphase filter network PPN and then adding; s4: and the two paths of I/Q delay are added to generate an FBMC signal. The invention can reduce the calculation complexity of the FBMC transmitter and obtain the FBMC waveform with lower PAPR.

Description

Method for realizing filter bank multicarrier transmitter
Technical Field
The invention belongs to the technical field of mobile communication, and relates to a method for realizing an FBMC transmitter with low PAPR.
Background
The FBMC technology, as one of the 5G candidate waveforms, can meet the overall requirements of future mobile communications. FBMC is based on subcarrier filtering, where strict orthogonality between subcarriers is not required. The subcarrier spacing can be flexibly set, so that effective utilization of scattered spectrum resources becomes feasible. In order to avoid interference between FBMC subcarriers and large spectrum leakage, the FBMC baseband adopts OQAM modulation and individual filtering of each subcarrier, but at the same time, the complexity of a transceiving system is increased, so how to reduce the complexity of the FBMC system becomes one of research hotspots. Another topic widely discussed in the waveform research process is PAPR performance, and low PAPR can improve the working efficiency of a power amplifier, which can directly translate into larger cell coverage or lower terminal power consumption. PAPR is one of the key factors affecting the performance of FBMC systems.
Patent [ CN106878220A ] proposes to divide the constellation sequence of the modulated FBMC symbol into an I-way FBMC symbol sequence and a Q-way FBMC symbol sequence, where the even carriers and the odd carriers of the I-way FBMC symbol sequence are combined into a frequency domain sequence of the I-way FBMC symbol, and the even carriers and the odd carriers of the Q-way FBMC symbol sequence are combined into a frequency domain sequence of the Q-way FBMC symbol. And combining the frequency domain sequence of the I path of FBMC symbols and the frequency domain sequence of the Q path of FBMC symbols, and performing inverse Fourier transform on the combined frequency domain sequence to obtain a time domain sequence of the modulated FBMC symbols. The scheme adopts a frequency domain expansion mode to realize the FBMC waveform, but the FFT point number can be expanded by K times.
Patent CN105681241A proposes an algorithm for reducing PAPR of FBMC system. The algorithm maps a symbol to be sent into a real number symbol, obtains the real number symbol after phase rotation, modulates the real number symbol after each phase rotation, obtains a time domain signal after modulation, calculates each time domain signal after modulation to obtain a phase rotation factor vector corresponding to the minimum, and takes the phase rotation factor vector as an optimal phase rotation factor vector. The phase factor adopted by the algorithm is a traditional phase factor, real part data and imaginary part data of the FBMC signal respectively correspond to an optimal phase twiddle factor vector, and although the PAPR of the signal is reduced, the calculation complexity is high.
Therefore, the invention adopts the joint algorithm to carry out PAPR suppression on the FBMC waveform, and reduces the calculation complexity.
Disclosure of Invention
In view of the above, the present invention is directed to an implementation method of an FBMC transmitter, by which the computational complexity of the FBMC transmitter can be reduced and an FBMC waveform with a low PAPR can be obtained.
In order to achieve the purpose, the invention provides the following technical scheme:
a method for realizing a filter bank multicarrier transmitter comprises the following steps:
s1: performing interleaved Quadrature (OQAM) Modulation on the transmitted data stream, then grouping data blocks of the data stream with Modulation numbers, and performing IFFT respectively;
s2: designing a Peak to Average Power Ratio (PAPR) suppression module, wherein the module mainly realizes optimal phase search, and after obtaining an optimal phase factor, the module is respectively multiplied by data blocks after each group of IFFT;
s3: designing a PPN module, inputting each optimized data block into the PPN and then adding the data blocks;
s4: and the two paths of I/Q delay are added to generate an FBMC signal.
Further, the step S1 specifically includes the following steps:
s11: carrying out OQAM modulation on the transmitted data stream;
s12: after the data stream is subjected to OQAM modulation, carrier allocation and training sequence insertion are carried out, and a serial-parallel conversion process is completed;
s13: calling a carrier classifier module, setting the number of FFT points of a parameter, and setting the total number of subcarriers according to the parameter;
s14: setting training sequence parameters to generate a synchronous word I and a synchronous word II;
s15: setting parameters of occupied sub-carriers to realize sub-carrier mapping;
s16: after the above parameters are set, the data stream is multiplied by a linear phase matrix, and then IFFT transformation is performed.
Further, the OQAM modulation specifically includes the following steps:
s111: completing QAM mapping on the transmitted data stream;
s112: taking a real part and an imaginary part of the mapped QAM complex symbol to obtain a data vector D;
s113: multiplying the resulting data vector D by the vector θk,mCarrying out the alternate between the virtual and the real in the frequency domaink,mThe values are shown as follows:
Figure BDA0001715413390000021
where k is a subcarrier index and m is a symbol index.
Further, the step S2 specifically includes the following steps:
the PAPR suppression module performs PAPR suppression on the FBMC waveform by combining a partial transmission sequence algorithm and an MBJO-PTS algorithm of a novel phase factor, wherein an input data block is m modulated OQAM symbols, the input data block is divided into v groups, and the v groups are expressed by vectors as follows:
Figure BDA0001715413390000022
wherein m is a symbol mark, and m is a symbol mark,
Figure BDA0001715413390000023
represents the kth group of data of the mth symbol, and after the group of data is subjected to IFFT, the sequence of the signals is represented as:
Figure BDA0001715413390000031
wherein k is a subcarrier label, and M represents the number of IFFT points;
the phase factor matrix for symbol m is represented as:
Figure BDA0001715413390000032
wherein
Figure BDA0001715413390000033
The phase factor of the kth group, representing the symbol m, satisfies the following equation:
Figure BDA0001715413390000034
Figure BDA0001715413390000035
where ω is the number of discrete phases, and after each group is multiplied by a phase factor, the mth symbol sequence is represented as:
Figure BDA0001715413390000036
wherein the content of the first and second substances,
Figure BDA0001715413390000037
and
Figure BDA0001715413390000038
the first half and the second half of the kth group data after IFFT are respectively expressed, namely:
Figure BDA0001715413390000039
Figure BDA00017154133900000310
Figure BDA00017154133900000311
and
Figure BDA00017154133900000312
phase factors respectively representing the first half and the second half of the kth group, the phases of the first half and the second half thereof being respectively determined by
Figure BDA00017154133900000313
And
Figure BDA00017154133900000314
expressed, both satisfy the following equation:
Figure BDA00017154133900000315
alpha is 0-0.5, and only the overlapping performance of the previous m data blocks is considered in the method when the mth data block is optimized; assuming that the phase factors of the first m-1 data blocks have been determined, the objective function of optimizing the mth data block is
Figure BDA00017154133900000316
Subject to
Figure BDA00017154133900000317
Figure BDA00017154133900000318
bm＝bm+1＝…＝bM-1＝0
Where K is a superposition factor, PqThe value of the PAPR of the q-th segment is expressed by the following formula:
Figure BDA0001715413390000041
where L represents the total number of transmitted symbols, n represents the discrete time domain index, SnRepresents the time domain value sent at the time n, E { | Sn|2And represents the mean value of the time domain energy sent at the time n.
Further, the step S3 specifically includes: obtaining an optimal phase factor through the step S2, and multiplying the optimal phase factor by each group of data blocks respectively to obtain parallel M paths of data, wherein the total group number is v; let the time domain impulse response length of the filter be LPShifting and extracting the M groups of extracted values at intervals of M, wherein the number of each group of extracted values is K; after performing convolution operation on each optimized M paths of data and K sampling values of M groups of data respectively, performing M times of sampling on each path of data, and performing delay addition to obtain data with the data length of K x M; after repeating the step v times, overlapping the obtained data to obtain I-path data; the magnitude of the M-point DFT integral operation is regarded as being proportional to M log2M, and the frequency domain spreading algorithm needs to spread the data amount by K times during the carrier mapping, so the total calculation amount is about KM × log2(KM); in the polyphase structure, the data stream is first subjected to M-point IFFT directly, which calculates the total amount of M × log2And M, the calculation complexity is lower.
Further, the step S4 specifically includes the following steps: step S3 is repeated to obtain Q channels of data, and half of the delay symbol period is added to the I channel to generate an FBMC waveform with a lower PAPR.
The invention has the beneficial effects that: the method is different from the traditional PTS algorithm, the influence of the first m-1 symbols is considered when the mth symbol is optimized, and a novel phase factor is adopted for optimization, so that the PAPR is effectively inhibited, and the side information of a transmitter is not required to be additionally increased. Meanwhile, the FBMC symbol is generated by combining the PPN structure, so that the computational complexity is reduced.
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In order to make the object, technical scheme and beneficial effect of the invention more clear, the invention provides the following drawings for explanation:
fig. 1 is a block diagram of an FBMC transmitter according to the present invention;
FIG. 2 is a comparative graph of Complementary Cumulative Distribution Function (CCDF) curves for different algorithms;
fig. 3 is a comparison graph of algorithm complexity.
Detailed Description
Preferred embodiments of the present invention will be described in detail below with reference to the accompanying drawings.
The invention mainly solves the problem that the FBMC waveform has higher PAPR, and the method comprises the following steps: modulating the transmitted data stream by OQAM; grouping the modulated data stream into data blocks, wherein each group of data is uniformly divided into two parts according to the sequence of subcarriers, and the two parts are respectively multiplied by different phase factors, and the phase of each group of data meets the alpha-time relation; carrying out optimal phase search, determining an optimal phase factor of each group, and considering the influence of the first m-1 symbols when the mth symbol is optimized; and inputting PPN into each optimized data block and then adding. The invention combines the PPN structure and the Partial Transmission Sequence (Improved Partial Transmission Sequence, Improved-PTS) algorithm of the novel phase factor and the Partial Transmission Sequence (Multi-Block-Joint-Optimization Partial Transmission Sequence, MBJO-PTS) algorithm jointly optimized by multiple data blocks to carry out PAPR inhibition on the FBMC waveform, can effectively inhibit the PAPR, does not need to additionally increase the sideband information of a transmitter, and reduces the calculation complexity.
As shown in fig. 1, the FBMC transmitter implementation method specifically includes the following steps:
s1: data streams are generated and QAM modulated, defining the parity of the number of symbols. The parity subcarrier characteristic is defined in terms of the number of subcarriers. And taking a real part and an imaginary part of the obtained QAM complex symbol. Multiplying the obtained real number by a vector thetak,mAnd carrying out frequency domain alternate between virtual and real. The imaginary and real characters of the carrier mapping of the even symbols are opposite to those of the odd symbols, and constitute the imaginary and real phases of the time domain. The processing flow is divided into three steps:
s11: completing QAM mapping on the input bit stream;
s12: carrying out real part and imaginary part taking on the mapped QAM complex symbol to obtain a data vector D;
s13: multiplying the resulting data vector D by the vector θk,mCarrying out the alternate between the virtual and the real in the frequency domaink,mThe values are shown as follows:
Figure BDA0001715413390000051
where k is a subcarrier index and m is a symbol index.
S2: after the data stream is subjected to OQAM modulation, carrier allocation and training sequence insertion are carried out, and a serial-parallel conversion process is completed. And calling a carrier classifier module, setting the number of FFT points of the parameter, and setting the total number of the subcarriers according to the parameter. And setting the training sequence parameters to generate a synchronous word I and a synchronous word II. And setting parameters of occupied subcarriers to realize subcarrier mapping. After the above parameters are set, the data stream is multiplied by a linear phase matrix, and then IFFT transformation is performed.
S3: and finishing the steps, and enabling the data stream to enter a PAPR restraining module. The module realizes the following flow;
s31: generating a phase factor matrix, wherein the phase factor matrix comprises all combination conditions of phase factors of two data blocks, and searching the phase factors in the subsequent processing flow in the matrix;
s32: traversing the phase factor matrix for two data partitions of each symbol, and correspondingly multiplying the two data partitions by symbol data;
s33: and traversing m FBMC-OQAM symbols in sequence and calculating the minimum value of the sum of PARP values of the first n symbols, wherein the n value is increased by 1 each time. The PARP values of the respective partitions of the first n symbols are calculated to obtain the sum of the PARP values of the first n symbols. Here, the division segment length of each symbol is the number of sample points of IFFT. The PARP values of the first n symbols corresponding to the phase factor combination can be obtained through the processing;
s34: the optimal phase factors corresponding to the first n symbols are obtained by screening the PARP value minimum value of the first n symbols, and finally the optimal phase factors corresponding to the input m FBMC-OQAM symbols can be obtained along with the sequential increase of the n values;
as shown in fig. 2, for the conventional Partial Transmission Sequence (PTS) method and the PTS method using the novel phase factor, since the overlapping property of FBMC symbols is not considered, the PAPR performance is not improved within the threshold (10-13), even a certain peak value is increased. The MBJO-PTS algorithm and the combined algorithm (Multi-Block-Joint-Optimization Partial Transmission Sequence with New Factor, MBJO-New Factor-PTS) used by the invention do not cause peak value proliferation, and the PAPR improvement performance is more obvious. The combined algorithm is better than the MBJO-PTS algorithm in improving the PAPR performance, and the experimental parameters are shown in the table 1.
TABLE 1 Primary simulation parameters
Figure BDA0001715413390000061
For the entire FBMC-OQAM signal, (L) is requiredP+1) MN real number multiplication-filtered signals, where N is the number of transmitted symbols, LPIs the length of the filter. After the signal is divided into v groups, each group is IFFT modulated, requiring v (L) in totalP+ M/2) M complex multiplications, and (v-1) (L)P+ M/2) M complex additions, each complex multiplication containing 4 real multiplications and 2 real additions; each complex addition contains 2 real additions. With respect to the conventional PTS algorithm, it is,each packet has a common value of ωvSpecies combinations are assigned, requiring v (L)P+ M/2) complex multiplications, and (v-1) (L)P+ M/2) complex additions. In general, the time and hardware resources required for multiplication are larger than for addition, so the real multiplication computation complexity for the conventional PTS algorithm is (4vN (L)P+M/2)(N+ωv)+2(LP+1) MN), for the joint algorithm, each data segment phase combination has ωvSeed, need vLPComplex multiplication, and (v-1) LPComplex addition, and the number of data segments is (N + K), so the real multiplication computation complexity of the joint algorithm is (4v (L)P+NM-M/2)(N+ωv)+2(LP+1) MN). As shown in fig. 3, the computational complexity of the joint algorithm is lower than that of the conventional PTS algorithm.
S4: and multiplying the obtained optimal phase factor by the data after IFFT, and processing by a PPN module to finish signal filtering. An FIR filter time domain impulse response matrix h is generated, the length of which is the product of the overlap factor and the number of IFFT points. The processing flow is divided into three steps:
s41: setting a tap coefficient matrix of a parameter filter;
s42: setting parameters of IFFT points M and an overlap factor coefficient K, extracting the impact response function of the filter by M times to obtain M groups, wherein each group contains K data, and then convolving each path of data after IFFT with the M groups of K data respectively;
s43: filling zero to each group of K data by M times of interpolation, and finally carrying out delay addition;
s5: and repeating the steps to obtain Q paths of data, and adding half of the delay symbol period with the I path to generate an FBMC waveform with lower PAPR.
Finally, it is noted that the above-mentioned preferred embodiments illustrate rather than limit the invention, and that, although the invention has been described in detail with reference to the above-mentioned preferred embodiments, it will be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the scope of the invention as defined by the appended claims.

Claims (5)

1. A method for implementing a filter bank multicarrier transmitter, the method comprising the steps of:
s1: performing interleaved Quadrature (OQAM) Modulation on the transmitted data stream, then grouping data blocks of the data stream with Modulation numbers, and performing IFFT respectively;
s2: designing a Peak to Average Power Ratio (PAPR) suppression module, realizing optimal phase search, and multiplying the optimal phase factors by data blocks after IFFT respectively;
s3: designing a PPN module, inputting each group of optimized data blocks into a Polyphase filter Network (PPN) and adding the data blocks;
s4: the I/Q two paths of delay are added to generate a Filter Bank Multicarrier (FBMC) signal;
the step S2 specifically includes the following steps: the PAPR suppression module combines a Partial Transmission Sequence (Improved Partial Transmission Sequence, Improved-PTS) algorithm of a novel phase factor and a Partial Transmission Sequence (MBJO-PTS) algorithm of Multi-data Block Joint Optimization to perform PAPR suppression on an FBMC waveform, wherein an input data Block is m modulated OQAM symbols, the input data Block is divided into v groups, and the V groups are expressed by vectors as:
Figure FDA0002882483970000011
wherein m is a symbol mark, and m is a symbol mark,
Figure FDA0002882483970000012
and k-th group data representing the m-th symbol, wherein after the group data is subjected to IFFT, the sequence of the signal is represented as:
Figure FDA0002882483970000013
wherein k is a subcarrier label, and M represents the number of IFFT points;
the phase factor matrix for symbol m is represented as:
Figure FDA0002882483970000014
wherein
Figure FDA0002882483970000015
A phase factor of a k-th group representing an m-th symbol satisfies the following equation:
Figure FDA0002882483970000016
Figure FDA0002882483970000017
where ω is the number of discrete phases, and after each group is multiplied by a phase factor, the mth symbol sequence is represented as:
Figure FDA0002882483970000018
wherein the content of the first and second substances,
Figure FDA0002882483970000019
and
Figure FDA00028824839700000110
the first half and the second half of the kth group data after IFFT are respectively expressed, namely:
Figure FDA0002882483970000021
Figure FDA0002882483970000022
Figure FDA0002882483970000023
and
Figure FDA0002882483970000024
phase factors respectively representing the first half and the second half of the kth group, the phases of the first half and the second half thereof being respectively determined by
Figure FDA0002882483970000025
And
Figure FDA0002882483970000026
expressed, both satisfy the following equation:
Figure FDA0002882483970000027
alpha is 0-0.5, and only the overlapping performance of the previous m data blocks is considered in the method when the mth data block is optimized; assuming that the phase factors for the first m-1 data blocks have been determined, the objective function to optimize the mth data block is:
Figure FDA0002882483970000028
Subject to
Figure FDA0002882483970000029
Figure FDA00028824839700000210
bm＝bm+1＝…＝bM-1＝0
where K is a superposition factor, PqThe value of the PAPR of the q-th segment is expressed by the following formula:
Figure FDA00028824839700000211
where L represents the total number of transmitted symbols, n represents the discrete time domain index, SnRepresents the time domain value sent at the time n, E { | Sn|2And represents the mean value of the time domain energy sent at the time n.
2. The method as claimed in claim 1, wherein the step S1 specifically includes the following steps:
s11: carrying out OQAM modulation on the transmitted data stream;
s12: after the data stream is subjected to OQAM modulation, carrier allocation and training sequence insertion are carried out, and a serial-parallel conversion process is completed;
s13: calling a carrier classifier module, setting the number of FFT points of a parameter, and setting the total number of subcarriers according to the parameter;
s14: setting training sequence parameters to generate a synchronous word I and a synchronous word II;
s15: setting parameters of occupied sub-carriers to realize sub-carrier mapping;
s16: after the above parameters are set, the data stream is multiplied by a linear phase matrix, and then IFFT transformation is performed.
3. The method as claimed in claim 1 or 2, wherein said OQAM modulation specifically comprises the following steps:
s111: completing QAM mapping on the transmitted data stream;
s112: taking a real part and an imaginary part of the mapped QAM complex symbol to obtain a data vector D;
s113: multiplying the resulting data vector D by the vector θk,mCarrying out the alternate between the virtual and the real in the frequency domaink,mThe values are shown as follows:
Figure FDA0002882483970000031
where k is a subcarrier index and m is a symbol index.
4. The method according to claim 1, wherein the step S3 specifically includes: obtaining an optimal phase factor through the step S2, and multiplying the optimal phase factor by each group of data blocks respectively to obtain parallel M paths of data, wherein the total group number is v; let the time domain impulse response length of the filter be LPShifting and extracting the M groups of extracted values at intervals of M, wherein the number of each group of extracted values is K; after performing convolution operation on each optimized M paths of data and K sampling values of M groups of data respectively, performing M times of sampling on each path of data, and performing delay addition to obtain data with the data length of K x M; after repeating the step v times, overlapping the obtained data to obtain I-path data; in a polyphase structure, the data stream is first subjected directly to an M-point IFFT, which is computed to a total of about M × log2 M。
5. The method as claimed in claim 4, wherein the step S4 specifically includes the following steps: step S3 is repeated to obtain Q channels of data, and half of the delay symbol period is added to the I channel to generate an FBMC waveform with a lower PAPR.
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