CN108448898A - LLC based on phase shifting angle feedforward is without sensor synchronous rectification control method - Google Patents

LLC based on phase shifting angle feedforward is without sensor synchronous rectification control method Download PDF

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CN108448898A
CN108448898A CN201810272896.7A CN201810272896A CN108448898A CN 108448898 A CN108448898 A CN 108448898A CN 201810272896 A CN201810272896 A CN 201810272896A CN 108448898 A CN108448898 A CN 108448898A
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moment
current
control
frequency
phase shifting
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CN108448898B (en
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张兴
颜瑄
赵文广
刘闯
王付胜
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Hefei University of Technology
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Hefei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0025—Arrangements for modifying reference values, feedback values or error values in the control loop of a converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses a kind of LLC based on phase shifting angle feedforward without sensor synchronous rectification control method.Utilize relationship when reaching identical setting voltage between synchronous rectification driving and mixing control lower switch frequency and phase shifting angle, accurate synchronous rectification is realized by optimizing algorithm, phase shifting angle feedforward control is introduced, solves the problems, such as that the hysteresis quality of output voltage response leads to optimizing algorithm control lag.Control method disclosed by the invention realizes synchronous rectification under conditions of not increasing system hardware cost, further improve the efficiency of LLC resonant converter, solves the hysteresis quality of optimizing algorithm control by phase shifting angle feedforward control, transformer secondary when power supply and load sudden change is avoided to generate circulation, to influence system safety operation.

Description

LLC based on phase shifting angle feedforward is without sensor synchronous rectification control method
Technical field
The present invention relates to a kind of synchronous rectification control methods applied to LLC resonant converter.
Background technology
When LLC resonant converter is applied to the occasion of low-voltage, high-current output, the rectifier loss of transformer secondary will become Must be very important, in order to further increase system effectiveness, need a kind of effective LLC synchronous rectification schemes.And LLC resonant transformations The synchronous rectifier driving situation of device is complex, it is difficult to directly obtain accurate synchronous rectification drive signal.In order to realize essence True synchronous rectification, people are made that various effort, document " Wang J, Lu B.Open loop synchronous rectifier driver for LLC resonant converter[J].2013:2048-2051.”(Wang J,Lu B. Open loop synchronous rectifier driving [J] .2013 2048-2051 page for LLC resonance converters) provide a kind of synchronization of open loop Method for rectifying, the relationship preset between synchronous rectifier driving drives with main switch under different switching frequencies are synchronous to realize Rectification.This method realizes that relatively simple but synchronous rectification precision is not high.
Document " Feng W, Lee F C, Mattavelli P, et al.A Universal Adaptive Driving Scheme for Synchronous Rectification in LLC Resonant Converters[J].IEEE Transactions on Power Electronics,2012,27(8):3775-3781.”(Feng W,Lee F C, Mattavelli P, et al. are directed to universal adaptive drive scheme [J] .IEEE magazines of LLC resonant converter synchronous rectification, The 27th phase of August in 2012, page 3775 to page 3781) on the basis of detecting synchronous rectifier drain-source voltage it is further added by an electricity Press comparator, by constantly adjust synchronous rectifier driving make comparator output pulse width for 0 come reach accurately synchronize it is whole Stream.The detection device added needed for this method is more, and cost is higher.
Document " Hong L, Ma H, Wang J, et al.An efficient algorithm strategy for synchronous rectification used in LLC resonant converters[C]//Industrial Electronics Society,IECON 2016-,Conference of the IEEE.IEEE,2016:2452-2456.” (highly effective algorithm strategy [C] the industrial electros of Hong L, Ma H, Wang J, et al. for the synchronous rectification of LLC resonance converters Son, IECON 2016, IEEE meetings, 2016 page 2452 to page 2456) using flow through the electric current of synchronous rectifier with it is synchronous Relationship between rectifying tube driving realizes synchronous rectification.These methods also need addition current detection circuit, increase volume Outer hardware cost.
Document " Liu Heping, Li Jinlong, Miao Zhiru.Sensorless synchronous rectifier control strategy used in LLC resonant circuit[J].Electric Machines and Control,2014,18(5):(Liu's peace, Li Jinlong, Miao Yiru wait .LLC resonance circuits same without sensor to 49-55. " Walk rectification control method [J] Electric Machines and Controls, the 18th phase of May in 2014, page 49 to page 55) it is stagnant using Optimal gradient Ring comparison method realizes synchronous rectification, and using dynamic decoupling lifting system performance, this method is without adding additional detection circuit.But Decoupling precision in text is not high, does not consider the case where gain maximum changes when load variation, is not accounted in Uncoupled procedure The hysteresis quality of output voltage response.
In conclusion there is also following disadvantages for the prior art：
1) method of additional sensor is needed to increase the hardware cost of system.
2) the method decoupling precision without sensor is not high, does not consider the case where gain maximum changes when load variation, It is the case where hysteresis quality of output voltage response is not accounted in Uncoupled procedure leads to synchronous rectification control lag, prominent in input voltage In the case of increasing or load dump, transformer secondary will produce larger circulation, be unfavorable for system safety operation.
Invention content
In order to realize that accurate effective synchronous rectification, the present invention propose base under conditions of not increasing system hardware cost In the LLC that phase shifting angle feedovers without sensor synchronous rectification control method, can be realized accurately without adding additional sensor Synchronous rectification.
The object of the present invention is achieved like this, and the present invention provides a kind of LLC based on phase shifting angle feedforward without sensor Synchronous rectification control method includes the acquisition of LLC resonant converter output voltage, and steps are as follows：
Step 1, the controller of LLC resonant converter is initialized, controller is phased using frequency displacement is determined after initialization System, switching frequency initial value fs0It is fixed as resonant frequency fr, phase shifting angle initial value θ0For 180 degree；
Step 2, if current time is k, the output voltage U of current k moment LLC resonant converter is acquiredo(k), calculating is worked as Preceding k moment output voltage error eU(k) and current k moment error differential value deU(k)；
Step 3, if current controller is missed using frequency displacement phase control is determined according to the current k moment output voltage that step 2 obtains Poor eU(k), current k moment phase shift angle increment △ θ (k) are obtained by pid calculation, calculates current k moment phase shifting angle θ (k) and current k moment duty ratio D (k)；If current controller uses frequency control, the current k moment obtained according to step 2 defeated Go out voltage error eU(k), current k moment switch periods increment △ T are obtained by pid calculations(k), current k is calculated Moment switch periods Ts(k) and current k moment switching frequency fs(k)；
Step 4, if current k moment controllers are implemented to control using frequency displacement phase control is determined, according to step 4.1；If when current k It carves controller and uses frequency control, implement to control according to step 4.2；
Step 4.1, if current k moment controllers, which use, is determined frequency displacement phase control, when first determining whether the current k that step 3 obtains Carve whether phase shifting angle θ (k) is less than 0：
If θ (k)<0, controller switchs to use frequency control, current k timing synchronizations rectification driving high level time tSR(k) It is set as Tr/ 2, TrFor harmonic period；
If θ (k)>0, (k-1) moment phase shifting angle θ (k-1) in controller is first updated to current k moment phase shifting angle θ (k), Whether the current k moment phase shift angle increment △ θ (k) that judgment step 3 obtains again are less than 2 degree, and optimizing algorithm is used if less than 2 degree Synchronous rectification driving is adjusted, otherwise by current k timing synchronizations rectification driving high level time tSR(k) it is set as current k The corresponding time T of moment duty ratio D (k)D(k)；
Step 4.2, if current k moment controllers use frequency control, first determine whether that the current k moment that step 3 obtains opens Close frequency fs(k) whether it is less than resonant frequency frIf fs(k)<fr, controller switchs to use fixed-frequency control, if fs(k)>fr, will control (k-1) moment switching frequency f in device processeds(k-1) it is updated to current k moment switching frequency fs(k), synchronous rectification drives high level Time is set as half of harmonic period Tr/2；
Step 5, reboot step 2.
Preferably, resonant frequency f described in step 1rIt is determined by following formula：
Wherein, LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.
Preferably, current k moment output voltage errors e described in step 2U(k) and current k moment error differential value deU (k) it is determined by following formula：
deU(k)=eU(k)-eU(k-1)
Wherein,For output voltage given value, eU(k-1) it is (k-1) moment output voltage error.
Preferably, current k moment phase shift angle increment △ θ (k), current k moment phase shifting angle θ (k), current k described in step 3 Moment duty ratio D (k), current k moment switch periods Ts(k) and current k moment switching frequency fs(k) it is determined by following formula：
△ θ (k)=Kpθ(eU(k)-eU(k-1))+KiθeU(k)+Kdθ(eU(k)-2eU(k-1)+eU(k-2))
θ (k)=θ (k-1)+△ θ (k)
△Ts(k)=Kps(eU(k)-eU(k-1))+KiseU(k)+Kds(eU(k)-2eU(k-1)+eU(k-2))
Ts(k)=Ts(k-1)+△Ts(k)
Wherein, KpθTo determine the proportionality coefficient of pid calculation in frequency displacement phase control, KiθTo determine in frequency displacement phase control The integral coefficient of pid calculation, KdθTo determine the differential coefficient of pid calculation in frequency displacement phase control, KpsFor The proportionality coefficient of pid calculation, K in frequency controlisFor the integration system of pid calculation in frequency control Number, KdsFor the differential coefficient of pid calculation in frequency control, θ (k-1) is (k-1) moment phase shifting angle, Ts(k-1) it is (k-1) moment switch periods, eU(k-1) it is (k-1) moment output voltage error, eU(k-2) it is that (k-2) moment output voltage misses Difference.
Preferably, current k moment harmonic periods T described in step 4rTime T is corresponded to current k moment duty ratioD(k) by Following formula determines：
Wherein LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.
Preferably, the process of optimizing algorithm described in step 4 is：
1) the output voltage U of current k moment LLC resonant converter is seto(k) reach output voltage given value U0* and stabilization The Rule of judgment for detecting the current k moment afterwards is：It is current k moment switching frequency f under frequency controls(k), determine under frequency displacement phase control For current k moment phase shifting angle θ (k)；
2) work as tSR(k-1)>tSR(k-2) when, then,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to increase synchronous rectification driving high level time, even tSR(k) =tSR(k-1)+a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) reduces synchronous rectification and drives high level time, even tSR(k)=tSR (k-1)-a；
2) work as tSR(k-1)<tSR(k-2) when,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to reduce synchronous rectification driving high level time, even tSR(k) =tSR(k-1)-a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) increases synchronous rectification and drives high level time, even tSR(k)=tSR (k-1)+a；
Wherein, a is change step, tSR(k-1) it is that (k-1) timing synchronization rectification drives high level time, tSR(k-2) it is (k-2) timing synchronization rectification drives high level time.
The advantageous effect of the present invention compared with the existing technology is：
1. compared with the method for needing sensor, this method does not need additional sensor, reduces system hardware cost.
2. compared with the method for no sensing, this method decouples precision height, and the hysteresis quality for solving output voltage response causes The problem of synchronous rectification control lag, phase shifting angle feedforward control ensures that transformer secondary not will produce circulation, to ensure that system is pacified Row for the national games.
Description of the drawings
Fig. 1 is the circuit diagram of the present invention.
Fig. 2 is frequency control oscillogram.
Fig. 3 is to determine frequency displacement phase control oscillogram.
Fig. 4 is the control structure figure of the present invention.
Specific implementation mode
Clear, complete description is carried out to technical scheme of the present invention below in conjunction with attached drawing.
Fig. 1 is a kind of circuit arrangement of the present invention.UinFor input power, VT1-VT4For transformer primary side switching tube, CrFor Resonant capacitance, LrFor resonant inductance, LmFor magnetizing inductance, TrFor high frequency transformer, SR1And SR2For synchronous rectifier, C is filtering Capacitance, R are load.When using the frequency control in control is mixed, switching tube VT1With VT4Break-make simultaneously, switching tube VT2With VT3Break-make simultaneously, the switching tube complementation conducting of same bridge arm, each switching tube duty ratio is fixed as after ignoring dead time 50%, by changing switching frequency fsTo adjust output voltage.Corresponding oscillogram is as shown in Fig. 2, wherein Ug1,4For switching tube VT1With VT4Drive signal, Ug2,3For switching tube VT2With VT3Drive signal, irFor resonance current, imFor exciting current, Uds2,3For Switching tube VT2With VT3Drain-source voltage, Uds1,4For switching tube VT1With VT4Drain-source voltage, isr1To flow through synchronous rectifier SR1Electricity Stream, isr2To flow through synchronous rectifier SR2Electric current.When using mixing control in when determining frequency displacement phase control, same bridge arm switching tube Complementation conducting, each switching tube duty ratio is fixed as 50% after ignoring dead zone, and output voltage is adjusted by changing phase shifting angle θ. Corresponding oscillogram is as shown in figure 3, wherein Ug1For switching tube VT1Drive signal, Ug2For switching tube VT2Drive signal, Ug3To open Close pipe VT3Drive signal, Ug4For switching tube VT4Drive signal, uabFor transformer primary side bridge arm mid-point voltage, UinFor input electricity Source.Fig. 4 is the control structure figure of the present invention, wherein UoFor output voltage actual value,For output voltage given value, eUFor output Voltage error value, θ are phase shifting angle, TsFor switch periods.
Referring to Fig. 1, Fig. 2, Fig. 3 and Fig. 4, specific implementation process of the invention is as follows：
Step 1, the controller of LLC resonant converter is initialized, controller is phased using frequency displacement is determined after initialization System, switching frequency initial value fs0It is fixed as resonant frequency fr, phase shifting angle initial value θ0For 180 degree.The resonant frequency frBy following Formula determines：
Wherein, LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.In the present embodiment, resonant frequency Lr= 22.4uH, resonant capacitance Cr=450nF.
Step 2, if current time is k, the output voltage U of current k moment LLC resonant converter is acquiredo(k), calculating is worked as Preceding k moment output voltage error eU(k) and current k moment error differential value deU(k)。eU(k) and deU(k) true by following formula It is fixed：
deU(k)=eU(k)-eU(k-1)
Wherein,For output voltage given value, eU(k-1) it is (k-1) moment output voltage error.
Step 3, if current controller is missed using frequency displacement phase control is determined according to the current k moment output voltage that step 2 obtains Poor eU(k), current k moment phase shift angle increment △ θ (k) are obtained by pid calculation, calculates current k moment phase shifting angle θ (k) and current k moment duty ratio D (k)；If current controller uses frequency control, the current k moment obtained according to step 2 defeated Go out voltage error eU(k), current k moment switch periods increment △ T are obtained by pid calculations(k), current k is calculated Moment switch periods Ts(k) and current k moment switching frequency fs(k).Current k moment phase shift angle increment △ θ described in step 3 (k), current k moment phase shifting angle θ (k), current k moment duty ratio D (k), current k moment switch periods Ts(k) and the current k moment Switching frequency fs(k) it is determined by following formula：
△ θ (k)=Kpθ(eU(k)-eU(k-1))+KiθeU(k)+Kdθ(eU(k)-2eU(k-1)+eU(k-2))
θ (k)=θ (k-1)+△ θ (k)
△Ts(k)=Kps(eU(k)-eU(k-1))+KiseU(k)+Kds(eU(k)-2eU(k-1)+eU(k-2))
Ts(k)=Ts(k-1)+△Ts(k)
Wherein, KpθTo determine the proportionality coefficient of pid calculation in frequency displacement phase control, KiθTo determine in frequency displacement phase control The integral coefficient of pid calculation, KdθTo determine the differential coefficient of pid calculation in frequency displacement phase control, KpsFor The proportionality coefficient of pid calculation, K in frequency controlisFor the integration system of pid calculation in frequency control Number, KdsFor the differential coefficient of pid calculation in frequency control, θ (k-1) is (k-1) moment phase shifting angle, Ts(k-1) it is (k-1) moment switch periods, eU(k-1) it is (k-1) moment output voltage error, eU(k-2) it is that (k-2) moment output voltage misses Difference.In the present embodiment, Kpθ=0.02, Kiθ=0.015, Kdθ=0.006, Kps=0.032, Kis=0.025, Kds=0.01, a =30ns.
Step 4, if current k moment controllers are implemented to control using frequency displacement phase control is determined, according to step 4.1；If when current k It carves controller and uses frequency control, implement to control according to step 4.2；
Step 4.1, if current k moment controllers, which use, is determined frequency displacement phase control, when first determining whether the current k that step 3 obtains Carve whether phase shifting angle θ (k) is less than 0：
If θ (k)<0, controller switchs to use frequency control, current k timing synchronizations rectification driving high level time tSR(k) It is set as Tr/ 2, TrFor harmonic period；
If θ (k)>0, (k-1) moment phase shifting angle θ (k-1) in controller is first updated to current k moment phase shifting angle θ (k), Whether the current k moment phase shift angle increment △ θ (k) that judgment step 3 obtains again are less than 2 degree, and optimizing algorithm is used if less than 2 degree Synchronous rectification driving is adjusted, otherwise by current k timing synchronizations rectification driving high level time tSR(k) it is set as current k The corresponding time T of moment duty ratio D (k)D(k)；
Step 4.2, if current k moment controllers use frequency control, first determine whether that the current k moment that step 3 obtains opens Close frequency fs(k) whether it is less than resonant frequency frIf fs(k)<fr, controller switchs to use fixed-frequency control, if fs(k)>fr, will control (k-1) moment switching frequency f in device processeds(k-1) it is updated to current k moment switching frequency fs(k), synchronous rectification drives high level Time is set as half of harmonic period Tr/2。
Current k moment harmonic periods T described in step 4.1rTime T is corresponded to current k moment duty ratioD(k) by following public affairs Formula determines：
Wherein LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.
The process of optimizing algorithm described in step 4.1 is：
1) the output voltage U of current k moment LLC resonant converter is seto(k) reach output voltage given value U0* and stabilization The Rule of judgment for detecting the current k moment afterwards is：It is current k moment switching frequency f under frequency controls(k), determine under frequency displacement phase control For current k moment phase shifting angle θ (k)；
2) work as tSR(k-1)>tSR(k-2) when, then,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to increase synchronous rectification driving high level time, even tSR(k) =tSR(k-1)+a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) reduces synchronous rectification and drives high level time, even tSR(k)=tSR (k-1)-a；
2) work as tSR(k-1)<tSR(k-2) when,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to reduce synchronous rectification driving high level time, even tSR(k) =tSR(k-1)-a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) increases synchronous rectification and drives high level time, even tSR(k)=tSR (k-1)+a；
Wherein, a is change step, in the present embodiment, a=30ns.tSR(k-1) it is that (k-1) timing synchronization rectification drives High level time, tSR(k-2) it is that (k-2) timing synchronization rectification drives high level time, output voltage given value
Step 5, reboot step 2.

Claims (6)

1. a kind of LLC based on phase shifting angle feedforward is without sensor synchronous rectification control method, including LLC resonant converter output electricity The acquisition of pressure, which is characterized in that steps are as follows：
Step 1, the controller of LLC resonant converter is initialized, controller is opened using frequency displacement phase control is determined after initialization Close frequency initial value fs0It is fixed as resonant frequency fr, phase shifting angle initial value θ0For 180 degree；
Step 2, if current time is k, the output voltage U of current k moment LLC resonant converter is acquiredo(k), when calculating current k Carve output voltage error eU(k) and current k moment error differential value deU(k)；
Step 3, if current controller, which uses, determines frequency displacement phase control, the current k moment output voltage error e obtained according to step 2U (k), current k moment phase shift angle increment △ θ (k) are obtained by pid calculation, calculates current k moment phase shifting angle θ (k) With current k moment duty ratio D (k)；If current controller uses frequency control, electricity is exported according to the current k moment that step 2 obtains Press error eU(k), current k moment switch periods increment △ T are obtained by pid calculations(k), the current k moment is calculated Switch periods Ts(k) and current k moment switching frequency fs(k)；
Step 4, if current k moment controllers are implemented to control using frequency displacement phase control is determined, according to step 4.1；If the current k moment is controlled Device processed uses frequency control, implements to control according to step 4.2；
Step 4.1, if current k moment controllers first determine whether that the current k moment that step 3 obtains moves using frequency displacement phase control is determined Whether phase angle theta (k) is less than 0：
If θ (k)<0, controller switchs to use frequency control, current k timing synchronizations rectification driving high level time tSR(k) it sets For Tr/ 2, TrFor harmonic period；
If θ (k)>0, (k-1) moment phase shifting angle θ (k-1) in controller is first updated to current k moment phase shifting angle θ (k), then sentence Whether the current k moment phase shift angle increment △ θ (k) that disconnected step 3 obtains are less than 2 degree, using optimizing algorithm to same if less than 2 degree Step rectification driving is adjusted, otherwise by current k timing synchronizations rectification driving high level time tSR(k) it is set as the current k moment The corresponding time T of duty ratio D (k)D(k)；
Step 4.2, if current k moment controllers use frequency control, first determine whether that the current k moment that step 3 obtains switchs frequency Rate fs(k) whether it is less than resonant frequency frIf fs(k)<fr, controller switchs to use fixed-frequency control, if fs(k)>fr, by controller In (k-1) moment switching frequency fs(k-1) it is updated to current k moment switching frequency fs(k), synchronous rectification drives high level time It is set as half of harmonic period Tr/2；
Step 5, reboot step 2.
2. the LLC according to claim 1 based on phase shifting angle feedforward is without sensor synchronous rectification control method, feature exists In resonant frequency f described in step 1rIt is determined by following formula：
Wherein, LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.
3. the LLC according to claim 1 based on phase shifting angle feedforward is without sensor synchronous rectification control method, feature exists In current k moment output voltage errors e described in step 2U(k) and current k moment error differential value deU(k) by following formula It determines：
Wherein,For output voltage given value, eU(k-1) it is (k-1) moment output voltage error.
4. the LLC according to claim 1 based on phase shifting angle feedforward is without sensor synchronous rectification control method, feature exists In current k moment phase shift angle increment △ θ (k) described in step 3, current k moment phase shifting angle θ (k), current k moment duty ratio D (k), current k moment switch periods Ts(k) and current k moment switching frequency fs(k) it is determined by following formula：
△ θ (k)=Kpθ(eU(k)-eU(k-1))+KiθeU(k)+Kdθ(eU(k)-2eU(k-1)+eU(k-2))
θ (k)=θ (k-1)+△ θ (k)
△Ts(k)=Kps(eU(k)-eU(k-1))+KiseU(k)+Kds(eU(k)-2eU(k-1)+eU(k-2))
Ts(k)=Ts(k-1)+△Ts(k)
Wherein, KpθTo determine the proportionality coefficient of pid calculation in frequency displacement phase control, KiθTo determine ratio in frequency displacement phase control The integral coefficient of integral differential operation, KdθTo determine the differential coefficient of pid calculation in frequency displacement phase control, KpsFor frequency conversion The proportionality coefficient of pid calculation, K in controlisFor the integral coefficient of pid calculation in frequency control, Kds For the differential coefficient of pid calculation in frequency control, θ (k-1) is (k-1) moment phase shifting angle, Ts(k-1) it is (k-1) Moment switch periods, eU(k-1) it is (k-1) moment output voltage error, eU(k-2) it is (k-2) moment output voltage error.
5. the LLC according to claim 1 based on phase shifting angle feedforward is without sensor synchronous rectification control method, feature exists In current k moment harmonic periods T described in step 4rTime T is corresponded to current k moment duty ratioD(k) it is determined by following formula：
Wherein LrFor resonant inductance inductance value, CrFor resonant capacitance capacitance.
6. the LLC according to claim 1 based on phase shifting angle feedforward is without sensor synchronous rectification control method, feature exists In the process of optimizing algorithm described in step 4 is：
1) the output voltage U of current k moment LLC resonant converter is seto(k) reach output voltage given value U0 *And it is detected after stablizing Currently the Rule of judgment at k moment is：It is current k moment switching frequency f under frequency controls(k), determine under frequency displacement phase control to be current K moment phase shifting angle θ (k)；
2) work as tSR(k-1)>tSR(k-2) when, then,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to increase synchronous rectification driving high level time, even tSR(k)=tSR (k-1)+a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) reduces synchronous rectification and drives high level time, even tSR(k)=tSR(k- 1)-a；
2) work as tSR(k-1)<tSR(k-2) when,
If fs(k)>fs(k-1) or θ (k)>θ (k-1) continues to reduce synchronous rectification driving high level time, even tSR(k)=tSR (k-1)-a；
If fs(k)<fs(k-1) or θ (k)<θ (k-1) increases synchronous rectification and drives high level time, even tSR(k)=tSR(k- 1)+a；
Wherein, a is change step, tSR(k-1) it is that (k-1) timing synchronization rectification drives high level time, tSR(k-2) it is (k-2) Timing synchronization rectification drives high level time.
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