CN106067823B - System and method for MEMS sensor - Google Patents

Abstract

According to one embodiment, a sensor circuit includes a sigma-delta type analog-to-digital converter (ADC), a dither clock coupled to the sigma-delta type ADC, and a supply voltage circuit coupled to the sigma-delta type ADC. The sigma-delta ADC is configured to be coupled to a low frequency transducer, and the dither clock is configured to control the sigma-delta ADC based on the dither clock signal.

Description

System and method for MEMS sensor

This application claims the benefit of U.S. provisional application No. 62/150,027 filed on 20/4/2015, which is incorporated by reference in its entirety.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is also directed to the following co-pending and commonly assigned U.S. patent applications, which also claim the benefit of U.S. provisional application No. 62/150,027 filed 4/20/2015: serial number 15/074,510 entitled "System and Method for a Capacitive Sensor" (attorney docket No. INF 2015P 50066 US) filed on day 18/3/2016, serial number 15/074,649 entitled "System and Method for a MEMS Sensor" (attorney docket No. INF 2015P 50162US01) filed on day 18/3/2016, and serial number _____ entitled "System and Method for a MEMS Sensor" (attorney docket No. INF 2015P 50088 US) filed on day 23/3/2016, the entire contents of which are incorporated by reference into this application.

Technical Field

The present invention relates generally to electronic circuits, and in particular embodiments, to systems and methods for microelectromechanical systems (MEMS) interface circuits.

Background

Microelectromechanical Systems (MEMS), which typically include miniaturized various electrical and mechanical components, are manufactured from a variety of materials and manufacturing methods and are used in a variety of applications. These applications include automotive electronics, medical devices, and intelligent portable electronic devices such as cellular phones, Personal Digital Assistants (PDAs), hard drives, computer peripherals, and wireless devices. In these applications, MEMS can be used as sensors, actuators, accelerators, switches, micro-mirrors and many other devices. MEMS are also used in ambient pressure measurement systems to measure absolute or differential ambient pressure.

Various attributes may be considered when designing a system that uses a MEMS device as a sensor, including, for example, resolution and temperature sensitivity. Any ringing noise and energy loss caused by mechanical resonance of the MEMS device may also be considered. In some systems, such mechanical resonances may generate oscillations in response to an excitation signal, and these oscillations may have energy losses characterized by a quality factor (Q). A larger Q represents a smaller rate of energy loss relative to the stored energy of the resonator, whereby the mechanical oscillation vanishes more slowly. A smaller Q represents a greater rate of energy loss relative to the stored energy of the resonator, whereby the mechanical oscillation vanishes more rapidly.

Disclosure of Invention

According to one embodiment, a sensor circuit includes a sigma-delta analog-to-digital converter (ADC), a dithered clock (dithered clock) coupled to the sigma-delta ADC, and a supply voltage circuit coupled to the sigma-delta ADC. The sigma-delta ADC is configured to be coupled to a low frequency transducer, and the dither clock is configured to control the sigma-delta ADC based on the dither clock.

Drawings

For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram illustrating a pressure measurement device including a MEMS-based sensor in accordance with an embodiment of the present invention;

FIGS. 2a to 2c show a MEMS capacitor array layout and a cross section of a MEMS capacitor in a bridge configuration;

FIG. 3 is a graph showing waveforms for a differential response signal with ringing noise and for a square wave excitation signal, in accordance with an embodiment of the present invention;

FIG. 4 illustrates an embodiment of input and output waveforms of a MEMS capacitor sensor configured in a bridge configuration;

FIG. 5 illustrates an exemplary slope control circuit;

FIG. 6 illustrates an exemplary digital pressure measurement system utilizing a slope control circuit;

FIG. 7 shows a flow chart of an exemplary method;

FIG. 8 is a block diagram illustrating an ADC that samples a differential response signal using a pseudo-random sampling clock jitter (jitter) in accordance with an embodiment of the present invention;

FIG. 9 is a graph illustrating relative howling errors at a range of values for a resonant frequency of a MEMS-based sensor, in accordance with an embodiment of the present invention;

FIG. 10 is a block diagram illustrating a circuit for generating a variable clock signal with pseudo-random jitter according to an embodiment of the present invention;

FIG. 11 is a flow chart illustrating a measurement method according to an embodiment of the present invention;

FIGS. 12a and 12b show MEMS capacitor array schematics and layouts of different sizes and locations;

FIG. 13 is a graph showing the output waveform of a square wave excitation signal driving MEMS capacitor sensors of the same size and resonant frequency;

FIG. 14 is a graph showing the output waveform of a square wave excitation signal driving MEMS capacitor sensors having different sizes and resonant frequencies;

FIG. 15 is a graph showing the frequency spectrum of output waveforms from MEMS capacitor sensors having different sizes and resonant frequencies;

FIG. 16 is a graph showing ringing amplitude of output waveforms from MEMS capacitor sensors of the same size and resonant frequency and MEMS capacitor sensors of different sizes and leakage frequencies;

FIG. 17 illustrates a system block diagram of an exemplary MEMS pressure sensor system;

FIG. 18 shows a schematic block diagram of yet another exemplary MEMS pressure sensor system;

fig. 19a and 19b show waveform diagrams of exemplary noise signals generated in a sigma-delta type analog-to-digital converter (ADC);

fig. 20 shows a waveform diagram of a noise signal generated in a sigma-delta type analog-to-digital converter (ADC) without a jittered clock and with a jittered clock;

fig. 21a and 21b show schematic block diagrams of an exemplary sigma-delta type analog-to-digital converter (ADC); and

FIG. 22 shows a block diagram of an exemplary method of operation for a sensor.

Corresponding numerals and symbols in the various drawings generally indicate corresponding parts, unless otherwise indicated. The drawings are drawn to clearly illustrate the relevant aspects of the preferred embodiments and are not necessarily drawn to scale. To more clearly illustrate certain embodiments, letters indicating variations in the same structure, material, or process steps may follow the figure number.

Detailed Description

The making and using of the preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the invention.

Under specific conditions (system and method for performing measurements of a capacitive pressure measurement system using a MEMS-based sensor), the invention is described with reference to a preferred embodiment. Other embodiments may be applied to other sensor systems, such as piezoresistive sensor systems. Some of the various embodiments described herein include a capacitive MEMS pressure sensor, interface circuitry, a sigma-delta type analog-to-digital converter (ADC) for the MEMS pressure sensor interface circuitry, noise in the interface circuitry, and jittered clocks for the sigma-delta type ADC and the interface circuitry. In other embodiments, the various aspects may also be applied to other applications involving any type of transducer system according to any manner known in the art.

Capacitive MEMS pressure transducers use the pressure difference between two regions to adjust a variable capacitance structure and generate an output signal proportional to the pressure difference. In one particular application, a differential output capacitive MEMS pressure transducer uses two variable capacitance structures to generate a differential output that varies according to a measured pressure. In various embodiments, the signal output from the pressure transducer is an analog signal. The analog signal may be amplified and converted to a digital signal.

In an embodiment of the present invention, a capacitive MEMS pressure sensor is operated by introducing a periodic excitation signal at a first port of the MEMS pressure sensor and monitoring an output of the MEMS pressure sensor at a second port of the MEMS pressure sensor. A pressure measurement is then made by determining the amplitude of the signal at the second port of the MEMS sensor. One problem faced in such systems is the under-damped response of the MEMS-based sensor due to mechanical resonance within the MEMS-based sensor, which in some circumstances can lead to measurement errors due to the ringing nature of the output signal. In various embodiments disclosed herein, systems and methods of measuring such an underdamped system are disclosed.

In a first embodiment, to reduce errors caused by an under-damped response of the MEMS pressure sensor to the excitation signal, the slope of the excitation signal is reduced to attenuate harmonics of the under-damped response that may stimulate the MEMS pressure sensor. In some embodiments, the excitation signal is defined or generated such that sharp edges are reduced or eliminated, and in particular embodiments, slope reduction is achieved by generating a dual-slope integrated triangular waveform from a square wave input signal and integrating the signal again to generate a periodic waveform with smooth transitions between a first signal level and a second signal level. In some embodiments, the slope of the first square wave signal is controlled by a delay locked loop to synchronize the edge of the reduced slope excitation pulse with the upcoming clock signal.

In a second embodiment, the output of a MEMS-based sensor is measured using a jittered sampling clock. By dithering the sampling time relative to the under-damped response of the MEMS, a series of measurements can be made in which the under-damped components of the MEMS response are averaged. More specifically, the period of the variable clock signal is changed at regular intervals corresponding to the switching frequency. At a switching time after each such regular interval, the period of the variable clock signal is increased relative to the minimum period by a pseudo-randomly determined period adjustment amount. Equivalently, the frequency of the variable clock signal is reduced by a frequency shift corresponding to the period adjustment amount. The switching frequency of the variable clock signal is designed to be close to the mechanical resonance frequency of the MEMS-based sensor. The sampling clock signal is obtained by dividing the frequency of the variable clock signal. To extend the resonant ringing noise of the MEMS-based sensor output, the output is digitally sampled at pseudo-random variable intervals according to a sampling clock signal. The multiple digital samples are then filtered and combined to suppress wideband ringing noise.

In a third embodiment, the MEMS pressure sensors are implemented using an array or MEMS pressure sensors having variable dimensions such that each MEMS pressure sensor resonates at a different frequency. Thus, when the MEMS pressure sensor is stimulated by an excitation signal, the amplitude of the howling is reduced at each moment due to the various resonant responses being out of phase with each other. By sampling the output of the MEMS sensor as the coherent cancellation of the individual MEMS pressures reduces the amplitude of the impulse response, more accurate measurements can be made.

In a fourth embodiment, an oversampling analog-to-digital converter (ADC) is used to monitor the output of the MEMS sensor. To mitigate idle tones (idle tones) in an oversampling ADC, a jittered clock may be used to operate the oversampling ADC. In some embodiments, the jittered clock signal may be generated according to a second embodiment disclosed herein.

Fig. 1 shows an exemplary pressure measurement device 100 including a sensor 103. The sensor 103 is coupled to the output of the excitation signal generator 102. The excitation signal generator 102 generates an alternating excitation signal which is supplied to the sensor 103, which generates an analog measurement signal consisting of two excitation response signals. Each of these excitation response signals is generated by oscillation of the resonance (which may be, for example, a mechanical resonance) of the sensor 103. In one embodiment, the sensor 103 has an underdamped response.

Referring again to fig. 1, the sensor 103 comprises a capacitive bridge having two bridge segments 105. Each bridge segment 105 comprises a voltage dependent capacitor 111 in series with a reference capacitor 109, and outputs a response signal of the sensor 103 at a center tap between the voltage dependent capacitor 111 and the reference capacitor 109. Reference capacitor 109 has C_rAnd the capacitance C of the voltage-dependent capacitor 111_sRelatively stable with pressure. In one embodiment, the pressure sensitive capacitor 111 is implemented using one or more high-Q MEMS elements, respectively, included in the sensor 103, and the MEMS elements have a resonant frequency f_rIs mechanically resonant. In some embodiments, C is used_rRelative to C_sThe reference capacitor 109 is implemented with a capacitor selected in a known relationship to temperature variation.

Sense amplifiers 104 coupled to the outputs of the sensors 103 amplify these sensor response signals. An analog-to-digital converter (ADC)106 coupled to the output of the sense amplifier 104 then samples the difference between the amplified sensor response signals to provide digital samples. A filter 108 coupled to the ADC 106 combines several of these digital samples over a time interval to generate a single pressure measurement sample. In some embodiments, filter 108 is a low pass filter that averages digital samples. In other embodiments, the filter 108 combines the digital samples using a more complex algorithm, which may include, for example, selecting samples with a median value, discarding outlier samples prior to averaging, and so forth.

In some embodiments, each of excitation signal generator 102, sensor 103, sense amplifier 104, ADC 106, and filter 108 of pressure measurement device 100 are included in a single Integrated Circuit (IC), and the IC has a volume of less than 10 cubic millimeters. In other embodiments, multiple ICs may be included in pressure measurement device 100.

Fig. 2 a-2 c illustrate exemplary embodiments of MEMS sensors that may be used to implement the sensor 103 shown in fig. 1. As shown, in fig. 2a, to include a fixed capacitor C_rAnd a variable capacitor C_sThe MEMS sensor is arranged in a bridge structure. In one embodiment, variable capacitor C_sAre implemented using an array of MEMS sensors and the fixed capacitor is implemented using an array of fixed capacitors designed to track the nominal capacitance of the array of MEMS sensors.

Fig. 2b shows an exemplary layout 210 of the MEMS capacitor sensor array 200. As shown, the layout includes fixed capacitive cells 212 and 218 for implementing the capacitance Cs and MEMS sensor cells 214 and 216 that are sensitive to pressure. As shown, this layout 210 structure may be arranged to avoid gradient mismatch between all four capacitors in the bridge. In one embodiment, the MEMS sensor unit 214 may be implemented using MEMS sensor structures known in the art, while the fixed capacitive unit may be implemented using a MEMS sensor unit that is motion disabled. By using similar physical structures for the MEMS sensor units 214 and 216 and the fixed capacitive units 212 and 218, good matching between the MEMS sensor units 214 and 216 and the fixed capacitive units 212 and 218 can be achieved over process and temperature variations. In some embodiments, movement of the fixed capacitive units 212 and 218 is prevented, for example, by not opening pressure ports during processing of the semiconductor components or by adding mechanical motion stops within the MEMS structure.

Fig. 2c shows a cross section of the MEMS sensor cell 220 alongside the exemplary NMOS and PMOS transistors. As shown, the MEMS sensor cell is formed with a polysilicon diaphragm that serves as a top electrode 222 and a fixed counter electrode 224 to form the sensor cell. A vacuum chamber 226 is provided between the top electrode and the fixed counter electrode. The top electrode 222 is not covered to allow any pressure application and cause a change in the capacitor value. It should be understood that cross-section 220 is but one of many examples of suitable MEMS elements that may be used in embodiments of the present invention.

Fig. 3 shows waveforms for an exemplary square wave excitation signal 302 and a differential response signal 304. The square wave excitation signal is one embodiment of an alternating excitation signal that may be generated by the excitation signal generator 102 of FIG. 1. In other embodiments, any alternating excitation signal may be used, including, for example, a sinusoidal signal, a triangular signal, or a composite signal, among others.

Referring again to FIG. 3, differential response signal 304 represents the difference between the response signals at the outputs of sensor 103 of FIG. 1. Since the MEMS element or elements that make up the pressure sensitive capacitor 111 are high Q, the mechanical resonance of these MEMS elements is under-damped and the mechanical oscillations slowly disappear. At each of the sampling times t1, t2, and t3, howling noise is introduced in the differential response signal 304 relative to the ideal response signal 306 of the sensor formed by the ideal capacitor.

First embodiment

In the first embodiment, the slope of the excitation 302 is reduced to avoid unduly stimulating the resonance condition of the MEMS sensor 103. Fig. 4 shows a comparison between a square wave excitation signal 402 and a waveform excitation signal 404 for stimulating the MEMS sensor at the input port Vex. As shown, the output waveform 406 represents high amplitude ringing due to the inclusion of the steeply sloped square wave input 402 at the rising and falling edges, because the resonance stimulated by the square wave excitation signal 402 with steep rising and falling edges causes resonance. On the other hand, the output waveform 408 represents a response from the waveform excitation signal 404 and exhibits very little ringing due to smooth edges at the rising and falling portions of the input and transition regions to flat regions. The smooth edges at the rising and falling portions of the excitation signal 404 reduce resonance effects due to the high Q factor of the MEMS sensor and capacitor and provide a smoother output waveform 408.

In one embodiment, the slope of the excitation signal is controlled in the time domain by using cascaded integrators to control the rising and falling behavior of the excitation signal while maintaining a stable output voltage between edge transitions. Thus, the first integration produces a triangular edge, while the second integration produces an edge having a second order or parabolic shape. The output of the second integration is used to drive the MEMS sensor 103 arranged in a bridge configuration. Such an embodiment shapes to reduce the amplitude of the generated harmonics and reduce the ringing seen at the output of the MEMS sensor 103. It should be understood that the cascade integrator approach is only one example of many possible exemplary systems and methods that may be used to control the slope of the excitation signal. In some embodiments, the stimulus signal has a time period during which the signal value is stable, e.g., it has a fixed reference voltage. The time period during which the time period is stable may also be referred to as a "flat region". During this time period, the sensor output signal and the sense amplifier output signal are also stable and may be sampled, for example, by the ADC 106 shown in fig. 1.

FIG. 5 illustrates an exemplary excitation pulse generation system 500 comprising a timing control circuit 502, a charge pump 504, a first integrating capacitor C1, a second integrator and wave shaping circuit 508 for controlling the coupling of the second integrator output to C_LoadSwitch 510, comparator 512, phase detector 514, and second charge pump 516 that drives loop filter capacitor C2. The timing control circuit 502 accepts an input square wave signal or clock from the pulse generator 501 and generates two switch control signals for driving the charge pump 504. The two switch control signals selectively enable the switch connecting the charging or discharging current source to the integrating capacitor C1. The square wave clock signal is integrated across the integrating capacitor C1 and a sigma-delta type waveform is generated. The sigma-delta type waveform is buffered by buffer amplifier 506. The buffered triangular waveform is further integrated to generate a waveform containing a very low energy content at the resonant frequency of the MEMS capacitor. The second integral smoothes sharp edges present in the square wave as well as edges of the triangular waveform. It should be appreciated that a square wave with sharp edges contains a large number of high frequency components that can stimulate a resonant response within the MEMS sensor 103. The endpoints of the integration waveform are controlled to ensure that the edges of the slope control output signal are synchronized with the input clock signal. Opening 510 is configured to couple load capacitance C_LoadCoupled to the output of the integrator and wave shaping circuit 508 or to the supply voltage VDD (typically a temperature stable and low noise reference voltage) of the circuit block. In some embodiments, the supply voltage is not satisfied at the second integrator and beamformer 508Under voltage, the switch 510 is used to load the capacitor C_LoadCoupled to the supply voltage VDD.

As shown in FIG. 5, the slope control stimulus signal is passed through a comparator 512 to form a signal that can be used to adjust the phase difference between the input clock and the stimulus signal. The phase at the output of comparator 512 is compared with the phase of the input clock signal via phase detector 514, generating two control signals that activate two switches in charge pump circuit 516. The switch connects the charging or discharging current source to the loop filter capacitor C2. The voltage at capacitor C2 is an indication of the phase difference between the stimulus signal and the input clock signal. The loop filter capacitor C2 converts the instantaneous phase difference into an analog voltage. This voltage is used to control the amplitude of the charge and discharge currents while generating a sigma-delta type waveform. In one embodiment, excitation pulse generation system 500 may be implemented in a single Integrated Circuit (IC).

Fig. 6 shows an exemplary digital pressure measurement system 600 that includes an exemplary excitation signal generator 602, a capacitive pressure sensor 604 (including sense amplifier 104), a temperature sensor 606, a multiplexer 608, an analog-to-digital converter (ADC)610, digital signal processing 612, a digital core 614, a digital interface 616, a voltage regulator 618, a memory interface 620, a cell memory calibration coefficient 622, and a FIFO (first-in-first-out) 624. The excitation signal generator 602 provides a slope control excitation signal to the capacitive pressure sensor 604 according to the embodiments described above. The multiplexer 608 selects the measurement from the temperature sensor 606 or the capacitor sensor 604 and sends to the ADC circuit 610 for digital conversion of the measurement. The ADC output is then passed through a digital signal processing unit 612 for further filtering and mathematical calculations. Digital core 614 and digital interface 616 are part of an internal processor that converts temperature and pressure measurements into 24-bit digital words. Calibration coefficients 622 store calibration values for each individual pressure sensor for use in measurement correction. The FIFO 624 stores a plurality of temperature and pressure measurements during the low power mode. Storage interface 620 provides these values to digital core 614. Embodiments also include an internal voltage regulator for providing power to internal circuitry.

In one embodiment, the digital pressure measurement system 600 may be implemented using a single integrated circuit and/or a combination of integrated circuits and/or discrete components. It should be understood that system 600 is only one of many example systems in which an exemplary excitation signal generator may be implemented.

FIG. 7 shows a flow diagram of an exemplary method 700 of controlling the slope of a MEMS capacitor. In step 702, a first integration is performed on a first input signal. In one embodiment, the first input signal is a square wave signal. Next, in step 704, a second integration is performed on the first integrated output signal. In one embodiment, the first integration output is a triangular waveform and the second integration is performed for wave shaping. In some embodiments, when the output signal reaches the reference voltage (Vdd) or a certain amount of time passes, the output remains constant (eventually the output is switched to Vdd) until the falling edge process is triggered. Next, in step 706, the output of the second integration is used to drive a MEMS capacitor bridge, where the bridge has two sections and each section consists of one voltage dependent capacitor and one reference capacitor. Next, in step 708, the phase of the input signal and the output of the second integration are synchronized. In one embodiment, a phase detector is used to synchronize the phases. Next, in step 710, a sense amplifier connected to the common point of the voltage dependent capacitor and the reference capacitor is used to measure the transient capacitor change. Finally, in step 712, an A/D conversion is performed on the output of the sense amplifier to calculate the pressure.

According to various embodiments, a circuit or system may be configured to perform a particular operation or action by having hardware, software, firmware, or a combination thereof installed on the system that in operation cause the system to perform the action. One general aspect includes a method of performing a measurement with a capacitive sensor, the method comprising: generating a periodic excitation signal, the periodic excitation signal comprising a series of pulses; smoothing edge transitions of a series of pulses to form a shaped periodic excitation signal; providing a shaped periodic excitation signal to a first port of a capacitive sensor; and measuring a signal provided through the second port of the capacitive sensor. Other embodiments of this aspect include corresponding circuits and systems configured to perform the actions of the methods.

Implementations may include one or more of the following features. The method further comprises the following steps: an output measurement value is determined based on the measurement signal. In the method, the determined output measurement comprises a pressure measurement. In the method, smoothing the edge transition includes generating a first slope signal based on the generated periodic excitation signal to form a slope excitation signal. In the method, smoothing the edge transition further comprises: the slope signal is integrated to form a shaped periodic excitation signal. The method further comprises the following steps: the slope of the first slope signal is adjusted based on a timing difference between the shaped periodic excitation signal and the periodic excitation signal. In the method, generating the first slope signal comprises: the capacitor is charged with a first current source and discharged with a second current source. The method further comprises the following steps: adjusting a slope of the first slope signal based on a timing difference between the shaped periodic excitation signal and the periodic excitation signal, wherein adjusting the slope includes adjusting currents of the first current source and the second current source. The method further comprises the following steps: a timing difference between the shaped periodic excitation signal and the periodic excitation signal is determined. In the method, determining the timing difference includes using a phase detector. In the method, the capacitive sensor comprises a MEMS sensor. In the method, the MEMS sensor includes a sensor bridge having: a first branch having a first MEMS pressure sensor and a first capacitor; and a second branch having a second MEMS pressure sensor and a second capacitor. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

Yet another general aspect includes a system, comprising: a stimulus generator configured to be coupled to the first port of the capacitive sensor, the stimulus generator comprising a pulse generator and a pulse smoothing circuit coupled to an output of the pulse generator, wherein an output of the pulse smoothing circuit is configured to be coupled to the first port of the capacitive sensor. Other embodiments of this aspect include corresponding circuits and systems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. The system also includes a readout circuit configured to be coupled to the second port of the capacitive sensor. In the system, the readout circuit includes an a/D converter configured to be coupled to the second port of the capacitive sensor. In the system, the readout circuit is configured to determine a response of the capacitive sensor based on a signal from a second port of the capacitive sensor. The system also includes a capacitive sensor. In the system, the capacitive sensor comprises a MEMS sensor. In the system, the MEMS sensor includes a sensor bridge having: a first branch having a first MEMS pressure sensor and a first capacitor; and a second branch having a second MEMS pressure sensor and a second capacitor. In the system, the pulse smoothing circuit includes a ramp generator having an input coupled to an output of the pulse generator. In the system, the ramp generator includes a first current source and a second current source coupled to the first capacitor. In the system, the pulse smoothing circuit further comprises an integrator coupled to an output of the ramp generator, wherein an output of the integrator is coupled to an output of the pulse smoothing circuit. The system also includes a phase detector having a first input coupled to the output of the pulse generator and a second input coupled to the output of the integrator, wherein the output of the phase detector is configured to control the slope of the signal at the output of the ramp generator. The system also includes a charge pump coupled to an output of the phase detector and a capacitor coupled to an output of the charge pump. In this system, the slope of the signal at the output of the ramp generator is based on the voltage across the second capacitor. In this system, the excitation generator is provided on an integrated circuit. In this system, the capacitive sensor is further provided on the integrated circuit. In the system, the pulse smoothing circuit includes: a first integrator coupled to an output of the pulse generator; and a second integrator coupled to an output of the first integrator, wherein an output of the second integrator is coupled to an output of the pulse smoothing circuit. The system also includes a phase detector having a first input coupled to the output of the pulse generator and a second input coupled to the output of the integrator, wherein the output of the phase detector is configured to control the slope of the first integrator. In the system, the first integrator includes a plurality of current sources coupled to an integration capacitor, and controlling the slope of the first integrator includes adjusting the currents of the plurality of current sources based on the output of the phase detector. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

Advantages of some implementations of the first embodiment include: the ability to reduce the ringing effect when the capacitive MEMS is enabled via the excitation signal. The amount of ringing is a function of the excitation signal waveform and the resonant frequency of the MEMS capacitor.

Second embodiment

In a second embodiment, the ADC 106 changes its internal sampling clock signal by pseudo-random jitter to mitigate the effects of ringing noise. This pseudo-random jitter is provided by varying the timing of the rising and/or falling edges of the sampling clock from which ADC 106 derives its timing reference. Thus, the systematic ringing error of the output of the sensor 103 is thus converted into a broadband signal, which the filter 108 can suppress, for example by averaging a number of digital samples to form each combined measurement sample.

Fig. 8 illustrates an exemplary ADC 800 that may be used as the ADC 106 of fig. 1 to generate a clock signal with pseudo-random jitter. The ADC 800 includes a variable clock generator 804, a frequency divider 806, and a sampling unit 808.

The variable clock generator 804 generates a variable clock signal having pseudo-random jitter. The variable clock generator 804 switches the period T of the variable clock signal_perTo include the pseudo-random jitter in the variable clock signal. T is_perEqual to the minimum period T_{per_min}Plus at T_perIs adjusted by delta T_per. The sampling unit 808 receives the lower clock frequency through the clock divider 806. For example, in the embodiment of FIG. 8, the clock frequency is lower by an integer N times, such that the effective sampling clock has a duration T_swWherein the duration T is_swΔ T equal to N times_per. In one embodiment, T is selected_swSo that it is randomly sampled for at least one period 1/f_resA MEMS signal having a resonant behavior at an upper extended point in time, wherein f_resIs the resonant frequency of the MEMS sensor, and the resonant period T_resIs f_resThe reciprocal of (c). As an example, for f_res＝5MHz，T_resWhen N is 8 and 200ns, let Δ T_perLess than 200ns (preferably less than 200ns/4), which is still high enough for T to be_swGreater than 200 ns. In alternative embodiments, the clock divider 806 may have another division ratio and/or the MEMS sensor may have a different resonant frequency.

Variable clock generator 804 is controlled in a feedback loop by clock controller 820 to stabilize the average period T of the variable clock signal_{per_avg}. The clock controller 802 is provided with a reference oscillator signal by a reference oscillator 803, which may be an oscillating crystal or any other form of stable electronic oscillator 803. In one embodiment, the clock controller 802 may include a phase-locked loop. In one embodiment, the clock controller 802 may provide the variable clock generator 804 with a period other than the period T_{per_min}Then the frequency is scaled (scale) by the variable clock generator 804. The variable clock generator 804 provides a clock feedback signal to the clock controller 802.

A frequency divider 806 is coupled to the output of the variable clock generator 804 and generates a sampling period T_sam(which is the period T)_perN times) of the sampling clock signal. Thus, the sampling clock signal also includes pseudo-random jitter.

An input of the sampling unit 808 is coupled to an output of the frequency divider 806 to receive the sampling clock signal. The sampling unit 808 also has inputs that receive the two amplified sensor response signals output from the sense amplifier 104 of fig. 1. The sampling unit 808 generates samples by sampling the difference between the sensor response signals, and generates samples every T_samThis sampling is performed in seconds. The samples are quantized in one or more subsequent stages (not shown) of the ADC. In one embodiment, the ADC is a sigma-delta converter and the quantization stage further comprises an additional loop filter that filters the output of the sampling unit.

Fig. 9 shows a graph plotting the relative ringing error at the output of the filter 108 of fig. 1. The maximum period offset of the sampling clock is 150 nanoseconds, 16,384 digital samples are averaged to form each measurement sample, and the switching frequency f is_swIs 5120 kHz. Impulse noise is calculated relative to the non-resonant response signal of the sensor formed by the ideal capacitor. When the resonant frequency is equal to the switching frequency, the relative ringing error is minimized and increases in either direction as the resonant frequency changes away from the switching frequency.

Fig. 10 shows a block diagram of an exemplary variable clock generator 804. The variable clock generator 804 includes a counter 1002, a demultiplexer 1004, an LFSR1006, and an oscillator 1008.

The counter 1002 has a counter reset input that receives a clock control signal from the clock controller 802. In the exemplary embodiment of fig. 10, the frequency of the clock control signal is 160 kHz. The counter 1002 also has a counter clock input coupled to the output of the oscillator 1008 to receive a variable clock signal. In the embodiment of fig. 10, the counter 1002 is a 3-bit counter that generates a counter signal representing a count value that is incremented from 0 to 7 in synchronization with a variable clock signal received from the oscillator 1008, and resets the count value to 0 by a rising edge of a counter reset input. The most significant bit of the counter signal is set to the feedback control signal of the clock controller 802. The feedback control signal is one eighth of the frequency of the variable clock signal.

The demultiplexer 1004 also has an input coupled to the output of the counter 1002 to receive a counter signal. The demultiplexer 1004 turns on or off the binary LFSR enable signal based on the count value. The demultiplexer 1004 also turns on or off the first binary input of the AND gate 1010 based on the value of the counter signal.

In the embodiment of fig. 10, the demultiplexer turns on the LFSR enable signal when the count value is 0 or 1, and otherwise the demultiplexer 1004 turns off the LFSR enable signal. When the count value is 3 or 4, the demultiplexer turns on a first input of the AND gate 1010, otherwise the demultiplexer 1004 turns off the AND gate 1010. Since the count value may take any of the eight possible values, the LFSR enable signal is turned on only during the first quarter of the variable clock cycle of oscillator 1008, AND the first input of AND gate 1010 is turned on only during the next quarter of the oscillator clock cycle.

The LFSR1006 includes an enable input that receives the LFSR enable signal from the demultiplexer 1004. The LFSR1006 also has a reset input that receives a clock control signal from the clock controller 802 as an LFSR reset signal. The LFSR1006 also includes a clock input that receives a variable clock signal output from the oscillator 1008. Based on the LFSR enable signal and the LFSR reset signal, the LFSR1006 generates a pseudo-random sequence that is synchronized with the variable clock signal. In some embodiments, LFSR1006 is a fibonacci LFSR. In other embodiments, LFSR1006 is a galois LFSR. In still other embodiments, any pseudo-random sequence generator known in the art (including a non-linear feedback shift register) may be used in place of LFSR 1006.

In the embodiment of FIG. 10, LFSR1006 is a 17-bit LFSR that outputs an LFSR state signal that represents the two-bit binary state of LFSR 1006. The LFSR1006 provides the LFSR state signal bit by bit to the second binary input of the AND gate 1010. Based on the LFSR state signal AND the signal at the first input of AND gate 1010, AND gate 1010 generates a sequence select signal that is also a two-bit binary sequence.

The clock generator 804 also includes a D flip-flop 1012 that receives the frequency selection signal from the AND gate 1010, AND also receives a variable clock signal output from the oscillator 1008. D flip-flop 1010 also has an output coupled to an input of oscillator 1008, and D flip-flop 1012 provides a frequency selection signal to oscillator 1008 bit by bit in synchronization with the variable clock signal.

Oscillator 1008 generates a variable clock signal where oscillator 1008 varies based on a frequency selection signal provided by D flip-flop 1012. The oscillator 1008 has an oscillation frequency f_oscThe maximum value of which is T_{per_min}The reciprocal of (shown in fig. 8). Based on the two-bit binary value of the frequency selection signal received by the oscillator every two cycles (i.e., every two oscillations) of the variable clock signal, oscillator 1008 frequencies its oscillationRate f_oscEither to its previous value or to reduce its oscillation frequency by a frequency shift. Reducing the frequency of oscillator 1008 by this frequency shift corresponds to going to T_{per_min}Adjustment of the addition period Δ T_perTo obtain the period T of the variable clock signal_per。

In the embodiment of fig. 10, oscillator 1008 changes its frequency shift by an amount that is the inverse of 50 nanoseconds at each switching instant, and the total frequency shift at any instant is a Δ T of 0, 50, 100, or 150 nanoseconds_perThe reciprocal of (c). Maximum frequency f of oscillator_oscEqual to 1280kHz, which corresponds to a minimum period of the variable clock signal of 781.25 nanoseconds.

In one example, the frequency of the variable clock generator 804 is eight times the frequency of the sampling clock (i.e., N-8), such that each period T of the sampling clock is followed by a period T_samThe period frequency shift is determined to be four times. In this case, the oscillator 1008 may sample each period T of the clock signal_samDuring which its frequency shift is changed from the previous value at each of the four switching instants. In this example, the sampling clock period T_samIs eight times the minimum period of the variable clock signal of 781.25 nanoseconds, which is 6250 nanoseconds. In this example, T_samCorresponds to a maximum sampling clock frequency of 160 kHz. When applied to a maximum Δ T of 150 nanoseconds_perAt the corresponding frequency shift, the oscillator has a maximum period T of 931.25 nanoseconds_per. Due to T_samIs the maximum T_perEight times or 7450 nanoseconds, so in this example it corresponds to a maximum sampling clock frequency of 134,2 kHz. Delta T applied over long time intervals_perThe expected average of (d) would be 75 nanoseconds. The desired average is determined by averaging 0, 50, 100 and 150 nanoseconds, which is Δ T_perThe value is chosen pseudo-randomly. Thus, T_perThe desired average of (A) is 856.25 nanoseconds, which in this example corresponds to an average sampling clock period T of 6850 nanoseconds_samAnd an average sampling frequency of 146.0 kHz.

Fig. 11 is a flow chart illustrating an exemplary measurement method. The method starts in step 1102. In step 1104, the sensor 103 generates a response signal in response to the excitation signal. In step 1106, the counter signal is incremented in synchronization with the variable clock signal. In step 1108, an LFSR enable signal is determined based on the count value of the counter signal. In step 1110, an m-bit LFSR state signal is determined in synchronization with the variable clock signal based on the LFSR enable signal and the clock control signal. In step 1112, a frequency selection signal is determined based on the LFSR state signal and the count value of the counter signal.

In step 1114, a flow determination is made based on whether a new frequency shift different from the previous frequency shift is selected by the frequency selection signal. If a new frequency shift has been selected, flow continues in step 1118, where the frequency of the variable clock signal is switched according to the selected frequency shift. Otherwise, flow continues in step 1116 where the last (last) frequency of the variable clock signal is maintained. Then, in either case, the flow continues in step 1120, where the frequency of the variable clock signal is down-converted by a factor of N to obtain the sampling clock signal. In step 1122, the response signal generated by the sensor 103 is sampled according to the sampling clock signal.

In step 1124, a flow determination is made based on whether enough samples have been collected to perform an averaging operation. This necessary number of samples may be based on design settings, for example. If not enough samples are collected, flow continues in step 1125 where another flow determination is made based on whether the clock signal has a rising edge. If no rising edge is detected, flow continues in step 1106.

If a rising edge is detected in step 1125, flow continues in step 1127 and the counter is reset to 0. The flow then continues in step 1108.

If enough samples for averaging are collected in step 1124, then the process continues in step 1126, where the samples are averaged together to arrive at a combined pressure measurement sample. The method then ends in step 118.

According to various embodiments, a circuit or system may be configured to perform a particular operation or action by having hardware, software, firmware, or a combination thereof installed on the system that in operation cause the system to perform the action. One general aspect includes a method of measurement, comprising: generating, by a sensor comprising a microelectromechanical system (MEMS) element, a response signal in response to an excitation signal; generating a sampling clock signal according to the pseudorandom jitter; sampling the response signal according to a sampling clock signal to determine a plurality of digital samples; and combining the plurality of digital samples to form a measurement sample. Other embodiments of this aspect include corresponding circuits and systems configured to perform the actions of the methods.

Implementations may include one or more of the following features. In the method: generating the sampling clock signal includes generating a variable clock signal having a variable clock frequency, which is switched according to a switching frequency; and the period of the sampling clock signal is an integer multiple of the period of the variable clock signal. In the method, the excitation signal comprises a square wave; and the MEMS element comprises a first voltage dependent capacitor. In the method, generating the variable clock signal comprises: an LFSR state signal is generated from a variable clock signal and a reference oscillator signal by a Linear Feedback Shift Register (LFSR). In the method, generating the variable clock signal further comprises: generating a counter signal from the variable clock signal and the reference oscillator signal; generating an LFSR enabling signal according to the counter signal; and generating a frequency selection signal according to the LFSR state signal and the counter signal; generating the LFSR state signal further according to an LFSR enable signal; and generating the variable clock signal is further based on the frequency select signal. In the method, the sensor further comprises a capacitive bridge comprising a first bridge segment and a second bridge segment; the second bridge segment comprises a second voltage-dependent capacitor and a third capacitor; the first bridge segment comprises a first voltage-dependent capacitor and a fourth capacitor; and the response signal comprises an output signal of the first bridge segment and an output signal of the second bridge segment. In the method, combining the plurality of digital samples includes averaging the plurality of digital samples. In the method, the switching frequency is not less than 0.9 times the mechanical resonance frequency of the MEMS element; and the switching frequency is not more than 1.1 times the mechanical resonance frequency of the MEMS element. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

One general aspect includes a measurement circuit comprising a sensor, wherein the sensor comprises a microelectromechanical system (MEMS) element, the circuit configured to: generating a response signal in response to the excitation signal; generating a sampling clock signal according to the pseudorandom jitter; sampling the response signal according to a sampling clock signal to determine a plurality of digital samples; and combining the plurality of digital samples to form a measurement sample. Other embodiments of this aspect include corresponding circuits and systems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. The circuitry is further configured to: a variable clock signal having a variable clock frequency is generated, which is switched according to a switching frequency, wherein a period of the sampling clock signal is an integer multiple of a period of the variable clock signal. In the circuit, the excitation signal comprises a square wave; and the MEMS element comprises a first voltage dependent capacitor. The circuit also includes a Linear Feedback Shift Register (LFSR) configured to generate an LFSR state signal from the variable clock signal and the reference oscillator signal. In the circuit, the circuit is further configured to: generating a counter signal from the variable clock signal and the reference oscillator signal; generating a variable clock signal according to the frequency selection signal; generating an LFSR enabling signal according to the counter signal; and generating a frequency selection signal according to the LFSR state signal and the counter signal; and the LFSR is further configured to generate an LFSR state signal according to the LFSR enable signal. In the circuit, the sensor further comprises a capacitive bridge comprising a first bridge segment and a second bridge segment; the second bridge segment comprises a second voltage-dependent capacitor and a third capacitor; the first bridge segment comprises a first voltage-dependent capacitor and a fourth capacitor; and the response signal comprises an output signal of the first bridge segment and an output signal of the second bridge segment. The circuit is also configured to average the plurality of digital samples to form measurement samples. In the circuit, the switching frequency is not less than 0.9 times the mechanical resonance frequency of the MEMS element; and the switching frequency is not more than 1.1 times the mechanical resonance frequency of the MEMS element. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

One general aspect includes a measurement device comprising: a micro-electro-mechanical system (MEMS) element; an analog-to-digital converter (ADC) coupled to an output of the MEMS element, the ADC including a pseudo-random sequence generator and a first oscillator including an input coupled to an output of the pseudo-random sequence generator; and a filter including an input coupled to the output of the ADC. Other embodiments of this aspect include corresponding circuits and systems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. In the apparatus, the ADC includes a frequency divider coupled between an output of the first oscillator and an input of the filter. The apparatus also includes a square wave generator including an output coupled to an input of the MEMS element, wherein the MEMS element includes a first pressure sensitive capacitor. In the device, the pseudo-random sequence generator further comprises a Linear Feedback Shift Register (LFSR). In the apparatus, the pseudo random sequence generator further comprises a counter and a logical network; the counter includes a counter reset input coupled to the output of the reference oscillator and a counter clock input coupled to the first oscillator output; the LFSR includes an enable input coupled to the output of the counter, an LFSR reset input coupled to the reference oscillator output, an LFSR clock input coupled to the first oscillator output, and an LFSR output coupled to the first oscillator input; and the logic network includes a first logic input coupled to the counter output, a second logic input coupled to the LFSR output, a first drain output coupled to the enable input of the LFSR, and a second logic output coupled to the first oscillator input. The device further comprises a capacitive bridge, wherein the capacitive bridge comprises a first bridge section and a second bridge section, the second bridge section comprising a second voltage dependent capacitor and a third capacitor; the first bridge segment includes a first voltage-dependent capacitor and a fourth capacitor. In the apparatus, the filter comprises a low pass filter. In this device, the device occupies a volume of no more than 10 cubic millimeters. The apparatus may also include a measurement system and method using a micro-electromechanical system with sampling jitter. Embodiments of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

The illustrated embodiment of the invention has the advantage of suppressing narrow-band noise caused by resonance. An exemplary system may use, for example, pseudo-random sampling clock jitter to increase the width of the noise band so that it may be more easily filtered out.

Third embodiment

In a third embodiment, the MEMS pressure sensors are implemented using an array or MEMS pressure sensors having variable dimensions such that each MEMS pressure sensor resonates at a different frequency. Thus, when the MEMS pressure sensor is stimulated by the excitation signal, the amplitude of the howling can be reduced at each time instant because the respective resonant responses are out of phase with each other. This is the case, for example, when the individual resonance signals are added by electrically connecting the sensors in parallel. To reduce ringing caused by the under-damped response of the MEMS pressure sensor, an array of MEMS capacitive pressure sensors is designed. Each capacitive pressure sensor is designed with different dimensions such that the harmonic frequency for each capacitive element is different from the harmonic frequencies of the other capacitive elements in the array. When stimulated with a square wave excitation signal, each capacitive sensor element will ring itself with a different resonant frequency and attenuate harmonics that may initiate an under-damped response of the MEMS pressure sensor.

Fig. 12a shows a schematic diagram of a pressure sensitive MEMS array 1202, and fig. 12b shows a corresponding layout 1204 of a MEMS array comprising 20 parallel connected cells. Each unit cell is designed with different dimensions to have different resonance frequencies. For example, as shown in FIG. 12a, the first MEMS element has a size of 55.050 μm per side, the second MEMS element has a size of 55.500 μm per side, and the last MEMS element has a size of 63.975 μm per side, with an overall spread of +/-7.5%. It should be understood that the example of fig. 12a and 12b is but one of many possible exemplary embodiments. In alternative embodiments of the invention, the total expansion may be different and how the device sizes are allocated may also be different.

Since each MEMS element has a different size, each MEMS element has a different resonant frequency. Thus, each MEMS element rings itself at a different frequency when stimulated by an input excitation signal. At a particular time period, the howling amplitude may be small due to coherent cancellation, or the howling amplitude may be large due to coherent cancellation. In various embodiments, ringing is reduced as compared to components having MEMS of equal size. In one embodiment, the size of the MEMS elements is selected such that the resonant frequency of the MEMS elements is expanded such that the output of the MEMS elements can be measured and/or sampled during an appropriate time period in which coherent cancellation occurs, thereby reducing or minimizing measurement errors due to the howling response of the MEMS elements.

In embodiments of the present invention, the overall geometric spread between the smallest and largest MEMS sensor cells and how the cell sizes are assigned may be determined by performing simulations to determine how the size spread for a particular MEMS sensor cell affects the reduction in ringing of the composite response. In one particular embodiment, the variation in the size of each cell is limited to within 7.5%. However, this is only an example, and in alternative embodiments of the invention, the limit of dimensional change may be greater or less than 7.5%.

FIG. 13 shows a graph of an output waveform using a MEMS pressure sensor, where each MEMS cell has the same size and the same resonant frequency. The horizontal axis represents time in milliseconds, and the vertical axis represents the normalized value of the output. Waveform 1300 shows the normalized amplitude of the output howling when stimulated with a square wave input excitation signal. The waveform represents a large settling time and a large amplitude of the howling.

Fig. 14 shows a graph of an output waveform using a MEMS pressure sensor, in which each unit cell has a different size and the resonance frequency is expanded. The horizontal axis represents time in seconds, and the vertical axis represents the normalized value of the output. As shown, the settling time of the time response is faster than the embodiment represented by the diagram of fig. 13 (where all MEMS elements are the same size). In one embodiment, the ADC may sample the output of the MEMS pressure sensor at a time when the ringing response is minimal. In the particular embodiment represented by the diagram of FIG. 14, such time may comprise, for example, between about 2 μ s and about 4 μ s or between 6 μ s and 9 μ s when the normalized howling response is less than 0.1%. In alternative embodiments, other settling times and sampling periods may be used depending on the particular embodiment and its specifications. Thus, in various embodiments, the MEMS resonant frequency and the sampling clock frequency may vary with time or temperature or supply voltage without losing the advantages of the present exemplary method.

Fig. 15 shows a diagram of the frequency spectrum of an output waveform in which the resonance frequency is spread. The horizontal axis represents frequency and the vertical axis represents amplitude. In one embodiment, the resonant frequency extends between 4.5MHz and 6.2 MHz.

FIG. 16 shows a comparative graph including two analog waveforms representing the ringing of the output voltage. Waveform 1602 is an embodiment that includes an array of MEMS pressure sensors, where each unit cell has the same dimensions and resonant frequency; and waveform 1604 is an embodiment that includes an array of MEMS pressure sensors, where each unit cell has a different size. Waveform 1602 shows a peak-to-peak chirp amplitude of approximately 6uV, and waveform 1604 shows a peak-to-peak chirp amplitude of approximately 1 uV.

According to various embodiments, a circuit or system may be configured to perform a particular operation or action by having hardware, software, firmware, or a combination thereof installed on the system that in operation causes the system to perform the action. One general aspect includes a method for performing measurements using a microelectromechanical system (MEMS) sensor, wherein the MEMS sensor comprises MEMS sensors coupled in a bridge structure, wherein a plurality of the MEMS sensors comprise different resonant frequencies, the method comprising: applying an excitation signal to the first port of the bridge structure, wherein each of the plurality of MEMS sensors is stimulated by the excitation signal; measuring a signal at the second port of the bridge structure; and determining a measurement value based on the measurement signal. Other embodiments of this aspect include corresponding circuits and systems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. In the method, the MEMS sensor comprises a MEMS pressure sensor and the measurement comprises a pressure. In the method, the bridge structure comprises: a first branch having a first set of MEMS sensors coupled to a first capacitor; and a second branch having a second set of MEMS sensors coupled to a second capacitor. In the method, each of the plurality of MEMS sensors includes a different size dimension. In this approach, the size dimension may vary by about +/-7.5%. In this method, the size is uniformly distributed. In the method, measuring the signal at the second port of the bridge structure includes performing an a/D conversion. In this method, the transient response of the bridge structure includes ringing at different resonant frequencies, and the ringing includes a time interval of coherent constructive and a time interval of coherent destructive cancellation. In the method, measuring the signal at the second port of the bridge structure comprises: the signal at the second port of the bridge structure is measured during the interval of coherent cancellation. In the method, measuring the signal further includes sampling the signal during an interval of coherence cancellation. In the method, measuring the signal further comprises performing a/D conversion of the signal during the interval of coherence cancellation. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

One general aspect includes a system, comprising: a micro-electromechanical system (MEMS) sensor array comprising a bridge comprising a first bridge section and a second bridge section, wherein the first bridge section comprises a first pressure sensitive MEMS sensor coupled to a first reference MEMS capacitor, wherein the first pressure sensitive MEMS sensor comprises a first array of a plurality of MEMS sensors having different resonant frequencies. Other embodiments of this aspect include corresponding circuits and systems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. In the system, the second bridge section includes a second pressure sensitive MEMS sensor coupled to a second reference MEMS capacitor, where the second pressure sensitive MEMS sensor includes a second array of a plurality of MEMS sensors having different resonant frequencies. In the system, a plurality of MEMS sensors of a first array are coupled in parallel; and the plurality of MEMS sensors of the second array are coupled in parallel. In this system, the first array of the plurality of MEMS sensors is rectangular. In this system, the plurality of MEMS sensors of the first array all have different sizes. In this system, the different dimensions include different lengths. In this system, the different lengths may have a variation of about +/-7.5%. The system further comprises: an excitation generator having an output coupled to a first port of the MEMS sensor array; and a measurement circuit having an input coupled to the second port of the MEMS sensor array. In this system, the measurement circuit includes an a/D converter. The system also includes a filter coupled to an output of the a/D converter. In this system, the filter comprises a low pass filter. Implementations of the described techniques may include hardware, methods or processes, or computer software on a computer-accessible medium.

Advantages of some embodiments include the ability to reduce the effect of ringing when capacitive MEMS arrays are designed with MEMS cells of different sizes and resonant frequencies.

Fourth embodiment

In a fourth embodiment, an oversampling analog-to-digital converter (ADC) is used to monitor the output of the MEMS sensor. To mitigate idle tones in the oversampling ADC, a jittered clock is used to operate the oversampling ADC. In some embodiments, the jittered clock signal may be generated according to the second embodiment disclosed herein.

According to various embodiments, the conversion of the analog signal into the digital domain is performed using a sigma-delta type analog-to-digital converter (ADC). Each exemplary sigma-delta ADC includes a feedback and a reference power supply. For pressure sensing, the measurement signal is typically at a very low frequency near DC. For example, pressure sensing may measure an input signal of 0 to 10 Hz. The inventors have determined that idle tones present in a sigma-delta type ADC interact multiplicatively with noise in the reference voltage source to produce an error component at DC. In various embodiments, a sigma-delta type ADC is provided with a dither clock to spread the noise component and reduce or remove the error component at DC. In such embodiments, the jittered clock is used as a system clock for interface circuitry, where the interface circuitry includes, for example, a voltage supply circuit, a sigma-delta ADC, an output filter, or other components.

FIG. 17 illustrates a system block diagram of an exemplary MEMS pressure sensor system 1700, which includes a MEMS pressure sensor 1702, an output circuit 1704, a dithered clock 1706, and a power circuit 1708. According to eachIn an embodiment, MEMS pressure sensor 1702 measures physical pressure P_MESConversion to analogue signal A_MES. MEMS pressure sensor 1702 receives a voltage reference V from power supply circuit 1708_REFThe power supply circuit 1708 may generate a switching voltage as the voltage reference V based on the jittered clock signal CLK from the jittered clock 1706_REF. Power supply circuit 1708 may also provide a voltage reference V to output circuit 1704_REF. Output circuit 1704 receives analog signal A_MESAnd based on the analogue signal A_MESGenerating a digital pressure signal D_MES。

In various embodiments, output circuit 1704 operates to amplify, convert, and filter analog signal A_MESTo generate a digital pressure signal D_MES. In such an embodiment, the output circuit includes a sigma-delta type ADC that converts the analog signal a based on a sampling time controlled by a jittered clock signal CLK from a jittered clock 1706_MESConversion to digital pressure signal D_MES. Operation of a jittered clock based sigma-delta ADC may reduce or eliminate the passing of idle tones and voltage reference V from the sigma-delta ADC_REFA DC noise component generated by the interference of the noise in (b).

Fig. 18 shows a schematic block diagram of yet another exemplary MEMS pressure sensor system 1800 that includes a differential output MEMS pressure transducer 1802 and an Application Specific Integrated Circuit (ASIC)1803, the ASIC1803 further including an output circuit 1804, a dithered clock 1806, and a voltage reference supply 1808. The output circuit 1804 includes an amplifier 1810, an incremental sigma-delta ADC 1812, and a decimation filter 1814. MEMS pressure sensor system 1800 may be an implementation of MEMS pressure sensor system 1700 described above with reference to fig. 17.

According to various embodiments, a differential output capacitive MEMS pressure transducer 1802 converts a physical pressure signal to a differential analog output (including analog signals A + and A-). Differential output capacitive MEMS pressure transducer 1802 includes variable capacitive structures 1820 and 1824 connected to reference capacitive structures 1822 and 1826 as capacitive bridges, outputting analog signals a + and a-from the center node of each branch of the capacitive bridge as shown. In such embodiments, reference capacitive structure 1822 and reference capacitive structure 1826 may be formed from conductive structures separated by dielectric spacers, i.e., forming parallel plates, where the separation of the conductive structures is fixed and does not change in response to pressure changes. The variable capacitive structure 1820 and the variable capacitive structure 1824 are formed from conductive structures separated by a separation distance, wherein the separation of the conductive structures is dependent on a pressure applied to the conductive structures. For example, variable capacitive structure 1820 and variable capacitive structure 1824 may each include a deflectable diaphragm formed over a sealed cavity over a substrate with a conductive extension region under the diaphragm. In such embodiments, the diaphragm of the variable capacitance structure may deflect due to a pressure differential between the outer surface and the sealed cavity. This deflection affects the capacitance between the diaphragm and the conductive extension region, which is measured at the electrical contacts of the diaphragm and the conductive diffusion region. The reference capacitive structure 1822 and the reference capacitive structure 1826 may each have a similar structure, with the cavities filled with a dielectric spacer material. In other embodiments, many types of capacitive pressure sensors may be used for the differential output capacitive MEMS pressure transducer 1802, including, for example, a capacitive comb drive structure, a multiple plate release capacitor plate structure, or other capacitive MEMS structure.

In various embodiments, a differential analog output comprising analog signals a + and a-is provided from the differential output capacitive MEMS pressure transducer 1802 to an amplifier 1810, which amplifier 1810 amplifies the differential signal and provides an amplified analog electrical signal proportional to the measured physical pressure to an incremental sigma-delta ADC 1812. In other embodiments, the incremental sigma-delta ADC 1812 may be any type of sigma-delta ADC. In a particular embodiment, the incrementing sigma-delta ADC 1812 operates for a set duration or a number of samples before ending the operation, and is therefore referred to as incrementing. Such an embodiment may reduce power consumption. The incremental sigma-delta ADC 1812 begins operation (e.g., waking up) at a fixed time delay or in response to pressure changes above a threshold level. In some embodiments, the ADC 1812 powers up for a certain period of time, e.g., as determined by a target accuracy setting, and then turns off until the next conversion is requested.

In various embodiments, the incremental sigma-delta ADC 1812 operates according to the dithered clock signal CLK to generate a digital output signal proportional to an input amplified analog electrical signal proportional to the measured physical pressure from the differential output capacitive MEMS pressure transducer 1802. The incremental sigma-delta ADC 1812 includes a feedback mechanism that continuously adjusts the digital output signal. Further description of two exemplary sigma-delta ADCs is provided below with reference to fig. 21a and 21 b.

In various embodiments, the digital output signal from the incremental sigma-delta ADC 1812 may have a high bit rate. For example, the incremental sigma-delta ADC 1812 may include a sampling rate on the order of 1000 or 10000 times higher than the expected sampling rate, i.e., an oversampling rate. In a particular embodiment, the incremental sigma-delta ADC 1812 may output a digital signal based on a sampling rate of 160kHz (which corresponds to 160,000 samples per second). For such a system, the expected digital output signal may be only 10 Hz. In such an embodiment, decimation filter 1814 reduces the 160kHz signal to 10Hz and outputs a digital output signal D at a frequency of 10Hz_OUTWhich is proportional to the measured physical pressure signal from the differential output capacitive MEMS pressure transducer 1802. Thus, the decimation filter 1814 reduces the bit rate by a factor of 16,000. In other embodiments, decimation filter 1814 may reduce the bit rate by other factors.

According to various embodiments, the dither clock 1806 provides a dither clock signal CLK to the incremental sigma-delta ADC 1812 for controlling a sampling rate of the sigma-delta ADC. In various embodiments, the jittered clock signal CLK may also be provided to a voltage reference supply 1808, an amplifier 1810, or a decimation filter 1814. The jittered clock 1806 generates a jittered clock signal CLK having a jittered or random period. Typically, the clock signal is generated with a fixed or constant period, e.g., comprising a constant ringing or logic high duration and a constant falling or logic low duration. In the case of jittered clock 1806, the rising or logic high duration and the falling or logic low duration are adjusted. In such embodiments, the adjustment of the jittered clock signal CLK may be random or pseudo-random. Thus, the jittered clock signal CLK is generated to intentionally include a large number of clock beats with varying rising or logic high durations or varying falling or logic low durations.

In various embodiments, the voltage reference supply 1808 provides a voltage reference V to the incremental sigma-delta ADC 1812_REFThereby providing power to the ADC. The voltage reference power supply 1808 may also provide a reference voltage to the differential output capacitive MEMS pressure transducer 1802 to bias the capacitive structure. Specifically, voltage reference supply 1808 provides a positive reference voltage V + and a negative reference voltage V-. In some embodiments, the voltage reference power supply 1808 provides a pulsed reference voltage to the differential output capacitive MEMS pressure transducer 1802. In such an embodiment, the voltage reference power supply 1808 may include a cutoff switch to switch the reference voltage provided to the differential output capacitive MEMS pressure transducer 1802.

In various embodiments, the differential output capacitive MEMS pressure transducer 1802 and the ASIC1803 are formed on separate wafers or dies. In other embodiments, the differential output capacitive MEMS pressure transducer 1802 and the ASIC1803 are formed on the same wafer or die, such as a single Integrated Circuit (IC) die.

Fig. 19a and 19b show waveform diagrams of exemplary noise signals generated in a sigma-delta ADC. Fig. 19a shows a plot 1901 plotting a Fast Fourier Transform (FFT) illustrating the output of a sigma-delta type ADC, which also shows idle tones 1912 and 1914 at approximately 65kHz and 95kHz, respectively. As shown, the idle tone is an unwanted bit sequence at a particular frequency in the output that is not based on the input signal of the sigma-delta ADC. These idle tones are a product of the feedback mechanism in sigma-delta ADCs and may be present when the input signal to the ADC is not "busy". In some cases, idle tones may be particularly strong when the ADC uses a one-bit (2-level) equalizer. This case is applicable to an exemplary pressure sensor that uses a sigma-delta ADC with a one-bit equalizer for good linearity.

Fig. 19b shows a curve 1900 representing the number of bits (which represents the approximation accuracy) of a sigma-delta type ADC. Curve 1900 is generated by introducing an unwanted sine wave at the reference voltage of the sigma-delta ADC to determine the robustness of the sigma-delta ADC with respect to supply voltage ripple. The frequency of the sine wave is from 0 to 160kHz and the integrated noise is measured. Thus, the ADC is most sensitive to interference at Vref (which has the same frequency as the idle tone). Since the idle tone frequency varies with the DC input level of the sigma-delta ADC (which in some embodiments may be the measured pressure from the MEMS sensor), the sensitivity to interference varies with the ADC input. This behavior is problematic in some embodiments because it is difficult to predict whether an idle tone causes a problem in a particular application. In this exemplary sigma-delta type ADC, there is a dramatic decrease in accuracy of approximately 65kHz and 95kHz at points 1902 and 1904 corresponding to the frequencies of idle tones 1912 and 1914, respectively, shown in fig. 19 a.

As briefly described above, the inventors have determined that idle tones present in a sigma-delta ADC interact multiplicatively with noise in the reference voltage supply to produce an error component at DC. Accordingly, idle tones 1902 and 1904 may interact with noise in the reference voltage provided to the sigma-delta ADC. As described above with reference to fig. 17 and 18, various embodiments include providing a jittered clock signal CLK to a sigma-delta ADC. In such embodiments, the spikes of the idle tones (such as idle tones 1902 and idle tones 1904) are spectrally dispersed and the noise component at DC that is generated from the idle tones in combination with noise in the reference voltage supply is reduced or removed. An exemplary noise curve is shown in fig. 20.

Fig. 20 shows waveform diagrams of noise signals generated in the sigma-delta ADC without a jittered clock (curve 2000) and with a jittered clock (curve 2001), showing a comparison of the resolution with respect to the reference voltage disturbance frequency. As shown, an exemplary sigma-delta ADC with a standard non-jittered clock (curve 2000) exhibits a loss of performance at points 2002 and 2004 corresponding to idle tones 1912 and 1914 shown in fig. 19 above. However, an exemplary sigma-delta ADC with a jittered clock does not substantially lose noise performance at idle tone frequencies. As shown in curve 2001, the loss of noise performance is significantly reduced at 65kHz and 95kHz idle tone frequencies. In such embodiments, clock jitter may spread the spectrum of the idle tones and reduce or remove the DC noise component generated by the multiplicative interaction of the idle tones and noise in the reference voltage supply.

Fig. 21a and 21b show schematic block diagrams of exemplary sigma- delta ADCs 2100 and 2101. Fig. 21a shows a discrete-time sigma-delta ADC 2100 comprising a sampling switch 2102, a loop filter 2104, a comparator 2106, a digital-to-analog converter (DAC)2108 and an adder 2110. According to various embodiments, a discrete-time sigma-delta ADC 2100 receives an analog input signal a at a sampling switch 2101_IN. Analog input signal A_INMay be an amplified analog signal received from an amplifier coupled to the capacitive MEMS pressure transducer, such as from amplifier 1810 described above with reference to fig. 18. Thus, the analog input signal A_INFor example, may be proportional to the measured physical pressure signal.

In various embodiments, the sampling switch 2102 is controlled by a jittered clock signal CLK, which may be provided from the jittered clock 1706 or 1806 described above with reference to fig. 17 and 18. The jittered clock signal CLK causes the sampling switch 2102 to turn on and off according to a sampling rate equal to the frequency of the jittered clock signal CLK. Thus, sampling switch 2102 generates sampled analog input signal SA_INAnd is supplied to the loop filter 2104 via the adder 2110. By sampling, the analog signal is no longer a continuous signal, but a discretely sampled analog input signal SA_IN. In such an embodiment, loop filter 2104 may be implemented as a Low Pass Filter (LPF) to remove high frequency components. In some embodiments, loop filter 2104 is implemented as an integrator.

According to various embodiments, after filtering in the loop filter 2104, the sampled and filtered analog input signal is provided to a comparator 2106, which comparator 2106 compares the input signal to a threshold value. For example, the threshold may be 0V. Based on the comparison, the comparator 2106 provides a digital output signal D_OUT. Digital output signal D_OUTThe bit stream of (1) and the analog input signal A_INAnd (4) in proportion. Further, a digital output signal D is provided by DAC 2108_OUTReturning to adder 2110. In such an embodiment, the voltage reference V is provided to DAC 2108_REFSuch as from above with reference to fig. 17 and fig. 18, or a voltage reference supply 1808.

As discussed herein, the feedback loop of some sigma-delta ADCs may generate an idle tone and at a voltage reference V_REFMay be present with noise. In such an ADC, these two error sources may be combined by DAC multiplication to form a DC error component. In various embodiments, the introduction of the jittered clock signal CLK from the jittered clock spreads the frequency of the idle tones and reduces or removes the DC error component. Adder 2110 compares the recovered analog output of DAC 2108 with the sampled analog input SA_INCombine to provide feedback for improved performance.

Fig. 21b shows a continuous-time sigma-delta ADC 2101 comprising a loop filter 2105, a clocked comparator 2112, a DAC 2108 and an adder 2110. According to various embodiments, the continuous-time sigma-delta ADC 2101 operates as described above with reference to the discrete-time sigma-delta ADC 2100 of fig. 21a, with the sampling switch 2102 removed, the loop filter 2104 replaced with a loop filter 2105, and the comparator 2106 replaced with a clocked comparator 2112. In such an embodiment, the analog input signal a is provided by adder 2110_INIs provided to the loop filter 2105. Loop filter 2105 may operate as described with reference to loop filter 2104, but is configured to receive analog input signal A_INInstead of sampling the analog input signal SA_INOf the discrete sampled signals.

In various embodiments, the timing comparator 2112 compares the filtered analog input signal with a threshold voltage and provides a conversion result using the jittered clock signal CLK to generate the digital output signal D_OUT. In some embodiments, the threshold voltage may be 0 volts, VDD/2, and/or other threshold voltages. In such an embodiment, the jittered clock signal CLK determines the sampling rate of the continuous-time sigma-delta ADC 2101. The DAC 2108 provides feedback through the adder 2110, as described above with reference to fig. 21 a.

In such an embodiment, providing the jittered clock signal CLK to the timing comparator 2112 provides the same advantages as described above with reference to the jittered clock signal CLK in the other figures.

Fig. 22 shows a block diagram of an exemplary method of operation 2200 for a sensor. The method of operation 2200 includes steps 2202 + 2212. According to various embodiments, step 2202 comprises converting the pressure signal to an electrical signal. The pressure signal may be measured and converted using a capacitive MEMS pressure transducer. Step 2204 includes generating an amplified electrical signal by amplifying the electrical signal. For example, the electrical signal may be amplified by a differential input amplifier. Step 2206 includes generating a jittered clock signal. In such an embodiment, a jittered clock is included in the sensor system to generate a jittered clock signal. In step 2207, a reference voltage is provided to the sigma-delta ADC.

According to various embodiments, step 2208 includes converting the amplified electrical signal to a digital signal using a sigma-delta ADC, wherein the sigma-delta ADC is operated with a sampling time controlled by the jittered clock signal generated in step 2206. In various embodiments, steps 2202-2208 may be reconfigured and performed in other orders, and the method of operation 2200 may be modified to include additional steps.

According to various embodiments, a circuit or system may be configured to perform a particular operation or action by having hardware, software, firmware, or a combination thereof installed on the system that in operation causes the system to perform the action. According to one embodiment, a sensor circuit includes: a sigma-delta type analog-to-digital converter (ADC) configured to be coupled to a low frequency transducer; a dither clock coupled to the sigma-delta ADC and configured to control the sigma-delta ADC based on the dither clock signal; and a supply voltage circuit coupled to the sigma-delta ADC. Other embodiments include corresponding systems and apparatuses, each configured to perform various exemplary methods.

In various embodiments, the sensor circuit further comprises a low frequency transducer, wherein the low frequency transducer comprises a microelectromechanical system (MEMS) pressure transducer. In some embodiments, the MEMS pressure transducer comprises a differential capacitance MEMS pressure transducer. In still other embodiments, the sensor circuit further comprises an amplifier having an input coupled to the MEMS pressure transducer and an output coupled to the sigma-delta ADC. In such embodiments, the amplifier may comprise a differential input amplifier having a first input coupled to the first capacitive structure of the MEMS pressure transducer and a second input coupled to the second capacitive structure of the MEMS pressure transducer.

In various embodiments, the supply voltage circuit is further coupled to a jittered clock and configured to operate based on the jittered clock signal. In some embodiments, the supply voltage circuit includes a clocked charge pump circuit configured to operate based on a jittered clock signal.

In various embodiments, the sensor circuit includes a filter coupled to a sigma-delta ADC. The filter may comprise a decimation filter. In some embodiments, the sigma-delta ADC comprises a sampling sigma-delta ADC including a sampling switch having a sampling time based on a jittered clock signal, a loop filter coupled to the sampling switch, a comparator coupled to the loop filter, and a negative feedback loop coupled to inputs of the comparator and the loop filter. In still other embodiments, the sigma-delta ADC comprises a continuous-time sigma-delta ADC comprising a loop filter, a clocked comparator coupled to the loop filter and configured to operate based on a dithered clock signal, and a negative feedback loop coupled to inputs of the clocked comparator and the loop filter.

According to one embodiment, a method of operating a sensor comprises: receiving an electrical signal from a low frequency transducer; generating a jittered clock signal; and converting the electrical signal to a digital signal using a sigma-delta analog-to-digital converter (ADC) operating based on the dithered clock signal. Other embodiments include corresponding systems and apparatuses, each configured to perform various exemplary methods.

In various embodiments, the method further comprises: the pressure signal is converted to an electrical signal at the MEMS pressure transducer. The method may further comprise: the electrical signal is amplified before being converted into a digital signal. In some embodiments, the method further comprises: generating a reference voltage; and providing the reference voltage to the sigma-delta ADC. In such embodiments, generating the reference voltage may include clocking the charge pump circuit using a jittered clock signal.

In various embodiments, the method further comprises filtering the digital signal using a decimation filter. In some embodiments, converting the electrical signal to a digital signal using a sigma-delta ADC operating based on a jittered clock signal comprises: generating a sampling signal by sampling the electrical signal with a sampling time based on the jittered clock signal; generating a filtered signal by filtering the sampled signal at a loop filter; generating a digital signal by comparing the filtered signal with a threshold; and generating a feedback signal for the loop filter based on the digital signal.

In various embodiments, converting an electrical signal to a digital signal using a sigma-delta ADC operating based on a jittered clock signal comprises: generating a filtered signal by filtering the electrical signal at a loop filter; generating a digital signal by comparing the filtered signal with a threshold value based on the jittered clock signal; and generating a feedback signal for the loop filter based on the digital signal.

According to one embodiment, a microelectromechanical system (MEMS) capacitive pressure sensor system includes a differential output MEMS capacitive pressure sensor including a first reference capacitive structure, a first variable capacitive structure configured to vary a first capacitance value with reference to a first pressure signal, a first output coupled between the first reference capacitive structure and the first variable capacitive structure, a second reference capacitive structure, a second variable capacitive structure configured to vary a second capacitance value with reference to a second pressure signal, and a second output coupled between the second reference capacitive structure and the second variable capacitive structure. The MEMS capacitive pressure sensor system further comprises: a differential amplifier coupled to first and second outputs of the differential output MEMS capacitive pressure sensor; a sigma-delta type analog-to-digital converter (ADC) coupled to an output of the differential amplifier; and a jittered clock coupled to the sigma-delta ADC and configured to control a sampling time of the sigma-delta ADC using the jittered clock signal. Other embodiments include corresponding systems and apparatuses, each configured to perform various exemplary methods.

In various embodiments, the MEMS capacitive pressure sensor system further comprises a supply voltage circuit coupled to the sigma-delta ADC. In some embodiments, the supply voltage circuit includes a charge pump circuit coupled to the jittered clock and configured to be clocked with the jittered clock signal. In still other embodiments, the sigma-delta ADC comprises an incremental sigma-delta ADC. In additional embodiments, the first pressure signal and the second pressure signal are the same pressure signal.

In some embodiments, a MEMS pressure transducer having a sigma-delta ADC that operates according to a dithered clock signal generated by a dithered clock is particularly advantageous. In this particular embodiment, as described above, absolute pressure measurements or very low frequency pressure measurements are particularly affected by DC noise from idle tones and reference voltage noise. Thus, these particular embodiments include the advantage of reducing noise or reducing error components at DC and very low frequency measurements, which results in improved sensitivity or greater resolution.

Still further advantages of some embodiments include having a very robust sensor that is less affected by interference at the sensor supply node, particularly not affected by tone interference having the same or similar frequency as the ADC idle tone.

While the invention has been described with reference to the illustrated embodiments, the description is not intended to be limiting. Various modifications and combinations of the illustrative embodiments, as well as other embodiments of the invention, will be apparent to persons skilled in the art upon reference to the description. It is therefore intended that the appended claims encompass any such modifications or embodiments.

Claims (23)

1. A sensor circuit, comprising:

a sigma-delta type analog-to-digital converter configured to be coupled to a microelectromechanical systems MEMS pressure transducer;

a jittered clock generator coupled to the sigma-delta type analog-to-digital converter, the jittered clock generator configured to generate a jittered clock signal and configured to control the sigma-delta type analog-to-digital converter based on the jittered clock signal; and

a supply voltage circuit having an input coupled to the dithered clock generator, a first output configured to be coupled to a capacitive structure of the MEMS pressure sensor, and a second output configured to be coupled to a reference voltage input of the sigma-delta type analog-to-digital converter, the supply voltage circuit configured to apply a pulsed reference voltage to the capacitive structure of the MEMS pressure transducer as a function of the dithered clock signal.

2. The sensor circuit of claim 1, further comprising the MEMS pressure transducer.

3. The sensor circuit of claim 2, wherein the MEMS pressure transducer comprises a differential capacitance MEMS pressure transducer.

4. The sensor circuit of claim 2, further comprising: an amplifier having an input coupled to the MEMS pressure transducer and an output coupled to the sigma-delta type analog-to-digital converter.

5. The sensor circuit of claim 4, wherein the amplifier comprises: a differential input amplifier having a first input coupled to a first capacitive structure of the MEMS pressure transducer and a second input coupled to a second capacitive structure of the MEMS pressure transducer.

6. The sensor circuit of claim 1, wherein the supply voltage circuit comprises a clocked charge pump circuit configured to operate based on the jittered clock signal.

7. The sensor circuit of claim 1, further comprising a filter coupled to the sigma-delta type analog-to-digital converter.

8. The sensor circuit of claim 7, wherein the filter comprises a decimation filter.

9. The sensor circuit of claim 1, wherein the sigma-delta type analog-to-digital converter comprises a sampling sigma-delta type analog-to-digital converter comprising:

a sampling switch having a sampling time based on the jittered clock signal;

a loop filter coupled to the sampling switch;

a comparator coupled to the loop filter; and

a negative feedback loop coupled to the comparator and an input of the loop filter.

10. The sensor circuit of claim 1, wherein the sigma-delta type analog-to-digital converter comprises a continuous-time sigma-delta type analog-to-digital converter comprising:

a loop filter;

a timing comparator coupled to the loop filter and configured to operate based on the jittered clock signal; and

a negative feedback loop coupled to the timing comparator and an input of the loop filter.

11. A method of operating a sensor, the method comprising:

receiving an electrical signal from a microelectromechanical system (MEMS) pressure transducer;

generating a jittered clock signal;

generating a pulse reference voltage according to the jittered clock signal;

applying the pulsed reference voltage to a capacitive structure of the MEMS pressure transducer; and

converting the electrical signal to a digital signal using a sigma-delta type analog-to-digital converter operating based on the jittered clock signal.

12. The method of claim 11, further comprising: converting, at the MEMS pressure transducer, a pressure signal into the electrical signal.

13. The method of claim 12, further comprising: amplifying the electrical signal before converting the electrical signal into the digital signal.

14. The method of claim 11, further comprising:

generating a reference voltage; and

providing the reference voltage to the sigma-delta type analog-to-digital converter.

15. The method of claim 14, wherein generating the reference voltage comprises: the charge pump circuit is clocked using the jittered clock signal.

16. The method of claim 11, further comprising: a decimation filter is used to filter the digital signal.

17. The method of claim 11, wherein converting the electrical signal to the digital signal using the sigma-delta type analog-to-digital converter operating based on the jittered clock signal comprises:

generating a sampling signal by sampling the electrical signal with a sampling time based on the jittered clock signal;

generating a filtered signal by filtering the sampled signal at a loop filter;

generating the digital signal by comparing the filtered signal to a threshold; and

generating a feedback signal for the loop filter based on the digital signal.

18. The method of claim 11, wherein converting the electrical signal to the digital signal using the sigma-delta type analog-to-digital converter operating based on the jittered clock signal comprises:

generating a filtered signal by filtering the electrical signal at a loop filter;

generating the digital signal by comparing the filtered signal to a threshold based on the jittered clock signal; and

based on the digital signal, a feedback signal for the loop filter is generated.

19. A microelectromechanical systems MEMS capacitive pressure sensor system, comprising:

a differential output MEMS capacitive pressure sensor, comprising:

a first reference capacitor structure having a first reference capacitance,

a first variable capacitance structure configured to change a first capacitance value with reference to a first pressure signal,

a first output coupled between the first reference capacitance structure and the first variable capacitance structure,

a second reference capacitor structure having a first reference capacitance,

a second variable capacitance structure configured to change a second capacitance value with reference to a second pressure signal, an

A second output coupled between the second reference capacitive structure and the second variable capacitive structure;

a differential amplifier coupled to the first and second outputs of the differential output MEMS capacitive pressure sensor;

a sigma-delta type analog-to-digital converter coupled to an output of the differential amplifier;

a jittered clock generator coupled to the sigma-delta type analog-to-digital converter, the jittered clock generator configured to generate a jittered clock signal and configured to control a sampling time of the sigma-delta type analog-to-digital converter using the jittered clock signal; and

a supply voltage circuit having an input coupled to the jittered clock generator, a first output coupled to the first variable capacitance structure and the second variable capacitance structure, the supply voltage circuit configured to generate a pulsed reference voltage at the first output based on the jittered clock signal.

20. The microelectromechanical systems MEMS capacitive pressure sensor system of claim 19, wherein the supply voltage circuit further comprises a second output coupled to the sigma-delta type analog-to-digital converter.

21. The MEMS capacitive pressure sensor system of claim 20, wherein the supply voltage circuit comprises a charge pump circuit coupled to the jittered clock and configured to be clocked with the jittered clock signal.

22. The microelectromechanical systems MEMS capacitive pressure sensor system of claim 21, wherein the sigma-delta type analog-to-digital converter comprises an incremental sigma-delta type analog-to-digital converter.

23. The MEMS capacitive pressure sensor system of claim 19, wherein the first pressure signal and the second pressure signal are the same pressure signal.

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