CN105991490B

CN105991490B - Signal sending and receiving method, system and device based on filter bank

Info

Publication number: CN105991490B
Application number: CN201510083155.0A
Authority: CN
Inventors: 喻斌; 孙鹏飞; 朱大琳
Original assignee: Beijing Samsung Telecommunications Technology Research Co Ltd; Samsung Electronics Co Ltd
Current assignee: Beijing Samsung Telecommunications Technology Research Co Ltd; Samsung Electronics Co Ltd
Priority date: 2015-01-12
Filing date: 2015-02-15
Publication date: 2020-07-10
Anticipated expiration: 2035-02-15
Also published as: EP3245772A1; EP3245772A4; CN105991490A

Abstract

The application discloses a signal sending method based on a filter bank, which comprises the following steps: a transmitter acquires pre-equalization configuration information indicating whether to start pre-equalization; when the pre-equalization configuration information indicates that pre-equalization is started, the transmitter generates pre-equalization parameters according to a pre-equalization mode, performs pre-equalization operation on a transmission signal according to the pre-equalization parameters, and then performs filter bank-based modulation. The application also discloses a filter bank-based signal receiving method, and a corresponding transmitter, a receiver, a communication method and a communication system. By the method and the device, flexibility can be provided, and performance can be guaranteed under the condition of not cutting tailing when network load is high, so that the spectrum efficiency of the FBMC system is maximized, influence of tailing can be avoided by changing the configuration of the subframe when the network load is low, and the user service quality can be guaranteed as much as possible.

Description

Signal sending and receiving method, system and device based on filter bank

Technical Field

The present application relates to the field of wireless communication technologies, and in particular, to a method, a system, and an apparatus for transmitting and receiving a signal based on a filter bank.

Background

With the rapid development of the information industry, and in particular the growing demand from the mobile internet and internet of things (IoT), future mobile communication technologies present unprecedented challenges, as reported ITU-R M [ imt. beyond 2020.TRAFFIC ] according to the international telecommunications union ITU, it is expected that mobile TRAFFIC growth will grow nearly 1000 times in the year 2010 (4G era), the number of user device connections will also exceed 170 billion, as massive IoT devices gradually penetrate into mobile communication networks, the number of connected devices will be more dramatic by 2020, to cope with this unprecedented challenge, the communications industry and academia have developed a broad fifth generation mobile communication technology research (5G) facing the 2020 era, the ITU report ITU-R M [ imt. VISION ] has now discussed the framework and overall goals of the future 5G, with the need for 5G, application scenarios and important performance indicators, and detailed statements of the IoT-5 [ IMT. VISION ] in discussing the framework and overall goals of the detailed reporting of the ITU-3683, the detailed requirements of the ITU-365G, the detailed performance indicators, detailed reports of the ITU-3683, the relevant spectrum-supporting the relevant system experiences, the issues of supporting the consistent and the relevant latency.

Modulation waveform and multiple Access are important bases of Wireless communication air interface (air-interface) design, and are not exceptional in 5G, currently, OFDM is widely applied to broadcast Audio and Video fields and civil communication systems, such as long Term Evolution (L on Term Evolution, L TE) systems corresponding to the third generation mobile Partnership Project (3 GPP) protocol, Digital Video Broadcasting (DVB, european) and Digital Audio Broadcasting (Digital Video Broadcasting, DVB) and Digital Audio Broadcasting (Digital Broadcasting, DAB), high speed Digital transmission (DAB), high speed Digital band coding (OFDM) and Digital Audio Broadcasting (OFDM) systems, which are well known to be capable of achieving Wireless transmission by making use of OFDM Frequency domain Modulation, OFDM Frequency Division Multiplexing (OFDM) and OFDM Frequency domain Modulation, so that the Wireless broadband Wireless transmission (OFDM) system can achieve high Frequency bandwidth Modulation and Frequency domain Modulation by making use of OFDM Frequency domain Modulation, OFDM Frequency Division, Modulation Frequency Division, Modulation, and demodulation of multiple Access to make it greatly reduce Interference, and demodulation of OFDM, so that Wireless broadband Wireless transmission bandwidth, Wireless transmission, Wireless broadband Wireless transmission, Wireless broadband, Wireless transmission, Wireless broadband, Wireless transmission, Wireless broadband, Wireless transmission, Wireless broadband, Wireless transmission, Wireless broadband, Wireless transmission, Wireless broadband, Wireless transmission, Wireless:

(1) the spectrum utilization rate can be greatly reduced in a 5G low-delay transmission scenario by adding the CP to resist the ISI, because the low-delay transmission will greatly shorten the symbol length of the OFDM, and the length of the CP is only limited by the impulse response of the channel, so the ratio of the length of the CP to the symbol length of the OFDM is greatly increased, and such overhead causes a very large loss of spectrum efficiency, which is unacceptable.

(2) Strict time synchronization requirements cause great signaling overhead required by closed-loop synchronization maintenance in a 5G IoT scene, and a strict synchronization mechanism causes a frame structure to be inelastic and cannot well support different synchronization requirements of multiple services.

(3) The OFDM employs rectangular pulse shaping (rectangular pulse) to cause large out-of-band leakage, because such waveform causes the sidelobe roll-off in the frequency domain to be slow, which is why the OFDM is very sensitive to frequency offset (CFO), while 5G will have a lot of requirements for flexible access/sharing of fragmented spectrum, and the out-of-band leakage of the OFDM greatly limits the flexibility of spectrum access or requires a large frequency domain guard band to reduce the spectrum utilization rate.

These disadvantages are mainly caused by their inherent characteristics, and although the influence of these disadvantages can be reduced by taking certain measures, the complexity of system design increases and the problem cannot be solved fundamentally.

Because of this, some new waveform Modulation techniques (based on multi-Carrier Modulation) are considered within the scope of 5G as described in ITU's report ITU-R M. [ imt. simple techon L OGY trees ], wherein, a Filter Bank based multi-Carrier (FBMC: Filter Bank Multiple Carrier) Modulation technique becomes one of the hot research objects, because it provides freedom of design of a shaping Filter (protocol Filter), a good Time/frequency focusing (TF L) Filter can be used to pulse-shape a transmission waveform so that a transmission Signal can exhibit a variety of superior characteristics, including no need of CP to combat ISI to improve spectral efficiency, low out-of-band leakage to well support flexible fragmented spectral access, and a good fragmented frequency Offset, a more typical mc system generally uses a technique called Modulation Offset Quadrature Amplitude Modulation (qam) to maximize spectral efficiency, which is also referred to as OFDM symbol qam, and OFDM symbol, which is also referred to the early term qam 3650.

FBMC has some good characteristics that OFDM does not have, and thus has gained attention in 5G research, but its inherent disadvantages make its application in wireless communication systems more challenging, and these challenges that need to be solved are being continuously researched. One significant problem is that the filter used by FBMC causes a long tail effect (tail effect) of the time domain waveform, which is also called transition period problem (transition period problem). When the uplink is transmitted based on the short data block, if the data block length extension contains the tailing to avoid the overlapping of the tailing and other data blocks, the number of symbols transmitted in the effective time is reduced, and the spectrum efficiency is greatly reduced, so that the FBMC is only suitable for long data transmission, on the contrary, if the data block length does not contain the tailing, the tailing part is overlapped with other data blocks, the bad processing can cause great interference, and the spectrum efficiency can be limited. The conventional method is to truncate the tail part so as to avoid overlapping with other data blocks, but truncation of the waveform causes signal distortion and also affects spectral efficiency, and the truncated signal spectrum causes spreading and increases inter-carrier-interference (ICI) interference, and such truncation is not an effective method.

To sum up, to improve the competitiveness of FBMC in 5G candidate technology, we need to solve their own deficiencies besides utilizing and developing their advantages, and for the service mode of scattered short data block transmission (scattered Access) in 5G, especially in IoT scenario, we need to solve the problem caused by the tail effect of FBMC by an effective method.

Disclosure of Invention

The technical problem to be solved by the invention is to solve the tailing problem in an FBMC system during uplink data block-based transmission, and no effective method for reducing the influence caused by tailing exists at present. Therefore, the application provides a communication system based on a filter bank and a signal sending and receiving method thereof, provides effective configurable pre-equalization processing, can ensure the performance under the condition of not cutting off tailing, and maximizes the spectrum efficiency of an FBMC system.

The application provides a signal sending method based on a filter bank, which comprises the following steps:

a transmitter acquires pre-equalization configuration information indicating whether to start pre-equalization;

when the pre-equalization configuration information indicates that pre-equalization is started, the transmitter generates pre-equalization parameters according to a pre-equalization mode, performs pre-equalization operation on a transmission signal according to the pre-equalization parameters, and then performs filter bank-based modulation.

Preferably, the pre-equalization configuration information is included in scheduling control signaling acquired by the transmitter.

Preferably, the transmitter further obtains subframe type information used when pre-equalization is turned on and off from the scheduling control signaling, where the subframe type information indicates the number of transmission symbols in a subframe and the transmission time of a first symbol.

Preferably, the pre-equalization manner is predefined or included in pre-equalization configuration information.

Preferably, the pre-equalization method includes: both amplitude and phase are pre-equalized, or only phase is pre-equalized.

Preferably, when the pre-equalization manner is to pre-equalize both the amplitude and the phase, the generating the pre-equalization parameter according to the pre-equalization manner includes: and generating a pre-equalization parameter according to the acquired channel state information and a pre-equalization mode, wherein the real number or the imaginary part of an equivalent channel frequency response synthesized by the pre-equalization parameter and the channel frequency response approaches to 0, and the modulus of the equivalent channel frequency response approaches to 1.

Preferably, when the pre-equalization manner is to pre-equalize only the phase, the generating the pre-equalization parameter according to the pre-equalization manner includes: and generating a pre-equalization parameter according to the acquired channel state information and a pre-equalization mode, wherein the equivalent channel frequency response synthesized by the pre-equalization parameter and the channel frequency response is that the real number or the imaginary part approaches to 0, and the mode of the equivalent channel frequency response approaches to the mode of the channel frequency response.

Preferably, the channel state information is obtained through channel reciprocity.

Preferably, when the pre-equalization manner is to pre-equalize only the phase, the generating the pre-equalization parameter according to the pre-equalization manner includes: and acquiring a phase indication through feedback of the receiver, and generating pre-equalization parameters according to the acquired phase indication.

Preferably, the transmitter obtains phase information of a quantized channel frequency response according to the phase indication, and generates a frequency phase response coefficient on the allocated bandwidth as a frequency domain pre-equalization parameter according to the phase information.

Preferably, the method further comprises: the frequency domain is divided into a plurality of sub-bands, the phase indication is indicated based on the sub-bands, the number of the sub-bands is more than or equal to 1, and the transmitter generates frequency domain pre-equalization parameters on the allocated bandwidth according to the corresponding relation between the phase indication and the sub-bands.

Preferably, the transmitter obtains the phase information of the quantized channel impulse response according to the phase indication, and transforms the phase information into a corresponding frequency phase response coefficient as a frequency domain pre-equalization parameter.

Preferably, the method further comprises: the quantized phase information of the channel impulse response is the phase information of the strongest path in the multi-path channel, or the phase information synthesized by averaging the phases of the X paths with the strongest energy in the multi-path channel, wherein X is larger than 1.

Preferably, the feedback is obtained by adding a phase indication bit field or a special feedback channel in the uplink scheduling grant signaling.

The application also provides a signal receiving method based on the filter bank, which comprises the following steps:

the receiver judges whether to carry out post-equalization on the demodulation signal based on the filter bank according to the pre-equalization configuration information indicated to the transmitter; the pre-equalization configuration information indicates whether to start pre-equalization;

when the pre-equalization configuration information indicates that pre-equalization is started and the pre-equalization mode is pre-equalization on both amplitude and phase, the receiver does not perform post-equalization on the demodulated signal based on the filter bank;

when the pre-equalization configuration information indicates to start pre-equalization and the pre-equalization mode is to perform pre-equalization only on the phase, the receiver performs post-equalization on the demodulation signal based on the filter bank according to the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response;

and when the pre-equalization configuration information indicates that the pre-equalization is not performed, the receiver performs amplitude and phase post-equalization on the demodulated signal based on the filter bank according to the estimated channel frequency response.

Preferably, the performing post-equalization according to the equivalent channel frequency response synthesized by the pre-equalization parameter and the channel frequency response includes: the receiver directly estimates the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response by receiving the reference signal which adopts the pre-equalization and is used for channel estimation, or the receiver indirectly estimates the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response according to the pre-equalization mode adopted by the transmitter by performing channel estimation on the reference signal which does not adopt the pre-equalization, wherein the equivalent channel frequency response is used for post-equalization of the demodulated signal based on the filter.

The present application further provides a transmitter based on a filter bank, including: control signaling processing module, pre-equalization switch, information source, pre-equalization parameter generation module and send synthesis filter bank, wherein:

the control signaling processing module is used for acquiring pre-equalization configuration information, and the pre-equalization configuration information indicates whether to start pre-equalization;

when the pre-equalization configuration information indicates to perform pre-equalization, the control signaling processing module controls the pre-equalization switch to be connected with the pre-equalization parameter generation module, and a real number signal generated by the information source is modulated by the sending synthesis filter bank after being subjected to pre-equalization processing by the pre-equalization parameter generated by the pre-equalization parameter generation module;

when the pre-equalization configuration information indicates that pre-equalization is not performed, the control signaling processing module controls the pre-equalization switch to be switched on to transmit the synthesis filter bank, and the real number signal generated by the information source is directly modulated by the transmit synthesis filter bank.

Preferably, the pre-equalization configuration information is obtained from the scheduling control signaling by the control signaling processing module.

Preferably, the pre-equalization parameter generating module generates the pre-equalization parameter according to a pre-equalization mode; the pre-equalization method is predefined or included in the pre-equalization configuration information.

The present application further provides a filter bank-based receiver, including: the device comprises a receiving analysis filter bank, a post-equalization switch, a post-equalization parameter generation module and a real part operation module; wherein:

the post-equalization switch judges whether to perform post-equalization on the demodulation signal of the receiving analysis filter bank according to pre-equalization configuration information indicated to the transmitter; the pre-equalization configuration information indicates whether to start pre-equalization;

when the pre-equalization configuration information indicates to start pre-equalization and the pre-equalization mode is to pre-equalize the amplitude and the phase, the post-equalization switch switches on the receiving analysis filter bank and the real part acquiring module;

when the pre-equalization configuration information indicates to start pre-equalization and the pre-equalization mode is only to pre-equalize phases, the post-equalization switch sends the pre-equalization mode to the post-equalization parameter generation module, the receiving analysis filter bank and the post-equalization parameter generation module are switched on, the post-equalization parameter generation module generates equalization parameters, demodulated signals of the receiving analysis filter bank are subjected to amplitude post-equalization processing and then sent to the real part acquisition module;

when the pre-equalization configuration information indicates that pre-equalization is not performed, the post-equalization switch is switched on the receiving analysis filter bank and the post-equalization parameter generation module, the post-equalization parameter generation module generates equalization parameters, amplitude and phase post-equalization processing is performed on demodulation signals of the receiving analysis filter bank, and then the demodulation signals are sent to the real part acquisition module.

The configurable pre-balanced OQAM signal sending scheme provides flexibility for network scheduling, can ensure the spectral efficiency as much as possible by pre-balancing the transmitted signals under the condition of high network load, and can avoid the influence of tailing by changing the configuration of subframes when the network load is low, thereby ensuring the user service quality as much as possible.

Drawings

FIG. 1 is a schematic diagram of signal generation of FBMC/OQAM;

FIG. 2(a) is a schematic diagram of the tail of FBMC/OQAM based on burst transmission;

FIG. 2(b) is a schematic diagram of data block waveforms without overlap;

FIG. 2(c) is a schematic diagram of partial overlap of data block waveforms;

FIG. 3 is a schematic diagram of an FBMC/OQAM system employing the present application;

fig. 4 is a schematic diagram of pre-equalized OQAM signal transmission and reception according to a first embodiment of the present application;

fig. 5 is a schematic diagram illustrating transmission and reception of an OQAM signal without pre-equalization according to a first embodiment of the present application;

fig. 6 is a schematic diagram of transmitting and receiving pre-equalized OQAM signals according to the second embodiment of the present application.

Detailed Description

In order to make the objects, technical solutions and advantages of the present application more apparent, the present application is further described in detail below by referring to the accompanying drawings and examples.

By using a modulation mode FBMC based on a filter bank technology, a signal waveform with better Time-Frequency focusing can be obtained, for example, based on prototype filter functions such as Isotropic Orthogonal Transform Algorithm (IOTA), Extended Gaussian Function (Gaussian) and European PHYDYAS, etc. FBMC performs pulse shaping (pulse shaping) on a signal of each subcarrier by using a shaping filter with good Time/Frequency L equalization (35TF 84), which makes:

1) the FBMC can greatly inhibit ISI (inter-symbol interference) caused by multipath without a CP (physical channel), not only can bring higher spectral efficiency and energy efficiency compared with OFDM (orthogonal frequency division multiplexing) energy, but also can obtain good receiving robustness under a larger time error, thereby allowing transmission of non-strict synchronization;

2) due to good frequency focusing, the FBMC can transmit signals in extremely narrow frequency resources and keep very low out-of-band leakage, so that inter-carrier interference (ICI) caused by Doppler or phase noise and the like can be well inhibited.

Therefore, the FBMC has great potential in the scenes of cognitive radio, fragmented frequency band access, non-synchronous transmission and the like.

In order to obtain the highest spectral efficiency at FBMC, it is necessary to use Offset Quadrature Amplitude Modulation (OQAM) technique called FBMC/OQAM or OFDM/OQAM, hereinafter abbreviated as OQAM. In OQAM, a QAM symbol is divided into two signals, which are alternately modulated to real and imaginary parts of a subcarrier, respectively, and transmitted by a time-interleaving method. At the receiving end, if there is no influence of the channel, the real part and the imaginary part of the signal on each subcarrier are alternately extracted, and the transmitted signal can be recovered. Fig. 1 is a schematic diagram of signal generation of FBMC/OQAM, and the functions of the blocks can be easily understood from the following formula expression of the OQAM signal. The baseband equivalent of the multi-carrier FBMC/OQAM signal in time series (continuous-time) can be expressed by the following equation (1):

wherein: (.)_m,nRepresents a frequency-time point (frequency-time point);

a_m,na real modulation signal transmitted on the mth subcarrier of the nth symbol, that is, a pulse amplitude modulation symbol (PAM);

a_m,nfor symbol periods τ 2 τ₀A complex QAM (quadrature Amplitude modulation) symbol

The real or imaginary value of, e.g.

And

respectively taking real and imaginary parts;

j is an imaginary symbol, j^m+nRepresenting alternating between real and imaginary;

m is an even number and represents the number of subcarriers;

z is the sent symbol set;

ν₀is the subcarrier spacing;

τ₀symbol period, τ, of OQAM₀＝τ/2＝1/(2v₀)；

g is a prototype filter function, whose time-domain impulse response length is typically K times τ, which results in partial overlap of the time-domain waveforms of adjacent (2K-1) symbols, so K is also commonly referred to as the overlap factor of the filter (OverlappingFactor);

g_m,n(t) is unmodulation a_m,nThe overall synthesis filter function (synthesis filter).

It can be seen that the symbol rate of OQAM is 2 times of the conventional OFDM symbol rate and no cyclic prefix CP is attached, whereas the amount of information per OQAM symbol is half of that of conventional OFDM because the modulation of OQAM is real.

The real-domain orthogonality of OQAM is achieved by designing the prototype filter function g such that the Inner Product (Inner Product) of the synthesis filter function at transmission and the analysis filter function at reception satisfies or approximately satisfies equation (2), i.e. the prototype filter needs to satisfy:

wherein, represents the complex conjugate,

in order to carry out the operation of the solid-obtaining part,<·|·>represents the inner product, if m is m', n is n_m,m'＝1，_n,n'Otherwise 0, i.e. if m ≠ m 'or n ≠ n', the inner product is a pure imaginary term, which is used for descriptive convenience

And (4) showing. It is clear that the signals between different sub-carriers and different symbols cause by themselves purely imaginary interference, and thus a FBMC/OQAM modulated signals (t) passing through a distortion-free channel, the received signal is filtered by a transmit Synthesis Filter (SF) g_m,n(t) matched receive analysis Filter Bank (AF)

The original transmitted real signal a can be processed by simple operation according to the formula (3)_m,nPerfect Reconstruction (PR), η₀Is a noise term, followed by a complex QAM signal

The original data can be demodulated.

In practice, however, the channel is not distortion-free, and it is well known that channels typically exhibit frequency-domain selective fading due to multipath and/or time-selective fading due to doppler, simply expressing the channel impulse response as

H (norm) is the amplitude variation,

for phase variation, H is used assuming that the channel is constant over a unit symbol period and a unit subcarrier bandwidth_m,nRepresenting the frequency response (frequency response) of channel h (t) on subcarrier m at symbol n, the received signal can be represented as:

the complex channel will destroy the real-domain orthogonality of OQAM, and the demodulation a is used_m0,n0For example, as shown in equation (5), directly for y_m0,n0A cannot be reduced by performing the operation of the solid-taking part_m0,n0And additional operations must be performed to eliminateOr to minimize H_m,nDisruption of real-number domain orthogonality.

Above_Im0,n0In relation to the design of the focusing characteristic of the prototype filter, so called intrinsic interference (i.e. g-function), the characteristic of the g-function of the prototype filter is such that only the peripheral finite (m, n) ≠ (m0, n0) symbols of the intrinsic interference terms (m0, n0) have a non-negligible effect, not all, i.e. a pure imaginary number

Only at a limited (m, n) ≠ (m0, n0) is the absolute value of large and non-negligible, the others are very small, being approximately zero or zero, and the latter part is completely negligible with respect to noise and only those symbols of the peripheral limit need to be concerned. The values of the peripheral limited (m, n) ≠ (m0, n0) which cannot be ignored are used as a set omega_Δm,ΔnGiven by { (p, q), | p-m0| ≦ Δ m, | q-n0| ≦ Δ n } (this set corresponds to the time-frequency focality of the prototype filter function), Im0, n0 may be further rewritten as:

from equations (5) and (6), it is clear that the elimination ξ is to be achieved by simple frequency domain channel equalization_m0,n0To reduce a_m0,n0The following equation (7) is satisfied, that is, although a time-varying multipath channel is passed, the condition is satisfied at Ω_Δm，ΔnThe local range represented by the required channel is correlated, that is, the prototype filter is designed to match the channel characteristics, and the better the time-frequency focusing is, the corresponding omega is_Δm，ΔnThe smaller the range, the easier condition (7) is satisfied.

H_mo,no≈H_p,q，(p,q)∈Ω_Δm，Δn(7)

Thus, the signal after the reception filter shown in equation (5) can be approximated to the form of equation (8):

then, once the channel frequency response at (m0, n0) is equalized, the transmitted a can be further processed by taking the real part_m0,n0And reducing to obtain the final product.

In summary, although FBMC/OQAM symbols are partially overlapped in time domain and frequency domain subcarriers are not complex domain orthogonal like OFDM, the focusing design of the OQAM real-virtual alternation and prototype filter enables the influence of the channel to be eliminated by simple frequency domain equalization when the demodulated signal output satisfies the condition of formula (7), thereby restoring the real domain orthogonality, and then the transmitted signal can be demodulated by taking the real part. A typical OQAM signal equalization detection can be represented by equation (9), where SINR represents the signal to interference plus noise ratio in the presence of multi-user interference.

| | be a modulo operation

For downlink or uplink single user transmission, as long as the prototype filter design can match the Delay Spread (Delay Spread) and/or Doppler Frequency Spread (Doppler Frequency Spread) of the channel, the subcarrier spacing and data transmission time are reasonably set, and the condition in formula (7) can be basically satisfied, so that ICI and ISI can be ignored. However, the applicant of the present invention considers that the condition of equation (7) is difficult to satisfy for the uplink multi-user data block transmission of FBMC/OQAM, because the channels that the signals of the uplink different users reach the base station are generally uncorrelated, so there is overlapping set Ω adjacent to the time domain of different users in the frequency domain_ΔnIt is difficult to consider { | q-n0| ≦ Δ n } as relevant, which is why the tail of FBMC/OQAM needs an efficient scheme for special handling during uplink multi-user data block transmission.

The tailing problem is explained first. Communication system resources are typically in accordance withThe time-frequency is divided into time-frequency units (resource grid), and then these smallest time-frequency units are grouped into a basic resource block that is easy to schedule. For example, the concept that the Transmission Time Interval (TTI) or the subframe (also called burst) in the communication standard is transmitted in the basic resource block in time; in the frequency domain, the basic resource block concept is defined by the size of the bandwidth, for example, several sub-carriers, and the data blocks of the end users are mapped to the resource block for transmission. OFDM adopts rectangular waveform, so that all symbols are arranged next to each other on a resource block without overlapping, and FBMC/OQAM adopts prototype filter with longer length, although every tau₀For example, the waveform of the prototype filter on subcarrier 0 shown in fig. 2(a) has data subframe burst1 for the data block transmitted by the first user, data subframe burst 2 for the data block transmitted by the second user, data subframe burst 3 for the data block transmitted by the third user, all of which have a length of L_BComprising a plurality of OQAM symbols, the tail of each burst comprising a length of L_TThe length of the tail depends on the length of a prototype filter, and the tail causes the overlapping of the waveforms of the data blocks, so that the interference is needed to be solved; fig. 2(b) shows that the waveforms of the data blocks of the user do not overlap, and it is necessary to reduce the number of OQAM symbols to be transmitted in each data block, which is wasteful; fig. 2(c) shows the case where the data block waveforms of the users are partially overlapped, which can be considered as a compromise in practical design. Also, if the waste is avoided by tail truncation, waveform distortion is caused and ICI is additionally introduced, which requires a good assessment of achievable spectral efficiency and is prudent.

In view of the above problems, the present invention provides an FBMC/OQAM system capable of configuring pre-equalization and a signal transmitting and receiving method thereof, fig. 3 shows a schematic diagram of the present invention, and the following describes the working principle of the FBMC/OQAM system of the present invention with reference to fig. 3:

a) the control signaling processing unit (302) obtains scheduling control signaling, also called scheduling grant signaling (scheduling grant), from the network scheduler (301), the scheduling grant signaling at least including resource allocation and mapping information and pre-equalization configuration information. Preferably, the resource allocation and mapping information is used to indicate the time-frequency position of the resource block used when transmitting data, the number of OQAM symbols on a unit resource block or subframe burst, and the mapping of the transmission data to the resource block; the pre-equalization configuration information is used to indicate whether to start pre-equalization when data is transmitted on the corresponding resource, and further, if the pre-equalization is started, a pre-equalization mode may be configured, or the pre-equalization mode may be predefined. The pre-equalization method may include: pre-equalizing both amplitude and phase based on Channel Reciprocity (Channel Reciprocity), or pre-equalizing only phase based on Channel Reciprocity, or pre-equalizing only phase based on limited feedback (limited feedback); the pre-equalization mode correspondingly determines a pre-equalization parameter W_pre. Further, the number of OQAM symbols on a unit resource block or a subframe burst is set to be different when pre-equalization is configured and when pre-equalization is not configured, namely: the subframe type information adopted when pre-equalization is turned on and off can be configured, and the subframe type information indicates the number of transmitted symbols in a subframe and the transmission time of the first symbol. Different subframe types can be set, the lengths of the tails extending out of the subframes on the different subframe types are different, the method is realized by setting the number of OQAM symbols transmitted on a unit subframe, and the preset equalization and the non-preset equalization are configured to respectively use different subframe types.

b) The signal source (303) prepares data to be transmitted according to the allocated resources, the data being a real signal, using a_m,nAnd (4) showing.

c) A pre-equalization switch (304) determines whether to output a real signal a to 303 according to pre-equalization configuration information obtained from 302_m,nPre-equalization is performed before OQAM modulation. When configured without starting pre-equalization, the real signal is converted from serial/parallel and then sent directly to the synthesis filter bank (306) for OQAM modulation, wherein the output s (t) can be represented by formula (1); when the pre-equalization is configured to be started, a pre-equalization parameter generating unit (305) generates pre-equalization parameters (pre-equalization coefficients) W according to the acquired channel state information and the configured pre-equalization mode_preFirst, the pre-equalization parameter pair a_m,nPreprocessing, performing OQAM modulation at 306 to generate OQAM signal s (t), so that the signal s (t) can approach to satisfy the real number domain orthogonality of OQAM after passing through the channel, i.e. performing pre-equalization on the channel, thereby making their combined equivalent channel frequency response

With little or negligible real or imaginary part, i.e.

Wherein

Denotes a real number, ≈ denotes an equality or approximation, H is a channel frequency response; or to make the equivalent channel frequency response of different users on adjacent bursts

Are as close as possible to reduce interference from trailing overlap. Wherein:

when the pre-equalization mode is pre-equalization of both amplitude and phase, it is generally necessary to obtain more complete channel state information based on channel reciprocity for generating the pre-equalization parameter W_preSo that the real or imaginary part of the equivalent channel frequency response of the pre-equalization parameter and the channel frequency response is close to 0 and the modulus of the equivalent channel frequency response is close to 1, i.e. the channel frequency response is equal to

When the pre-equalization mode is to pre-equalize only the phase, the actual system can obtain more complete channel state information based on the channel reciprocity for generating the pre-equalization parameter W_preOr baseObtaining phase adjustment information for generating pre-equalization parameters W from finite feedback_preThe former may make the real or imaginary part of the equivalent channel frequency response of the composite of the pre-equalization parameter and the channel frequency response approach 0, and the mode of the equivalent channel frequency response approaches the mode of the channel frequency response, i.e. the mode of the channel frequency response

Therefore, the method can greatly eliminate the damage of the channel to the orthogonality of the OQAM real number domain to obtain good performance, and the phase of the equivalent composite channel between different users can be close to the same as far as possible only through quantized phase adjustment because the practical system can only allow limited feedback overhead, so that the damage of the channel to the orthogonality of the OQAM real number domain is reduced as far as possible through post-equalization, and only suboptimal performance is obtained.

S (t) after pre-equalization can be expressed by equation (10). Wherein the Channel state information used for pre-equalization may be obtained through feedback or Channel Reciprocity (Channel Reciprocity).

d) The reception analysis filter bank (307) performs OQAM demodulation on the received OQAM signal r (t) which has passed through the channel, and outputs y_m,nAnd (4) showing.

e) The determining module 308 determines whether to perform post-equalization on the signal output by 307 according to pre-equalization configuration information indicated to the transmitter by the network scheduler. If pre-equalization is configured to be performed during data transmission and the pre-equalization is performed in such a way that both the amplitude and the phase of the channel are pre-equalized, then the output y of 307 is not equalized_m,nPost-equalization is performed and the real-part-taking operation is performed directly at 310 to obtain the estimated transmitted signal

When pre-equalization is configured at the time of transmitting data in such a manner that only the phase is pre-equalized, the output y of 307 is subjected to pre-equalization_m,nAlso needs to perform post-equalization

In the post-equalization, the post-equalization parameter is generated and equalized according to the equivalent channel frequency response H & lt- & gt synthesized by the estimated pre-equalization parameter and the channel frequency response, and the used post-equalization parameters (equalization coefficients) are W^a _aftRepresents, the equivalent channel frequency response

The estimation can be directly estimated and obtained by the receiver receiving the reference signal for channel estimation after pre-equalization is carried out by adopting the pre-equalization parameters the same as the data, or the receiver carries out channel estimation on the reference signal which does not adopt the pre-equalization firstly and then indirectly estimates and obtains the reference signal according to the pre-equalization mode adopted by the transmitter, and the equalized signal goes to 310 to carry out real part operation, thereby obtaining the estimated sending signal

When there is no pre-equalization configured in sending data, then output y to 307_m,nPerforming conventional channel equalization on amplitude and phase, wherein the equalization parameter is W^b _aftMeaning that the equalized signal is passed to 310 for the operation of the real part to obtain the estimated transmission signal

The channel state information required for post-equalization can be obtained by known pilot signals used for channel estimation.

The technical solution of the present application is further described in detail by several preferred embodiments.

Detailed description of the preferred embodiment

In practical implementations, the baseband signal is a directly generated discrete-time (dft) form, which is a sampling sequence of a continuous signal at strict sampling points, i.e., the sampling frequency satisfies the sampling theorem. The discrete baseband signal of the FBMC/OQAM system can be described by equation (11), where the variable t is converted to the sampling time i in equation (1):

wherein, L_gIs the length of the prototype filter, typically L_gKM, or KM-1 or KM +1, K being an overlap factor, is to ensure a causal system. In particular, the present invention does not relate to a specific prototype filter design, and the prototype filter waveform illustrated in fig. 2(a) is merely an example, and is a prototype filter function approximately satisfying equation (2) and having time domain tap coefficients of

In this embodiment, let us say that the FBMC/OQAM system is a Time Division Duplex (TDD) system, and the parameters are the number M of subcarriers 512 and the subcarrier spacing v₀15khz, sampling frequency 7.68Mhz, repetition factor K4, length of prototype filter L_g2048 sample samples (samples), and regarding the Burst type, we set two different types, namely, Extended Burst (EB) and Non-Extended Burst (NEB), respectively, the Burst lengths of EB and NEB are both 0.4 milliseconds (ms), that is, 3072 samples, and it is further illustrated that the EB and NEB lengths may be preset differently, or the system may support multiple subframe lengths, which is set to be the same for convenience of explanation_B3072s amples, and one tail before and after each burst, the length of each tail being equal to

Each EB comprises 12 symbols with tails extending beyond the burst, whereas the burst type in fig. 2(b) is NEB, with burst length also L_B3072s amples, but each burst includes 6 symbols as set out in the figure, since 6 symbols are reduced, the tail is within the burst and does not extend beyond the burst. Please note that we can also set the burst type as shown in FIG. 2(c), i.e. each burst contains 8 symbols, allowing a part of the tail to be extendedOutside of burst, the specific burst length and number of symbols depends on overall considerations of system design, since the interference of the tail is already small.

The scheduling selection of the multi-user scheduler in this embodiment includes two choices shown in table one:

watch 1

For convenience of description, we assume that M subcarriers are all used for user data transmission and omit the resource allocation and mapping process.

When the configurable pre-equalization FBMC/OQAM TDD system is configured with pre-equalization on (scheduling option a), the baseband signal transmission and reception are respectively:

the transmitter knows that the signal sent by the transmitter needs to be pre-equalized from the scheduling control signaling sent by the network scheduler, adopts an EB structure to send the signal, and generates a frequency domain pre-equalization parameter W according to the channel reciprocity of TDD_preThe transmitted signal is pre-equalized and then subjected to OQAM modulation, and fig. 4 shows a specific implementation diagram of transmitting and receiving an OQAM signal with pre-equalization. Referring to fig. 4:

at a transmitting end, 401 is a module for converting complex QAM modulation symbols of OQAM into real PAM signals, 402 is a pre-equalization module, 403 is a module for implementing a synthesis filter bank in which the OQAM modulation adopts Inverse Fast Fourier Transform (IFFT) plus a Polyphase filter (Polyphase Filtering), and 404 is a parallel/serial conversion module. The OQAM signal with pre-equalization can be represented by equation (12), and as shown in fig. 4, the signal to be transmitted on each subcarrier is multiplied by the parameter W before IFFT_preThe pre-equalization parameter is to avoid the time-varying multipath channel from destroying the real number domain orthogonality of OQAM, so that the transmitter can obtain the frequency response on the subcarrier m at the symbol n according to the channel reciprocity estimation

Generated according to equation (13), equation (13) may also be referred to as a matching predictorThe equalization algorithm, i.e. taking the conjugate of the estimated channel and then being the square of the modulus of the estimated channel.

| | be a modulo operation (13)

At the receiving end, 405 is a serial/parallel conversion module, 406 is a receive synthesis filter bank implementation module matched to 403, 407 is a real part operation module, 408 is a real to complex conversion module, which operates as the inverse of the operation corresponding to module 401. The OQAM signal r (l) received by the receiver and passing through the channel can be represented by (14):

as can be understood from equation (14), in the case where the channel estimation error is less than a certain range,

following actual channel H_m,nCan be considered equal to each other, then the resultant equivalent channel frequency response

The receiver's 406 module thus demodulates the received r (l) to y as the output at (m0, n0)_m0,n0For example, as shown in equation (15), it is obvious that according to the design of the prototype filter, the latter term can be considered as an imaginary number, so the receiver directly sends the output y of the slave module 406 according to the pre-equalization information configured to the transmitter before the network scheduler_m,nThe signal can be transmitted by inputting the signal into the module 407 for real part operation

Obviously, through the foregoing description of the pre-equalization scheme, it can be seen that even if different users use EB-based multi-user time division multiplexing shown in fig. 2(a), the tail will not cause serious interference because the channel does not satisfy formula (7), that is, as long as the transmission signal of each user is pre-equalized according to the channel, the tail can be allowed to extend beyond the burst, and transmission in the form of EB can ensure that the spectrum efficiency is maximized.

Of course, the pre-equalization is based on the condition that the channel estimation is accurate, and the performance can be ensured. When the network load is relatively low, in order to guarantee Quality of service (Qos) of each user, the network may configure FBMC/OQAM signal transmission (scheduling option B) based on NEB without pre-equalization, at this time, the transmission of the OQAM signal is as shown in fig. 5, compared with fig. 4, one less pre-equalization module is used, one more post-equalization module is used as shown in 501, and the transmitter transmission signal is in the form shown in formula (11). As shown in fig. 2(b), because the user uses NEB transmission and there is no overlap between different users, there is no need to consider whether the channel between different users satisfies the condition of formula (7), and for a single user, the overlap signal on the burst of the user can satisfy the condition of formula (7), so each user receiver only needs to synthesize the output y of the filter bank for the user receiver_m,nEqualization is performed to restore the transmitted signal, and the output at (m0, n0) is y_m0,n0For example, the form is equation (8), then the transmitter's 501 module is simply based on the estimated channel

The equalization parameters are generated, and at this time, the post-equalization mode is the traditional channel equalization, and may be Minimum Mean Square Error (MMSE) equalization or matched filter equalization, etc., and the matched filter equalization is taken as an example and set

Then using the parameters to y_m,nThe equalization is carried out and the balance is carried out,the operation of the real part can restore the transmitted signal, i.e.

For reference, in this embodiment, the configurable pre-equalization OQAM signaling scheme provides flexibility for network scheduling, and may ensure spectrum efficiency as much as possible under a high network load, and may also ensure user service quality as much as possible when the network load is low. For the case of a short burst, for example, in this embodiment, the burst is 0.4ms, the length of the tail is about 0.1167ms, the occupation ratio is very large, and if the tail overlap is to be completely avoided, a large spectrum efficiency loss is caused, so it is generally considered that the FBMC system is not suitable for transmission based on a short burst, and the technical bias before can be completely broken through by pre-equalization, so that the application range of the FBMC/OQAM is further expanded.

Detailed description of the invention

In this embodiment, we consider an FBMC/OQAM system as Frequency Division Duplex (FDD). In the same way as the first embodiment, burst is also divided into EB and NEB, and the parameter setting is the same as the first embodiment. The uplink transmission scheduling selection of the scheduler in this embodiment includes two options shown in table two:

watch two

In this embodiment, it is assumed that the transmitter may perform beamforming through multiple transmit antennas so that the transmitted beam has good directivity, and thus, the energy maximum path is in the quasi-beam (boresight) direction and most of the energy is concentrated therein, and the channel impulse response (time domain) estimated by the receiver when the receiver receives the energy maximum path can be simply represented as

The amplitude of | h | (norm),

for the phase variation coefficient caused by propagation delay, the receiver obtains the feedback channel phase indication on the feedback channel

It is a pair

The middle θ is quantized and transmitted on the feedback channel using limited bits, for example, in the case of 360-degree channel direction, 6 bits and 8 bits can characterize the channel direction information with granularity of 5.625 degrees and 1.5 degrees, which is sufficient for most of the spatial resolution of beamforming. Even if the maximum path of the energy has a reflection path, and if the receiving beam forming exists, a plurality of main paths capable of being received can be equivalent to a path received at an angle, so that the maximum path of the energy is used

To simplify the characterization.

When configured with pre-equalization on (scheduling option C), the baseband signal transmission and reception are:

the transmitter knows from the scheduling control signaling sent by the network scheduler that the signal sent by the transmitter needs to be pre-equalized and adopts an EB structure to send the signal for only pre-equalizing the phase.

The transmitter obtains the quantization of the receiver feedback through the feedback channel according to the indication of the network scheduler

To obtain

(time domain) the transmitter transmits the channel phase information

Conversion to frequency-phase response coefficients of the channel over (m, n)

The frequency domain transform is represented below by a Discrete Fourier Transform (DFT), note: equivalently, the feedback channel here can also be generated directly by directly feeding back the frequency-phase response of the channel

Further, the feedback channel may be used as a part of a conventional uplink scheduling grant signaling (uplink scheduling request), that is, a phase indication bit field is added; or a downlink feedback channel can be newly set up for the base station to feed back the channel phase indication to the terminal.

A transmitter according to

Generating frequency domain pre-equalization parameters W_preFig. 6 shows a specific implementation diagram of pre-equalizing OQAM signal transmission and reception according to this embodiment. Compared with fig. 4, the equalization module 601 generates pre-equalization parameters in a different manner, while the post-equalization module 602 is added, and other modules are the same as the corresponding modules of fig. 4. The transmitter generated pre-equalized OQAM signal and the pre-equalization parameters generated by the 601 block can be expressed by equation (12) and the following (16), respectively, in the first embodiment:

the OQAM signal r (l) received by the receiver and passing through the channel can be represented by equation (17):

according to equation (17), if there is a main path where energy is very concentrated, the sum of estimation errors in channel direction informationIn the case that the quantization error is smaller than a certain range, the estimated channel direction and the direction of the actual channel are similar, and thus the phase is similar, so that the resultant equivalent channel frequency response approaches the real number, that is, the imaginary part is small and can be ignored,

the receiver demodulates r (l) with the output at (m0, n0) as y_m0,n0For example, we can know y of the synthesis filter bank_m,nIs approximately orthogonal in the real number domain, as shown in equation (18), the second term in the equation can be considered to be an imaginary number.

The receiver knows that the transmitter has only pre-equalized the phase, so the receiver's 602 module expresses y in the form shown in equation (18) based on the estimated channel frequency response_m,nPerforming post-equalization according to the synthesized equivalent channel frequency response, and directly estimating to obtain the equivalent channel frequency response by a receiver by receiving a reference signal for channel estimation adopting pre-equalization processing

Or the receiver acquires by receiving a reference signal without pre-equalization

And receiver is aware of transmitter usage

Thereby indirectly knowing

I.e. post-generation equalization parameters

Then heel y_m,nMultiplying, and then operating the real part to obtain the hairEmission signal

It can be seen that pre-equalization and post-equalization process the phase and amplitude, respectively, which can reduce the feedback requirement. Furthermore, feeding back channel direction information (time domain) instead of directly feeding back channel phase in frequency domain can further reduce feedback overhead, but only the transmitter needs to generate channel phase in frequency domain by fourier transform, but this operation is very low in complexity.

When configured as pre-equalization off (scheduling option D), the processing is the same as that of scheduling option B of embodiment one, and the description is not repeated here.

Detailed description of the preferred embodiment

In this embodiment, we consider an FBMC/OQAM system as Frequency Division Duplex (FDD). In the same way as the first embodiment, burst is also divided into EB and NEB, and the parameter setting is the same as the first embodiment. The uplink transmission scheduling selection of the scheduler in this embodiment includes two selections shown in table three:

watch III

In this embodiment, assuming that the transmitter uses an omnidirectional transmitting antenna to pass through a time-varying multipath channel, the channel impulse response at time t can be simply expressed by equation (19):

wherein f is_DnThe Doppler shift for the nth diameter; theta_nPhase shift caused by transmission delay of the nth path; tau is_nIs the relative time delay of the nth path relative to the first arrival path, and C_n(t) the complex gain of the nth path; is a Dirac Delta function. As can be seen from the formula, for such multipath channel with energy dispersion, the time domain channel state information is directly fed back to reconstruct the channel impulse response phase informationIt is impractical and the overhead is particularly large. But according to the multipath channel delay expansion, the channel can be divided into different related sub-bands in the frequency domain, each sub-band can assume that the frequency response of the channel is basically the same, so that the frequency phase response of the channel can be quantized from the angle of the frequency domain based on sub-band feedback, each sub-band can feed back a phase indication, and the phase indication can be used for reconstructing the frequency phase response coefficient of each corresponding sub-carrier on the sub-band

Utilization of subband reconstruction based channel frequency phase response coefficients may be used to generate pre-equalization parameters for use over an allocated bandwidth

For example, we can transform the channel impulse response of the multipath channel to the frequency domain, which is simply expressed by equation (20):

where | h (f) | represents the frequency-amplitude response (also called amplitude-frequency response),

representing the frequency phase response (also called phase frequency response),

representing the corresponding complex frequency phase response coefficient. Dividing the frequency domain into N subbands according to the channel correlation bandwidth, and dividing the frequency-phase influence into k segments

For each segment of frequency f_kLet us consider that

Is linearly variable so that we can use a quantized phase indication

To perform a limited feedback, from which phase indication we can reconstruct the band f belonging to_kFrequency phase variation coefficient of each subcarrier in the inner

The parameters are used to generate pre-equalization parameters W_pre。

For example, when configured as pre-equalization on (scheduling option E), the baseband signal transmission and reception are:

The transmitter obtains the phase indication of the sub-band from the feedback channel according to the indication of the network scheduler

The transmitter progressively reconstructs the frequency-phase response coefficients from the subbands, according to the conditions of the subbands comprised by the allocated bandwidth

And concatenating the frequency-phase response coefficients of each sub-band according to the sub-band positions to obtain the frequency-phase variation system of all sub-carriers in the allocated bandwidth

Further, the feedback channel may be used as a part of a conventional uplink scheduling grant signaling (uplink scheduling request), that is, an indication of an added phase

A bit field; or a downlink feedback channel can be newly set up for the base station to feed back the channel state information to the terminal.

A transmitter according to

Generating frequency-domain pre-equalization parameters, e.g.

And pre-equalizing the transmitted signal and then carrying out OQAM modulation. Whether the receiver receives the OQAM signal r (l) passing through the channel is expressed by equation (17):

combining the condition of the formula (7) and the formula (17), as long as the phase variation coefficients of the pre-equalization parameters on the adjacent symbols and the equivalent channel frequency response synthesized by the channel frequency response approach to the same value, the interference influence caused by the channel destroying the orthogonality of the OQAM real number domain can be reduced, that is, the interference influence is reduced

Wherein Δ_m,nIs a value that varies little between adjacent symbols. The receiver demodulates r (l) with the output at (m0, n0) as y_m0,n0For example, as shown in equation (21):

based on

Based on equation (6), we can see that the interference between adjacent symbols can be significantly reduced by further post-equalization. We can further approximate equation (21) as the following equation (22):

it is clear that the receiver knows that the transmitter has only pre-equalized the phase, so the 602 module of the receiver is based on the estimated equivalent channel frequency response

For y in the expression form shown in formula (22)_m,nThe transmitted data can be restored according to the post-equalization, and the receiver can obtain the equivalent channel frequency response by receiving the reference signal for channel estimation which adopts the pre-equalization processing

Thereby generating post-equalization parameters

For post-equalising operations, i.e. with y_m,nMultiplying, and then operating the real part to obtain the transmitting signal

When configured as pre-equalization off (scheduling option F), the processing is the same as that of scheduling option B of embodiment one, and the description is not repeated here.

Based on the foregoing specific embodiments, the present application provides a method for sending a signal based on a filter bank, including:

Corresponding to the above method, the present application provides a filter bank based transmitter, comprising: control signaling processing module, pre-equalization switch, information source, pre-equalization parameter generation module and send synthesis filter bank, wherein:

Based on the foregoing specific embodiment, the present application further provides a filter bank based signal receiving method, including:

Corresponding to the above method, the present application provides a filter bank based receiver, comprising: the device comprises a receiving analysis filter bank, a post-equalization switch, a post-equalization parameter generation module and a real part operation module; wherein:

when the pre-equalization configuration information indicates to start pre-equalization and the pre-equalization mode is only to pre-equalize phases, the post-equalization switch sends the pre-equalization mode to the post-equalization parameter generation module, the receiving analysis filter bank and the post-equalization parameter generation module are switched on, the post-equalization parameter generation module generates equalization parameters, demodulation signals of the receiving analysis filter bank are subjected to post-equalization processing according to equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response, and then the demodulation signals are sent to the real part obtaining module;

when the pre-equalization configuration information indicates that pre-equalization is not performed, the post-equalization switch is switched on the receiving analysis filter bank and the post-equalization parameter generation module, the post-equalization parameter generation module generates equalization parameters, amplitude and phase post-equalization processing is performed on demodulation signals of the receiving analysis filter bank according to estimated channel frequency response, and then the demodulation signals are sent to the real part acquisition module.

The above description is only exemplary of the present application and should not be taken as limiting the present application, as any modification, equivalent replacement, or improvement made within the spirit and principle of the present application should be included in the scope of protection of the present application.

Claims

1. A filter bank-based signal transmission method, comprising:

when the pre-equalization configuration information indicates to start pre-equalization, the transmitter generates pre-equalization parameters according to a pre-equalization mode, performs pre-equalization operation on a transmission signal according to the pre-equalization parameters, and then performs filter bank-based modulation;

when the pre-equalization configuration information indicates that pre-equalization is not performed, the transmitter performs filter bank-based modulation on the transmission signal which is not subjected to pre-equalization;

the pre-equalization configuration information is contained in a scheduling control signaling acquired by a transmitter; the transmitter also acquires subframe type information adopted when pre-equalization is started and closed from the scheduling control signaling, wherein the subframe type information indicates the number of sending symbols in a subframe and the sending time of a first symbol.

2. The method of claim 1, wherein:

the pre-equalization method is predefined or included in pre-equalization configuration information.

3. The method of claim 2, wherein:

the pre-equalization mode comprises the following steps: both amplitude and phase are pre-equalized, or only phase is pre-equalized.

4. The method of claim 3, wherein:

when the pre-equalization mode is to pre-equalize both the amplitude and the phase, the generating pre-equalization parameters according to the pre-equalization mode includes: and generating a pre-equalization parameter according to the acquired channel state information and a pre-equalization mode, wherein the real number or the imaginary part of an equivalent channel frequency response synthesized by the pre-equalization parameter and the channel frequency response approaches to 0, and the modulus of the equivalent channel frequency response approaches to 1.

5. The method of claim 3, wherein:

when the pre-equalization mode is to pre-equalize the phase only, the generating the pre-equalization parameter according to the pre-equalization mode includes: and generating a pre-equalization parameter according to the acquired channel state information and a pre-equalization mode, wherein the equivalent channel frequency response synthesized by the pre-equalization parameter and the channel frequency response is that the real number or the imaginary part approaches to 0, and the mode of the equivalent channel frequency response approaches to the mode of the channel frequency response.

6. The method according to claim 4 or 5, characterized in that:

the channel state information is obtained through channel reciprocity.

7. The method of claim 3, wherein:

when the pre-equalization mode is to pre-equalize the phase only, the generating the pre-equalization parameter according to the pre-equalization mode includes: and acquiring a phase indication through feedback of the receiver, and generating pre-equalization parameters according to the acquired phase indication.

8. The method of claim 7, wherein:

and the transmitter acquires the phase information of the quantized channel frequency response according to the phase indication and generates a frequency phase response coefficient on the allocated bandwidth as a frequency domain pre-equalization parameter according to the phase information.

9. The method of claim 8, wherein:

the method further comprises the following steps: the frequency domain is divided into a plurality of sub-bands, the phase indication is indicated based on the sub-bands, the number of the sub-bands is more than or equal to 1, and the transmitter generates frequency domain pre-equalization parameters on the allocated bandwidth according to the corresponding relation between the phase indication and the sub-bands.

10. The method of claim 7, wherein:

and the transmitter acquires the phase information of the quantized channel impulse response according to the phase indication and converts the phase information into a corresponding frequency phase response coefficient as a frequency domain pre-equalization parameter.

11. The method of claim 10, wherein:

the method further comprises the following steps: the quantized phase information of the channel impulse response is the phase information of the strongest path in the multi-path channel, or the phase information synthesized by averaging the phases of the X paths with the strongest energy in the multi-path channel, wherein X is larger than 1.

12. The method of claim 7, wherein:

and acquiring the feedback in a phase indication bit field or a special feedback channel added in the uplink scheduling authorization signaling.

13. A filter bank based signal receiving method, comprising:

14. The method of claim 13, wherein:

the post-equalization according to the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response comprises: the receiver directly estimates the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response by receiving the reference signal which adopts the pre-equalization and is used for channel estimation, or the receiver indirectly estimates the equivalent channel frequency response synthesized by the pre-equalization parameters and the channel frequency response according to the pre-equalization mode adopted by the transmitter by performing channel estimation on the reference signal which does not adopt the pre-equalization, wherein the equivalent channel frequency response is used for post-equalization of the demodulated signal based on the filter.

15. The method according to claim 13 or 14, characterized in that:

16. A filter bank based transmitter, comprising: control signaling processing module, pre-equalization switch, information source, pre-equalization parameter generation module and send synthesis filter bank, wherein:

17. The transmitter of claim 16, wherein:

the pre-equalization configuration information is acquired from the scheduling control signaling by the control signaling processing module.

18. The transmitter according to claim 16 or 17, characterized in that:

a pre-equalization parameter generation module generates the pre-equalization parameters according to a pre-equalization mode; the pre-equalization method is predefined or included in the pre-equalization configuration information.

19. A filter bank based receiver, comprising: the device comprises a receiving analysis filter bank, a post-equalization switch, a post-equalization parameter generation module and a real part operation module; wherein:

20. The receiver of claim 19, wherein: