CN105490604A - Predictive control method of three-phase four-switch induction motor variable-frequency speed control system - Google Patents

Abstract

The invention discloses a predictive control method of a three-phase four-switch induction motor variable-frequency speed control system. The method includes the steps of: A. measuring three-phase current, two direct current side capacitor voltages and motor rotating speed through a current sensor, a voltage sensor and a speed sensor in an induction motor drive system; B. calculating voltage vectors corresponding to four switch combinations (00, 01, 11, 10) through the measured direct current capacitor voltages; C. estimating stator and rotor flux linkage of an induction motor according to the measured phase current and rotating speed; D. predicting absolute values of stator flux linkage, predicted torque and predicted direct voltage corresponding to the four voltage vectors; E. calculating four voltage vector cost functions: subtracting a reference value from predicted values and taking absolute values, multiplying each item by weight coefficients and adding together; and F. applying a switch state corresponding to a voltage vector which enables the cost function to be minimum to an inverter. The method is suitable for various variable-frequency speed control systems driven by a three-phase four-switch.

Description

Predictive control method for three-phase four-switch induction motor variable-frequency speed control system

Technical Field

The invention belongs to the field of variable frequency speed regulation of induction motors, and particularly relates to a predictive control method of a variable frequency speed regulation system of an induction motor under a three-phase four-switch power converter topology.

Background

The frequency conversion speed regulation system taking the induction motor as a main body is widely applied to the fields of aerospace, military, industry and the like, and a power converter of the system consists of six-switch three-phase fully-controlled power electronic devices. Among them, the converter has high energy density, the power electronic devices are relatively fragile, once a certain power tube of the converter has an open circuit or short circuit fault, the whole system loses the capability of normal operation, and even has disastrous results.

With the increasing requirements on the safety and reliability of the variable frequency speed control system, the real-time fault-tolerant control is highly emphasized, however, most variable frequency speed control systems are not provided with redundancy backup, so that a three-phase four-switch topological structure without redundancy backup is more concerned. In numerous patents and documents for control strategies of three-phase four-switch variable-frequency speed control systems, the general methods are divided into two categories: one is that the direct current capacitor voltage is assumed to be constant, a control algorithm is designed on the basis, and because one phase of the motor is directly connected to a capacitor neutral point, the flow of phase current can cause the fluctuation and drift of the capacitor voltage, so the method can not be used in a practical system; the other type is to design a control algorithm aiming at the voltage fluctuation and the drift of the capacitor, but the algorithm is generally an open-loop control strategy, and the dynamic performance of the speed regulating system is poor.

Disclosure of Invention

The invention provides a prediction control method under a three-phase four-switch power converter topology, aiming at overcoming the defects of the control strategy of the existing three-phase four-switch variable frequency speed control system. The method can realize high-performance flux linkage and torque closed-loop control under the condition of capacitor voltage fluctuation, and can also inhibit the drift of the capacitor voltage without a pulse width modulator and coordinate transformation. The method is suitable for various variable frequency speed regulation systems driven by three-phase four switches.

In order to achieve the above object, the present invention provides a predictive control method for a variable frequency speed control system of an induction motor under a three-phase four-switch power converter topology, wherein the method comprises:

(1) three-phase current i is respectively measured by the existing current Hall sensor, voltage Hall sensor and photoelectric coded disc speed sensor in the induction motor driving system_a ^k,i_b ^k,i_c ^kTwo DC side capacitor voltages U_C1 ^k,U_C2 ^kAnd motor speed ω;

(2) by measuring two DC-side capacitor voltages U_C1 ^k,U_C2 ^kCalculating correspondence of four switch combinationsVoltage vector V₁,V₂,V₃,V₄Value at the present moment and based on the measured three-phase currents i_a ^k,i_b ^k,i_c ^kSignal calculation of current vectorWherein the four switch combinations are 00,10,11, 01;

(3) by measuring motor speed omega and current vectorEstimating stator flux linkageAnd rotor flux linkage ${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k;$

(4) Predicting all voltage vectors V from a motor model and an inverter model₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torque

(5) Using predicted individual voltage vectors V₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torqueCalculating a cost function, and taking a voltage vector which minimizes the cost function as an optimal voltage vector;

(6) and (3) applying a switching signal corresponding to the optimal voltage vector, wherein the corresponding relation between the voltage vector and the switching signal is the same as that in the step (2).

The invention has the advantages that through accurate modeling of the motor model, the closed-loop high-performance control of the magnetic field, the torque and the rotating speed of the induction motor can be realized under the condition of capacitor voltage fluctuation, and meanwhile, the balanced control of three-phase current under the topology of the asymmetric power electronic converter is realized. In order to ensure the reliability of the system, the invention restrains the drift of the capacitor voltage. The invention directly outputs the switch signal without a pulse width modulator. All variables are completed under the stator coordinate system without coordinate transformation. The control structure is simple and easy to understand and is easy to realize physically.

Drawings

FIG. 1 is a schematic diagram of an induction motor drive system and its basic structure for use in the method of the present invention;

FIG. 2 is a control schematic block diagram of the method of the present invention;

fig. 3 is a control flow chart of a model predictive control method of a three-phase four-switch driven induction motor according to the present invention.

Detailed Description

In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

As shown in fig. 1, the present invention relates to a power converter structure of an induction motor variable frequency speed control system and a connection model of an induction motor. Two phases of the three phases of the motor are connected with a normal switch bridge arm, and the third phase is connected with a capacitance neutral point at a direct current side.

As shown in fig. 2, a control block diagram related to the present invention and a connection block diagram of an induction motor system driven by a three-phase four-switch are described with reference to fig. 2, which illustrates the principle of the technical solution adopted in the present invention:

in order to realize a high-performance closed-loop control strategy, a traditional proportional-integral controller is adopted for speed outer loop control to obtain a given value of torque, a model prediction controller is adopted for current inner loop control, and the scheme comprises three steps of flux linkage estimation, torque, flux linkage and capacitance voltage prediction and cost function optimization.

First, the current induction motor stator and rotor flux linkage is estimated using an induction motor current model. The current stator flux linkage estimation method based on the current model is adopted, the current stator flux linkage and the current rotor flux linkage can be measured and estimated by using the rotating speed and the phase current, and the influence of the capacitor voltage fluctuation on the flux linkage estimation is overcome.

Secondly, measuring the real-time voltage of the capacitor by using a voltage sensor, calculating the accurate value of the current voltage vector, and then substituting four voltage vectors corresponding to the current four switch states into the model one by using a mathematical model of the induction motor to predict the stator flux linkage, the stator current, the electromagnetic torque and the capacitor voltage of the next sampling period under different voltage vectors.

And finally, respectively subtracting the predicted absolute value of the flux linkage, the electromagnetic torque and the capacitor voltage from the reference value, calculating the absolute value, multiplying the absolute value by corresponding weight coefficients, adding the weight coefficients to obtain a cost function, wherein the four voltage vectors correspond to the four cost function values, and applying a switching signal corresponding to the minimum voltage vector of the cost function values to the inverter.

Three-phase current of the motor is obtained from the motor through the Hall sensor, and a rotating speed signal is measured from the photoelectric coded disc speed sensor. From the power converter, the capacitor voltage is obtained by a voltage hall sensor. The above variables are used as input quantities of the control system to participate in system control. The control system directly outputs discrete switch signals, and the control structure is simplified. The control system is divided into an inner control ring and an outer control ring: the outer ring is a traditional PI regulator, closed-loop control of the rotating speed is realized, and torque setting is generated through the speed regulator; the inner ring is a model prediction controller, closed-loop control of motor torque and flux linkage is realized, and simultaneously, the suppression of capacitor voltage drift is also realized in the inner ring.

As shown in fig. 3, which is a control flow chart of a predictive control method of a three-phase four-switch driven induction motor model according to the present invention, as shown in the figure, the method includes:

1. the initialization initializes the cost function g to a sufficiently large value.

2. Three-phase current i is respectively measured by the existing current Hall sensor, voltage Hall sensor and photoelectric coded disc speed sensor in the induction motor driving system_a ^k,i_b ^k,i_c ^kTwo DC side capacitor voltages U_C1 ^k,U_C2 ^kAnd motor speed ω;

3. by measuring two DC-side capacitor voltages U_C1 ^k,U_C2 ^kCalculating voltage vectors V corresponding to four switch combinations₁,V₂,V₃,V₄Value at the present moment and based on the measured three phasesCurrent i_a ^k,i_b ^k,i_c ^kSignal calculation of current vectorWherein the four switch combinations are 00,10,11, 01; the voltage vector calculation method is shown in table 1.

TABLE 1

The calculation method of the current vector is as follows:

${{\stackrel{\→}{i}}_s}^k={i_a}^k+j\·\frac{\sqrt{3}}{3}({i_a}^k+2\×{i_b}^k)---\left(0.1\right)$

where the superscript k is the sampling instant.

4. By measuring motor speed omega and current vectorEstimating stator flux linkageAnd rotor flux linkage ${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k;$

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k=\frac{{\mathrm{\τ}}_r}{T_s(1-\mathrm{j\ω}\·{\mathrm{\τ}}_r)}\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^{k-1}+\frac{L_m}{1-\mathrm{j\ω}\·{\mathrm{\τ}}_r}\·{{\stackrel{\→}{i}}_s}^k---\left(0.2\right)$

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^k=k_r\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k+{\mathrm{\σL}}_s\·{{\stackrel{\→}{i}}_s}^k---\left(0.3\right)$

Wherein L is_s，L_m，L_rRespectively stator inductance, excitation inductance and rotor inductance, R_sR_rRotor resistance and stator resistance, respectively. T is_sIs the sampling time, k_r＝L_m/L_rIs the mutual inductance coefficient of the rotor,is the magnetic leakage coefficient, τ_r＝L_r/R_rIs the rotor time constant.

5. Predicting all voltage vectors V from a motor model and an inverter model₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torque

Predicted current vector for next time instantThe following were used:

${{\stackrel{\→}{\hat{i}}}_s}^{k+1}=(1+\frac{T_s}{{\mathrm{\τ}}_{\mathrm{\σ}}})\·{{\stackrel{\→}{i}}_s}^k+\frac{T_s}{{\mathrm{\τ}}_{\mathrm{\σ}}+T_s}\·\{\frac{1}{R_{\mathrm{\σ}}}\·((\frac{k_r}{{\mathrm{\τ}}_r}-j\·k_r\·\mathrm{\ω})\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k+{{\stackrel{\→}{v}}_s}^k)\}---\left(0.4\right)$

wherein,is the equivalent resistance, L_σ＝σL_sIs the leakage inductance, tau, of the motor_σ＝σL_s/R_σ，Which represents a vector of voltages that is, ${{\stackrel{\→}{v}}_s}^k=\{V_1,V_2,V_3,V_4\}.$

the stator flux linkage and the electromagnetic torque are predicted based on the predicted current vector.

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}={{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^k+T_s\·{{\stackrel{\→}{v}}_s}^k-R_s{T}_s\·{{\stackrel{\→}{i}}_s}^k---\left(0.5\right)$

${{\hat{T}}_e}^{k+1}=\frac{3}{2}p\·\mathrm{Im}\{{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}\·{{\stackrel{\→}{i}}_s}^{k+1}\}---\left(0.6\right)$

Wherein p is the induction machine pole pair number, and Im is represented by the imaginary part of a complex number.

For a four-switch three-phase topological structure, the upper and lower tubes cannot be directly connected, so that the switch combination S_bS_cCan only take values of 00,10, 01, 11, capacitance current i_dc1,i_dc2Is taken as follows

i_dc1 ^k＝i_b ^k·S_b+i_c ^k·S_c(0.7)

i_dc2 ^k＝i_b ^k·(1-S_b)+i_c ^k·(1-S_c)

Predicted value U of capacitor voltage_C1(k+1)，U_C2(k +1) is

${{\hat{U}}_{C1}}^{k+1}={U_{C1}}^k-(1/C)\·{i_{\mathrm{dc}1}}^k\·T_s$ (0.8)

${{\hat{U}}_{C2}}^{k+1}={U_{C2}}^k+(1/C)\·{i_{\mathrm{dc}2}}^k\·T_s$

6. Using predicted individual voltage vectors V₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torqueCalculating a cost function, and taking a voltage vector which minimizes the cost function as an optimal voltage vector;

designed cost function control rate g_iThe following were used:

$g_i=\frac{|T_e^{*}-{\left({\hat{T}}_e^{k+1}\right)}_i|}{T_{e_{\mathrm{nom}}}}+{\mathrm{\λ}}_0\frac{\left|\right|\left|{{\stackrel{\→}{\mathrm{\ψ}}}_s}^{*}\right||-|\left|{\left({{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}\right)}_i\right|\left|\right|}{{\left|\right|{\stackrel{\→}{\mathrm{\ψ}}}_s\left|\right|}_{\mathrm{nom}}}+{\mathrm{\λ}}_{\mathrm{dc}}\frac{|{\left({{\hat{U}}_{C1}}^{k+1}\right)}_i-{\left({{\hat{U}}_{C2}}^{k+1}\right)}_i|}{V_{\mathrm{dc}}},i\∈\left\{\mathrm{1,2,3,4}\right\}---\left(0.9\right)$

whereinRespectively the rated torque and the rated flux linkage of the induction machine,for the purpose of the torque giving,for the given absolute value of the flux linkage, | - | symbol is the solving absolute value, | | | - | is the solving absolute value of the complex quantity, and is obtained according to the motor nameplate parameter. Lambda [ alpha ]₀，λ_dcAll the parameters are adjustable parameters, and the parameters are obtained by a trial-and-error method, so that the overall performance of the system is optimal. Subscripti represent the parameters calculated from the four voltage vectors, respectively.

7. And applying the switching signal corresponding to the optimal voltage vector. Wherein the corresponding relation between the voltage vector and the switching signal is the same as that in the step (2). G to minimize cost function_iThe voltage vector of (2) is regarded as the optimal voltage vector among the four voltage vectors, the switch combination corresponding to the optimal voltage vector is applied, and the corresponding relation is shown in table 1, thereby realizing the optimal control of the system.

8. Repeating 1-7 at the next moment to obtain the optimal voltage vector at the next moment.

It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.

Claims (7)

1. A predictive control method for a variable-frequency speed control system of an induction motor under a three-phase four-switch power converter topology is characterized by comprising the following steps:

(1) three-phase current i is respectively measured by the existing current Hall sensor, voltage Hall sensor and photoelectric coded disc speed sensor in the induction motor driving system_a ^k,i_b ^k,i_c ^kTwo DC side capacitor voltages U_C1 ^k,U_C2 ^kAnd motor speed ω;

(2) by measuringTwo dc side capacitor voltages U of magnitude_C1 ^k,U_C2 ^kCalculating voltage vectors V corresponding to four switch combinations₁,V₂,V₃,V₄Value at the present moment and based on the measured three-phase currents i_a ^k,i_b ^k,i_c ^kSignal calculation of current vectorWherein the four switch combinations are 00,10,11, 01;

(3) by measuring motor speed omega and current vectorEstimating stator flux linkageAnd rotor flux linkage ${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k;$

(4) Predicting all voltage vectors V from a motor model and an inverter model₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torque

(5) Using predicted individual voltage vectors V₁,V₂,V₃,V₄Corresponding capacitance electricityPress and pressStator flux linkageAnd electromagnetic torqueCalculating a cost function, and taking a voltage vector which minimizes the cost function as an optimal voltage vector;

(6) and (3) applying a switching signal corresponding to the optimal voltage vector, wherein the corresponding relation between the voltage vector and the switching signal is the same as that in the step (2).

2. The method of claim 1, wherein step (2) is performed by measuring the capacitance voltage U_C1 ^k,U_C2 ^kCalculating voltage vectors V corresponding to four switch combinations₁,V₂,V₃,V₄The current time value is specifically:

voltage vector V corresponding to switch combination 00₁＝2·U_C2 ^k/3；

Voltage vector corresponding to switch combination 10 $V_2=({U_{C2}}^k-{U_{C1}}^k)/3-j\·\sqrt{3}({U_{C1}}^k+{U_{C2}}^k)/3;$

Voltage vector corresponding to switch combination 11 $V_3=({U_{C2}}^k-{U_{C1}}^k)/3+j\·\sqrt{3}({U_{C1}}^k+{U_{C2}}^k)/3;$

Voltage vector V corresponding to switch combination 01₄＝-2·U_C1 ^k/3。

3. A method according to claim 1 or 2, characterized in that in step (2) the measured three-phase currents i are used as a basis_a ^k,i_b ^k,i_c ^kSignal calculation of current vectorThe value of (d) is calculated according to the following formula:

${{\stackrel{\→}{i}}_s}^k={i_a}^k+j\·\frac{\sqrt{3}}{3}({i_a}^k+2\×{i_b}^k).$

4. a method according to claim 1 or 2, characterized in that step (3) is performed by measuring the motor speed ω and the current vectorEstimating stator flux linkageAnd rotor flux linkageThe method specifically comprises the following steps:

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k=\frac{{\mathrm{\τ}}_r}{T_s(1-\mathrm{j\ω}\·{\mathrm{\τ}}_r)}\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^{k-1}+\frac{L_m}{1-\mathrm{j\ω}\·{\mathrm{\τ}}_r}\·{{\stackrel{\→}{i}}_s}^k,$

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^k=k_r\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k+{\mathrm{\σL}}_s\·{{\stackrel{\→}{i}}_s}^k,$

wherein L is_s，L_m，L_rRespectively stator inductance, excitation inductance and rotor inductance, R_sR_rRespectively rotor resistance and stator resistance, T_sIs the sampling time, k_r＝L_m/L_rIs the mutual inductance coefficient of the rotor,is the magnetic leakage coefficient, τ_r＝L_r/R_rIs the rotor time constant.

5. The method according to claim 1 or 2, wherein all voltage vectors V are predicted in step (4) based on the motor model and the inverter model₁,V₂,V₃,V₄Corresponding capacitor voltageThe method specifically comprises the following steps:

predicted value U of capacitor voltage_C1(k+1)，U_C2(k +1) is

${{\hat{U}}_{C1}}^{k+1}={U_{C1}}^k-(1/C)\·{i_{\mathrm{dc}1}}^k\·T_s$

${{\hat{U}}_{C2}}^{k+1}={U_{C2}}^k+(1/C)\·{i_{\mathrm{dc}2}}^k\·T_s,$

Wherein the capacitance current i_dc1,i_dc2Is taken as follows

i_dc1 ^k＝i_b ^k·S_b+i_c ^k·S_c

i_dc2 ^k＝i_b ^k·(1-S_b)+i_c ^k·(1-S_c)，

S_b,S_cIndicating the switch state.

6. The method according to claim 1 or 2, wherein all voltage vectors V are predicted in step (4) based on the motor model and the inverter model₁,V₂,V₃,V₄Corresponding stator flux linkageAnd electromagnetic torqueThe method specifically comprises the following steps:

predicting a current vector at a next time instantThe following were used:

${{\stackrel{\→}{\hat{i}}}_s}^{k+1}=(1+\frac{T_s}{{\mathrm{\τ}}_{\mathrm{\σ}}})\·{{\stackrel{\→}{i}}_s}^k+\frac{T_s}{{\mathrm{\τ}}_{\mathrm{\σ}}+T_s}\·\{\frac{1}{R_{\mathrm{\σ}}}\·((\frac{k_r}{{\mathrm{\τ}}_r}-j\·k_r\·\mathrm{\ω})\·{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_r}^k+{{\stackrel{\→}{v}}_s}^k)\},$

wherein,is the equivalent resistance, L_σ＝σL_sIs the leakage inductance, tau, of the motor_σ＝σL_s/R_σ，Which represents a vector of voltages that is, ${{\stackrel{\→}{v}}_s}^k=\{V_1,V_2,V_3,V_4\};$

predicting stator flux linkage based on predicted current vectorAnd electromagnetic torque

${{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}={{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^k+T_s\·{{\stackrel{\→}{v}}_s}^k-R_s{T}_s\·{{\stackrel{\→}{i}}_s}^k,$

${{\hat{T}}_e}^{k+1}=\frac{3}{2}p\·\mathrm{Im}\{{{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}\·{{\stackrel{\→}{i}}_s}^{k+1}\},$

Wherein p is the induction machine pole pair number, and Im is represented by the imaginary part of a complex number.

7. A method according to claim 1 or 2, characterized in that in step (5) use is made of the predicted individual voltage vectors V₁,V₂,V₃,V₄Corresponding capacitor voltageStator flux linkageAnd electromagnetic torqueCalculating a cost function, specifically:

g according to a cost function_iCalculating each voltage vector separatelyQuantity V₁,V₂,V₃,V₄The value of the corresponding cost function is determined,

$g_i=\frac{|T_e^{*}-{\left({\hat{T}}_e^{k+1}\right)}_i|}{T_{e_{\mathrm{nom}}}}+{\mathrm{\λ}}_0\frac{\left|\right|\left|{{\stackrel{\→}{\mathrm{\ψ}}}_s}^{*}\right||-|\left|{\left({{\stackrel{\→}{\widehat{\mathrm{\ψ}}}}_s}^{k+1}\right)}_i\right|\left|\right|}{{\left|\right|{\stackrel{\→}{\mathrm{\ψ}}}_s\left|\right|}_{\mathrm{nom}}}+{\mathrm{\λ}}_{\mathrm{dc}}\frac{|{\left({{\hat{U}}_{C1}}^{k+1}\right)}_i-{\left({{\hat{U}}_{C2}}^{k+1}\right)}_i|}{V_{\mathrm{dc}}},i\∈\left\{\mathrm{1,2,3,4}\right\}$

whereinRespectively the rated torque and the rated flux linkage of the induction machine,for the purpose of the torque giving,for the given absolute value of the flux linkage, | - | symbol is the solving absolute value, | | | - | is the solving absolute value of the complex quantity, obtained according to the motor nameplate parameter, λ₀，λ_dcAll the parameters are adjustable parameters, the parameters are obtained through a compact test method, the overall performance of the system is optimal, and subscript i indicates the parameters calculated by the four voltage vectors respectively.