CN104040982B - The mechanism of correction is detracted for I/Q and detracts measurement using the transmitter for offseting local oscillator - Google Patents
The mechanism of correction is detracted for I/Q and detracts measurement using the transmitter for offseting local oscillator Download PDFInfo
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Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
- H04L27/36—Modulator circuits; Transmitter circuits
- H04L27/362—Modulation using more than one carrier, e.g. with quadrature carriers, separately amplitude modulated
- H04L27/364—Arrangements for overcoming imperfections in the modulator, e.g. quadrature error or unbalanced I and Q levels
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
- H04L27/38—Demodulator circuits; Receiver circuits
- H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
- H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
- H04L27/3863—Compensation for quadrature error in the received signal
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0016—Stabilisation of local oscillators
Abstract
The present invention is described for reducing the mechanism that I/Q is detracted in communication equipment.Transmitter performs precorrection before digital i/q signal is converted into simulation i/q signal to digital i/q signal, to pre-compensate for I/Q detractions, I/Q detractions will then be introduced in digital-to-analogue conversion, I/Q modulation or in order to produce in the other processing procedures transmitted signal and occurred.Receiver receives transmission signal, digital i/q signal is produced according to the transmission signal and filtering is performed to digital i/q signal, to correct I/Q detractions at multiple frequency offsets.In addition, also disclosing the mechanism for measuring transmitter and/or receiver I/Q detractions, the alternative manner of transmitter I/Q detractions, and the method that measuring receiver I/Q is detracted are measured using shared local oscillator or using the local oscillator deliberately offset.Also disclose for calculating I/Q detractions, for calculating between transmitters and receivers the DC attributes of signal path and for converting the method that I/Q is detracted by linear system according to the complex signal sampled.
Description
Technical field
The present invention relates to field of signal processing, and more specifically, it is related to for receiving device or sends I/Q in equipment
The system and method for detracting measurement and the correction of (impairment).
Background technology
Transmitter receive complex digital signal I (n)+jQ (n), the complex digital signal is converted into analog signal I (t)+
JQ (t), and utilize the I/Q modulators up-conversion analog signal.The signal of up-conversion is sent on channel.It is desirable that
Being supplied to the pure complex exponential tone (tone) of I/Q modulators will cause pure pitch to be sent.But, among reality, transmitter
In I/Q detraction will cause I channel and Q channel that there is different gains and different phase shifts.Among other things, this distortion is dark
Show that transmitted signal there will be undesirable energy in the frequency of the negative equal to pitch frequency.Dependent on communication standard, this
Planting undesirable " mirror image " causes planisphere (constellation diagram) or artificial noise floor (artificial
Noise floor) on potential distortion.The problem of receiver has similar.When receiver passes through the pure audio quilt under frequency f
During stimulation, except the energy under frequency f, the complex signal occurred in the output of the i/q demodulator of receiver will also include being in
Frequency-f undesirable signal energy.In both cases (transmitters and receivers), all due to I channel and Q signal it
Between gain and phase imbalance and cause difficulty.Thus, exist and subtract to the I/Q in transmitter and/or receiver can be corrected
The demand of the mechanism of damage.
In addition, in order to realize to the I/Q high-quality corrections detracted, it is necessary to the high-quality measurement that can be detracted using I/Q.But
It is that mass measurement is likely difficult to obtain.For example, the I/Q detractions of measurement transmitter, which are related to guide sender to receiver, sends letter
Number.I/Q detraction of the receiver based on its signal estimation transmitter received.But, the i/q demodulator its own of receiver
I/Q detraction destroy the estimation.In addition, the signal path between the I/Q modulators of transmitter and the I/Q demodulators of receiver
Also distortion is introduced to the estimation.Thus, there is the demand to following mechanism：It can estimate or measure transmitter and/or receiver
I/O detraction mechanism, can accurately measure implied in sampled signal I/Q detraction mechanism, can determine signal path
The mechanism of attribute and can predict I/Q detraction how by the mechanism of the system changeover of such as signal path.
The content of the invention
Among other things, this patent discloses the mechanism of the I/Q that can compensate in transmitter and/or receiver detractions.With
Come the parameter that performs compensation calculated based on the I/Q measured values detracted or estimate.For example, for compensate transmitter (or
Receiver) the parameters of I/Q detractions be that measured value or estimate based on those detractions are calculated.Any of technology is all
It can be detracted for the I/Q for the tandem compound for measuring or estimating transmitter or receiver or transmitters and receivers, including but
It is not limited to techniques disclosed herein.
In one embodiment, following operation can be related to for the I/Q of the compensated receiver system and methods detracted.
Analog input signal is received from transmitting medium.I/Q demodulation is performed to analog input signal, to produce analog in-phase
(I) signal and orthogonal (Q) signal of simulation.Then, simulation I signal and simulation Q signal are digitized, and are believed with producing digital I respectively
Number and digital Q signal.Digital iota signal and digital Q signal are filtered according to the 2x2 matrixes of digital filter, to produce filtering
Digital iota signal and the digital Q signal that has filtered.(filtering can be in such as FGPA programmable hardware element or such as
Performed in software in ASIC special digital circuit or on a processor, etc..) digital filter 2x2 matrixes at least
Partially compensate for the I/Q detractions of the receiver in a frequency range.The frequency of at least one diagonal components of the 2x2 matrixes
Response is the measurement of the I/Q detractions based on the function as frequency and calculated as the measurement of the function of the negative of frequency.
(measurement of the I/Q detractions of receiver can be obtained by any of method.This document is described for obtaining many of this measurement
The method of kind.) in addition, the frequency response of at least one non-diagonal component of the 2x2 matrixes is the survey based on the function as frequency
Amount and calculated as the measurement of the function of the negative of frequency.
In some embodiments, it can be assumed that receiver on positive frequency I/Q detraction and receiver negative frequency it
On I/Q detraction be functional dependence.(A) in a kind of such embodiment, the frequency response of 2x2 matrixes can be counted as follows
Calculate.The frequency response of at least one diagonal components of the 2x2 matrixes under optional frequency f can be based only upon the I/Q under frequency f
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of detraction is calculated.In addition, under frequency f
2x2 matrixes at least one non-diagonal component frequency response can be based only upon under frequency f I/Q detraction measurement (or
Person is alternatively, the measurement for the I/Q detractions being based only upon under frequency-f) calculate.(B) in another such embodiment,
Assuming that gain imbalance be even number and assume phase it is crooked be odd number.Then, two non-diagonal components of 2x2 matrixes can
It is arranged to zero；One of diagonal components can correspond to pure straight-through wave filter (that is, cell frequency is responded)；And in any frequency
The frequency response of another diagonal components under rate f can be based only upon the measurement of the I/Q detractions under frequency f (or as replacing
In generation, it is based only upon the measurement of the detractions of the I/Q under the frequency-f) calculate.(C) in another such embodiment, 2x2 matrixes
Two diagonal components can correspond to pure straight-through wave filter；One of non-diagonal component can be arranged to zero；And it is in office
The frequency response of another non-diagonal component under meaning frequency f can be based only upon the I/Q detractions under frequency f measurement (or
Alternatively, the measurement for the I/Q detractions being based only upon under frequency-f) calculate.
In another embodiment, for receiver is configured to compensated receiver at least in part I/Q detract be
System and method can be related to following operation.
Receive the measurement (or from memory access) of the I/Q detractions of receiver on a frequency band.Based on the measurement,
Calculate the 2x2 matrixes of digital filter.The 2x2 matrixes of digital filter are calculated, are subtracted with the I/Q to the receiver on the frequency band
Damage and realize at least partly compensation.The frequency response of at least one diagonal components of 2x2 matrixes is based on the function as frequency
Measurement and calculated as the measurement of the function of the negative of frequency.In addition, the frequency of at least one non-diagonal component of 2x2 matrixes
Rate response is the measurement of the function of the measurement based on the function as frequency and the negative as frequency to calculate.Then, number
Word circuit is programmed to implement the 2x2 matrixes of digital filter.When such programming, digital circuit is configured to mend at least in part
Repay the I/Q detractions of the receiver on the frequency band.Digital circuit can in a variety of forms in any one realize.For example, digital
Circuit can be by programmable hardware element or by such as ASIC special digital circuit or by processor response in program
The execution of instruction is realized.(digital circuit can be incorporated as a part for receiver or be used as another system (example
Such as master computer or controller board) a part).
In another embodiment, it can be related to for operating transmitter so as to realize that I/Q detracts the system and method compensated
And following operation.
Receive digital inphase (I) signal and digital quadrature (Q) signal.Digital iota signal and digital Q signal are according to digital filtering
The 2x2 matrixes of device are filtered, with the digital Q signal for producing the digital iota signal filtered He having filtered.The 2x2 of digital filter
Matrix pre-compensates for the I/Q detractions of the transmitter in a frequency range at least in part.Diagonal point of at least one of 2x2 matrixes
The frequency response of amount be based on the function as frequency I/Q detraction measurement and as frequency negative function measurement come
Calculate.(measurement of the I/Q detractions of transmitter can be obtained by any of method.This document describes to be used to obtain this
A variety of methods of measurement.) moreover, the frequency response of at least one non-diagonal component of 2x2 matrixes is based on the letter as frequency
Several measurement and calculated as the measurement of the function of the negative of frequency.Then, the digital iota signal and numeral Q filtered is believed
Number analog form is converted into, to obtain corresponding simulation I signal and simulation Q signal.I/Q modulation can be to simulation I signal
Performed with simulation Q signal, to produce the analog signal modulated.
In some embodiments, it can be assumed that I/Q detraction and transmitter of the transmitter in positive frequency are in negative frequency
I/Q detractions are functional dependences.(A) in a kind of such embodiment, the calculating of the 2x2 matrixes of digital filter can be as follows
Simplify.The frequency response of at least one diagonal components of the 2x2 matrixes under optional frequency f in the frequency range can only base
Counted in the measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of the I/Q detractions under frequency f
Calculate.In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes under frequency f can be based only upon under frequency f
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions is calculated.(B) another
In such embodiment, it is assumed that gain imbalance be even number and assume phase it is crooked be odd number.Then, two of 2x2 matrixes
Non-diagonal component can be arranged to zero；One of diagonal components can correspond to pure straight-through wave filter, and (that is, cell frequency rings
Should)；And the frequency response of another diagonal components under optional frequency f can be based only upon the I/Q detractions under frequency f
Measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) calculate.(C) another such
In embodiment, two diagonal components of 2x2 matrixes can correspond to pure straight-through wave filter；One of non-diagonal component can be with
It is arranged to zero；And another frequency response of non-diagonal component under optional frequency f can be based only upon the I/Q under frequency f
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of detraction is calculated.
In another embodiment, the I/Q detractions for transmitter being configured to compensate transmitter at least in part are
System and method can be related to following operation.
Receive the measurement (or from memory access) of the I/Q detractions of transmitter in a frequency range.Based on this
The 2x2 matrixes of survey calculation digital filter.The 2x2 matrixes of digital filter are calculated, to be realized extremely to the I/Q of transmitter detractions
Small part is pre-compensated for.The frequency response of at least one diagonal components of 2x2 matrixes be measurement based on the function as frequency and
It is used as the survey calculation of the function of the negative of frequency.In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes is
Measurement based on the function as frequency and as frequency negative function survey calculation.Then, digital circuit is compiled
Journey is to realize the 2x2 matrixes of digital filter.When such programming, digital circuit is configured to pre-compensate for transmitter at least in part
I/Q detraction.
In another embodiment, for operating transmitter to be realized so as to be detracted to I/Q of the transmitter under given frequency f
The system and method for at least partly compensation can be related to following operation.
Receive digital inphase (I) signal and digital quadrature (Q) signal.Digital iota signal and digital Q signal are according to constant
2x2 matrixes are converted, to produce resulting number I signal and resulting number Q signal.(in other words, including digital iota signal sum
The vector signal of word Q signal and the 2x2 matrix multiples).Resulting number I signal and digital Q signal are converted into analog form, with
Just corresponding simulation I signal and simulation Q signal are obtained.I/Q modulation is performed to simulation I signal and simulation Q signal, to produce modulation
Analog signal.2x2 matrix configurations are the I/Q detractions pre-compensated at least in part under frequency f.Corresponding to 2x2 matrixes
First constant of one diagonal element is that the measurement based on the I/Q detractions under frequency f and the I/Q under frequency-f are detracted
Measure to calculate.In addition, second constant corresponding to an off-diagonal element of 2x2 matrixes is based under frequency f
Measurement and survey calculation under frequency-f.
In another embodiment, for determining and (that is, measuring) that it is following dynamic that the method for I/Q detractions of transmitter can be related to
Make.
This method is related to one group of operation of execution.This group operation includes：(a) the complex exponential tone under frequency f is guided to be carried
Supply transmitter；(b) precompensation conversion is supplied to the pre-compensation circuit of transmitter, wherein pre-compensation circuit is configured to refer to multiple
Number tone applies precompensation conversion, to obtain the complex signal that have adjusted, wherein precompensation alternate arrangement is that the I/Q of transmitter is subtracted
The current estimation damaged is pre-compensated for, and the complex signal that wherein transmitter is configured to have adjusted sends signal, wherein connecing
Receive the complex signal that device is configured to receive the transmission signal and catch the sampling for sending signal received by representing；(c) base
Original I/Q detractions are calculated in the complex signal sampled；(d) original I/Q detractions are converted, to determine that the I/Q converted is detracted,
Wherein described conversion removes the measured I/Q detractions of receiver from original I/Q detractions；(e) from the I/Q converted
Detraction removes the current estimation of signal path, is detracted with the I/Q for obtaining path compensation, wherein signal path is included from transmitter
I/Q modulators to the path of the demodulator of receiver；And the I/Q detractions of (f) based on path compensation update the I/Q of transmitter
The current estimation of detraction.(architectural framework of receiver is depended on, demodulator can be i/q demodulator or not be I/Q demodulation
Device).
In another embodiment, for determining that the method that the I/Q of transmitter is detracted can be related to following action.
The local oscillator (LO) of local oscillator (LO) and receiver that this method can include configuration transmitter is quilt
Public reference is phase-locked to, and causes the LO of receiver frequency to subtract the LO of transmitter frequency equal to (for example, complete etc.
In) amount Δ LO.
This method can also include performing one group of operation, and wherein this group operation includes：(a) the multiple finger under frequency f is guided
Number tone is provided to transmitter；(b) precompensation conversion is supplied to the pre-compensation circuit of transmitter, wherein pre-compensation circuit is matched somebody with somebody
It is set to and applies precompensation conversion to complex exponential tone, to obtain the complex signal that have adjusted, wherein precompensation alternate arrangement is to hair
The current estimation for sending the I/Q of device to detract is pre-compensated for, and the complex signal that wherein transmitter is configured to have adjusted is sent
Signal, wherein receiver are configured to answering for the sampling for receiving the transmission signal and catching the transmission signal received by representing
Signal；(c) Δ LO makes the complex signal frequency displacement of sampling according to quantity, to obtain the signal of frequency displacement；(d) letter based on the frequency displacement
Number calculate original I under frequency f/Q detractions；(e) the current of signal path is removed from the original I under frequency f/Q detractions to estimate
Meter, is detracted with the I/Q for obtaining the path compensation under frequency f, wherein signal path include from the I/Q modulators of transmitter to
The path of the demodulator of receiver；And the I/Q detractions of (f) based on the path compensation under frequency f, update under frequency f
The current estimation of the I/Q detractions of transmitter.(architectural framework of receiver is depended on, demodulator can be i/q demodulator or not
It is i/q demodulator).
In another embodiment, for determining and (that is, measuring) that it is following dynamic that the method for I/Q detractions of receiver can be related to
Make.
This method can be related to guide input signal and be provided to receiver, and wherein the input signal is included in displacement frequency
Isolation tone under f and the invalid interval (void interval) being included in around displacement frequency-f.(in a kind of embodiment
In, receiver includes the calibration tone generator for being configured to generate input signal).Receiver is configured to demodulate input signal, with
Just the complex signal sampled is obtained.Displacement frequency f and-f are the displacements of the local oscillator frequencies relative to receiver.
This method can also relate to calculate the I/Q detractions of the receiver under frequency f based on the complex signal sampled.
This method can also relate to the action for repeating to guide and calculate to the value of the frequency f across assigned frequency band.
This method can also relate to store the I/Q detractions of the receiver of the value for these frequencies f in memory.
In another embodiment, for the side for the I/Q detractions for estimating to associate with the sampled complex produced by receiver
Method can be related to following action.
Equipment is directed to stimulate receiver using stimulus signal, and the stimulus signal has the isolation under displacement frequency f
Tone and the invalid interval under displacement frequency-f.(displacement frequency f and-f are the positions of the local oscillator frequencies on receiver
Move.The complex signal sampled can be the baseband signal that receiver is produced).Calculated for the I component of sampled complex under frequency f
Discrete time Fourier transformed value CI.The discrete time Fourier conversion under frequency f is calculated for the Q component of sampled complex
Value CQ.The gain of sampled complex under frequency f is unevengIt is to be based on value CIAnd CQMagnitude calculation.Gain imbalance g
At least the gain including receiver is uneven.The phase of sampled complex under frequency f is crookedIt is to be based on value CIAnd CQ's
Phase calculation, wherein phase is crookedAt least the phase including receiver is crooked.
In another embodiment, for estimating signal between the I/Q modulators of transmitter and the i/q demodulator of receiver
The method of the DC scalings in path can be related to following operation.In order to promote this method of estimation, the output of transmitter can be such as
The input of receiver is coupled to via cable.
Transmitter is directed is fed as input to I/Q modulators zero-signal.The first response signal is received, first sound
Induction signal has responded to capture from i/q demodulator in the offer zero-signal.Transmitter is directed normal again non-zero is equal to
The constant signal of amount is fed as input to I/Q modulators.Receive the second response signal, second response signal have responded in
The constant signal is provided and captured from i/q demodulator.First response signal is averaged, to obtain the first average value, and
And second response signal be averaged, to obtain the second average value.Calculate the difference of the second average value and the first average value.Based on this
Difference and the multiple constant of the non-zero calculate DC scalings.In addition, the DC rotations of signal path can be based on the difference phase and this is non-
Zero answers the phase of constant to calculate.DC is scaled and DC rotational energies are used for removing signal road from the I/Q detractions measured in receiver
The influence in footpath, to obtain the estimation that the I/Q of transmitter is detracted.
In above-mentioned DC scales/rotated a kind of alternative embodiment of method of estimation, transmitter does not have (or with negligible
) local oscillator leakage.(this is probably following situation, for example, when transmitter has different from direct converting system framework
During other RF architectural frameworks).Therefore, it is possible to omit the transmission of zero-signal, the seizure of the first response signal, the meter of the first average value
Calculation and the calculating of difference.Then, DC scalings are calculated based on the second average value and the multiple constant of non-zero.DC rotations are flat based on second
The phase and non-zero of average answer the phase of constant to calculate.
In another embodiment, detract and count for the I/Q based on the multiple input (that is, I/Q input to) in electronic system
The method calculated in the I/Q detractions of the multiple output (that is, I/Q output to) of electronic system can include following operation.
Frequency spectrum A (f) is calculated according to following formula,
Wherein H (f) is the frequency spectrum of the Linear system model of electronic system, and wherein g (f) is the gain injustice in multiple input
Weighing apparatus, whereinIt is that phase in multiple input is crooked.Frequency spectrum B (f) is calculated according to following formula.
Calculate frequency spectrum A (f) and B (f) sums and frequency spectrum A (f) and B (f) difference.Real and imaginary parts based on this and value with
And the gain that the real and imaginary parts of the difference are calculated in multiple output is uneven crooked with phase.
In some embodiments, the electronic system modeled by spectrum H (f) is from the I/Q modulators of transmitter to receiver
Demodulator signal path reversion, for example, as herein in a variety of different ways describe as.In electronic system
Gain in multiple input is uneven and phase is crooked can represent the input in demodulator (or instead, in demodulator
Output) in gain it is uneven and phase is crooked.Gain in the multiple output of electronic system is uneven and phase is crooked can be with
Representative is uneven crooked with phase in the gain of the output of I/Q modulators.
This document describes communication equipment and for reducing the correlating method that I/Q is detracted in the signal used by the communication equipment
Various embodiments.According to a kind of embodiment, receiving device can receive through communication media and send signal, and can be to being connect
The transmission signal received performs I/Q demodulation, to produce a pair of simulation I (same to phase) and Q (orthogonal) signal.Receiving device can be performed
The analog-to-digital conversion of each in I signal and simulation Q signal is simulated, to produce corresponding digital iota signal and digital Q signal.Produced
Raw digital iota signal and digital Q signal can have the I/Q caused by I/Q demodulation and/or analog-to-digital conversion and/or other processing
Detraction.Receiving device is configurable to perform digital iota signal and digital Q signal broadband I/Q detraction corrections, is subtracted with correcting I/Q
Damage.I/Q detraction corrections in broadband can compensate uneven and unbalance in phase the frequency of gain in digital iota signal and digital Q signal
Rate associated change, for example, digital iota signal sum can be compensated at multiple frequency shift (FS)s of the instant bandwidth across receiving device
Gain imbalance and unbalance in phase in word Q signal.
The correction of broadband I/Q detractions is performed to digital iota signal and digital Q signal can include believing digital iota signal and numeral Q
It is one or more in number to be filtered, to produce resulting number I signal and resulting number Q signal.Resulting number I signal and numeral
Q signal represents the signal corrected.In some embodiments, resulting number I signal is constantly equal to digital iota signal, and number of results
Word Q signal be by digital iota signal and digital Q signal it is one or more be filtered it is one or more corresponding to obtain
Filtering signal and generated by the way that the one or more signal filtered is added.In other embodiments,
Resulting number Q signal is constantly equal to digital Q signal, and resulting number I signal is by digital iota signal and digital Q signal
It is one or more to be filtered to obtain one or more corresponding signals filtered and by the one or more filter
The signal of ripple is added to generate.In other other embodiments, resulting number I signal be by digital iota signal and
It is one or more in digital Q signal to be filtered to obtain one or more corresponding signals filtered and by this
Individual or multiple signals filtered are added to generate；And resulting number Q signal is by believing digital iota signal and numeral Q
One or more signals being filtered to obtain one or more filtering additional accordingly and by this in number
The signal of individual or multiple additional filtering is added to generate.
In further embodiments, by multiple known test signals be supplied to receiving device and measure in response to
The known test signal and the I/Q that is introduced by receiving device is detracted, calibration system (or receiving device itself) can be determined
Control information.(in one embodiment, receiving device can include the calibration tone generator of generation known test signal).It is wide
It is uneven uneven with phase that band I/Q detraction corrections can compensate gain in digital iota signal and digital Q signal using control information
The frequency dependence change of weighing apparatus.
In some embodiments, calibration system can be in off-line calibration stage and on-line operation stages operating.Perform offline
Calibration phase can include providing multiple known test signals to receiving device, measure in response to the known test signal
And detracted by the I/Q that receiving device is introduced and control information is determined based on measured I/Q detractions.Perform on-line operation rank
Section can include receiving transmission signal through communication media, and the transmission signal received is performed I/Q demodulation to produce simulation I signal
Perform analog-to-digital conversion to produce digital iota signal and numeral with simulation Q signal, to each in simulation I signal and simulation Q signal
Q signal, and I/Q detraction corrections in broadband are performed to digital iota signal and digital Q signal.I/Q detraction corrections in broadband can be used
Gain in digital iota signal and digital Q signal is uneven and phase is uneven to compensate for the control information determined in the off-line calibration stage
The frequency dependence change of weighing apparatus.
In some embodiments, the off-line calibration stage can be energized and perform in response to receiving device.In some implementations
In example, receiving device can be in response to determining the off-line calibration stage to complete automatically into the on-line operation stage.In some implementations
In example, receiving device can be in response to determining that receiver is not busy with the transmission signal that processing is received in the on-line operation stage
And it is switched to the off-line calibration stage from the on-line operation stage automatically.In some embodiments, the off-line calibration stage can be in response to
User inputs and started.
According to other embodiments, the digital I to be sent (same to phase) and Q (orthogonal) signal can be received by sending equipment.Send
Equipment can perform broadband I/Q detraction precorrection to digital iota signal and digital Q signal.Perform the dynamic of broadband I/Q detraction precorrection
Work can be related to be filtered to produce resulting number I signal and result to one or more in digital iota signal and digital Q signal
Digital Q signal, so that it is uneven and unbalance in phase to pre-compensate for the gain being then introduced into transmission signal building-up process
Frequency dependence changes.Transmission signal can be synthesized using resulting number I signal and resulting number Q signal.
The action of synthesis transmission signal can include performing digital-to-analogue conversion to resulting number I signal and resulting number Q signal,
To produce simulation I signal and simulation Q signal, and using I signal and simulation Q signal execution I/Q modulation is simulated, to produce transmission
Signal.Resulting number I signal and resulting number Q signal can be caused to one or more in the digital-to-analogue conversion and I/Q modulation
Uneven and unbalance in phase the frequency dependence change of gain is pre-compensated for.
In some embodiments, resulting number I signal is constantly equal to digital iota signal, and resulting number Q signal is by right
It is one or more in digital iota signal and digital Q signal to be filtered to obtain one or more corresponding filtering signals and lead to
Cross and the one or more filtering signal is added to generate.In other embodiments, resulting number Q signal is constantly equal to digital Q
Signal, and resulting number I signal is by being filtered to one or more in digital iota signal and digital Q signal to obtain
One or more corresponding filtering signals and generated by the way that the one or more filtering signal is added.Other
In embodiment, resulting number I signal is by being filtered to one or more in digital iota signal and digital Q signal to distinguish
Obtain one or more filtering signals and generated by the way that the one or more filtering signal is added；And resulting number
Q signal is by being filtered to one or more in digital iota signal and digital Q signal to obtain one or more add respectively
Filtering signal and generated by the way that the one or more additional filtering signal is added.
In further embodiments, by providing multiple known digital test signals to transmission equipment and measuring
The I/Q introduced in response to the known test signal by transmission equipment is detracted, and calibration system can determine control information.It is wide
Band I/Q detraction precorrection can produce resulting number signal using control information.
In some embodiments, sending equipment can operate in off-line calibration stage and on-line operation stage.Offline school
The quasi- stage can include providing multiple known test signals to transmission equipment, measure in response to the known test signal
The I/Q detractions introduced by transmission equipment, and control information is determined based on the I/Q detractions measured.
In some embodiments, the off-line calibration stage can be energized and perform in response to sending equipment.In some implementations
In example, sending equipment can be in response to determining to the off-line calibration stage to complete automatically into the on-line operation stage.In some realities
Apply in example, send equipment can in response to determine to transmitter to be not busy with sending signal in the on-line operation stage and it is automatic from
The on-line operation stage is switched to the off-line calibration stage.In some embodiments, the off-line calibration stage can input in response to user
And start.
The on-line operation stage can include receiving the digital iota signal and digital Q signal to be sent, and to digital iota signal
Broadband I/Q detraction precorrection is performed with digital Q signal.Performing the action of broadband I/Q detraction precorrection can use in offline school
The control information determined in the quasi- stage is filtered to one or more in digital iota signal and digital Q signal, to produce result
Digital iota signal and resulting number Q signal, so that it is uneven to pre-compensate for the gain being then introduced into transmission signal building-up process
Change with the frequency dependence of unbalance in phase.Transmission signal can be closed using resulting number I signal and resulting number Q signal
Into.
According to another embodiment, measuring system can include receiving device and equipment under test.Receiving device can be configured
To receive the transmission signal including the measurement data collected from equipment under test, I/Q is performed to the transmission signal received
Demodulation is to produce simulation I (same to phase) and Q (orthogonal) signal, and the modulus of each performed in simulation I signal and simulation Q signal turns
Change and corrected with producing digital iota signal and digital Q signal, and broadband I/Q detractions being performed to digital iota signal and digital Q signal.It is wide
Uneven and unbalance in phase the frequency dependence change of gain in digital iota signal and digital Q signal can be compensated by detracting correction with I/Q
Change.
In further embodiments, measuring system can also include sending equipment.Transmission equipment is configurable to receive and wanted
The digital iota signal and digital Q signal of transmission.Digital iota signal and digital Q signal can specify that the letter of equipment under test to be sent to
Breath.Equipment is sent to be also configured as performing digital iota signal and digital Q signal broadband I/Q detraction precorrection.Perform broadband I/
The action of Q detraction precorrection can be related to be filtered to produce result to one or more in digital iota signal and digital Q signal
Digital iota signal and resulting number Q signal, so that it is uneven to pre-compensate for the gain being then introduced into signal building-up process is sent
Change with the frequency dependence of unbalance in phase.Sending equipment can be passed using resulting number I signal and the synthesis of resulting number Q signal
Defeated signal, and transmission signal is sent to equipment under test.
Brief description of the drawings
When considering the following specifically describes in conjunction with the following drawings, it can obtain and the present invention is best understood from.
Figure 1A illustrates a kind of possible application of compensation method disclosed herein, wherein mobile device 10 and/or wireless
Transmitting-receiving station 15 applies digital pre-compensation to the signal transmitted by them and/or the signal that they are received is applied to be mended after numeral
Repay.
Figure 1B illustrates the alternatively possible application of compensation method disclosed herein, and wherein tester 20 is sent to it
Signal to tested receiver 25 applies digital pre-compensation, to remove the influence of its I/Q detractions.
Fig. 1 C illustrate the yet further possibility application of compensation method disclosed herein, wherein tester 35 to it from
The signal that tested transmitter is received applies digital post-compensation, to remove the influence of its I/Q detractions.
Fig. 2A illustrates for operating receiver to realize that a kind of the of method of at least part I/Q detraction compensation implements
Example.
Fig. 2 B illustrate to be arranged for carrying out a kind of embodiment of the receiver of at least part I/Q detraction compensation.
Fig. 3 is illustrated for receiver to be configured so that the method that receiver can compensate I/Q detractions at least in part
A kind of embodiment.
A kind of embodiment for the method that Fig. 4 illustrates for operating transmitter to realize at least part I/Q detraction compensation.
Fig. 5 illustrates to be arranged for carrying out a kind of embodiment of the transmitter of at least part I/Q detraction compensation.
Fig. 6 is illustrated for transmitter to be configured so that the method that transmitter can compensate I/Q detractions at least in part
A kind of embodiment.
Fig. 7 illustrates to be configured to provide for a kind of embodiment of the system of I/Q detraction compensation.I/Q detractions are modeled as completely
Appear on Q channel.
Fig. 8 illustrates to be configured to provide for another embodiment of the system of I/Q detraction compensation.I/Q detractions have been modeled as
It is complete to occur on the i channel.
Fig. 9 illustrates to be configured to provide for the yet another embodiment of the system of I/Q detraction compensation.I/Q detractions are modeled as
Partly appear on two channels.
Figure 10 illustrates for operating receiver to realize the I/Q detractions under frequency f the side of at least partly compensation
A kind of embodiment of method.
Figure 11 illustrates to be configured to realize that one kind of at least partly receiver of compensation is real to the I/Q detractions under frequency f
Apply example.
Figure 12 is illustrated for receiver to be configured so that receiver can be realized extremely to the I/Q detractions under frequency f
A kind of embodiment of the method for small part compensation.
Figure 13 illustrates for operating transmitter to realize the I/Q detractions under frequency f the side of at least partly compensation
A kind of embodiment of method.
Figure 14 illustrates to be configured to realize that one kind of at least partly transmitter of compensation is real to the I/Q detractions under frequency f
Apply example.
Figure 15 illustrates the system for being appeared in the complex exponential tone stimulation of the complex exponential tone of system output and distortion,
Wherein distortion is characterized by gain is uneven and phase is crooked.
Figure 16 illustrates wherein gain imbalance and the crooked system fully appeared on Q channel of phase.
Figure 17 illustrates a kind of embodiment of the system for performing detraction compensation at a single frequency.
Figure 18 illustrates the 2x2 system models for performing I/Q detraction compensation.
Figure 19 illustrates wherein embodiments of the impairment model G prior to compensation model H.
Figure 20 A illustrate that wherein impairment model G follows the embodiment after compensation model H.
Figure 21 on having frequency response U (f) and V (f) a pair of digital filters to illustrate to be used for compensation model H respectively
A kind of embodiment.
Figure 22 illustrates Figure 21 improvement figure, and wherein U and V are represented with its even segments and odd number part.
Figure 23 illustrates the equivalently represented of Figure 22 systems, wherein strange frequency spectrum B and D followed by Hilbert with converting
Corresponding even frequency spectrum is replaced.
The response for the system input corresponding to two that Figure 24 A illustrate Figure 23 with 24B.
Figure 25 provides the equation derived respectively from Figure 24 A and 24B.
Figure 26 A and 26B illustrate the polar plot (phasor diagram) corresponding to Figure 25 equatioies.
Figure 27 is provided detracts information regulation compensation spectrum A, E according on I/QB, C and EDA kind of embodiment equation.
Figure 28 illustrates to represent the 2x2 models H of the I/Q detractions of system.
Figure 29 illustrates a kind of model H embodiment on frequency U and V.
Figure 30 illustrates Figure 29 improvement figure, and wherein U and V are represented with their even segments and odd number part.
Figure 31 illustrates the equivalently represented of Figure 30 systems, wherein strange frequency spectrum B and D is replaced with corresponding even frequency spectrum, Zhi Houshi
Hilbert is converted.
Figure 32 A and 32B illustrate Figure 31 system to two responses accordingly inputted.
Figure 33 gives the equation derived respectively from Figure 32 A and 32B.
Figure 34 A and 34B illustrate the polar plot corresponding to Figure 33 equatioies.
Figure 35 gives the matrix equality derived from Figure 34 A and 34B polar plot.
Figure 36 gives the solution to Figure 35 matrix equalities.
Figure 37 is illustrated
A kind of embodiment of system.
Figure 38 illustrate LO leakages vector A, the DC vector B deliberately injected and they and C.
Figure 39 illustrates to correspond respectively to vectorial A, B and C response vector A ', B ' and C '.
Figure 40 illustrates a kind of embodiment of the method for calculating DC mapping values for signal path.
Figure 41 illustrates the system with frequency response H (f), and the system is by crooked with gain imbalance g (f) and phaseInput signal sinput(f, t) is stimulated and produced crooked with gain imbalance g ' (f) and phaseOutput
Signal soutput(f,t)。
Figure 42 provides the equation derived from Figure 41.
Figure 43 illustrates to be used for convert a kind of embodiment of the method for I/Q detractions by linear system H (f).
Figure 44 illustrates a kind of embodiment for the method that the I/Q for determining transmitter is detracted.
Figure 45 illustrates that the local oscillator (intentionally-displaced) using intentional displacement determines to send
A kind of embodiment of the method for the I/Q detractions of device.
Figure 46 illustrates a kind of embodiment for the method that the I/Q for determining receiver is detracted.
Figure 47 illustrates a kind of embodiment of the method for the I/Q detractions for estimating to associate with complex signal.
Figure 48 illustrates a kind of embodiment of the system for measuring transmitter and/or receiver I/Q detractions, and wherein this is
System includes the transmitters and receivers that its local oscillator frequencies deliberately offset by.
Figure 49 illustrates the frequency for the signal being received by the receiver in response to transmitter to the transmission in 31MHz tone
Spectrum.Local oscillator frequencies high 6MHz of the local oscillator frequencies of transmitter than receiver.Thus, in the frequency spectrum received
In, tone appears in 37 MHz.
Figure 50 illustrates the frequency spectrum received after the I/Q detractions of receiver are removed.
Figure 51 illustrates the frequency spectrum of Figure 50 after frequency displacement.
Figure 52 illustrates the frequency spectrum of the frequency displacement in the case of without the detraction of receiver is removed first.
Figure 53 A illustrate the vectorial calibration correction 5310 of single-point, are the vectorial damage model 5320 of two point afterwards.
Figure 53 B show Figure 53 A improvement figure, and the wherein vectorial calibration correction of single-point is determined by constant α and β, and
Wherein the destruction of two point vector is by constant A, EB, C and EDDetermine.
Figure 54 illustrates the polar plot corresponding to Figure 53 B right hand portions (that is, on the right of dotted line).
Figure 55 A illustrate to include the receiver of receiver wave filter 5525 and i/q demodulator 5530.
The system that Figure 55 B illustrate to include the transmitters and receivers being coupled together.The system may be used to determine hair
The I/Q of device and/or receiver is sent to detract.
Figure 55 C are illustrated along three points from the I/Q modulators of transmitter to the path of the i/q demodulator of receiver
The tone under frequency f at place and the relative magnitude of the mirror image under-f.
Figure 56 A illustrate the rate of convergence of the function as value evaluated error.
Figure 56 B illustrate the rate of convergence of the function as rotation (phase) evaluated error.
Figure 57 introductions are used for the complex amplitude α of tone and by crooked by gain imbalance g (f) and phase
And the complex amplitude β for the mirror image that the complex signal of distortion is carried notation.
Figure 58 A and 58B derive crooked according to gain imbalance g (f) and phaseCharacterize the equation of tone and mirror image.
Figure 59 illustrates gain not according to Q channel signal (" Q is actual ") relative to the distortion of I channel signal (" I references ")
Balance g (f) and phase is crooked
Figure 60 and 61 is shown for phase and quadrature signal component (that is, " I references " and " Q is actual " letter for Figure 59
Number) magnitude spectrum.
Figure 62 illustrates to be used to calculate local oscillator leakage, signal amplitude, gain imbalance, mirror according to a kind of embodiment
As suppressing and the crooked LabVIEW graphic packages of phase.
Figure 63 illustrate to receive the data that are calculated by programmable hardware element (for example, FPGA of receiver) and according to
The data calculate LO leakages, amplitude gain imbalance and the crooked LabVIEW graphic packages (VI) of phase.
Figure 64 and 65 shows the rectangular window letter of the public sample rate with different acquisition length and with 120MHz
The figure of several amplitude frequency spectrums.
Figure 66 illustrates that there is its complex input signal I/Q to detract gin(ω) andAnd its complex output
G is detracted with I/Qout(ω) andSystem model.
Figure 67 is given according to input I/Q detractions gin(ω) andWith output I/Q detractions gout(ω) andCarry out regulation Figure 66 frequency response function U (ω) and V (ω) equation.
Figure 68 is illustrated can be real for the one kind for the computer system 6800 for performing any means embodiment described herein
Apply example.
Although the present invention easily has various modifications and alternate forms, it is shown in the drawings as an example and herein
Its specific embodiment is described in detail.It is understood, however, that accompanying drawing and detailed description are not meant to the present invention to be limited to institute
Disclosed particular form, on the contrary, the present invention, which is intended to cover, belongs to spirit and scope of the present invention defined by the appended claims
All modifications, equivalent and alternative arrangement.It should be pointed out that the various pieces title in detailed description below is used for the purpose of
Organize rather than mean for limiting claim.
Embodiment
Term
The following is the nomenclature of term used in this application.
Any one in storage medium-various types of memory devices or storage facilities." memory is situated between term
Matter " is intended to include：Install medium, such as CD-ROM, diskette 1 05 or belt-type apparatus；Computer system memory is deposited at random
Access to memory, DRAM, DDR RAM, SRAM, EDO RAM, Rambus RAM etc.；Nonvolatile memory, such as flash memory,
Magnetic medium (such as hard disk driver), or optical storage device；Register, or other similar types memory component, etc..Storage
Device medium can include other types of memory and combinations thereof.In addition, storage medium can be located therein configuration processor
In first computer, or the network connection through such as internet can be located to the different second computers of the first computer
In.In the latter case, second computer can provide the programmed instruction to be performed to the first computer.Term " memory
Medium " can include may reside within two or more storages in diverse location (such as the different computers through network connection)
Device medium.
Programmable hardware element-include various hardware devices, the hardware device is included through many of programmable interconnection connection
Individual programmable function blocks.Example includes FPGA (field programmable gate array), PLD (programmable logic device), FPOA (scene can
Programming object array), and CPLD (complicated PLD).Programmable function blocks can from fine granularity (combinational logic or look-up table) to
Coarseness (ALU or processor core) changes.Programmable hardware element can also be referred to as " reconfigurable logic ".
Any one in computer system-various types of calculating or processing system, including personal computer system
(PC), large computer system, work station, the network equipment, internet device, personal digital assistant (PDA), system for TV set,
Computing system, or miscellaneous equipment or equipment combination.In general, term " computer system " can be defined widely
To include any equipment (or combination of equipment) of the processor with least one instruction of the execution from storage medium.
Local oscillator (LO)-be configured to the circuit that generation is in the cyclical signal of assigned frequency and amplitude.The cycle
Property signal can be pure sinusoid, and its frequency and/or amplitude can be programmable.The cyclical signal can be or
Phase or Frequency Locking be can not be to another cyclical signal.
Embodiments of the invention can any one central realization in a variety of manners.For example, in some embodiments, this
Invention can be implemented as computer implemented method, computer-readable storage medium, or computer system.In other realities
Apply in example, the present invention can be realized using the hardware device of one or more such as ASIC custom design.Implement other
In example, the present invention can be realized using one or more such as FPGA programmable hardware element.
In some embodiments, computer readable memory medium may be configured such that its storage program instruction and/or
Data, wherein, if programmed instruction is executed by a computer system, cause computer system to perform a kind of method, for example herein
Described any embodiment of the method, the either any combination of method embodiments described herein or any side as described herein
The random subset of method embodiment, or this subset any combination.
In some embodiments, computer system is configurable to include processor (or one group of processor) and memory
Medium, wherein storage medium storage program are instructed, and wherein processor is configured to read from storage medium and configuration processor refers to
Order, wherein programmed instruction can perform, with realize various embodiments of the method described herein any one (or, it is as described herein
Times of any combination of embodiment of the method, the either random subset of any embodiment of the method as described herein or this subset
Meaning combination).Computer system can in a variety of manners in any one realize.For example, computer system can be personal
Computer (with any one among its various way of realization), work station, computer (computer on a card) on card,
Special-purpose computer (application-specific computer in a box) in box, server computer, client
Hold computer, portable equipment, tablet PC, wearable computer (wearable computer), etc..
In some embodiments, one group of computer being distributed across a network be configurable to segmentation perform computational methods (for example,
Any one of method disclosed herein embodiment) work.In some embodiments, the first computer is configurable to connect
Receive the signal of O-QPSK modulation and catch the sample of the signal.Sample can be sent to second by the first computer by network
Computer.Second computer can be according to any embodiment of the method or method embodiments described herein as described herein
The random subset or any combination of this subset of any combination or any embodiment of the method as described herein, to sample
This progress is operated.
Figure 1A illustrates a kind of (among many possible applications) possible application of inventive concept as described herein.It is mobile
Equipment 10 (for example, mobile phone) wirelessly communicates with radio transceiver station 15.Mobile device 10 can include number as described herein
Word precorrection, to improve the quality of the signal transmitted by it, i.e. to correct in it transmits hardware (for example, being modulated in its I/Q
In device) so-called " I/Q detractions ".Similarly, the digital post-equalization of signal application that radio transceiver station 15 can be received to it,
To correct the I/Q detractions in it receives hardware (for example, in its i/q demodulator).In addition, radio transceiver station and mobile device
Identical precorrection and post-equalization can be applied with role exchange, i.e. for transmission in the opposite direction.
Figure 1B illustrates the alternatively possible application of inventive concept as described herein.Test sender 20 is received to tested
Device 25 sends signal.Test sender 20 can perform digital pre-calibration as described herein and be detracted to correct the I/Q of its own, and
Therefore the quality of its transmission is improved.For example, due to the use of digital pre-calibration, test sender 20 can suppress to realize to mirror image
Higher standard.Thus, the distortion (for example, I/Q is detracted) measured in the signal that receiver is captured can be attributed to connect
Receive the defect of device.
Fig. 1 C illustrate the more alternatively possible application of inventive concept as described herein.Test receiver 35 is received by quilt
Survey the signal that transmitter 30 is sent.Test receiver corrects the I/Q detractions of its own using digital post-equalization as described herein.
Thus, compared to not having post-equalization situation, receiver can meet the higher standard suppressed for mirror image.Therefore, in receiver
Any distortion (for example, I/Q is detracted) measured in the signal captured can clearly point to the defect of transmitter.
Broadband bearing calibration for receiver
In one group of embodiment, the method 100 for the I/Q detractions of compensated receiver in a frequency range can be related to
And the operation shown in Fig. 2A.
110, receiver can receive analog input signal.Analog input signal can be received from transmission medium.Transmission
Medium is the medium for the transmission for allowing signal energy.For example, transmission medium can be free space, air, the earth or the earth
Some part on surface, power cable, fiber optic cables, the water body of such as ocean.
115, receiver can perform I/Q demodulation to analog input signal, to produce analog in-phase (I) signal and simulation
Orthogonal (Q) signal.The process of I/Q demodulation is well-known in the communications field.Generally, I/Q demodulation is related to hybrid analog-digital simulation input
Signal and a pair of orthogonal carrier wave.For example, the mixing can be explained according to drag：
I (t)=y (t) cos (ω t)
Q (t)=y (t) sin (ω t)
In some embodiments, simulation I signal and simulation Q signal can be construed to baseband signal, i.e. be used as complex baseband
The component of signal.In other embodiments, simulation I signal and simulation Q signal can be construed to intermediate frequency (IF) signal.
120, receiver can be digitized to simulation I signal and simulation Q signal, to produce digital iota signal respectively
And digital Q signal.(term " data signal " is sampled signal to be implied, rather than binary states signal (two-state
signal)).Thus, receiver can include a pair of analog-digital converters (ADC).
125, digital iota signal and digital Q signal can be filtered according to the 2x2 matrixes of digital filter, to produce filter
The digital iota signal of ripple and the digital Q signal filtered.Filtering can be related to according to following relation NEURAL DISCHARGE BY DIGITAL FILTER
2x2 matrixes (hij)：
IF(n)=h11(n)*I(n)+h12*Q(n)
QF(n)=h21(n)*I(n)+h22(n)*Q(n)
Wherein symbol " * " represents convolution.(it should be noted that in other places of this patent disclosure, dependent on specific
Situation, symbol " * " can refer to convolution or multiplication.As subscript, " * " represents complex conjugate.)
It is (all in a frequency range that the 2x2 matrixes of digital filter can compensate (or, at least partly compensate) receiver
As the wide signal of communication to transmitted by being enough to cover bandwidth or receiver instant bandwidth frequency range) in I/Q subtract
Damage.(process for being used to measure I/Q detractions is discussed in detail below in this patent disclosure.) in other words, digital filter
The ideal receiver for making the input-output behavior of receiver body more closely approximately be detracted without I/Q.In response to will be in optional frequency
Pure sinusoid tone under ω is applied to input, and preferable receiver will produce the signal I (n) that amplitude is equal and phase is separated 90 degree
With Q (n), i.e. uneven and crooked without phase without gain.
The 2x2 matrixes of digital filter can have with properties.The frequency of at least one diagonal components of 2x2 matrixes is rung
Should be based on frequency function I/Q detraction measurement and as frequency negative function I/Q detraction measurement
To calculate.If for example, utilizing gain imbalance function g (f) and the crooked function of phaseTo characterize I/Q detractions, wherein f
Covering frequence scope, then component h22(or component h11, or component h11And h22In each) frequency response can be based on letter
Number g (f), g (- f),WithTo calculate.
In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes is based on the I/Q of the function of frequency
The measurement of detraction and calculated as the measurement of I/Q detractions of the function of the negative of frequency.
The filtering for saying digital iota signal and digital Q signal is " according to the 2x2 matrixes of digital filter " being not intended to for execution
Imply：Receiver (or for realizing any equipment of filtering) must include realizing the simple (trivial that is multiplied with zero
Multiplication by zero) filter circuit (when the corresponding element identically vanishing of 2x2 matrixes) or realize with zero
Simple addition (trivial addition by zero) adder.As an example, if h12=0, then IF(n) can be only
Using a convolution circuit according to simplified expression formula IF(n)=h11(n) * I (n) are calculated.Similarly, if 2x2 matrixes
One-component is unit pulse in time n=0, then receiver need not include multiplier to perform simple convolutional (trivial
convolution).If for example, h11(n)=0, then IF(n) can only using a convolution unit and an adder according to
Expression formula IF(n)=I (n)+h12(n) * Q (n) calculate come simply.Thus, " according to 2x2 matrixes of digital filter " filtering
The complete 2x2 arrays of convolution circuit are not necessarily required in all cases.
In some embodiments, the digital iota signal filtered and the digital Q signal filtered can be used to recover information bit
Stream.Receiver (or another processing agency of such as master computer) can pass through the digital iota signal to having filtered and filtering
Digital Q signal perform symbolic solution and transfer to recover information bit stream.In symbol demodulation, vector signal (IF(n),QF(n)) can be with
Decimated, to determine the sequence of complex symbol, and each complex symbol may be mapped to given constellation
(constellation) immediate constellation point in (set of point in complex plane).The sequence of resulting complex points is determined
The stream of information bit.
In some embodiments, receiver includes Aristogrid, and the wherein Aristogrid performs above-mentioned digitlization and filtering
Action.Term " Aristogrid " will imply the instrument for being calibrated to known standard.For example, for I channel and Q channel, simulation
Relation between input and numeral output is all calibrated to known standard.
In some embodiments, receiver is the tester of such as vector signal analyzer (VSA).(term " vector letter
Number " be complex signal or i/q signal synonym).It is defeated that tester can receive simulation from transmitter (such as being tested transmitter)
Enter signal.Analog input signal is in response to receive the action that transmission signal is sent on transmission medium in transmitter.Survey
Test instrument is configurable to compensate the I/Q detractions of its own, but the I/Q of uncompensation transmitter is detracted.What is tested and measure
In the case of, it is critically important that can accurately measure and report the detraction of equipment under test rather than compensate the detraction of the equipment.Thus,
For tester, the measurement of the preferably I/Q detractions of (the detraction compensation of receiver is based on) receiver does not include hair
The I/Q of device is sent to detract.This patent disclosure describes the method for only testing receiver detraction.
In general tester is used for performing the test of equipment under test (DUT) or system under test (SUT) (SUT).Tester one
As for include one or more being used to be connected to SUT inputs and output.Input and output can be simulation, numeral,
Radio frequency etc., for example, in various voltage levels and frequency.In general tester is able to carry out one or more tests
Or feature.For example, tester is configurable to catch and analysis waveform, calculates measured power, generation and is being programmed
Frequency under tone, etc..Tester is generally also calibrated, to realize defined exact level on its I/O.Most
Afterwards, tester generally includes user interface, to provide how tester should operate.
In other cases, it is contemplated that the detraction of receiver compensation transmitter and the detraction of its own.Thus, numeral filter
The 2x2 matrixes of ripple device can be calculated based on the measurement of the I/Q detractions of transmitters and receivers combination.On based on being used as f's
The detraction of function and as-f function detraction come calculated frequency response same principle be also suitable herein.
In some embodiments, filtering operation 125 can be in such as FPGA programmable hardware element, or such as
Performed in the special digital circuit system of application specific integrated circuit (ASIC).Can be programmable hardware element or special digital circuit
System provides the identical sampling clock of driving ADC conversions.
In some embodiments, filtering operation 125 can be performed by processor response in the execution of programmed instruction.Processing
Device can be incorporated as a part for receiver, or one of another system as such as master computer or controller board
Point.
As described above, at least one diagonal components of 2x2 matrixes be based on the function as f I/Q detraction measurement and
Calculated as the measurement of the I/Q detractions of-f function.In some embodiments, " at least one is diagonal " should be construed to
" definitely one diagonal ", and 2x2 matrixes another diagonal components be discrete time unit impulse function (for example, when
Between value one at zero, in other local values zero).
As described above, at least one non-diagonal component of 2x2 matrixes is the measurement of the I/Q detractions based on the function as f
The measurement that is detracted with the I/Q of the function as-f is calculated.In some embodiments, " at least one non-diagonal " should be explained
For " a definitely non-diagonal ", and another non-diagonal component of 2x2 matrixes is null function.
Constraint between receiver detraction in frequency f and frequency-f
In some embodiments, it can be assumed that I/Q detraction and receiver of the receiver in positive frequency are in negative frequency
I/Q detractions are functional dependences.In a kind of such embodiment, the calculating of the 2x2 matrixes of digital filter can be simple as follows
Change.Frequency response under optional frequency f of one diagonal components of 2x2 matrixes in the frequency range can be based only upon in frequency
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions under f is calculated.For example,
If I/Q is detracted by gain imbalance function g (f) and the crooked function of phaseCharacterize, then component h22Frequency response H22
(f) can be based only upon g (f) measurement andMeasurement calculate, wherein f includes acquiring the frequency of measurement.In addition,
Frequency response of the one non-diagonal component of 2x2 matrixes under frequency f can be based only upon the measurement of the I/Q detractions under frequency f
(or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) is calculated.
In some embodiments, under frequency f I/Q detractions and the I/Q detractions under frequency-f are constrained for so that in f
Under I/Q detractions determined by the I/Q under-f is detracted, or the I/Q detractions under the frequency-f are subtracted by the I/Q under f
Damage to determine.For example, gain under frequency f is uneven and the gain imbalance under frequency-f can be constrained for it is equal,
And crooked and under frequency-f the phase of the phase under frequency f is crooked can be constrained for it is equal (or mutual negative
Number).
In some embodiments, it is assumed that gain imbalance be even number and assume phase it is crooked be odd number.In these implementations
In example, two non-diagonal components of 2x2 matrixes can be arranged to zero；One diagonal components can correspond to pure straight-through wave filter
(that is, cell frequency is responded)；And frequency response of another diagonal components under optional frequency f can be based only upon in frequency f
Under the measurements (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions calculate.
In some embodiments, two diagonal components of 2x2 matrixes can correspond to pure straight-through wave filter；One non-right
Angle component can be arranged to zero；And another frequency response of non-diagonal component under optional frequency f can be based only upon in frequency
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions under rate f is calculated.
It is configured to the receiver of broadband correction
In one group of embodiment, receiver 200 can be configured as shown in Figure 2 B.(receiver 200 can include
Above in conjunction with the random subset of the feature described in method 100).Receiver 200 can include i/q demodulator 210, digital unit
215 and digital circuit 220.
I/q demodulator 210 is configurable to receive analog input signal y (t) and I/Q solutions is performed to analog input signal
Adjust, to produce analog in-phase (I) signal and simulate orthogonal (Q) signal, be expressed as I (t) and Q (t).I/q demodulator can be from this
Ground pierce circuit receives a pair of orthogonal carrier wave.
Digital unit 215 is configurable to be digitized simulation I signal and simulation Q signal, to produce quilt respectively
It is expressed as I (n) and Q (n) digital iota signal and digital Q signal.Digital unit 215 can be received from clock generating circuit to be turned
Change clock.Digital unit includes I- channel ADC and Q- channel ADC, and each is driven by identical change over clock.
Digital circuit 220 is configurable to be entered according to the 2x2 logm word I signals and digital Q signal of digital filter
Row filtering (as described above), with the digital Q signal for producing the digital iota signal filtered He having filtered.The 2x2 squares of digital filter
Battle array is configurable to I/Q detraction of compensation (or, at least partly compensate) receiver in a frequency range.It is digital when utilizing
During the 2x2 matrixes programming of wave filter, digital circuit makes the behavior of receiver 200 more like mathematically preferable receiver, i.e. have
The receiver of ideal I/Q demodulators and ideal digital unit.
In some embodiments, digital circuit 220 is by programmable hardware element or such as ASIC special digital circuit
System is realizing (or, be used as programmable hardware element or a part for such as ASIC special digital circuit system).
In some embodiments, digital circuit 220 is configured as the processor of execute program instructions (or including at this
Device is managed, or is realized by the processor).In one embodiment, processor be such as master computer computer system or
A part for controller board.
In some embodiments, receiver 200 can include being used for by digital iota signal to having filtered and having filtered
Digital Q signal performs symbolic solution and transfers to recover the device of information bit stream.Recovery device can include following any one or more：
The processor that performs on the receiver, the processor performed on a host computer, in controller board (for example, together with receiver one
Rise be arranged on cabinet of instrument and meter (instrumentation chassis) on controller board) on perform processor, can compile
Journey hardware element, ASIC.
In some embodiments, receiver 200 is (or including) tester.To the concept of tester more than
Discussion.
Method for receiver to be configured to perform detraction correction
In one group of embodiment, the method 300 for configuring receiver can be related to the operation shown in Fig. 3.Method 300
It can be detracted for receiver being configured to the I/Q of compensated receiver at least in part.Method 300 can be rung by computer system
It should be realized in the execution of programmed instruction.(method 300 can include the random subset of features described above).
310, computer system can receive the measurement of I/Q detraction of the receiver on a frequency band.(" in a frequency
Take " refer to the measurement for measuring the multiple different frequencies being included in the frequency band, for example, these different frequencies are equably or uneven
The frequency band is covered evenly).Receiver can include i/q demodulator, a pair of analog-digital converters (ADC) and digital circuit, for example,
As described above.I/q demodulator is configurable to from analog input signal generation simulation I signal and simulation Q signal.ADC can match somebody with somebody
It is set to and simulation I signal and simulation Q signal is sampled, obtains digital iota signal and digital Q signal respectively.Digital circuit can
To be configured to be filtered digital iota signal and digital Q signal, with the digital Q for obtaining the digital iota signal filtered He having filtered
Signal.(to the discussion for the various approach for realizing digital circuit more than).
315, computer system can the 2x2 matrixes based on the survey calculation digital filter.Digital filter can be calculated
The 2x2 matrixes of ripple device, at least part compensation of I/Q detraction of the device on the frequency band is received to achieve a butt joint.2x2 matrixes are at least
The frequency response of one diagonal components be based on the function of frequency measurement and as frequency negative function survey
Measure to calculate.In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes is based on the survey of the function of frequency
Amount and calculated as the measurement of the function of the negative of frequency.
320, computer system can be programmed to digital circuit, to realize the 2x2 matrixes of digital filter, its
In, when being so programmed, digital circuit is configured as I/Q detraction of the compensated receiver on the frequency band at least in part.It is right
The action that digital circuit is programmed is related to 2x2 matrixes (or providing the parameter of those wave filters) transmission digital filter
The memory used to digital circuit or digital circuit.
Broadband bearing calibration for transmitter
In one group of embodiment, the behaviour shown in Fig. 4 can be related to for compensating the method 400 that the I/Q of transmitter is detracted
Make.
410, digital inphase (I) signal and digital quadrature (Q) signal can be received.Digital iota signal and digital Q signal can
To be interpreted complex valued signals I (n)+jQ (n) component.For example, being used as the result of the symbol-modulated according to given constellation, numeral
I signal and digital Q signal can carry one or more information bit streams.In some embodiments, digital iota signal and numeral Q believe
Number it can be construed to the component of complex-valued base-band signal or intermediate frequency (IF) signal.
415, digital iota signal and digital Q signal can be filtered according to the 2x2 matrixes of digital filter, to produce filter
The digital iota signal of ripple and the digital Q signal filtered.(filtering operation can be held by transmitter or some other agency
OK).Filtering operation can be related to the 2x2 matrixes (h come NEURAL DISCHARGE BY DIGITAL FILTER according to following relationij)：
IF(n)=h11(n)*I(n)+h12(n)*Q(n),
QF(n)=h21(n)*I(n)+b22(n)*Q(n).
The 2x2 matrixes of digital filter can pre-compensate for (or, at least partly pre-compensate for) transmitter in a frequency model
Place the I/Q detractions of (such as in the frequency range of width to the bandwidth for being enough to cover the signal of communication to be sent).
The 2x2 matrixes of digital filter can have with properties.The frequency of at least one diagonal components of 2x2 matrixes is rung
Should be based on frequency function I/Q detraction measurement and as frequency negative function I/Q detraction measurement
To calculate.If for example, I/Q detractions are by gain imbalance function g (f) and the crooked function of phaseTo characterize, wherein f coverings
The frequency range, then digital filter h22(or digital filter h11, or digital filter h11And h22In each)
Frequency response can based on g (f), g (- f),WithTo calculate.
In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes is based on the I/Q of the function of frequency
The measurement of detraction and calculated as the measurement of I/Q detractions of the function of the negative of frequency.
In the description of receiver 100, we carefully limit (qualify) " according to the 2x2 matrixes of digital filter "
The meaning being filtered.Those identicals, which are limited, is used herein will apply to transmitter compensation.
420, the digital iota signal filtered and the digital Q signal filtered can be converted into analog form by transmitter,
To obtain simulation I signal and simulation Q signal respectively.
425, transmitter can perform I/Q modulation to simulation I signal and simulation Q signal, to produce the simulation modulated
Signal.The analog signal modulated can be sent on transmission medium, for example, transmission medium as described above.Receiver can be with
The analog signal modulated is received, may be with noise disturbance and channel distortion form.
More than, we detract the 2x2 matrix descriptions of digital filter for the I/Q of " precompensation " transmitter.Because
I/Q detractions occur after the application of digital filter, especially in I/Q stage of modulating.Thus, 2x2 matrixes can be construed to
Using inverse distortion, this inverse distortion will be provided to the approximate of overall identical mapping together with ensuing distortion.
In some embodiments, filtering operation 415 can be in such as FPGA programmable hardware element (PHE) or all
Performed in such as special digital circuit system of application specific integrated circuit (ASIC).
In some embodiments, filtering operation 415 can be by processor (such as mainframe computer system or instrument and meter control
The processor of device plate) performed in response to the execution of programmed instruction.
In some embodiments, transmitter is tester (for example, AWG or vector signal generator).
Tester can send the analog signal modulated to receiver (such as being tested receiver).In the situation tested and measured
Under, it is critically important that tester, which corrects the detraction for detracting but not correcting receiver of its own,.Thus, in this case,
The I/Q that the measurement of the I/Q detractions of above-mentioned (precompensation of transmitter is based on) transmitter does not preferably include receiver is detracted.This
The method that patent disclosure describes to be used for only measurement transmitter detraction (detract and be clearly separated with receiver).
In some cases, can expectability pick up calibration the detraction and the detraction of its own of receiver.Thus, numeral filter
The 2x2 matrixes of ripple device can be calculated based on the measurement of the I/Q detractions of transmitters and receivers combination.On based on being used as f's
The detraction of function and it is also suitable herein as the same principle of the detraction calculated frequency response of-f function.
The constraint between transmitter detraction under frequency f and frequency-f
In some embodiments, it can be assumed that I/Q detraction and transmitter of the transmitter in positive frequency are in negative frequency
I/Q detractions are functional dependences.In a kind of such embodiment, the calculating of the 2x2 matrixes of digital filter can be simple as follows
Change.Frequency response under optional frequency f of at least one diagonal components of 2x2 matrixes in the frequency range can be based only upon
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions under frequency f is calculated.
If for example, I/Q detractions are by gain imbalance function g (f) and the crooked function of phaseCharacterize, then component h22Frequency ring
Should value H22(f) can be based only upon g (f) measurement andMeasurement calculate, wherein f includes having obtained the frequency of measurement.This
Outside, at least one frequency response of non-diagonal component under frequency f of 2x2 matrixes can be based only upon the I/Q detractions under frequency f
Measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) calculate.
In some embodiments, under frequency f I/Q detractions and the I/Q detractions under frequency-f are constrained for so that in f
Under I/Q detraction by under-f I/Q detract determine, or cause under frequency-f I/Q detraction by under f I/Q detract
It is determined that.For example, gain under frequency f is uneven and the gain imbalance under frequency-f can be constrained to it is equal, and
It (or alternatively, is mutual negative that crooked and under frequency-f the phase of phase under frequency f is crooked, which can be constrained to equal,
Number).
In some embodiments, it is assumed that gain imbalance be even number and assume phase it is crooked be odd number.Then, 2x2 squares
Two non-diagonal components of battle array can be arranged to zero；One diagonal components can correspond to pure straight-through wave filter (that is, cell frequency
Response)；And frequency response of another diagonal components under optional frequency f can be based only upon the I/Q detractions under frequency f
(or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) is measured to calculate.
In some embodiments, two diagonal components of 2x2 matrixes can correspond to pure straight-through wave filter；One non-right
Angle component can be arranged to zero；And another frequency response of non-diagonal component under optional frequency f can be based only upon in frequency
The measurement (or alternatively, measurement for the I/Q detractions being based only upon under frequency-f) of I/Q detractions under rate f is calculated.
The transmitter of configuration is corrected for broadband
In one group of embodiment, transmitter 500 can be configured as shown in figure 5.(transmitter 500 can be incorporated to
The random subset of feature described in upper contact method 400).Transmitter 500 can include digital circuit 510, digital analog converter
(DAC) unit 515 and I/Q modulators 520.
Digital circuit 510 is configurable to receive digital inphase (I) signal and digital quadrature (Q) signal, and utilizes number
The 2x2 logm word I signals and digital Q signal of word wave filter are filtered, to produce the digital iota signal filtered and filtering
Digital Q signal.(filtering can be performed as described in above in a variety of different ways).Digital iota signal and numeral Q believe
Number it can carry one or more information bit streams.
The 2x2 matrixes of digital filter can be calculated, (or, at least partly pre-compensate for) transmitter is one to pre-compensate for
I/Q detractions in individual frequency range.The frequency response of at least one diagonal components of 2x2 matrixes is based on the letter of frequency
The measurements of several I/Q detractions and calculated as the measurement of I/Q detractions of the function of the negative of frequency.In addition, 2x2 matrixes are extremely
The frequency response of a few non-diagonal component is based on the measurement of the I/Q detractions of the function of frequency and as the negative of frequency
The measurement of the I/Q detractions of several function is calculated.
Digital circuit 510 is said to be the I/Q detractions of " precompensation " transmitter, because I/Q detractions are after digital circuit
Occur in the transmitter stage, especially in I/Q modulators 520.Thus, digital circuit (passes through the 2x2 of NEURAL DISCHARGE BY DIGITAL FILTER
Matrix) predistortion is introduced to complex signal I (n)+jQ (n) so that will which is followed by the net effect of the predistortion subsequently detracted
It is similar to the preferable transmitter of no I/Q detractions.In other words, digital circuit applies inverse distortion, and this is against distortion and main body distortion
(subject distortion) is combined next approximate identical mapping and (that is, is constantly equal to the frequency response letter of unit function (unity)
Number).
DAC units 515 are configurable to the I and Q signal that have filtered to be converted into analog form, to obtain corresponding mould
Intend I signal and simulation Q signal.DAC units 515 can receive change over clock from clock generation unit.Digital circuit 510 can connect
Receive identical change over clock so that it with DAC units sample be converted into analog form (I (t), Q (t)) phase same rate give birth to
Pluralize sample (IF(n),QF(n))。
I/Q modulators 520 are configurable to perform I/Q modulation to simulation I signal and simulation Q signal, to produce modulation
Analog signal.The analog signal modulated can be sent to receiver by transmission medium.The concept of I/Q modulation is led in communication
It is well-known in domain.For example, I/Q modulation can be modeled by following formula：
X (t)=I (t) cos (ω t)-Q (t) sin (ω t)
=Re { (I (t)+jQ (t)) exp (j ω t) },
Wherein ω is carrier frequency.
In some embodiments, digital circuit 510 is by programmable hardware element or such as ASIC special digital circuit
System realization (or, the part for being used as programmable hardware element or such as ASIC special digital circuit system to realize).
In some embodiments, digital circuit 510 is arranged to the processor of execute program instructions (or including the processing
Device, or realized by the processor).In one embodiment, processor be such as mainframe computer system computer system or
A part for person's controller board.
In some embodiments, transmitter 500 can be tester.To test in the case of method 400 more than
The discussion of instrument.
Method for configuring transmitter for detraction correction
In one group of embodiment, the method 600 for configuring transmitter can be related to the operation shown in Fig. 6.Method 600
It can be detracted for transmitter being configured to compensate (or being introduced by transmitter) I/Q of transmitter at least in part.Method
600 can be performed by computer system in response to the execution of programmed instruction.
In 610, the measurement that computer system can be detracted with I/Q of the receiver transmitter in a frequency range.(" one
In individual frequency range " measurement that implies I/Q detractions is multiple frequencies in the frequency range (for example, equably or uneven
Cover the frequency of the frequency range evenly) under obtain).Transmitter can include digital circuit, a pair of digital analog converters (DAC)
And I/Q modulators.Digital circuit is configurable to be filtered digital iota signal and digital Q signal, to be filtered respectively
Digital iota signal and the digital Q signal that has filtered.This can be configured as to DAC the digital iota signal and filtering filtered
Digital Q signal be converted into analog form, respectively to obtain simulation I signal and simulation Q signal.I/Q modulators can match somebody with somebody
It is set to using I signal and simulation Q signal modulation carrying signal is simulated, to obtain the carrying signal modulated.The carrying modulated
Signal can be sent to receiver by transmission channel.
615, computer system can the 2x2 matrixes based on the survey calculation digital filter.Digital filter can be calculated
The 2x2 matrixes of ripple device, the precompensation detracted with I/Q of the receipts device in the frequency range that achieve a butt joint (or, it is at least partly pre- to mend
Repay).The frequency response of at least one diagonal components of 2x2 matrixes is based on the measurement of the function of frequency and as frequency
The measurement of the function of the negative of rate is calculated.In addition, the frequency response of at least one non-diagonal component of 2x2 matrixes can be based on
The measurement of the function of measurement as the function of frequency and the negative as frequency is calculated.
620, computer system can be programmed to digital circuit, to realize the 2x2 matrixes of digital filter, its
In, when being so programmed, digital circuit is configured as pre-compensating for the I/Q detractions in the frequency range at least in part.Logarithm
The action that word circuit is programmed be related to the digital filter parameter of wave filter (or regulation) be sent to digital circuit or
The parameter storage used by digital circuit.
In various embodiments, digital circuit can be programmable hardware element, application specific integrated circuit (ASIC), in program
The processor performed under the control of instruction, or its any combination.
The derivation of the digital filter of compensation is detracted for broadband
As described above, the 2x2 matrixes of digital filter can detract for the I/Q compensated in receiver or transmitter.It is (real
On border, transmitters and receivers can use matrix compensation, and each utilizes the 2x2 compensation matrixs of its own.Transmitter
Compensation matrix can be detracted to calculate based on the I/Q of transmitter, and the compensation matrix of receiver can be subtracted based on the I/Q of receiver
Damage to calculate).This part by derive 2x2 matrixes have the special shape shown in Fig. 7 in particular cases be used for digital filtering
The frequency response of device.
Because gain imbalance g and phase are crookedIt is measurement of correlation, therefore we have freely gain imbalance and phase
Position is crooked to be modeled as merely due to the distortion of a channel (I or Q), and one other channel is preferable.Fig. 7 represents gain is uneven
Weighing apparatus and phase it is crooked be all modeled as merely due on Q channel distortion selection.Fig. 8 illustrates opposite selection.(thus, frequency is rung
Answer H11And H12For realizing compensation, and H22=1 and H21=0).Gain imbalance can also be modeled as merely due to a letter
Amplitude distortion on road, is modeled as the phase distortion merely due on Relative channel phase is crooked.It is used as still another alternative side
Case, can be uneven gain and/or phase is crooked is modeled as due to the partial distortion on two channels, for example, as Fig. 9 is carried
Go out.Thus, digital compensation can utilize all four frequency response H11、H12、H21And H22To perform.Understanding based on Fig. 7's
After deriving below, it is very easy that those skilled in the art, which will be seen that an identical mathematical principle is applied to all other situation,
's.
Fig. 7 can be construed to the filtering operation performed by the digital circuit 220 of receiver or the numeral electricity by transmitter
The filtering operation that road 510 is performed.Thus, the compensation square of compensation matrix and receiver simultaneously suitable for transmitter is derived below
Battle array.
Although compensation is digital form to apply, to put it more simply, following derive will carry out table on continuous time t
Show.In order to realize compensation, we find frequency response U (ω) and V (ω), to cause, in frequency band (for example, on zero pair
The frequency band of title) in all frequencies omegas or at least can obtain detraction g (ω) andMeasurement selected frequency at,
The signal of distortionIt is transformed into signal cos (the ω t)+jsin (ω corrected
t).The gain that g (ω) corresponds to frequencies omega is uneven, andThe phase for corresponding to frequencies omega is crooked.Thus, we
Obtain equation：
Wherein " * " represents convolution, and wherein u (t) and v (t) are the pulses for corresponding respectively to frequency response U (ω) and V (ω)
Response.
By being replaced
Cos (θ)=(1/2) { exp (j θ)+exp (- j θ) }
Sin (θ)=(- j/2) { exp (j θ)-exp (- j θ) }
We obtain equation
Due to exp (j ω t) and exp (- j ω t) linear independent, we obtain following two equatioies：
Because equation (b) is set up to all ω, we can substitute ω with-ω, thus obtain below equation (b ').
Equation (a) and (b ') define unknown vector [U (ω), V (ω)]TIn matrix equality, its solution given by following formula
Go out：
It was observed that U (ω) and V (ω) all rely on respectively g (ω), g (- ω),With(digital filter
) this attribute of detraction information for depending under ω and-ω of frequency response is more suitable generally than with Fig. 7 Special matrix form
With.In fact, it is applied to any type of compensation matrix.It was additionally observed that U and V is that conjugation is symmetrical on frequency：U(-ω)
=U (ω)*And V (- ω)=V (ω)*, as desired by being entirely the wave filter of real number for its impulse response.
In order to simplify process of the design corresponding to frequency response U (ω) and V (ω) digital filter (impulse response), according to
It can be useful that those frequency responses are represented according to its even segments and odd number part：
U (ω)=A (ω)+B (ω)
A (ω)=(1/2) { U (ω)+U (- ω) }
B (ω)=(1/2) { U (ω)-U (- ω) }
V (ω)=C (ω)+D (ω)
C (ω)=(1/2) { V (ω)+V (- ω) }
D (ω)=(1/2) { V (ω)-V (- ω) }
In time domain, corresponding expression is：
U (t)=a (t)+b (t)
A (t)=(1/2) { u (t)+u (- t) }
B (t)=(1/2) { u (t)-u (- t) }
V (t)=c (t)+d (t)
C (t)=(1/2) { v (t)+v (- t) }
D (t)=(1/2) { v (t)-v (- t) },
Wherein u, a, b, v, c and d are the impulse response for corresponding respectively to frequency response U, A, B, V, C and D.
The expression formula derived more than for U (ω) and V (ω), then has：
Above expression formula can for based on measurement or estimation detraction function g andCalculated frequency response U and V.This
A little expression formulas are equally applicable to the post-compensation in receiver or the precompensation in transmitter.In other words, for precorrection I/Q
Detract g (f) andFrequency response U (ω) and V (ω) and the frequency response phase for those identical I/Q detractions of post-compensation
Together.
The frequency response U and V calculated can be for true using any one in various known filter design algorithms
Fixed corresponding impulse response u (n) and v (n).
Points for attention on the wave filter with strange frequency response
The given wave filter with strange frequency response B (ω), the essential fact is that by EB(ω)=jB (ω) sgn (ω) gives
The function E gone outB(ω) is even number and had the property that：
B (t) * x (t)=HT (eB(t)*x(t)).
Wherein HT is Hilbert transformation operators, and wherein b (t) corresponds to B (ω) impulse response, and x (t) is to appoint
Meaning input function, wherein sgn (ω) is 1 and is -1 when ω is less than zero when ω is more than zero.
If we the fact that be applied to from odd function B (ω) discussed above and D (ω), we will obtain
Corresponding even function：
On even number g (ω) and odd numberSpecial circumstances points for attention
In most cases, the uneven function of gain can be modeled as even number and the crooked function of phase can be modeled as very
Number, i.e. g (ω)=g (- ω) andUnder these constraints, U (ω)=0 and V (ω) are plural numbers.
On even number g (ω) and even numberSpecial circumstances points for attention
It is typically above complex value for expression formula derived from U (ω) and V (ω).But, when gain is uneven and phase is crooked
When function is even number, i.e. g (ω)=g (- ω) andThen U (ω) and V (ω) become real number value：
To the scalar matrix of post-equalization receiver detraction at a single frequency
In one group of embodiment, for operating the method 1000 of receiver (or operation include the system of receiver) can be with
It is related to the operation shown in Figure 10.
1010, receiver can receive analog input signal.Analog input signal can be received from transmission medium, example
Such as, as described above.
1015, receiver can perform I/Q demodulation to analog input signal, to produce analog in-phase (I) signal and mould
Quasiorthogonal (Q) signal, for example, as described above.
1020, receiver can be digitized to simulation I signal and simulation Q signal, to produce digital iota signal respectively
And digital Q signal.
1025, receiver can be according to constant 2x2 matrix c=(cij) digital iota signal and digital Q signal are converted, to produce
Raw resulting number I signal and resulting number Q signal.Conversion can be performed by the following matrix multiple of application：
Wherein IRAnd Q (n)R(n) resulting number I signal and resulting number Q signal are represented respectively.2x2 matrixes c can be configured
For the I/Q detractions of post-compensation (or, at least part post-compensation) measurement of receiver under specific frequency f.
Matrix c can have with properties.Diagonal element c11And c22In at least one can based on receiver in frequency f
Under the I/Q of measurement detract and calculate.For example, coefficient c22The value g (f) and/or the value of measurement measured can be used as's
Function is calculated, and wherein g is the uneven function of gain, andIt is the crooked function of phase.Similarly, off-diagonal element c12And c21
In at least one can based on receiver under frequency f the I/Q of measurement detract and calculate.For example, coefficient c21Survey can be used as
The value g (f) of amount and/or the value of measurementFunction calculate.In some embodiments, it is each in this four matrix elements
It is individual all similarly to calculate (namely based on the detraction measured under f).
For a kind of matrix c possible embodiment, referring to " performing traditional detraction at a single frequency to compensate " part.
Make cij(f) represent to be used for determine coefficient c according to the I/Q detractions under frequency fijFunction expression.Due to function
Expression formula cij(f) continuity on frequency f, therefore matrix c (f) is to matrix c (f+ Δs f) good approximations, as long as Δ f is sufficient
It is enough it is small just can be with.Thus, when receiver performs map function 1025 using matrix c (f), receiver is by the frequency around f
It is implemented around at least partly compensating.The quality of compensation will generally degrade with the increase of Δ f absolute values.
In some embodiments, analog input signal is pure sinusoid tone, for example, tone under frequency f or in frequency
Rate f+fLOUnder tone, wherein fLOIt is the frequency of the local oscillator of receiver.In other embodiments, analog input signal
It is the signal of communication for carrying binary message stream.
In some embodiments, matrix c has the adeditive attribute that one diagonal element is one.
In some embodiments, matrix c has the adeditive attribute that one off-diagonal element is zero.In some embodiments
In, matrix c has one of following special shape：
As described above, map function 1025 is " according to 2x2 matrixes " to perform.That limits phrase and does not mean that needs connect
Receiving device includes realizing the simple multiplier (or adder) for being multiplied (or being added with zero simple) with one.For example, in the above
In the first special shape provided, resulting number I signal is equal to digital iota signal：IR(n)=I (n).This is completely without meter
Calculate.I (n) inputs simply can be delivered to IR(n) export.
In one group of embodiment, receiver 1100 can be configured as shown in fig. 11.(receiver 1100 can be simultaneously
Enter the random subset above in conjunction with the feature described in method 1000).Receiver 1100 can include i/q demodulator 1110, numeral
Change unit 1115 and digital circuit 1120.
I/q demodulator 1110 is configurable to receive analog input signal, and I/Q solutions are performed to analog input signal
Adjust, to produce analog in-phase (I) signal and simulate orthogonal (Q) signal.Analog input signal can be received from transmission medium, as above
It is described.
Digital unit 1115 is configurable to be digitized simulation I signal and simulation Q signal, to produce number respectively
Word I signal and digital Q signal.
Digital circuit 1120 is configurable to 2x2 matrixings digital iota signal and digital Q signal according to constant, to produce
Raw resulting number I signal and resulting number Q signal.2x2 matrixes be configurable at least in part compensated receiver in specific frequency
I/Q detractions under rate f.Can be based on receiver in frequency f corresponding to first constant of first diagonal element of 2x2 matrixes
Under the I/Q of measurement detract and calculate.In addition, can be with corresponding to second constant of first off-diagonal element of 2x2 matrixes
The I/Q of measurement based on receiver under frequency f detracts to calculate.In some embodiments, each in this four constants
All similarly calculate (namely based on the detraction of the measurement under f).
In one group of embodiment, the method 1200 for configuring receiver can be related to the operation shown in Figure 12.Method
1200 can detract for receiver is configured to I/Q of the compensated receiver under given frequency f at least in part.Method 1200
It can be realized by computer system in response to the execution of programmed instruction.(method 1200 can include above in conjunction with Figure 10 and 11
The random subset of described feature).
1210, computer system can receive the I/Q detractions of measurement of the receiver under frequency f.Receiver can be wrapped
I/q demodulator, analog-to-digital conversion (ADC) unit and digital circuit are included, for example, as described by above in conjunction with Figure 10 and 11.I/Q is solved
Device is adjusted to be configurable to according to analog input signal generation simulation I signal and simulation Q signal.ADC units are configurable to mould
Intend I signal and simulation Q signal is sampled, to obtain digital iota signal and digital Q signal respectively.Digital circuit is configurable to
Digital iota signal and digital Q signal are converted, to obtain resulting number I signal and resulting number Q signal.(to realizing number more than
The discussion of the various approach of word circuit).
1215, computer system can detract the 2x2 matrixes of calculation constant based on the I/Q measured under frequency f.Can
To calculate the 2x2 matrixes, to realize at least part compensation of the I/Q detractions to being measured under frequency f.2x2 matrixes are at least
One diagonal components can detract to calculate based on the I/Q measured in frequency f.In addition, at least one non-diagonal of 2x2 matrixes
Component can detract to calculate based on the I/Q measured under frequency f.
1220, computer system can be programmed to digital circuit, to realize the 2x2 matrixes of constant, wherein, when this
When sample is programmed, digital circuit is configured to compensate the I/Q detractions measured under frequency f at least in part.Digital circuit is entered
The action of row programming, which is related to, to be sent to digital circuit or digital circuit 2x2 matrixes (or providing the information of the matrix) and is made
Memory.
Real matrix precorrection at a single frequency
In one group of embodiment, the method 1300 for compensating I/Q detraction of the transmitter under specific frequency f can be related to
Shown operation in fig. 13.
1310, digital inphase (I) signal and digital quadrature (Q) signal can be received (for example, with various differences as more than
Form description).In some embodiments, digital iota signal and digital Q signal can represent the complex exponential under frequency f together
Tone.In other embodiments, digital iota signal and digital Q signal can carry corresponding binary message stream.Digital iota signal
Can be the component of complex base band signal or complex intermediate frequency signal with digital Q signal.
1315, digital iota signal and digital Q signal can be according to the 2x2 matrix c=(ci of constantj) convert, to produce
Resulting number I signal and resulting number Q signal.(conversion can be performed by transmitter or certain other agency).Conversion can be by
Following matrix multiple is described：
Wherein IRAnd Q (n)R(n) resulting number I signal and resulting number Q signal are represented respectively.2x2 matrixes are configurable to
Precompensation (or, at least partly pre-compensate for) I/Q detraction of the transmitter under frequency f.Referring to above with respect to " precompensation " essence
Discussion.In brief, the application of conversion introduces inverse distortion, this inverse distortion and the distortion phase in ensuing transmitter stage
With reference to so that the input-output behavior of transmitter seems more preferable.It should be noted that being converted above with respect to " according to 2x2 matrixes "
The discussion of meaning be also suitable herein.
2x2 matrixes c can have with properties.Diagonal element c11And c22In at least one can be based under frequency f
The measurement of I/Q detractions and the measurement of the I/Q detractions under the frequency-f are calculated.For example, diagonal element c22G (f), g can be based on
(-f)、WithIn each the value measured calculate, wherein g is the uneven function of gain, andIt is phase
The crooked function in position.For example, diagonal element c22Can based on g (f), g (- f),WithIn each measure
Value calculate, wherein g is the uneven function of gain, andIt is the crooked function of phase.In addition, off-diagonal element c12And c21In
At least one can be calculated based on the measurement under frequency f and the measurement under frequency-f.In some embodiments, this four
Each in individual coefficient can be calculated based on the measurement under frequency f and the measurement under frequency-f.For matrix c's
A kind of possible embodiment, referring to " calculating real single-point vector calibration constant " part.
The detraction measured can be the detraction measured at output (that is, I/Q modulators) place of destruction, also, if
, will be different from the detraction if being measured in input.Alternatively, this method can be included defeated under+f and-f
Go out the input detraction for detracting and being transformed into only under+f, then using the input detraction only under+f according to simplified formula calculating square
Battle array constant.Conversion can be derived as follows.First, led using equation (7.9) and (7.10) based on the output detraction under+f and-f
Go out the dedicated expression formula to U (f) and V (f), wherein gin(f)=gin(- f)=1 andThen, base
Input detraction g is calculated in equation (7.7)in(f) andWherein gout(f)=1 and
Then, matrix constant can be based on gin(f) andTo determine, for example, according to relationWith
The quality for the compensation realized by operation 1315 will be limited by the quality of detraction measurement.This patent disclosure
Describe the mass measurement of the I/Q detractions for obtaining transmitter under any given frequency or in whole frequency range
Method.
Make cij(f) represent to be used for determine coefficient according to the I/Q detractions under frequency f and the I/Q detractions under frequency-f
cijFunction expression.Due to function expression cij(f) continuity on frequency f, therefore matrix c (f) is to matrix c (f+
Δ f) good approximation, just can be with as long as Δ f is sufficiently small.Thus, when transmitter uses matrix c (f) execution map functions 1315
When, the frequency around f is implemented around at least partly compensating by transmitter.The quality of compensation is generally by with Δ f absolute values
Increase and degrade.
1320, resulting number I signal and digital Q signal can be converted into analog form by transmitter, to obtain phase
The simulation I signal and simulation Q signal answered.
1325, transmitter can perform I/Q modulation to simulation I signal and simulation Q signal, to produce the simulation modulated
Signal, for example, as described above.
In some embodiments, matrix c has one of following special shape：
More than in the first special shape, constant c21And c22Value A (f), E can be based onB(f), C (f) and ED(f) count
Calculate, such as described in " calculate real single-point vector and calibrates constant " part, especially in equation (1.81) and (1.82).
In some embodiments, conversion 1315 can be in such as FPGA programmable hardware element or such as special
Performed in the special digital circuit system of integrated circuit (ASIC).Can be programmable hardware element or special digital circuit system
The identical sampling clock of driving ADC conversions is provided.
In some embodiments, conversion 1315 can be performed by processor response in the execution of programmed instruction.Processor
A part for a part for transmitter or another system as such as master computer or controller board can be incorporated as.
In one group of embodiment, transmitter 1400 can as shown in Figure 14 as configure.(transmitter 1400 can be wrapped
Include the random subset above in conjunction with the feature described in method 1300).It is mono- that transmitter 1400 can include digital circuit 1410, DAC
1415 and I/Q of member modulators 1420.
Digital circuit 1410 is configurable to receive digital inphase (I) signal and digital quadrature (Q) signal, and according to
The 2x2 matrixings digital iota signal and digital Q signal of constant, to produce resulting number I signal and resulting number Q signal.Numeral
Circuit 1410 can be realized by any one in various forms, for example, such as above in conjunction with method 1300 in a variety of different ways
Description.
DAC units 1415 are configurable to a resulting number I signal and resulting number Q signal is converted into analog form, with
Just simulation I signal and simulation Q signal are obtained respectively.
I/Q modulators 1420 are configurable to perform I/Q modulation to simulation I signal and simulation Q signal, to produce modulation
Analog signal.I/Q detraction of the 2x2 matrix configurations at least partly precompensation transmitter under frequency f.Corresponding to 2x2 matrixes
First diagonal element first constant can based under frequency f I/Q detraction measurement and the I/Q under frequency-f
The measurement of detraction is calculated.It can be based on corresponding to second constant of first off-diagonal element of 2x2 matrixes under frequency f
Measurement and measurement under frequency-f calculate.
The meaning of " detraction under frequency f "
Present disclosure is repeatedly using term " the I/Q detractions under frequency f ".No matter this term be used for transmitter,
Receiver still includes the tandem compound of transmitter, transmission path and receiver, and it is all included in frequency f within the meaning
Under I/Q caused by stimulating discussed system using complex exponential tone exp (j2 π ft)=cos (2 π ft)+jsin (2 π ft)
Detraction, as shown in Figure 15.The real number output and imaginary number output of system can be expressed as：
I/Q detractions under frequency f can include the gain imbalance g (f) being given by and phase is crooked
G (f)=gQ(f)/gI(f)
Herein, we are using by the use of I channel as to uneven and the crooked reference of phase the convention of gain.But,
Invention principle as described herein is equally applicable to any other reference convention.For example, it is also possible to just use opposite convention
(that is, select Q channel as to gain is uneven and the crooked reference of phase), or gain imbalance with reference to a channel and
The convention of the crooked reference one other channel of phase.
Because we are uneven to the gain between two channels of compensation and phase difference is interested, we can be increasing
Beneficial uneven and phase is crooked to be modeled as seeming all on the i channel or all on Q channel.For example, Figure 16 is illustrated
Latter is selected.Thus, Q channel output has form：
The physical consequence of I/Q detractions
The consequence of I/Q detractions under frequency f is that occur undesirable signal energy under frequency-f.In order to see this
Point, we analyze complex output as follows：
(we are switched to the π f of ω=2 from f, it is only for notation it is succinct).Thus, in response to stimulus signal exp (j
ω t), system is produced under frequencies omega with complex amplitude ATONEThe complex exponential tone of (ω) and the generation tool under frequency-ω
There is complex amplitude AIMAGEThe complex exponential tone of (ω).
Complex exponential tone under frequencies omega is usually simply referred as " tone ", and the complex exponential tone under frequency-ω
Commonly referred to as " mirror image ".As expected, as g (ω) → 1 andWhen, ATONE(ω) → 1 and AIMAGE(ω)
→0.Expect to allow g (ω) as close possible to one and allowAs close possible to zero.(assume herein for gain imbalance
Lineal scale.Gain imbalance can also represent by logarithmic scale, for example, in units of dB, in this case, 0dB generations
Table is without the unbalanced situation of gain.)
From described above, the tandem compound of two systems is readily visible, first has gain imbalance g1
(ω) and phase are crookedAnd second has gain imbalance g2(ω) and phase are crookedDo not provide
Net gain imbalance g (ω)=g1(ω)g2(ω) and net phase position are crooked(because the
Two systems are stimulated by pure complex exponential exp { j ω t }).Real relation is more complicated.
Mirror image suppresses (Image Rejection)
It is to complex amplitude A that mirror image, which suppresses,TONE(ω) and AIMAGEThe measurement of the relative magnitude of (ω).For example, according to one kind
Usual definition：
Mirror image suppression=20*log (AIMAGE|/|ATONE|).
Because | AIMAGE| typically smaller than | ATONE|, so mirror image suppresses to be typically negative.Mirror image suppresses more negative better.
Post-compensation and precompensation
The concept of post-compensation is related to the output for compensation block being coupled to the system that I/Q detractions are presented.Compensation block is configured to make
The tandem compound that must be followed by the system of compensation block show (or, approximation behavior goes out) have unit gain uneven and
The crooked ideal model of zero phase.When system is stimulated by the complex exponential tone under frequencies omega, generation can be modeled as by itDistortion complex signal, wherein g (ω) andIt is system under frequencies omega
I/Q detraction.Compensation block is operated to the complex signal of distortion, to generate the original complex exponential tone being equal under frequencies omega
The output signal corrected.Thus, compensation block be said to be " to compensate " or " post-compensation " system under frequencies omega I/Q detraction.
The broadband post-compensation of I/Q detractions means the post-compensation to the I/Q detractions under each frequencies omega in frequency range or frequency band.
The concept of precompensation, which is related to, to be placed on compensation block before system, i.e. the output coupling of compensation block to the defeated of system
Enter.Compensation block is configured so that the tandem compound for being followed by the compensation block of system shows (or, approximation behavior goes out) tool
There is unit gain uneven and the crooked ideal model of zero phase.In response to the complex exponential tone under frequencies omega, compensation block will
Produce the complex signal of predistortion.System receives the complex signal of predistortion and further makes the distorted signals (by introducing
I/Q is detracted), thus produce complex output.Compensation block generates the complex signal of predistortion so that plural defeated from system
Go out signal and be equal to the original complex exponential tone under frequencies omega.Thus, compensation block is said to be " compensation " or " precompensation " system and existed
I/Q detractions under frequencies omega.The broadband precompensation of I/Q detractions means under each frequency in frequency range or frequency band
I/Q detraction precompensation.
Tradition detraction compensation is performed at a single frequency
If in specific frequency ω0Under I/Q detraction post-compensation it is interested, it is possible to using have real constant α
With β Figure 17 block diagram.Pass through the appropriate selection of constant, the complex input signal of multilatedIt will be mapped to the output signal cos (ω corrected0t)+jsin(ω0T), such as institute
It is desired.Appropriate value is：
This compensation method is referred to herein as " traditional single-point compensation ".
Because gain imbalance g and phase are crookedOn the continuity of frequencies omega, thus real constant α and β will to
ω0I/Q detractions under neighbouring frequency realize that part is compensated, with from ω0Distance increase, compensate degrading quality.But,
Because g (ω0) it is typically different than g (- ω0) andIt is typically different thanSo for compensation in frequencies omega0
Under I/Q detraction appropriate value to (α, β) generally with for compensating in frequency-ω0Under I/Q detraction appropriate value to difference.
Thus, unfortunately, it generally can not find out simultaneously to ω0With-ω0The single value pair all worked.
It is directed in α and β derived above value in single frequency ω0Under I/Q detraction post-compensation ideally work
While, they can be also used in single frequency ω0Under I/Q detractions precompensation, its usual result is less preferable.
Although (it provides not ideal result, and various methods as described herein can use this precompensation, and this is partly
Because it is required no knowledge about in frequency-ω0Under I/Q detraction).In order to realize that the ideal of I/Q detractions at a single frequency is pre-
Compensation, referring to " calculating real single-point vector calibration constant " part.
I/Q detractions in broadband are balanced
Figure 18 depict will pass through this patent disclosure reuse system H basic model, for example, with represent by
The equalization filtering of receiver execution and the equalization filtering performed by transmitter.(equilibrium is herein used as the synonymous of I/Q detraction compensation
Word).
In the case where system H represents the equalization filtering of receiver, complex input signal I (t)+jQ (t) is represented by first
The signal of distortion that provides of system G, as illustrated in fig. 19.System G under frequency f by complex exponential in response to being believed
Number i (t)+jq (t)=exp (j2 π ft) stimulates and generated the signal of distortionGain imbalance g (f) and phase are crookedIt is system G in frequency
I/Q detractions under rate f.System G can represent the base band equivalent of receiver front end, i.e. receiver is from its RF input to I/Q
Part at the output of digital unit.Alternatively, in the I/Q detractions and the I/ of its own of expected receiver compensation transmitter
In the case that Q is detracted, system G can be represented from the input of the I/QDAC units of transmitter to receiver I/Q digital units
Output at path.
For all f in desired frequency band, inputs of the system H to distortion is operated, to produce correct defeated
Go out signal I'(t)+jQ'(t)=exp (j2 π ft).It is noted, however, that by { exp (j2 π ft)：F in the range of given frequency } provide
Set B constitute be restricted to by frequency band given frequency scope function space { x (t) } basis.Because being followed by H G
Tandem compound be identical mapping to basis set B each function, so, due to linear, it will limit letter to all frequency bands
Number x (t) identical mapping.
Equalizing system H can be realized by the digital circuit 220 of receiver, as described in above in a variety of different ways.
In the case where system H represents the equalization filtering of transmitter, we are construed to H to receive basic functionAlso, in response to the basic function, generation is as shown in fig. 20a
Precompensation complex signal I'(t)+jQ'(t)=exp (j2 π ft).It should be noted that byF in given frequency scope } the set X that provides also constitutes and is restricted to give by frequency band
The basis of the function space { x (t) } of frequency range.
The distortion by following system G of the signal of precompensation.System G generates the signal of distortionWherein g (f) andRepresent gains of the system G under frequency f not
Balance and phase are crooked.Because the tandem compound for being followed by G H is the identical mapping of each function on basis set X, institute
To be the identical mapping on all frequency band restricted function x (t) with it.Thus, referred to again when under any frequency f in frequency band
When number tone exp (j2 π ft) is stimulated, the tandem compound will produce identical complex exponential tone at its output, such as institute in Figure 20 B
Show.
System G can represent the base band equivalent of the RF front ends of transmitter, i.e. from the input of the DAC units of transmitter
Transmitter part to RF outputs.Alternatively, in the I/Q detractions of expected transmitter compensated receiver and the I/Q of its own
In the case of detraction, system G can be represented at the output of digital unit of the DAC of the transmitter input to receiver
Path.System H can be realized by digital circuit 510, as described in above in a variety of different ways.
Complex exponential is used through this analysis, because any band-limited signal can be expressed as multiple finger through Fourier analysis
Several overall superpositions.When relatively more same phase (I) channel and orthorhombic phase (Q) channel, I/Q detractions can include the injustice of gain
Weighing apparatus, and it is crooked due to undesirable quadrature hybrid to occur phase.(preferable 90 degree between crooked disturbance I and the Q channel of phase
Phase relation).Generally it is modeled as undesirable in quadrature hybrid although phase is crooked, it can also be modeled as I (t) and Q (t)
Phase between signal is crooked.In the case of two kinds discussed above, the input to distortion model G is all complex exponential signal.
Because I/Q detractions are relative, therefore we can assume that I/Q detractions fully appear in Q (t) outputs, and I (t) outputs are reasons
Think.Although can also make other it is assumed that still this assumes that following mathematical derivation will be simplified.
Equalizing system H can be by 2x2 frequency response matrix Hs (f)=(Hij(f)), or equivalently rung by real number value pulse
The 2x2 matrix h (t) answered=(hij(t)) model.But, it is identified above at distortion model G output how table
Show under the hypothesis of detraction, matrix H can be reduced to the structure shown in Figure 21, i.e. H11=1 and H (f)12(f)=0.In order to remember
The efficiency of method, we define U (f)=H21And V (f)=H (f)22(f).Thus,
I'(t)=I (t)
Q ' (t)=u (t) * I (t)+v (t) * Q (t),
Wherein u (t) and v (t) is the impulse response for corresponding respectively to U (f) and V (f).
Any real number value filter all necessarily has symmetrical magnitude responses and antisymmetric phase response.In other words, x
(t) it is that real number is implied for all f,
| X (f) |=| X (- f) |
Phase { X (- f) }=- Phase { X (f) }
Wherein X (f) is x (t) Fourier transform.Therefore, frequency response V (f) can not be under frequency f and-f using only
Vertical detraction correction.In a typical case, g (f) is different from g (- f), andWithIt is different.Thus, itself action
Wave filter V (that is, U identically vanishing) be not enough to provide the correction under f and-f.If target be only correct only positive frequency it
I/Q detractions in broadband upper or only on negative frequency, then wave filter V will be enough.(note：As long as detraction is constrained to g (f)=g
(- f) andAuto- V can just correct+f and-f and detract, and such as be proved in " addition constraint " part
).But, due to expecting the both sides of corrected spectrum, therefore introduce second wave filter U (f).Using from the another of in-phase component
One wave filter and the free degree being added to needed for the both sides that control Complex frequency is provided in quadrature channel.This be by
In in-phase component I (t)=cos (2 π ft) for frequency f is identical with-f and changes when the phase of quadrature phase component is becoming-f from f
The fact that 180 degree.
In order to solve U (f) and V (f), it is necessary to know their own output signal.In order to simplify mathematical derivation, U (f) and
V (f) is divided into their even segments and odd number part, as shown in Figure 22.Thus, A (f) and B (f) is U (f) even number
Part and odd number part, and C (f) and D (f) they are V (f) even segments and odd number parts.
Because any real number value filter all necessarily has symmetrical magnitude responses, therefore we can be by only to each
Frequency spectrum A, B, C and D positive frequency part solve to reduce complexity.But, negative frequency and positive frequency are subtracted in order to realize
Damage compensation, it is impossible to simply ignore input I (t)+jQ (t) corresponding to negative frequency.On the contrary, the odd symmetry dependent on SIN function
Property and cosine function even symmetry, we, which are inputted by the way that they are expressed as equivalent positive frequency, considers this input：
Thus, we will draw two equatioies to A, B, C and D positive frequency part,
First based on input
And second based on input
Wherein there is f for the two equatioies>0.
If wave filter is constrained for symmetrical impulse response, symmetrical magnitude responses and zero will be presented in wave filter
Phase response.It is such situation for wave filter a (t) and c (t).But, if the impulse response of wave filter is antisymmetry,
Then it will be presented symmetrical magnitude responses, but will be in the phase response now equal to-(pi/2) sgn (f).
Thus, antisymmetric impulse response is equivalent to the even pulse response for being followed by Hilbert conversion.For filtering
Device b (t) and d (t), is such situation.Therefore, what wave filter b (t) can be expressed as followed by Hilbert conversion (HT)
Even pulse responds eB(t), as shown in Figure 23.Similarly, what wave filter d (t) can be expressed as followed by Hilbert conversion
(HT) even pulse response eD(t)。EBAnd E (f)D(f) it is to correspond respectively to eBAnd e (t)D(t) frequency response.Now, it is original
Wave filter A, B, C and D definite output can be readily determined.Figure 24 A show four wave filters A, B, C and D in response to
Signal I1(t)+jQ1(t) output.Figure 24 B show four wave filters in response to signal I2(t)+jQ2(t) output.
In Figure 24 A and 24B each can (or non-negative f) be directly translated into A (f), E for positive fB(f)、C(f)
And ED(f) corresponding linear equality in.We use following notation：
g1(f)=g (f) for f ＞ 0
g2(f)=g (- f) for f ＞ 0
Figure 24 A and 24B sets forth equation (1.1) and (1.2), this figure 25 illustrates.Figure 26 A and 26B give
Corresponding polar plot.(recall the cos (2 π ft) in polar plot and be mapped to 1, and sin (2 π ft) is mapped to-j).
Vectorial floor projection provides below equation (1.3) in Figure 26 A；Upright projection provides equation (1.4).Similarly,
Vectorial floor projection provides equation (1.5) in Figure 26 B；Upright projection provides equation (1.6)：
This equation system is unknown vector (A, EB, C, ED) in 4x4 linear systems：
Wherein
And
Matrix P determination is given by：
Det (P)=w2+x2+y2+z2-2wy+2xz. (1.13)
As long as
It there is unique solution vector (A (f), EB(f), C (f), ED(f)).As an example, equation can not be crooked in phaseSolved when with gain imbalance g (f) being all entirely odd number.But, it is entirely odd number for gain imbalance g (f),
This is null(NUL) because gain imbalance generally for all f all close to one, or at least by some positive constant lower bound
It is fixed.
Utilize Cramer rules, it has been found that
A (f)=- 2 (wz+xy)/Det (P) (1.16)
EH(f)=(- w2-x2+y2+z2)/Det(P) (1.17)
C (f)=2 (x+z)/Det (P) (1.18)
ED(f)=2 (w-y)/Det (P) (1.19)
Equation (1.9) to (1.14) be substituted into equation (1.16) into (1.19) will produce equation (1.20) extremely
(1.23), figure 27 illustrates.
Addition constraint
In many cases, gain imbalance and the crooked approximate common constraint of phase.This part is typical existing to some
The condition in the real world simplifies equation (1.20) to (1.23).For optimal compensation, equation (1.20- can be used
1.23).But, if compensation performance can be loosened, calculating demand can be reduced by adding some constraints.
Situation 1：Strange phase is crooked
In the case where strange phase is crooked, i.e. for f ＞ 0,Equation (1.20) is extremely
(1.23) it is exclusively used in (specialize to)：
A (f)=0 (1.24)
EB(f)={ g2(f)-g1(f)}/{g1(f)+g2(f)} (1.25)
Situation 2：Even gain is uneven
In the case of even gain is unbalanced, i.e. for f ＞ 0, g (f)=g1(f)=- g2(f), equation (1.20) is extremely
(1.23) it is exclusively used in：
EB(f)=0 (1.29)
Situation 3：The strange crooked and even gain of phase is uneven
In the case of the crooked and even gain of strange phase is unbalanced, equation (1.20) to (1.23) is exclusively used in：
A (f)=0 (1.32)
EB(f)=0 (1.33)
Situation 4：The crooked and any gain of zero phase is uneven
In the case of the crooked and any gain of zero phase is unbalanced, equation (1.20) to (1.23) is exclusively used in：
A (f)=0 (1.36)
EB(f)={ g2(f)-g1(f)}/{g2(f)+g1(f)} (1.37)
C (f)=2/ { g2(f)+g1(f)} (1.38)
ED(f)=0. (1.39)
Situation 5：Arbitrary phase is crooked and unit gain is uneven
In the case of arbitrary phase is crooked and unit gain is unbalanced, equation (1.20) to (1.23) is exclusively used in：
EB(f)=0 (1.41)
Situation 6：Constant gain is uneven and phase is crooked
In the case where gain is uneven and the crooked function of phase is constant function, i.e. have g (f)=g for all f
AndEquation (1.20) to (1.23) is exclusively used in：
EB(f)=0 (1.45)
ED(f)=0. (1.47)
Wave filter is designed
In one embodiment, symmetric line phase FIR filterWithIt is to be based respectively on magnitude responses | A
(f) | and | C (f) | design, and antisymmetry linear phase FIR filterWithIt is to be based respectively on magnitude responses | B
(f) | and | D (f) | design.Note, for all f, | B (f) |=| EB(f) | and | D (f) |=| ED(f)|.Remez algorithms
Can be for designing these wave filters.Then, Figure 22 equalizing system can utilize wave filter
WithTo design.By creating four wave filters, each wave filter has symmetrical or antisymmetric filter valve (tap), and
And the wave filter shown in Figure 22 is summed, we can effectively match two arbitrary frequency response U (f) and V (f).(note
Meaning：Dependent on filter design tools, summation, which is actually likely to be, subtracts each other.It is used for creating anti-by many filter design tools
The definition of the Hilbert conversion of balanced-filter is different come the definition used by negating (negation) from us.The filtering
Device design tool is to phase usually using+(pi/2) sgn (f).
In another embodiment, symmetric line phase FIR filter WithIt is respectively
Based on magnitude responses | A (f) |, | EB(f) |, | C (f) | and | ED(f) |.Equally, Remez algorithms can be filtered for designing these
Ripple device.Then, Figure 23 equalizing system can utilize wave filterWithTo realize.
In also another embodiment, wave filterWithIt can be set based on frequency response U (f) and V (f)
Meter.Lp- model (Lp- norm) design method can be for value and phase response and V (f) value and phase based on U (f)
Respond to design these wave filters.Then, Figure 21 equalizing system can utilize wave filterWithTo realize.
Destroy I/Q detractions
As described above, Figure 15 illustrates the I/Q to detract, (that is, gain imbalance g (f) and phase are crooked) be incorporated into
Received complex exponential signal exp (j2 π ft) system.In general, the 2x2 frequency response matrix Hs for characterizing the system can
With from detraction function g (f) andExport.In order to simplify this derivation, we are the gain imbalance g (f) and phase system
It is crookedIt is modeled as fully appearing in Q channel output, as shown in Figure 28.This model causes in convenient use Figure 29
The special shape of shown matrix, wherein U (f) and V (f) correspond to real filter u (t) and v (t) frequency response.Ring
Answer U (f) that its even segments A (f) and odd number part B (f) sums can be expressed as, as shown in Figure 30.Similarly, V (f) can be with
It is expressed as its even segments C (f) and odd number part D (f) sums.Wave filter with strange frequency spectrum B (f) can be by being followed by
Hilbert conversion HT's has even frequency spectrum EB(f) subsystem represents, as shown in Figure 31.(referring to above " on tool
Have the attention of the wave filter of strange frequency response ").Similarly, the wave filter with strange frequency spectrum D (f) can be by being followed by
Hilbert conversion HT's has even frequency spectrum ED(f) subsystem is represented.Note, B and EBMagnitude responses be it is identical, | B
(f) |=| EB(f) |, just as D (f) and ED(f) magnitude responses.
We will be directed to A (f), EB(f), C (f) and ED(f) positive frequency part draws equation, because negative frequency part is
Determined by respective positive frequency part.One equation will come from inputs I using positive frequency1(t)+jQ1(t)=exp (j2 π
Ft) the stimulating system (for f ＞ 0), as shown in Figure 32 A.Another equation will come to be inputted using according to equivalent positive frequency
The following negative frequency input represented carrys out stimulating system (for f ＞ 0), as shown in fig. 32b.
I2(t)+jQ2(t)=exp (- j2 π ft)
=cos (- 2 π ft)+jsin (- 2 π ft)
=cos (2 π ft)-jsin (2 π ft)
Figure 33 shows two equatioies.Equation (1.48) is based on Figure 32 A.Equation (1.49) is based on Figure 32 B.
Dependent on following notation, Figure 34 A and 34B show corresponding polar plot：
g1(f)=g (f) for f ＞ 0
g2(f)=g (- f) for f ＞ 0
Polar plot provides below equation：
These equatioies include unknown A (f), EB(f), C (f) and ED(f) in the 4x4 matrix equalities (1.54) in, such as Figure 35
It is shown.Inverted by the coefficient matrix to 4x4, we are solved.The matrix equality (1.55) seen in Figure 36.It is followed：
Due to A, EB, C and EDIt is frequency f even function.Thus, their negative frequency part is provided by even symmetry.In addition,
Strange frequency response B (f) and D (f) is given by：
B (f)=- jEB(f) sgn (f) and
D (f)=- jED(f)sgn(f).
Special circumstances：The uneven and strange phase of even gain is crooked
Gain imbalance be even function and phase is crooked be odd function in the case of, i.e. g (f)=g (- f) andEquation (1.56) to (1.59) is exclusively used in：
A (f)=0
EB(f)=0
Special circumstances：The uneven and even phase of even gain is crooked
Gain it is uneven and phase is crooked be even function in the case of, i.e. g (f)=g (- f) and
Equation (1.56) to (1.59) is exclusively used in：
EB(f)=0
ED(f)=0.
Special circumstances：Constant gain is uneven and phase is crooked
Gain it is uneven and phase is crooked be constant function in the case of, i.e. g (f)=g andEquation
(1.56) it is exclusively used in (1.59)：
EB(f)=0 (1.61)
ED(f)=0. (1.63)
Calculate the mapping between Rx and Tx
In some embodiments, transmitter carries out predistortion to digital i/q signal, is detracted so as to the I/Q of compensator itself,
As being described in a variety of different ways as more than.In order to realize this compensation, it is necessary to have what the I/Q of transmitter was detracted estimates
Meter.The quality of compensation limits the quality estimated (degree matched with the fact).Although high-quality estimation is desired,
It is difficult to the I/Q detractions of direct measurement transmitter.On the contrary, measurement is secondhand, for example, utilizing reception as shown in Figure 37
Device.
Figure 37 shows that channel (for example, cable 3720 or wireless channel) is coupled to the transmitter of receiver 3725
3700.Transmitter can include digital compensation unit 3702, DAC units 3705, I/Q modulators 3710 and front end 3715.Compensation
Unit 3702 can perform precompensation (predistortion) to data signal I (n)+jQ (n), with obtain the signal I ' (n) pre-compensated for+
JQ ' (n), for example, as being described in a variety of different ways as more than.DAC units 3705 can turn the signal pre-compensated for
Change analog signal s (t)=I ' (n)+jQ ' (n) into.Analog signal s (t) can be up-converted to RF using I/Q modulators 3710.
The signal of up-conversion is adjusted by TX front ends 3715, to obtain transmission signal.Sending signal can be transported to by cable 3720
Receiver.
Receiver 3725 can include front end 3730, i/q demodulator 3735 and digital unit 3740.Front end 3830 can be with
Transmitted signal is received from cable 3720 and operation is performed to the signal received, to produce the signal that have adjusted.Regulation
Signal can by i/q demodulator to down coversion, to produce plural down-conversion signal.Plural down-conversion signal can be by numeral
Change unit 3740 to sample, to obtain the complex signal sampled.The complex signal sampled can for carry out I/Q detractions measurement.
In some embodiments, receiver is frequency spectrum analyser, for example, vector signal analyzer.
Understand that how associated with the I/Q detractions of transmitter measuring for the I/Q detractions got in receiver 3725 is very heavy
Want.They are differed.Because detracting and (such as being detracted in the I/Q of I/Q modulators) by including TX in the I/Q of transmitter
The signal path of front end 3715, cable 3720 and receiver front end 3730 fogs (obscure) (distortion).The signal path can
To be characterized by frequency response H (f)=m (f) exp (j θ (f)), wherein H (f) is plural number.Amplitude m (f) refers to herein
" scaling " of signal path under frequency f.Phase theta (f) refers to " rotation " of the signal path under frequency f herein.
The problem of being detracted according to the I/Q of the measurement estimation transmitter based on receiver is not inappreciable.Its solution party
Case is disclosed in this patent disclosure.(referring to the alternative manner of following discloses).The part of solution includes being directed to signal
Path responses function H (f) obtains initial estimation.This part will focus on the initial estimation that acquisition form is H (0), i.e. in DC
The frequency response of signal path under (zero frequency).H (0) amplitude m (0) is referred to as " the DC scalings " of signal path.H's (0)
Phase theta (0) is referred to as " the DC rotations " of signal path.
A kind of approach of the I/Q detractions of estimation transmitter relates to the use of frequency spectrum analyser and performs iterative process.(spectrum analysis
Device is arranged to measure the value of input signal and the equipment of frequency in the frequency range of instrument).Frequency spectrum analyser measures it
The I/Q detractions for the signal that demodulate, are then based on the measurement and are compensated in transmitter application.Measurement can only roughly approximate transmitter
I/Q detractions, but it is for realizing that at least partly compensation can be good enough.Then, the signal that frequency spectrum analyser demodulate to it
I/Q detractions carry out second and measure.This second measurement can for being adjusted to the compensation applied in transmitter, etc.
Deng.Measurement sequence can restrain, i.e. the gain imbalance measured can converge to one, and the phase measured is crooked to restrain
To zero, this indicates that appropriate compensation is realized in transmitter.Because frequency spectrum analyser does not catch phase information, in order to
Realize that convergence may require that successive ignition.
In some embodiments, the I/Q detractions of transmitter can be using can carry out phase measurement and can be measurement
PGC demodulation is determined to the measuring apparatus (such as vector signal analyzer) of transmitter.In this case, the I/Q of transmitter
Detraction can be measured using two in measuring apparatus or less determined.
Method described below is measured twice, but needs the I/Q demodulation of the I/Q modulators and receiver of transmitter
Device is locked together in frequency and (mutually circulated through the lock with common reference).Unlike other methods, this method is for synchronization
Excitation (spur) (that is, the excitation for being such as phase-locked to the LO of transmitter LO leakages) has repellence.Although this technology is any
Frequency can be used, but main apply is to determine that the DC scaling m (0) and DC of signal path rotate θ (0), to calibrate transmission
The LO leakage detractions of device.
In Figure 38, vectorial A=AI+jAQRepresent when transmitter is stimulated using constant zero-signal I ' (n)=Q ' (n)=0
The LO leakages of transmitter (term " vector " is herein used as the synonym of " plural number ").Vectorial A amplitude and phase represents LO and let out
The amplitude and phase of leakage.When this LO leakage signal is moved from the I/Q modulators of transmitter to the i/q demodulator of receiver,
It has scaled m (0) and have rotated θ (0) so that vectorial A is transformed into vectorial A ' in receiver.Referring to Figure 39.Vectorial A ' passes through
Complex signal for example the sampling captured from the output of i/q demodulator is averaging to measure.
Then, we stimulate transmitter using known non-vanishing vector B：
I ' (n)=BI
Q ' (n)=BQ.
(vectorial B need not be real number as shown in Figure 38.But, it is but needed for non-zero).This intentional application
LO leakage B be added to transmitter intrinsic LO leakage A on, to cause total leakage of transmitter to be vectorial C.(B selection
(mainly its value) can influence the degree of accuracy of measurement.Optimal size will depend on specific hardware.If too small, noise will
More influence measurement；If too big, hardware can be placed in the nonlinear area of operation).This total leakage signal experience
Identical scaling m (0) and rotation θ (0) when crossing signal path with it, to cause vectorial C to be transformed into vectorial C ' in receiver.
With reference to Figure 39, it was observed that vector C ' is vectorial A ' and B ' sums.Vectorial B ' is if vector B itself is incited somebody to action when crossing signal path
Obtained vector.
In receiver, vectorial C ' from the output of i/q demodulator for example, by capturing during being stimulated by vectorial B
The complex signal sampled is averaging to measure.Due to A ' and C ' all by measurement, it is known that therefore vector B ' can be by subtracting each other
To calculate.DC scaling m (0) and DC rotation θ (0) can be calculated from vectorial B ' and vector B：
Mapping=m (0) exp (j θ (0))=B '/B
Similarly, the inverse mapping of the influence in cancel message path can be determined from inverse expression formula：
Inverse mapping=exp (- j θ (0))/m (0)=B/B '
Then, LO leaks vector A and can calculated by the way that vectorial A ' is multiplied with inverse mapping.Among practice, keep B's
Value is desirable in the A order of magnitude.Vectorial B is not sent in itself but to send vector B and another signal K sums be also one
Good practice, wherein signal K has the energy bigger than vectorial B signal and the frequency content defined away from DC, because sending
The LO leakages of device are possible in instant bandwidth become with power.For example, signal K can be tone.
In some embodiments, the complex signal sampled is opened a window.If not application widget, the frequency to tone K has
Constraint.In addition to tone K, if also there are other signal tones, they are also possible to leak among measurement.Cause
And, if without using window, tone (deliberate or be not) is preferably tied to some frequencies, to avoid leakage.
For determining the method that the LO of transmitter is leaked
1. utilize constant zero signal stimulus transmitter.
2. measure the vectorial A ' produced in receiver.
3. stimulate transmitter using the multiple constant B of non-zero.
4. in receiver measurement vector C '.
5. vector A is leaked according to the LO that below equation calculates transmitter：
B '=C '-A ' (1.64)
InvMap=B/B ' (1.65)
A=A '*InvMap. (1.66)
Once calculating the LO leakage vector A of transmitter, transmitter just can be by transmitted signals below application
Translation vector-A leaks to remove (or essentially compensating for) LO.
I ' (n)=I (n)-AI
Q ' (n)=Q (n)-AQ
Except above-mentioned I/Q detracts precompensation, compensating unit 3702 can also apply this translation.For example, complex signal (I
(n), Q (n)) the 2x2 matrixes of digital filter can be limited by, to be pre-compensated for I/Q detractions, then it is translated, with
Just LO leakages are pre-compensated for.
In some embodiments, the calculating of DC mappings can include following extra calculating.As described in this article, such as
The evaluated error of fruit phase place is too big, then alternative manner can dissipate.In the case where phase is crooked greatly, this extra step
Can be for more accurately being estimated and restrain alternative manner：(1) calculate as has been described from RX to TX
Mapping.(2) the crooked measurement of line phase is entered.(3) " gain is changed by linear system using the mapping application method from #1
Uneven and phase is crooked " calculate.(4) phase calculated for #1 wheel measuring being added to #3 is crooked, more accurate to obtain
True rotation estimation.
For calculating the method that DC mappings and DC rotate for signal path
In one group of embodiment, method 4000 can be related to the action shown in Figure 40.Method 4000 can be for estimating
Count the DC scaling m (0) of the signal path between the I/Q modulators of transmitter and the demodulator of receiver.(method 4000 can be simultaneously
Enter the random subset of feature of the above described in " calculating the mapping between Rx and Tx " part).Method 4000 is described below
Serve as reasons " processing agency " perform.Processing agency can be any system of digital circuitry, for example, processor is (in program
Under the control of instruction perform), programmable hardware element, ASIC, or its any combination.
In some embodiments, receiver observes direct converting system framework, and demodulator is simulation i/q demodulator.
In other embodiments, receiver can observe the different systems for performing analog down and ensuing digital I/Q demodulation
Framework (for example, superhet architectural framework).Thus, in this case, demodulator be by digital circuit, for example,
In software in programmable hardware element, in special digital circuit system, on a processor, or its any combination.
4010, processing agency can guide transmitter to provide zero-signal as input to I/Q modulators.Zero-signal is normal
Measure zero-signal.Zero-signal can be available to the plural number input of the DAC units (for example, Figure 37 DAC units 3705) of transmitter
Digital zero signal.Thus, I ' (n)=0 and Q ' (n)=0.
4105, processing agency can receive in response to the action of zero-signal is provided and has been captured from demodulator the
One response signal.First response signal can be caught from the output of the ADC units of receiver.(see such as Fig. 2 B digitlization list
Member is 215).
4020, processing agency can guide transmitter to provide constant B=B multiple equal to non-zero to I/Q modulatorsI+jBQ's
Constant signal is used as input.Equally, the constant signal can be supplied to the plural number input of the DAC units of transmitter.Thus, I '
(n)=BIAnd Q ' (n)=BQ.In some embodiments, B is entirely real number, i.e. BQ=0.
4025, processing agency can receive what is captured in response to the action of offer constant signal from demodulator
Second response signal.Second response signal can be caught from the output of the ADC units of receiver.
4030, processing agency can be averaging the first response signal, to obtain the first average value, and second
Response signal is averaging, to obtain the second average value.Being averaging helps to reduce the noise in measuring.
4035, processing agency can calculate the difference of the second average value and the first average value, for example, according to expression formula：
The-the first average value of average value of difference=second
4040, processing agency can calculate DC scalings based on the difference and the multiple constant of non-zero, for example, as described above.Place
Reason agency can store DC scalings in memory.
In some embodiments, method 4000 can also include the phase of the multiple constant B of phase and non-zero based on the difference
The DC rotation θ (0) of signal path are calculated, for example, according to expression formula：
θ (0)=phase (difference)/phase (B)
In some embodiments, DC scalings and DC, which rotate, is used for removing signal road from the I/Q detractions measured in receiver
The influence in footpath, to obtain the estimation to the I/Q detractions of transmitter.
In some embodiments, signal path includes the cable coupling between transmitters and receivers.In other embodiments
In, signal path includes the wireless channel between transmitters and receivers.
As the alternative arrangement for the difference for calculating average value, processing agency can be instead by from the second response signal
Subtract the first response signal and carry out calculating difference signal, then the difference signal is averaging.Then, DC scalings can be flat based on this
Average and non-zero answer constant to calculate.
In one group of embodiment, computer system be used for estimate transmitter I/Q modulators and receiver demodulator it
Between signal path DC scaling m (0), the computer system include processor and memory.Memory storage programmed instruction, its
In, programmed instruction when being executed by a processor, makes processor：Guide transmitter to provide zero-signal to I/Q modulators and be used as input；
Receive the first response signal in response to providing the zero-signal and having been captured from demodulator；Transmitter is guided to be adjusted to I/Q
The constant signal that device offer processed is equal to the multiple constant of non-zero is used as input；Connect in response to providing the constant signal from demodulation
The second response signal that device is captured；First response signal is averaging, to obtain the first average value, and the second response believed
Number be averaging, to obtain the second average value；Calculate the difference of the second average value and the first average value；It is normal again based on the difference and non-zero
Amount calculates DC scalings.Programmed instruction can be incorporated to above system, method 4000 in parallel in " calculating the mapping between Rx and Tx " part
The random subset of described feature.
Gain is changed by linear system uneven crooked with phase
When calibrating transmitter or measuring the I/Q detractions of transmitter, the method for this part can be for from the I/Q of receiver
The measurement of detraction removes the influence of signal path between the I/Q modulators of transmitter and the i/q demodulator of receiver.Those influences
The influence of the front end of front end, transmission channel and the receiver of transmitter can be included.For example, the front end of transmitter can include pair
The RF wave filters that the frequency response of signal path is worked.Similarly, the front end of receiver can include the frequency to signal path
The RF wave filters that rate response is worked.
In some embodiments, the magnitude responses m (f) of signal path can be calibrated, and phase place θ (f) is not calibrated.
(calibration can be performed precompensation in transmitter by using digital circuit 510 and/or be held using digital circuit 220 in receiver
Row post-compensation is realized).The calculating of this part allow for the correct measurement of the I/Q detractions of transmitter, without calibrating letter first
The phase response in number path.
In some embodiments, the overall frequency response (including value and phase place) of signal is calibrated.
It is given to there is frequency response H (f) system and crooked with gain imbalance g (f) and phaseInput
Signal sinput(f, t), as shown in Figure 41, we will be allowed we determined that exporting s in systemoutputThe gain of (f, t) is not
Balance g ' (f) and phase is crookedEquation.We assume that input gain imbalance g (f) and input phase are crookedIt is complete
Appear in entirely on Q input channels.But, we can not make identical in system output simultaneously and assume.In general, output point
Amount I ' (t) and Q ' (t) will have form：
Then, output gain imbalance g ' (f) and output phase are crookedIt can be determined by following formula：
G ' (f)=gQ(f)/gI(f)
Dependent on soutput(f, t)=h (t)*sinputThe fact that (f, t), derive the equation for starting from providing in Figure 42
(1.60) to (1.62), wherein h (t) corresponds to H (f) impulse response.Equation 1.61 and 1.62 is implied：
It is equation (1.63) and the right-hand side of (1.64) respectively to define A (f) and B (f)：
Moreover, the left-hand side based on equation (1.63) and (1.64) defines w (f), x (f), y (f) and z (f)：
This is followed
W (f)+jx (f)+y (f)+jz (f)=A (f) (1.69)
W (f)-jx (f)-y (f)+jz (f)=B (f), (1.70)
And therefore
W (f)=(1/2) Re { A (f)+B (f) }
X (f)=(1/2) Im { A (f)-B (f) }
Y (f)=(1/2) Re { A (f)-B (f) }
Z (f)=(1/2) Im { A (f)+B (f) }
Note, if H (f) has even magnitude responses and strange phase response, i.e. H (- f)=H (f)*, then corresponding to H (f)
Impulse response be entirely real number.Therefore, under this special case, wave filter H (f) does not change the measurement of I/Q detractions：
W (f)=Re (H (f)), x (f)=Im (H (f))
How following method description changes when signal path transfer function H (f) value and phase are probably known
Generation ground measurement TX detractions.A part for the iteration measuring method in being directed to use with this section derived equation be based in reception
The I/Q detractions of the input (or alternatively, output) of the i/q demodulator of device calculate transmitter I/Q modulators it is defeated
The I/Q detractions gone out.In order to perform this calculating, frequency response H (f) is set equal to the estimation of the frequency response of signal path
It is inverse.The different estimations of signal path frequency response can be used in different situations.
I/Q detractions are converted by linear system H (f)
In one group of embodiment, method 4300 can be related to the operation shown in Figure 43.Method 4300 can be for base
In the plural number input z in electronic systemINI/Q detraction calculate electronic system plural number output zOUTI/Q detraction.Plural number is defeated
Enter is to include the input of in-phase channel and orthogonal channel.Equally, plural number output is to include the output of in-phase channel and orthogonal channel.
(method 4300 can include spy above described in " changing gain by linear system uneven crooked with phase " part
The random subset levied).Method 4300 can be performed by processing agency, as described above.
4310, processing agency can be according to expression formulaFrequency spectrum A (f) is calculated, wherein
H (f) is the frequency spectrum of the Linear system model of electronic system, and wherein g (f) is to input z in plural numberINGain it is uneven, whereinIt is to input z in plural numberINPhase it is crooked.
4315, processing agency can be according to expression formulaCalculate frequency spectrum B (f).
4320, processing agency can calculate frequency spectrum A (f) and B (f) sums, and frequency spectrum A (f) and B (f) difference, example
Such as, according to relation：
Sum (f)=A (f)+B (f),
Diff (f)=A (f)-B (f)
4325, processing agency can be calculated and be existed with the real number of value and the real number and imaginary part of imaginary part and the difference based on this
Plural number output zOUTGain it is uneven and phase is crooked.Especially, as described above, function gI(f)、gQ(f)、WithIt can be calculated based on this with the frequency spectrum of value and the frequency spectrum of the difference, then export z in plural numberOUTGain it is uneven
With phase is crooked can be based on gI(f)、gQ(f)、WithCalculating is adopted, as shown in Figure 41.Output gain is uneven
The useful information with the crooked composition of phase, this is partly because them can be for performing the compensation of I/Q detractions or calibrating, as herein
In in a variety of different ways describe.
It is uneven crooked with output phase that processing agency can store output gain in memory.
In some embodiments, the electronic system modeled by spectrum H (f) is from the I/Q modulators of transmitter to receiver
Demodulator signal path reversion, for example, as described in herein in a variety of different ways.In the plural defeated of electronic system
Enter zINUneven and the crooked input (or alternatively, output) that can represent demodulator of phase the gain of gain it is uneven
It is crooked with phase.In the plural number output z of electronic systemOUTGain it is uneven and phase is crooked can represent I/Q modulators
Output gain it is uneven and phase is crooked.
In some embodiments, receiver observes direct converting system framework, and demodulator is simulation i/q demodulator.
In other embodiments, receiver, which can be observed, performs analog down and the different system framves of subsequent digital I/Q demodulation
Structure (for example, superhet architectural framework).Thus, in this case, demodulator is realized by digital circuitry, example
Such as, in the software in programmable hardware element, in special digital circuit system, on a processor, or its any combination.
In some embodiments, processing agency can also include the inverse of the frequency spectrum for calculating signal path, to determine spectrum H
(f), for example, as described in herein in a variety of different ways.
In some embodiments, spectrum H (f) (or can be estimated determining based on the DC scalings of signal path and DC rotations
Meter), for example, according to relation
H (f)=exp-j θ (0))/m (0)
In some embodiments, processing agency can calculate DC scalings and DC rotations in the following manner：Zero-signal
It is fed as input to I/Q modulators；In response to providing the zero-signal the first response signal is caught from i/q demodulator；
Constant signal equal to the multiple constant of non-zero is fed as input to I/Q modulators；In response to the offer constant signal from I/Q
Demodulator catches the second response signal；First response signal is averaging, to obtain the first average value, and the second response believed
Number be averaging, to obtain the second average value；Calculate the difference of the second average value and the first average value；And based on the difference and non-zero
Multiple constant calculates DC scalings.
In some embodiments, processing agency can also measure the gain imbalance g (f) of electronic equipment at multiple frequencies
It is crooked with phase(for example, guiding gain imbalance g (f) and phase to electronic equipment crookedMeasurement).Electronics
Equipment can be the tandem compound of transmitter, receiver, or transmitters and receivers, such as retouch in a variety of different ways herein
State.
In some embodiments, processing agency can be programmable hardware element.In other embodiments, processing agency can
To be arranged to the processor of the execution of responder instruction and execution method 4300.
Detracted and determined using shared LO transmitter I/Q
In one group of embodiment, for determining that the method 4400 that the I/Q of transmitter is detracted can be related to shown in Figure 44
Action.(in addition, method 4400 can be included in " being used for the iterative technique for measuring Tx detractions " part, " utilize shared LO hair
The iterative estimate for sending device to detract " partly and in " utilizing iterative estimate-optimization of shared LO transmitter detraction " part is retouched
The random subset for the feature stated).Method 4400 can be by processing agency (for example, as described in a variety of different ways herein
Processing agency) formulate (enact).
4410, processing agency can perform one group of operation.This group operation can include operation as shown in Figure 44
4415 to 4440.
4415, the complex exponential tone that processing agency can be guided under frequency f is provided to transmitter.For example, processing
Agency can issue order, complex exponential tone is provided to transmitter (or being generated by transmitter).Frequency f can be construed to
Relative to the displacement frequency of the local oscillator frequencies of transmitter.Frequency f can be non-zero.
4420, processing agency can provide precompensation conversion to the pre-compensation circuit of transmitter.Pre-compensation circuit can be with
It is configured to convert complex exponential tone application precompensation, to obtain the complex signal that have adjusted.(for example, pre-compensation circuit can be
Fig. 5 digital circuit 510 or Figure 37 compensation circuit 3702).Precompensation conversion is configurable to pre-compensate for the I/Q of transmitter
The current estimation of detraction.Transmitter be configurable to based on the complex signal that have adjusted to send signal be transmitted, for example, such as with
On in a variety of different ways describe.Receiver is configurable to receive transmission signal and catches the transmission received by representing
The complex signal of the sampling of signal, for example, as more than in a variety of different ways described in.(action of " sampling " complex signal is related to
Sample its I component and its Q component of sampling.Thus, " complex signal sampled " includes the I signal sampled and the Q sampled letters
Number.)
4425, processing agency can calculate original I/Q detractions based on the complex signal sampled.For example, original I/Q detractions
Gain imbalance and phase that can be including the complex signal sampled be crooked.For the letter on how to calculate original I/Q detractions
Breath, is shown in " accurate measuring technique " part.
4430, processing agency can convert original I/Q detractions, to determine that the I/Q converted is detracted.Conversion can be from original
Beginning I/Q detractions remove the I/Q measured the detractions of receiver.For the more information on how to perform this conversion, portion is seen
Divide " removing receiver detraction from the output detraction measured ".
As the alternative arrangement to operation 4425 and 4430, processing agency can filter to the complex signal application numeral sampled
The 2x2 matrixes of ripple device, are detracted with the I/Q that measures for removing receiver, for example, such as above in conjunction with Fig. 2A, 2B and 3 and
Described in " I/Q detractions in broadband are balanced " and " wave filter design " part.By the 2x2 matrix applications of digital filter to sampling
Complex signal will produce the complex signal filtered.The complex signal filtered can detract for calculating the I/Q converted.
Method described in " accurate measuring technique " part can be used for determining that the I/Q converted subtracts based on the complex signal filtered
Damage.
4435, processing agency can remove the current estimation of signal path from the I/Q detractions converted, to obtain path
The I/Q detractions that compensate for, wherein signal path is included from the I/Q modulators of transmitter to the path of the demodulator of receiver.(letter
Number path estimation can be by using described in " changed by linear system gain is uneven and phase is crooked " part
Method is removed).The I/Q detractions of path compensation can represent the estimation that I/Q detractions are remained to transmitter, i.e. just 4420
The realized partial correction of precompensation conversion after they be for remaining detraction be " residual ".
In some embodiments, receiver can observe direct converting system framework, and demodulator is simulation I/Q demodulation
Device, in this case, the complex signal sampled can be by being digitized to the plural simulation output for simulating i/q demodulator
To catch.In other embodiments, receiver can observe different types of architectural framework, for example, superhet architectural framework.
Thus, receiver can generate the real number analog signal (for example, real number intermediate-freuqncy signal) for representing the transmission signal received.Real number
Analog signal can be digitized, to obtain the real number signal sampled.Then, the complex signal sampled can be given birth to by calculating
Into for example, the real number signal sampled by digitally mixing and numeral are sinusoidal orthogonal right, to be sampled respectively
Complex signal I and Q component.
4440, I/Q detractions current that the I/Q detractions that processing agency can be based on path compensation update transmitter is estimated
Meter, for example, the I/Q that compensate for by combinatorial path detracts detraction corresponding to what is currently estimated.
In some embodiments, method 4400 can include repeating this group operation, to determine the transmitter under frequency f
The convergent estimation (stable estimation) of I/Q detractions.(this convergent estimation is included in the I/Q detractions of the transmitter under frequency f
Measurement).This group operation can be repeated, until the quality measurements of the I/Q detractions based on path compensation are more than threshold value.
Convergent estimation can detract for compensating the I/Q of the transmitter under frequency f at least in part, for example, such as this
Described in a variety of different ways in text.
In some embodiments, the action of this group operation of above-mentioned repetition can be performed repeatedly with itself, to determine for frequency
Convergence estimate under rate f multiple different values.The action of this group operation of above-mentioned repetition is to determine that the convergence estimate under frequency f exists
Herein referred to as " measurement of the I/Q detractions of the transmitter under frequency f ".Therefore, it is possible to carry out multiple transmitter I/Q detractions
Measurement, so as to cover multiple frequency values.
In some embodiments, the plurality of frequency values are symmetrical on zero.Furthermore, it is possible to carry out the I/Q detractions of transmitter
Measurement, to cause frequency values accessed and absolute value does not reduce in the alternate mode of symbol, for example, as herein with it is various not
Described with mode.
In some embodiments, the local oscillator of transmitter and the local oscillator of receiver are phase-locked to identical frequency
Rate is with reference to (imply that Frequency Locking).
In some embodiments, at least for the measurement of first time transmitter I/Q detractions, the current estimation of signal path is
DC scalings and DC rotations based on signal path.
In some embodiments, DC scalings and DC rotations can be determined in the following manner：To transmitter provide zero to
Measure signal；Non-zero DC vector signals are provided to transmitter；And calculated based on the response of the first DC vectors and the 2nd DC vector responses
DC is scaled and DC rotations, wherein the first DC vector responses are in response to what is measured in null vector signal in receiver, wherein the
Two DC vector responses are in response to what is measured in non-zero DC vector signals in receiver.To on how to calculate DC scalings and DC
The more information of rotation, is shown in " calculating the mapping between RX and TX " part.
In some embodiments, at least the first diagonal element of form, the wherein matrix of the precompensation conversion with 2x2 matrixes
Element is calculated according to the current estimation of the I/Q detractions of the transmitter under frequency f and-f, and wherein at least the of the matrix
One off-diagonal element is calculated according to the current estimation of the I/Q detractions of the transmitter under frequency f and-f.
In some embodiments, the current estimation of signal path includes what the complex signal sampled was measured under frequency f
Amplitude.The amplitude can be measured as described in " accurate measuring technique " part.
In some embodiments, the current estimation of the signal path also complex signal including having sampled is measured in frequency f
Rotation.
The determination detracted using offset LO transmitter I/Q
In one group of embodiment, for determining that the method 4500 that the I/Q of transmitter is detracted can be related to shown in Figure 45
Action.(in addition, method 4500 can be included in appointing for the feature described in " being used for the iterative technique for measuring Tx detractions " part
Meaning subset).Method 4500 can by processing agency (for example, as more than in a variety of different ways described in processing agency) execution.
4510, processing agency can match somebody with somebody the local oscillator (LO) of transmitter and the local oscillator (LO) of receiver
Be set to be phase-locked to common reference and so that the LO of receiver frequency subtract transmitter LO frequencies be equal to non-zero amount Δ LO.
It can be positive or negative to measure Δ LO.
4520, processing agency can perform one group of operation So.This group of So can include operation 4525 to 4550, such as scheme
Shown in 45.
4525, the complex exponential tone that processing agency can be guided under frequency f is provided to transmitter.(frequency f can be with
It is construed to the displacement of the LO frequencies relative to transmitter).Complex exponential tone can be provided in digital form, for example, as more than with
Various different modes descriptions.In some embodiments, transmitter may be coupled to (or including) and is configurable to generate multiple finger
The programmable hardware element of number tone.In order to promote this generation, PHE can be to adopt used in the DAC units of receiver transmitter
Sample clock.
4530, processing agency can provide precompensation conversion to the pre-compensation circuit of transmitter.Pre-compensation circuit can be with
It is configured to apply complex exponential tone precompensation conversion, to obtain the complex signal that have adjusted.(for example, pre-compensation circuit can be with
It is Fig. 5 digital circuit 510 or Figure 37 compensating unit 3702).Precompensation conversion is configurable to subtract the I/Q of transmitter
The current estimation damaged is pre-compensated for.Transmitter is configurable to be transmitted to sending signal based on the complex signal that have adjusted
(or sending the transmission signal drawn from the complex signal that have adjusted), for example, as described in above in a variety of different ways.Receive
Device is configurable to receive the complex signal for sending signal and catching the sampling for sending signal that representative is received, for example, such as
Describe in a variety of different ways above.Transmitter can be sent to transmission signal on transmission channel (for example, cable), and
Receiver can receive from the channel and send signal.
4535, for example, the complex signal of sampling is multiplied by by using the discrete time complex exponential signal for operating in frequency Δ LO,
Processing agency can carry out the complex signal that frequency displacement is sampled by Δ LO according to quantity, to obtain frequency shift signal.
4540, processing agency can calculate original I/Q detractions under frequency f based on frequency shift signal.Original I/Q detractions
Gain imbalance g can be includedR(f) it is crooked with phase(begged for according to the process that complex signal calculates I/Q detractions above
By).
4545, processing agency can remove the current estimation of signal path from the original I under frequency f/Q detractions, with
The I/Q detractions of the path compensation under frequency f are obtained (for example, in " converting I/Q detractions by linear system " portion as more than
Divide or described in " changing gain by linear system uneven crooked with phase " part).Signal path can be wrapped
Include from the I/O modulators of transmitter to the path of the demodulator of receiver.Can be with the I/Q detractions of frequency f path compensation
Represent the estimation to the residual I/Q detractions of the transmitter under frequency f.
In some embodiments, receiver can observe direct converting system framework, and demodulator can be simulation I/Q
Demodulator, in this case, the complex signal sampled can be by entering line number to the plural simulation output for simulating i/q demodulator
Word is caught.In other embodiments, receiver can observe different types of architectural framework, for example, superhet system
Framework.Thus, receiver can generate the real number analog signal for representing the transmission signal received (for example, real number intermediate frequency is believed
Number).Real number analog signal can be digitized, to obtain the real number signal sampled.Then, the complex signal sampled can lead to
Calculating generation is crossed, for example, the real number signal sampled by digitally mixing and numeral are sinusoidal orthogonal right, with respectively
Obtain the I and Q component for the complex signal sampled.
4550, the I/Q detractions that processing agency can be based on the path compensation under frequency f are updated under frequency f
The estimation of the I/Q detractions of transmitter.
In some embodiments, method 4500 can include repeating group So to determine the transmitter under frequency f
The convergence estimate (or stable estimation) of I/Q detractions.(I/Q that this convergence estimate can be construed to the transmitter under frequency f subtracts
The measurement of damage).For example, operational group can be repeated, until the quality measurements of the I/Q detractions based on path compensation are more than threshold
Value.(mass measurement can be the negative that the mirror image under frequency f suppresses).Convergence estimate can be used for compensating at least in part
The I/Q detractions of transmitter under frequency f.Above-mentioned frequency displacement action can utilize the Phase Continuation between being successively repeated of operational group
Frequency shift signal perform.
In some embodiments, method 4500 can also this be repeated including (operational group So's) is performed a plurality of times, with true
It is scheduled on multiple different frequency values f (value for for example covering desired transmission (or communication) band) convergence estimate.
In some embodiments, operational group So is removed before being additionally may included in frequency displacement operation from the complex signal sampled
The I/Q measured the detractions of receiver under frequency f- Δs LO.The I/Q measured the detractions of receiver under frequency f- Δs LO
Can by using constant 2x2 matrix M=(mij) be multiplied by the complex signal sampled to remove, for example, according to relation：
Wherein I (n) and Q (n) represent the in-phase component and quadrature component for the complex signal sampled respectively.
In one embodiment, matrix M can have special shape
And constant m21And m22Following formula can be based on, according to the gain of the receiver under frequency f- Δs LO not
Balance gRX(f- Δ LO) and the phase of receiver are crookedIt is determined that：
Referring to the part of entitled " performing the compensation of tradition detraction in single frequency ".
In alternative embodiment, constant m21And m22Connecing under frequency f- Δs LO and its negative-(f- Δ LO) can be based on
The I/Q that measures for receiving device detracts to determine, such as described in " calculate real single-point vector and calibrates constant " part, and
And especially in equation (1.81) and (1.82).
In some embodiments, the I/Q detractions of receiver can be measured as a part for method 4500, i.e. in frequency
Measured before shifting based on the complex signal sampled.For example, operational group So can include existing based on the complex signal measurement sampled
The I/Q detractions of receiver under frequency f- Δs LO.A kind of technology for performing this measurement is related to：(a) calculate what is sampled
The discrete time Fourier transformed value C under frequency f- Δs LO of the I component of complex signalI；(b) complex signal sampled is calculated
Q component the discrete time Fourier transformed value C under frequency f- Δs LOQ；(c) it is based on value CIAnd CQMagnitude calculation frequency
Receiver gain under rate f- Δs LO is uneven；And (d) is based on value CIAnd CQReceiver of the phase calculation under frequency f- Δs LO
Phase is crooked.To the more information of the embodiment on this technology, " accurate measuring technique " part is seen.
In some embodiments, method 4500 is additionally may included in calculated value CIAnd CQBefore should to the complex signal sampled
Use time-domain window.Time-domain window can be any one in the non-unified window of rectangle (unification) window or various standards.It is right
On the more information used of rectangular window, " rectangular window optimization " part is seen.
In some embodiments, the measurement of above-mentioned receiver I/Q detractions and the estimation of transmitter I/Q detractions can be at least
It is partly parallel to perform.For example, in one embodiment, programmable hardware element (or being possible to polycaryon processor) can match somebody with somebody
It is set to the measurement that the frequency displacement operation with the complex signal to having sampled is performed in parallel receiver I/Q detractions.
In some embodiments, operational group can include the I/ for measuring the receiver under frequency f- Δs LO as described above
Q is detracted, and calculates the 2x2 matrixes of calibration constant based on the I/Q detractions measured as described above, then right before frequency displacement operation
The complex signal application 2x2 matrixes sampled.In other words, frequency displacement operation is applied to the modification obtained from the application of 2x2 matrixes
Complex signal (I ' (n), Q ' (n)).
In some embodiments, it is assumed that the I/Q detractions of receiver are before the execution of method 4500 in frequency interested
Take and measure.Thus, the 2x2 matrixes of digital filter can detract to design based on the I/Q of receiver, such as above in conjunction with figure
2A, 2B and 3 and described in " broadband I/Q detraction balanced " and " wave filter design " part.The operational group can include
To the operation for the 2x2 matrixes of complex signal NEURAL DISCHARGE BY DIGITAL FILTER sampled before frequency displacement operation.Then, resulting filter
The complex signal of ripple may by frequency displacement.
In some embodiments, form of the precompensation conversion with 2x2 matrixes, and matrix has at least one of matrix
The current estimation of I/Q detraction of the diagonal element based on the transmitter under frequency f and the I/Q detractions of the transmitter under frequency-f
Current estimation come the attribute calculated and I/Q of at least one off-diagonal element based on the transmitter under frequency f of matrix
The current estimation of detraction and the current estimation of the I/Q detractions of the transmitter under frequency-f are come the attribute that calculates.In some implementations
In example, each in four matrix elements is calculated by this way.
As described above, processing agency can remove the current estimation of signal path from the original I under frequency f/Q detractions,
Detracted with the I/Q for obtaining the path compensation under frequency f.In some embodiments, the current estimation of signal path can be wrapped
Include measuring and amplitude for the frequency shift signal under frequency f.In one embodiment, the current estimation of signal path can also be wrapped
Include the rotation measured of the frequency shift signal under frequency f.
In some embodiments, the current estimation of signal path can be based on signal path DC scalings and DC rotations.This
Plant the first time execution that estimation can be used at least group operation.
In some embodiments, method 4500 can also include determining DC scalings and DC rotations in the following manner：Xiang Fa
Device is sent to provide null vector signal；Non-zero DC vector signals are provided to transmitter；Based on the response of the first DC vectors and the 2nd DC vectors
Response calculates DC scalings and DC rotations, wherein the first DC vector responses are in response to what is measured in null vector signal in receiver,
Wherein the 2nd DC vector responses are in response to what is measured in non-zero DC vector signals in receiver.To on DC scalings and DC rotations
Determination more information, referring to " calculate Rx and Tx between mapping " partly and " be used for for single path computing DC mappings and
The method of DC rotations " part.
Determine the I/Q detractions of receiver
In one group of embodiment, for determining that the method 4600 that the I/Q of receiver is detracted can be included in shown in Figure 46
Operation.Method 4600 can be acted on behalf of by above-mentioned processing and performed.
4610, processing agency can guide input signal to be provided to receiver.In other words, processing agency can send out
Cloth order is so that input signal is provided to receiver (or being generated by receiver).Input signal can be included in displacement frequency
Isolation tone under f and the invalid interval around displacement frequency-f (that is, the interval for only including noise).(say tone given
Be under frequency " isolation " mean tone be except given frequency neighbouring frequency (for example, center given frequency frequency
In interval) noise outside sole energy source.If noise energy is too big, measurement quality will degrade.Tone is preferably this
Unique significant energy source in neighbouring frequency).Receiver is configurable to demodulate input signal, to obtain answering of having sampled
Signal, for example, as described in above in a variety of different ways.Displacement frequency f and-f can be shaken relative to the local of receiver
Swing the displacement of device frequency.
4615, processing agency can calculate the I/Q detractions of the receiver under frequency f based on the complex signal sampled.
4620, processing agency can be directed to across allocated frequency band (the input frequency band of the receiver of such as current selected or
The communication band of standardization) frequency values f repeat guide (4610) and calculating (4615) action.
4625, processing agency can store the receiver I/Q detractions for frequency f each value in memory.
In some embodiments, input signal is provided by transmitter, and the local oscillator frequencies of the transmitter are from receiver
Local oscillator frequencies offset a nonzero value, for example, as the above in a variety of different ways described in.
In some embodiments, input signal is provided by calibration tone synthesizer.Calibration tone synthesizer be arranged to for
Calibration other systems and the system for creating quality tone.In some embodiments, term " quality tone " is implied in amplitude, frequency
Rate, temperature or temporal stability.In one embodiment, receiver includes being easy to self-alignment calibration tone synthesizer.
In some embodiments, calculating the action of the I/Q detractions of the receiver under frequency f includes：Calculate what is sampled
The discrete time Fourier transformed value C under frequency f of the I component of complex signalI；The Q component for calculating the complex signal sampled exists
Discrete time Fourier transformed value C under frequency fQ；Based on value CIAnd CQReceiver of the magnitude calculation under frequency f gain
It is uneven；And based on value CIAnd CQReceiver of the phase calculation under frequency f phase it is crooked.
In some embodiments, method 4600 is additionally may included in described value CIAnd CQCalculating before to answering for having sampled
Signal application time-domain window, for example, as below described in " accurate measuring technique " part.
Measure the I/Q detractions associated with complex signal
In one group of embodiment, method 4700 can include the operation shown in Figure 47.Method 4700 can be for measurement
The I/Q that complex signal with the sampling that is produced by receiver is associated is detracted.Method 4600 can be by processing agency (for example, in journey
The computer system performed under the control of sequence instruction) perform.
4710, processing agency can utilize stimulus signal to stimulate receiver with guiding apparatus, and the stimulus signal has
Isolation tone under displacement frequency f and the invalid interval under displacement frequency-f.Displacement frequency f and-f can be construed on connecing
Receive the displacement of the local oscillator frequencies of device.The complex signal sampled can be in response in using stimulus signal stimulation action and
The baseband signal produced by receiver.
4715, processing agency can calculate the discrete time Fourier under frequency f of the I component for the complex signal sampled
Leaf transformation value CI。
4720, processing agency can calculate the discrete time Fourier under frequency f of the Q component for the complex signal sampled
Leaf transformation value CQ。
4725, processing agency can be based on value CIAnd CQSampling of the magnitude calculation under frequency f complex signal increasing
Beneficial imbalance g, the gain that wherein gain imbalance g includes receiver is uneven.
4730, processing agency can be based on value CIAnd CQSampling of the phase calculation under frequency f complex signal phase
Position is crookedWherein phase is crookedPhase including receiver is crooked.
In some embodiments, processing agency can be in described value CIAnd CQCalculating before should to the complex signal sampled
Use time-domain window.
It is in some embodiments calibration tone generator there is provided the equipment of input signal.
In some embodiments, equipment is transmitter, and the local oscillator frequencies of the transmitter are shaken from the local of receiver
Swing device frequency (intentionally) and offset by a non-zero amount.In a kind of such embodiment, the complex signal sampled receives frequency
Move, to remove the difference between local oscillator frequencies, in this case, gain imbalance g and phase are crookedCan portion
Ground is divided to be detracted dependent on the I/Q of transmitter.Especially, gain imbalance g and phase are crookedThe I/Q that transmitter can be represented subtracts
The distortion damage, introduced by (between the I/Q modulators of transmitter and the demodulator of receiver) signal path and receiver
The synthetic effect of I/Q detractions.In another such embodiment, the complex signal sampled is come from also not by above-mentioned frequency
The primary signal of the demodulator of shifting, therefore, gain imbalance g and phase are crookedIt can be construed to only include to be introduced by receiver
Detraction.
In some embodiments, method 4700 is additionally may included in calculated value CIAnd calculated value CQBefore to answering for having sampled
Signal application time-domain window, for example, as described below.
In some embodiments, receiver is vector signal analyzer.
In some embodiments, operation one or more of 4715 to 4730 can be performed by programmable hardware element.
In some embodiments, operation one or more of 4715 to 4730 can be held in special digital circuit system
OK.
In some embodiments, operation one or more of 4715 to 4730 can be by processor response in programmed instruction
Execution and perform.
Offset LO collimation techniques
Skew local oscillator (LO) method allows the survey that the I/Q for entering line receiver (RX) and transmitter (TX) simultaneously is detracted
Amount and carrier leak measurement.This method using can the LO of independent tuning be used for transmitters and receivers, for example, such as institute in Figure 48
Show.In some embodiments, transmitter LO step-length (step size) and/or receiver LO step-length substantially can be point
Number or integer.In some embodiments, the step-length of transmitter and/or receiver LO step-length should be the small of whole instant bandwidth
Percentage.
Transmitter includes I/Q modulators 4810 and front end 4815.Complex exponential tone under non-zero displacement frequency f is provided
To I/Q modulators 4810.I/Q modulators 4810 (are also referred to as " local oscillator signals ") using tone modulation carrying signal, with
Obtain the signal modulated.Carrying signal is provided by transmitter LO4805.The signal modulated is sent by transmitter, front 4815
Onto transmission medium (for example, cable 4820).
The front end 4830 of receiver receives the signal that have sent and adjusts received signal, to obtain what be have adjusted
Signal.I/q demodulator 4835 is using the signal that have adjusted by the receiver LO4840 carrying signals provided, so as to be had
There is the demodulated signal for the component for being expressed as RX I and RX Q.
As shown in Figure 49, RX and TX carrier waves are offset from one another to the hair by tone, the receiver mirror image of tone, tone is caused
The carrier leak of device mirror image, the carrier leak of transmitter and receiver is sent to be presented on different frequencies.Illustrated frequency spectrum is based on
In the demodulated signal of receiver.Transmitter produces tone under 31MHz.The frequency spectrum includes two different carrier leaks, one
It is due to the LO leakages of transmitter, another is due to the LO leakages of receiver.Two different primary mirrors of the frequency spectrum also including tone
Picture, one be due to transmitter I/Q detraction, another be due to receiver I/Q detraction.In addition, frequency spectrum includes transmitter
Mirror image receiver mirror image, and the carrier leak of transmitter receiver mirror image, be both due to the I/Q of receiver
Detraction.In this example, receiver carrier wave is arranged to 6MHz lower than transmitter carrier.This causes the mirror image of tone, transmitter
Leakage with transmitter shows the frequency in the high 6MHz of transmitter in receiver ratio.Then, in addition to receiver is leaked, due to
The result of the detraction of i/q demodulator, (tone, TX mirror images (TX Image) and Tx are leaked these three signals produced by transmitter
(TX Leakage)) in each after i/q demodulator have corresponding mirror image.
The frequency shift (FS) sent and received by knowing between LO and the tone produced before the modulator in transmitter
Frequency, the definite spectrum position of all detraction illusions can determine completely.If we make
FreqOffset=TxCarrierFrequency-RxCarrierFrequency, (1.75)
(as receiver is seen) frequency location of spectrum signature in the frequency spectrum then received is：
RxTone=TxTone+FreqOffset (1.76)
TxLeakage=FreqOffset (1.77)
TxImage=FreqOffset-TxTone (1.78)
RxImage=-TxTone-FreqOffset (1.79)
RxLeakage=0Hz (1.80)
RxImageofTxImage=TxTone-FreqOffset (1.81)
=RxTone-2FreqOffset
RxImageOfTxLeakage=-FreqOffset. (1.82)
The I/Q detractions and carrier leak of measuring receiver are with identical with conducted in " accurate measuring technique " part
Mode perform.But, measurement transmitter detraction relate in general to it is more because there is multiple things to consider.Measurement
The detraction of transmitter can be related to removing receiver detraction.Figure 50 shows what is received after the I/Q detractions of receiver are removed
Frequency spectrum.After that removal, frequency spectrum can be carried out frequency displacement by-FreqOffset, as shown in Figure 51.Now, the frequency that offset by
The frequency location of " tone " is identical with the original frequency f when transmitter is produced of tone in spectrum.In addition, the leakage of transmitter
(TXLeakage) and transmitter mirror image (TXImage) under correct frequency location (being-f and zero respectively), so as to once count
Calculate and remove rotation just using the algorithm found in " accurate measuring technique " part.(rotation can utilize " calculate RX and
Method described in mapping between TX " part is calculated).This algorithm will be provided to the I/Q detractions of transmitter and transmission
The estimation of the LO leakage vectors of device.As long as method signal path (including the transmitter of this I/Q detractions for being used to measure transmitter
Front end and receiver front end) there is even magnitude responses and strange phase response will work.In fact, situation is not this
Sample, and microvariations even in value or phase to measurement the problem of all can cause serious.Iterative algorithm eliminates this and asked
Topic.The iteration of iterative algorithm is related to the current estimation detracted based on transmitter and performs precorrection (for example, utilization " calculates real
The method of single-point vector calibration constant " part), and remove the optimal available of signal path from the detraction measured in receiver
Estimation (method for utilizing " changed by linear system gain is uneven and phase is crooked " part).Iterative algorithm is even at that
Having in the initial estimation detracted a bit also allows measurement transmitter detraction when error.
By carrying out the everything in above method in addition to removing receiver and detracting, the detraction for measuring transmitter can be with
Further optimization.It is illustrated that in Figure 52 without the frequency displacement frequency spectrum for removing receiver detraction first.By the way that these detractions are stayed in
In frequency spectrum, I/Q of the detraction measured under frequency f (that is, being in this example 31MHz) not exactly equal to transmitter subtracts
Damage, because the detraction of receiver makes measurement distortion.But, it can also be removed for removing the identical iterative algorithm of distortion of RF front ends
Go because receiver detracts the distortion caused.Although desirably removing the detraction of receiver more preferably, among actual this
Spent additional time during calibrating.
Constraint
Although this method is high expectations, because multiple measurements can be carried out concurrently, it is really with constraint.
Main constraint is that it can not be used for measurement amplitude, because the combination of its measuring receiver amplitude and transmitter amplitude, and is not being had
There is no any mode to separate both in the case of having other measurement.But, if receiver amplitude or transmitter amplitude are
Know, then both can be separated.In most cases, relative to frequency, amplitude ratio I/Q detraction changes are slow.Therefore, individually
Measurement process can be used for by than being measured for determining the coarse frequency step of step-length of I/Q detractions in instant bandwidth
Receiver amplitude or transmitter amplitude.Therefore, the overall measurement time including amplitude is still than alternative (alternative)
Faster.
Another minor issue on offseting LO methods is that it puts to constrain to calibration frequency planning.Dependent on LO shifted by delta
LO value, it is possible to be destroyed measurement under each measurement offset.As shown in Figure 49, there are seven energy in frequency spectrum
Measure the position occurred in response to the transmission of tone.The correct measurement of whole for to(for) transmitters and receivers is detracted, all
This seven signals must keep orthogonal, i.e. cannot occur without two signals in identical frequency location.If for example, connect
The LO of receipts device is arranged to 2.400GHz and the LO of transmitter is arranged to 2.39GHz, then measurement destruction is by transmitted base band
Tone occurs when being 4MHz, because this will definitely be placed on tone RX leakages (RX Leakage)；Occur in -4MHz, because
TX mirror images will be placed on RX leakages for this；Or occur in 8MHz, let out because RX mirror images (RX Image) will be placed on TX by this
Leakage.In order to avoid these problems, the tone that have sent (TxTone) can not be located at frequencies below：
{N*FreqOffset：N=-3, -2, -1.0,1,2,3 }
In addition, also bandwidth is limited.It is total can measuring tape it is wide be (TotalBW-LO_StepSize), and total symmetrical survey
Bandwidth is (TotalBW-2*LO_StepSize).This be LO step-lengths must be total instant bandwidth a part (preferably a small portion
Point) the reason for.If for example, instant bandwidth is 100MHz, and LO step-lengths only have 25MHz, then the 75MHz of bandwidth is theoretically
It is measurable.In fact, because we it is generally desirable to symmetrical bandwidth (that is, +/- 25MHz rather than -25MHz-50MHz), because
This our symmetrical can the wide only 50MHz of measuring tape.Further, since in decay (roll-off) effect of belt edge, there is less
Can measuring tape it is wide.
Calculate real single-point vector calibration constant
In view of in the case where understanding f and-f I/Q detractions, how this shown partially calculates for will be ideally in list if being
The constant (that is, ideally precompensation single frequency f I/Q detractions) of the real single-point calibration of individual position precorrection, such as in Figure 53 A
With indicate in 53B.Single-point vector calibration correction 5310 is before the vectorial damage model 5320 of two point.Thus, it is used as input quilt
It is supplied to the complex exponential tone under frequency f of the vectorial calibration correction of single-point to be pre-distorted, to produce complex signal cos (2 π ft)+j
Γsin(2πft+θ)
The signal of predistortion is by the further distortion of damage model 5320, so as to cause equal to original complex exponential tone
The output signal corrected.
According to " destruction I/Q detractions " part, we know how draw the 2x2 frequency responses that the I/Q of the system of representative is detracted
Matrix H.In the portion, it has been found that A (f), EB(f), C (f) and D (f) is (that is, by under f by " two point I/Q detractions "
I/Q is detracted and the I/Q detractions under-f) determine.In addition, according to " addition is constrained ", (that is, situation 6, wherein A and C are normal for part
Amount, and EBAnd EDIt is zero), known to the structure of single point correction.Utilize this information, it may be determined that real single-point calibration factor alpha and
β。
Given A (f), EB(f), C (f) and ED(f) these values, it is therefore an objective to it is determined that value α and β.Given two point I/Q detractions, i.e.
Gain unbalanced value g1(f)=g (f) and g2(f)=g (- f) and the crooked value of phaseAndA(f)、EB(f), C (f) and ED(f) according to known to equation (1.56) to (1.59).α and β value can root
Determined according to the Γ as shown in following formula and θ：
α=Γ sin (θ) (1.75)
β=Γ cos (θ) (1.76)
Using Figure 54 polar plot, equation (1.77) is obtained along the summation of x-axis, and along the summation of y-axis to obtain
Equation (1.78)：
CΓsin(θ)-EDΓ cos (θ)=- A (1.77)
CΓcos(θ)+EDΓ sin (θ)=1-EB (1.78)
We are dependent on the fact：
HT { sin (t) }=- cos (t),
HT { cos (t) }=sin (t),
Wherein HT represents that Hilbert is converted.Equation (1.77) and (1.78) are implied：
When need not solve α and β, solving Γ and θ needs the new gain and phase that teach that waveform to want true
Eliminate the influence of I/Q destructions with cutting.
It should be pointed out that the correction coefficient alpha and β that are provided by (1.81) and (1.82) in general with such as " in single frequency
α during traditional single-point is compensated described in rate execution tradition detraction compensation " part is different with β.(thus, when as precompensation,
That is, when in Figure 53 A and 53B, in general traditional single-point offset will provide dissatisfactory compensation).But, exist
Particular case of the two of which coefficient to conflict.As explained in " destruction I/Q detractions " part, when gain is uneven and phase
When the crooked function in position is even function, damage model value is reduced to：
EB(f)=0
ED(f)=0.
Thus, equation (1.81) and (1.82) will be specific to：
This is and traditional single-point compensation identical value used.
Iterative technique for measuring TX detractions
With reference now to Figure 55 A, the problem of I/Q of the amplitude-frequency response and receiver that measure receiving filter 5525 is detracted is able to
Simplify (relative to the correspondence problem of transmitter), because the I/Q from i/q demodulator 5530 is detracted in receiving filter 5525
Occur after distortion effect.For example, if pure pitch is to the input signal of RX path, the distortion of receiving filter will be only
Change the value and phase of single tone.Then, this pure pitch that have changed will be by i/q demodulator distortion, so as to produce
I/Q is detracted.When calibrating receiver, we can remove the I/Q detractions of receiver first, leave behind the amplitude and phase of wave filter
Position response effect, then, if desired, the just amplitude of correcting filter and phase distortion in additional step.
But, for transmitter, situation is not so.It is illustrated that what is combined for transmitters and receivers in Figure 55 B
Signal path.Transmitter includes I/Q modulators 5510 and transmitting filter 5515.In some embodiments, LO is sent and received
It is shared.When transmitter creates single tone, I/Q modulators 5510, which are introduced, sends I/Q detractions.Then, these detractions exist
Transmission signal path, cable are travelled through before finally reaching i/q demodulator and signal path is received.I/Q modulators export and
The measurement for the transmission I/Q detractions that this paths destruction between i/q demodulator input is obtained in receiver.In addition, demodulator
The measurement that further destruction is detracted in the transmitter I/Q that receiver is obtained of I/Q detractions.In alternative embodiment, receiver can
With based on alternative RF architectural frameworks (that is, different from direct converting system framework), to cause the I/Q of receiver to detract very
It is small, i.e. to be small enough to and ignore.
How the non-flat amplitude-frequency response being illustrated that in Figure 55 C in signal path, which destroys the I/Q seen in receiver, subtracts
The example of damage.Produced by I/Q modulators is actual I/Q detractions.Then, send signal path destroy them, be afterwards by
The phase place caused in the electric delay of cable, is another destruction caused by reception signal path afterwards.Except amplitude, phase
Position response also results in different but correlation (not shown in Figure 55 C) the problem of.
To initial inspection, it appears that preferable solution will characterize to believe between I/Q modulators and i/q demodulator first
The value and phase in number path.Then, by using in " gain being changed by wave filter uneven crooked with phase " part
Calculating, the I/Q detractions that the influence of signal path can be measured from receiver remove.But, give for detracting what is suppressed
Performance requirement, this is not a rational task.In order to realize that the more preferable mirror images of ratio -80dB suppress, the crooked needs of phase are less than
0.01 degree.Even in lower RF frequency, this also means that absolute phase must be stablized and measurable, than the degree of accuracy of picosecond
More preferably.In addition, the value and phase of signal of the I/Q detraction changes from modulator, such as " from the I/Q values detracted and phase
Described in position destruction " part and as expressed by Figure 58 A equation (4.9).Therefore, in order to determine signal path
Value and phase response, the I/Q detractions of transmitter will need to be known, and exactly our trials will for the I/Q of transmitter detractions
Measurement.
The more preferable method that definite I/Q detractions are determined by signal path is that solution is iterated.Give to signal path
The rough estimate of amplitude and phase and the estimation detracted to I/Q, definite I/Q detractions can be by enough iteration come really
It is fixed.(iteration can be performed using shared LO as described below or skew LO.In the case of shared LO, the I/Q of receiver subtracts
It is known to damage needs.In the case where offseting LO, receiver I/Q detractions need not be, it is known that while it is known that they can be helpful.
In both cases, the I/Q detractions of transmitter need not be previously known.They will be determined as the result of iteration).Iteration
Sum will be largely dependent upon initial estimation and performance standard.What is be listed below is for true to shared LO and skew LO
Determine the process of transmitter detraction.This process measurement calibration frequency position all in instant bandwidth, once for given
All measurements of instant bandwidth have all been completed, and just only these measurements are iterated.What is provided in the part on optimization is
Obtain identical result but in general need the process that have modified of less iteration.
Alternative manner step (general introduction)：
1. tune RX and TX LO.
2. measure RX detractions.
3. measure the mapping between RX and TX.
4. apply the detraction correction of estimation in TX.
5. generate tone in TX and measured in RX.
6. remove RX detractions from #5.
7. remove signal path estimation (for example, using mapping from #3).
8. combining the result of all iteration from #7, estimated with producing the detraction that have updated.
If 9. performance metric is acceptable, just going to #10；Otherwise just #4 is gone to be iterated.
10. couple each LO frequency repeat step #1 to #9.
Alternative manner step (description)
1. send and receive LO be tuned to first expect LO frequencies.If using shared LO (using identical LO or
Utilize the LO of two separation being locked together), then LO will be in identical frequency.In the case where offseting LO, LO is inclined each other
Move some known definite amount.In either case, it is lock phase to all ensure that all LO.For inclined on selection work
The more information of shifting amount, is shown in " constraint " subdivision of " skew LO methods calibration method " part.Also to remember to use in the measurements
Window.If without using window, just as done in " rectangular window optimization " part, then to ensure skew LO values office
It is limited to the frequency given in that part.
(2. being optional when using skew LO methods) is for each band bias internal frequency of transmitter to be measured, measurement
The gain of receiver is uneven and phase is crooked.This can be by using in " accurate measuring technique " measurement side specified in part
Method is realized.Due to making mirror image mirror image seem to be in different frequencies for receiving and sending using skew LO, therefore remove
It is crucial that detraction, which is received, unlike in the case of shared LO.In all known data sets, when LO is shifted by, this
Alternative manner convergence is planted, detraction is received without knowing.But, receive detraction and cause certain to destroy to sending detraction really.Cause
This, if they are too serious, then, when using skew LO, they will also result in this alternative manner diverging, rather than
Convergence.
3. a transmitter output is connected to receiver input.
(4. just for skew LO methods) is carried out the frequency spectrum of frequency displacement receiver by the amount equal to LO offsets.If for example, hair
Send the LO that the LO of device is located at 2.400GHz and receiver positioned at 2.404GHz, then make the positive 4MHz of spectrum offset.Frequency displacement must lock
LO is mutually arrived, the rotation estimation otherwise made in steps of 5 can not keep fixed.
5. the rotation between determining to receive and send by using the algorithm in " calculating the mapping between RX and TX " part
With scaling mapping.Because leakage can be sensitive to band internal power, therefore, for more preferable result, in instant bandwidth somewhere
Using tone.This mapping will be constant and can be repeated once LO is set.Thus, at least some embodiments
In, LO needs to be lock phase.When using skew LO methods, definite LO offsets are known.
If 6. this is #6 first time iteration, just not applying any correction (straight-through) in transmitter and proceeding to #7.
Otherwise, based on the measurement in #10 in transmitter application correcting filter.
7. for measurement position in each desired band, in transmitter application complex exponential tone, and by using " accurate
Computational methods in e measurement technology " part determine that original gain is uneven and phase is crooked in each frequency offset.
(8. being optional when using skew LO methods) for #7 in the value each measured, removed by mathematical way
The gain of receiver is uneven and phase is crooked.This can pass through " removing receiver from the output detraction measured to detract " part
Described in calculating carry out.This places the measurement of transmitter before the demodulator.Instead of step #8, another method be
Transmitted before step #7 in receiver application correcting filter (according to " broadband I/Q detract balanced " part) and by the correction
The waveform captured.This method is inaccurate, because due to limited wave filter valve, correcting filter is possible to unlike measurement
It is so accurate.
9. for the value each calculated in #8, by using " uneven askew with phase by linear system change gain
Conversion described in tiltedly " removes approximate known rotation, scaling, value and phase.Rotation and scaling are determined in step #5
's.After the first iteration, the estimation of value can also be determined.This approx sets measurement in the output of modulator.If
Measurement is definitely in the output of modulator, then we will not need this alternative manner.Need this alternative manner be because
We do not know the rotation in the path between the output of modulator and the input of demodulator, scaling, amount within the required degree of accuracy
Value and phase.
10. by finding out the product (when using lineal scale) of all gain imbalances and based on each frequency offset
The result of all iteration from #9 is combined with the crooked sum of all phases of LO combinations.If for example, measurement be -
Performed under 15MHz, -5MHz, 5MHz and 15MHz, then the measurement only obtained under -15MHz is combined to one from other iteration
Rise.When being moved to another LO in #13, this combination restarts so that the survey under -15MHz and LO=2.4GHz
Amount is not combined with the measurement under -15MHz and LO=2.6GHz.
11. according to the gain of each in-band frequency position that is being measured in #9 and being calculated by equation 4.15 not
Balance and the crooked mirror image that calculates of phase suppress.The minimum value calculated by finding out all mirror images to suppress determine it is worse in the case of it is whole
The mirror image of individual frequency band suppresses.
12. if the mirror image from #11 suppresses the performance metric needed for meeting, final gain is uneven and phase is askew
Tiltedly measurement is those calculated in step #10 and does not need more iteration for this LO frequency, otherwise by going to #6
Solution is iterated.
13. couple each LO frequency repeat step #1 to #11.
In one group of embodiment, the I/Q detractions of transmitter can be estimated according to the method provided in appendix A.
As a result
Figure 56 A and 56B show that a kind of improvement of each iteration of embodiment according to alternative manner (that is, restrains speed
Rate).In at least some embodiments, for value, alternative manner has the interval of convergence of [- 3dB, 3dB], for phase, tool
There is the interval of convergence of [- 30 degree, 30 degree].In these embodiments, if magnitude or phase have the mistake outside these intervals
Difference, then measuring sequence will dissipate.Figure 56 A and 56B show the convergence of each iteration for value error and phase error.
Optimization
How the description of this part optimizes above-mentioned iterative process to use less total collection and therefore use less calibration
Time.The problem of on above-mentioned iterative process is：Except calculating new wave filter between iterations, it is also to the list in instant bandwidth
Individual wide-band width measurement carries out multi collect.But, a single point is iterated by using the calibration of single-point vector to determine in fact
The detraction value on border, so that the sum of collection can greatly reduce.Then, by being stepped across band, the previous measurement position of detraction
Become the estimation of next measurement position.When detraction is not when band change is quickly, this is worked well with, thus to neighbouring reality
The estimation that actual value is provided.
By adding this optimization, it is desirable to create following frequency planning：
[Δ f/2,-Δ f/2,2* Δ f/2, -2* Δ f/2,3* Δ f/2, -3* Δ f/2 ..., N* Δ f/2,-N* Δ f/2]
For Integer N, wherein Δ f is the spacing of frequency measurement position in instant bandwidth.Because it is by using its neighbour
Produce the best estimate for the new point to be measured, this maximum benefit optimized.Because this method uses the real of transmitter
Single-point calibration, therefore its needs is on the information in tone locations and its detraction of mirror image.This is handed between positive and negative frequency
For the reason for.To the process of following numbering, this alternate frequency planning is also assumed that.
The alternative manner step (descriptive) of optimization：
1. send and receive LO be tuned to first expect LO frequencies.If using shared LO (using identical LO or
Utilize the LO of two separation being locked together), then LO will be in identical frequency.In the case where offseting LO, LO is inclined each other
Move some known definite amount.In either case, it is lock phase to all ensure that all LO.To on choosing work shift
The more information of amount, referring to " constraint " subdivision of " skew LO methods calibration method " part.Also to remember to use in the measurements
Window.If without using window, it is necessary to ensuring skew LO values office just as done in " rectangular window optimization " part
It is limited to the frequency given in that part.
(2. being optional when using skew LO methods) is for each band bias internal frequency of transmitter to be measured, measurement
The gain of receiver is uneven and phase is crooked.This can be by using in " accurate measuring technique " measurement side specified in part
Method is realized.Received due to making mirror image seem to be in different frequencies for receiving and sending using skew LO, therefore removing
Detraction is crucial unlike in the case of shared LO.It is this when LO is offset in all known data sets
Alternative manner is restrained, and detraction is received without knowing.But, receive detraction and cause certain to destroy to sending detraction really.Therefore, such as
Really they are too serious, then, when using skew LO, they will also result in this alternative manner diverging, rather than convergence.
3. a transmitter output is connected to receiver input.
(4. only for skew LO methods) presses the frequency spectrum of the amount frequency displacement receiver equal to LO offsets.If for example, sent
The LO of device is located at 2.400GHz and the LO of receiver is located at 2.404GHz, then makes the positive 4MHz of spectrum offset.Frequency displacement is phase-locked to
LO.(the rotation estimation otherwise made in steps of 5 can not keep fixed.)
5. rotation and contracting between determining to receive and send by using the algorithm in " calculating the mapping between RX and TX "
Projection is penetrated.Because leakage can be sensitive to band internal power, therefore, for more preferable result, in somewhere applying for instant bandwidth
Tone.This mapping should keep constant and can be repeated once LO is set.Thus, at least some embodiments,
LO is lock phase.When using skew LO methods, definite LO offsets are known.
If 6. this is the first time iteration of the #6 to this specific LO frequency, just not applying any correction in transmitter
(being only through) and proceed to #7.Alternatively, if this is the first time iteration of the #6 to this specific LO frequency, in step
Using the tone near 0Hz and using leakage (0Hz) information with being used in algorithm while the gain obtained is uneven in rapid #5
Weighing apparatus and the crooked information of phase all produce the initial estimation of detraction to tone and mirror image.Otherwise, using in " the real single-point of calculating
The calculating found in vector calibration constant " is based on following measurement in transmitter application single point correction (assuming that frequency provided above
Rate is planned).
If a. this be the #6 since #13 first time iteration, the estimation of optimal tone is in variable $ Previous_
Found in Impairments2.Otherwise, #10 currency is best estimate.
B. optimal mirror image estimation is found in variable $ Previous_Impairments1.
7. for current measurement position, in transmitter application complex exponential tone, and by using " accurate measuring technique "
Computational methods in part determine that original gain is uneven and phase is crooked to this specific in-band frequency offset.
(8. optional when using skew LO methods) for #7 in the value each measured, receiver is removed by mathematics
Gain it is uneven and phase is crooked.This can pass through institute in " the output detraction according to measuring removes receiver and detracted " part
The calculating of description is carried out.This sets the measurement of transmitter before the demodulator.Instead of step #8, another method is to pass through
Correction needed for (according to from " broadband I/Q detract balanced " part) is calculated is filtered before step #7 in receiver application correction
Device and the waveform captured by correction transmission.This method is inaccurate, because, due to limited wave filter valve, correction
Wave filter is possible to so accurate unlike measuring.
9. by using the conversion described in " changing gain by linear system uneven crooked with phase ", removed from #8
Approximate known rotation, scaling, value and phase.Rotation and scaling are determined in step #5.The good estimation of value can
To be found out by the same way with the good estimation that detraction is found out in step #6 using its neighbours' value.This approx exists
The output of modulator sets measurement.If measurement is definitely in the output of modulator, we will not need this alternative manner.
Need this alternative manner be because we do not know within the required degree of accuracy modulator output and demodulator input it
Between path rotation, scaling, value and phase.
10. by finding out the product (when using lineal scale) of all gain imbalances and based on each frequency offset
The result and variable $ Previous_ of all iteration from #9 are combined with the crooked sum of all phases of LO combinations
Impairments2.If for example, measurement is performed under -15MHz, -5MHz, 5MHz and 15MHz, only in -15MHz
The measurement of lower acquirement is grouped together from other iteration.When being moved to another LO in #13, this combination restarts,
So that the measurement in -15MHz and LO=2.4GHz is not combined with -15MHz and LO=2.6GHz measurement.
11. suppress by using the gain from #9 and equation 4.15 is uneven with the crooked information of phase to calculate mirror image.
If 12. mirror image from #11 suppresses the performance metric needed for meeting, for the final of current measurement position
Gain is uneven and the crooked measurement of phase be those calculated in step #10 and this LO frequency is not needed it is more repeatedly
Generation.Therefore, proceed to #13 and the value in variable $ Previous_Impairments1 is saved in $ Previous_
In Impairments2, and the current measurement of storage in variable $ Previous_Impairments1.Otherwise, by going to #6
Solution is iterated.
13. couple each in-band frequency measurement position repeat step #6 to #12.
14. couple each LO frequency repeat step #1 to #13 and remove all variables.
In some embodiments, the I/Q detractions of transmitter can be estimated using the skew LO as described in Appendix B.
In other embodiments, the I/Q detractions of transmitter can be estimated using the shared LO as described in appendix C.
From the I/Q values detracted and phase destruction
The export of this part is to understanding how I/Q detractions destroy the useful various equatioies of value and phase of signal.We will
See that form isSignal s (f, t) be included in sound under frequency f
Mirror image of the mediation under frequency-f.Figure 57 provides the notation of the amplitude for tone and mirror image.Including equation (4.8) extremely
(4.21) derivation is provided in Figure 58 A and 58B.Equation (4.11) provides the amplitude of tone | α | the result detracted for I/Q.Note
Meaning, if gain imbalance is equal to one and phase is crooked is equal to zero, the amplitude of tone does not change.In addition, once detracting
, it is known that mirror image suppresses directly can just calculate by using equation (4.15).
Accurate measuring technique
This part is described for accurately and quickly measuring value, phase, leakage, gain imbalance and the crooked side of phase
Method.Except measurement quality and speed, this method is additionally aided to be realized for the even more big FPGA accelerated that calculates.
This method is the S/R method then measured in input injection known signal in output.It is specific and
Speech, stimulation is pure complex exponential, and its frequency is equal to the frequency location for being used for expecting measurement.In some embodiments, this complex exponential
It is to be generated by calibration synthesizer or by the transmitter for being circulated back to receiver.For each frequency of complex exponential, quilt is responded
Digitize and handle, to determine corresponding measurement.The response data that the remainder discussion of this part has been digitized how by
Processing, to provide measurement interested.
When this processing is considered as in time domain, basic design is that each signal is mixed into DC, then using flat
Equal method obtains accurate result.In frequency domain, this can be regarded as the Windowing discrete time Fourier conversion of a small number of single-points
Calculating.This explanation and derivation it will be assumed that (its width is equal to acquisition length using rectangular window before DTFT is calculated
(acquisition length)).Windowing and its effect are discussed in more detail in next part " rectangular window optimization ".
Equation 6.1 describes the expection form of analog response.It is the multiple finger in given frequency f that this form, which is assumed to stimulate,
Number.Therefore the definition of equation 6.3 is not achievable for actual calculating with the DTFT infinitely supported.Equation 6.4 leads to
Cross and provide the DTFT with limited support using rectangular window.Value w represents the digitlization frequency on the standardization of interval [π, π]
Rate.Conversion from f to w is provided by w=2 π f/ sample rates.
The leakage of measurement signal need not offset and need only to be averaging, because its spectrum component position
In 0Hz.In order to measure value and phase to tone, it is multiplied by first by using complex exponential equal and opposite with pitch frequency
Frequency is mixed down to 0Hz plural tone.Then result is averaging in acquisition length.For under frequency interested
Single-point DTFT is taken to complex input signal, this is equally equivalent.
S [n]=ADC_Sampling (s (t, t)) (6.2)
AI=Re (Avg { s [n] exp (- jwn) }) (6.6B)
AQ=Im (Avg { s [n] exp (- jwn) }) (6.6C)
Alternatively, the phase of { s [n] } can be calculated according to following formula：
Calculate that gain is uneven and phase is crooked is related to the value and phase for independently finding out I and Q signal.For example, in figure
In 59, " Q is actual " signal is that (that is, (compare has 0.6 gain imbalance and 20 degree of phases askew to " I references " signal with in-phase signal
Oblique 26MHz signals.But, " Q expectations " track gives preferable orthogonal signalling, and it offsets 90 degree from in-phase signal.Pass through
The value and phase of in-phase component (" I references ") are measured, preferable orthogonal signalling can be by it relative to in-phase component just
The property handed over is determined.Then, by knowing the actual amplitudes and phase of orthogonal signalling (" Q is actual "), ideal quadrature signal and reality
The difference of orthogonal signalling can be determined.
It is illustrated that in Figure 60 and 61 for mutually with quadrature-phase component (that is, for " I references " signal in Figure 59
" Q actual signals ") value.Because complex signal s (t) each component is real-valued signal, therefore it is expected to have pair
The magnitude responses of title.In order to find out gain imbalance g (f), it is determined that the gain of each component of signal in the frequency location of tone,
Then Q-gain is removed with I gains, going out as given in equation 6.12.
Equation 6.8 to 6.11 illustrates how to calculate the value and phase of each composition.It is reason to follow hypothesis in-phase signal
Think and quadrature phase signal includes the convention of all detractions, and the detraction is to refer to and calculate relative to in-phase signal.(its
Its convention is also possible, as described in above in a variety of different ways.For example, orthogonal signalling similarly may select for ginseng
Examine).Therefore, it by finding out single-point DTFT is that each in I signal and Q signal is calculated that value and phase, which are,.Then, these
Value and phase are grouped together by equation 6.12 and 6.13, to determine the gain imbalance and phase of quadrature signal component
It is crooked.
In below equation, I (n, w) is I (t, w) sampled version, and Q (n, w) is Q (t, w) sampled version.
‖ I (w) ‖=| Avg { I (n, w) exp (- jwn) } | (6.8)
‖ Q (w) ‖=| Avg { Q (n, w) exp (- jwn) } | (6.10)
In alternative embodiment, ‖ I (w) ‖, Phase { I (w) }, ‖ Q (w) ‖ and Phase { Q (w) } can be calculated as below：
‖ I (w) ‖=Sum { I (n, w) exp (- jwn) } |/N. (6.8)
‖ Q (w) ‖=| Sum { Q (n, w) exp (- jwn) } |/N (6.10)
Wherein N is collection size.
Figure 62 is illustrated for calculating LO leakages, amplitude, uneven gain, mirror image suppression and the crooked software implementation of phase
Example (being write with LabVIEW graphical programming languages).
In some embodiments, it is to be performed by programmable hardware element (for example, FPGA of receiver) to following calculating
's.
Sum { Re (Q (n, w) exp (- jwn)) }
Sum { Im (Q (n, w) exp (- jwn)) }
Sum{Re(s[n])}
Sum{Im(s[n])}
Figure 63 shows reception by the FPGA summing values calculated and based on those summing values and acquisition length computation LO
Leakage, amplitude, gain imbalance and the crooked LabVIEW graphic packages (VI) of phase.(various computer systems as described herein
In any one can include be used for perform include the software infrastructure of computer program, wherein computer program for example
LabVIEW graphic packages).
Rectangular window optimizes
In some embodiments, non-rectangle window may apply to complex digital signal { s (n) }.Various normal window classes
Any one in type can be used.In other embodiments, no window is clearly applied to complex digital signal.But,
By only performing calculating to limited collection interval, impliedly using rectangular window.If our setting in frequency spectrum to tone
Put using frequency planning and constrain or judge that the measurement error calculated is acceptable, then no window needs clearly to be applied to
Complex digital signal.(thus, we can avoid the memory needed for memory window value, so that hardware utilizes minimum).
Otherwise, window should be used for measuring.This part will be discussed：, the derivation that constrains frequency planning and when without using window
If mouthful when without using affined frequency planning by the measurement error of generation.
The following is the derivation (that is, without clear and definite window) to rectangular window.Just for the sake of reference, equation 5.9 is to use
The closed-form solution for finite geometry series is provided in standard DTFT equation and equation 5.12.Rectangular window is in finite interval
It is defined as one and is zero in other places.Therefore, its DTFT is provided by 5.11.Utilize the geometric identies combining of equation 5.12, window
The DTFT of mouth can be simplified to equation 5.13.Finally, due to the Section 1 of 5.13 last expression formula has unit value,
Number amplitude is provided by equation 5.14 when therefore.
It should be noted that for pure pitch, the null value (null) in the tone that opens a window will occur in Ftone+/- N*
SampleRate/AcqLength, Ftone are pitch frequencies, AcqLength be complex digital signal collection in sample
Number, and the speed that the sample that SampleRate is complex digital signal is collected.It is furthermore noted that suppressing to calculate for mirror image, such as
Really we firmly believe that the tone of all generations exists only in SampleRate/AcqLength multiple, then will not have in the measurements
Any spectrum leakage.
Figure 64-65 shows the amplitude frequency spectrum with public sample rate 120MHz and different acquisition length | RECT (w) |
Two corresponding figures.First figure (Figure 64) corresponds to acquisition length 20.Second figure (Figure 65) corresponds to acquisition length 128.
It is general to derive
Given Figure 66 system model, we can be according to input I/Q detractions gin(ω) andAnd output I/Q
Detract gout(ω) andFor frequency response U (ω) and V (ω) derivation function form.In addition, we can be according to frequency
Respond U (ω) and V (ω) and input I/Q detractions derive output detraction.The two derivations all rely on following preliminary step.System
Model is implied：
Wherein u (t) and v (t) is the impulse response for corresponding respectively to U (ω) and V (ω).
Using the standard identities for cosine and SIN function, we obtain：
Provide following two in the middle coefficients for collecting item of exp (j ω t) and discretely in the coefficient of exp (- j ω t) middle term
Equation：
But, equation (7.8a) is applied to all ω.Therefore, we can replace ω with-ω, and obtain：
Equation (7.7) and (7.8b) define unknown vector [] U (ω), V (ω)]TIn 2x2 matrix equalities, its solution is by scheming
Equation (7.9) and (7.10) in 67 are provided.
Now, input detraction and wave filter U (ω) and V (ω) frequency response are given, we derive output detraction.Root
According to equation (7.7) and (7.8a) it can be seen that it is impossible to calculate output detraction, because problem is by overdetermination.But, due to U
(ω) and V (ω) they are real number value filters, therefore there is direct relation between their positive and negative frequency response, i.e. U (-
F)=U*(f) with V (- f)=V*(f).Therefore,
Receiver detraction is removed from the output detraction measured
In in this section, output detraction g is givenout(f) andAnd the intrinsic detraction g of systemsys(f) andWe derive detracts g for the input of computing systemin(f) andMethod.This method can for from
The detraction measured in the output (for example, output of i/q demodulator) of receiver removes the intrinsic detraction of receiver, to determine
In the detraction of the input (for example, input of i/q demodulator) of receiver.Give the frequency response U of the system model for Figure 66
(f) with V (f) and output detraction gout(f) andWe can calculate input detraction g since equation (7.7)in(f)
Withω is replaced to copy with f on frequency herein：
If we define
Then equation (7.14) can be expressed as more tersely：
Zin(f)={-jU (f)+Zout(f)}/V(f). (7.17)
We can be by using gin(f) be constantly equal to one,Identically vanishing, gout(f) gain equal to system is uneven
Weigh gsys(f) andPhase equal to system is crookedSpecific hypothesis from Figure 67 equation (7.9) and
(7.10) U (f) and V (f) are determined.Under these specific hypothesis, equation (7.9) and (7.10) are exclusively used in：
If we define
Then equation (7.15) and (7.16) can be expressed as：
U (f)=(j/2) { Zsys(-f)*-Zsys(f)} (7.21)
V (f)=(1/2) { Zsys(f)+Zsys(-f)*}. (7.22)
By the way that these expression formulas are substituted into equation (7.17), we obtain：
This computational methods as defined in equation (7.23) to (7.25) can for from receiver output (for example,
The output of i/q demodulator) the detraction g that measuresM(f) andRemove the intrinsic detraction g of receiverRX(f) andWith
Just the detraction g in the input (for example, input of i/q demodulator) of receiver is obtained as belowin(f) and
Additional embodiment is disclosed in the paragraph of following numbering.
1. a kind of method for operating receiver, this method includes：
Analog input signal is received from communication media；
I/Q demodulation is performed to analog input signal, to produce analog in-phase signal and analogue orthogonal signal；
Analog in-phase signal and analogue orthogonal signal are digitized, to produce digital inphase signal I (n) sums respectively
Word orthogonal signalling Q (n)；
Digital inphase signal I (n) and digital quadrature signal Q (n) are converted according to following formula, it is same to produce resulting number
Phase signals IR(n) with resulting number orthogonal signalling QR(n)
IR(n)=I (n),
QR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
Wherein HT represents that Hilbert is converted, wherein, design factor a, b, c and d receive device to achieve a butt joint in frequency f and-f
Under I/Q detraction at least part compensation, subtract wherein each coefficient is all based on the I/Q that measures of the receiver under frequency f
Damage and the I/Q that measures of the receiver under frequency-f detracts to calculate.
The method of 1B. paragraphs 1, wherein, it is used as the alternative arrangement to being given above expression formula, resulting number in-phase signal
IR(n) with resulting number orthogonal signalling QR(n) converted according to following formula：
IR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
QR(n)=Q (n)
2. the method for paragraph 1, wherein analog input signal are pure pitches.
3. the method for paragraph 1, wherein analog input signal are the signals of communication for carrying binary message stream.
4. a kind of receiver, including：
I/q demodulator, is configured to receive analog input signal, and I/Q demodulation is performed to analog input signal, to produce
Analog in-phase signal and analogue orthogonal signal；
Digital unit, is configured to be digitized analog in-phase signal and analogue orthogonal signal, to produce number respectively
Word in-phase signal I (n) and digital quadrature signal Q (n)；
Digital circuit, is configured to convert digital inphase signal I (n) and digital quadrature signal Q (n) according to following formula,
To produce resulting number in-phase signal IR(n) with resulting number orthogonal signalling OR(n)：
IR(n)=I (n),
QR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
Wherein HT represents that Hilbert is converted, wherein, design factor a, b, c and d, with least part compensated receiver in frequency
I/Q detractions under rate f and-f, wherein each coefficient is all based on the I/Q that measures detraction and reception of the receiver under frequency f
The I/Q that measures of the device under frequency-f detracts to calculate.
The receiver of 4B. paragraphs 4, wherein, it is used as the alternative arrangement to being given above expression formula, the same phase of resulting number
Signal IR(n) with resulting number orthogonal signalling QR(n) converted according to following formula：
IR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
QR(n)=Q (n)
5. the receiver of paragraph 4, wherein analog input signal are pure pitches.
6. the receiver of paragraph 4, wherein analog input signal are the signals of communication for carrying binary message stream.
7. a kind of method for operating transmitter, this method includes：
Receive digital inphase signal I (n) and digital quadrature signal Q (n)；
Digital inphase signal I (n) and digital quadrature signal Q (n) are converted according to following formula, it is same to obtain resulting number
Phase signals IR(n) with resulting number orthogonal signalling QR(n)：
IR(n)=I (n),
QR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
Wherein HT represents that Hilbert is converted, wherein, design factor a, b, c and d receive and send device at least partly to pre-compensate for
I/Q detractions under frequency f and-f, wherein each coefficient is all based on estimation and the hair of I/Q detraction of the transmitter under frequency f
The estimation of I/Q detraction of the device under frequency-f is sent to calculate；
Resulting number in-phase signal IR(n) with resulting number orthogonal signalling QR(n) analog form is converted into, so as to respectively
Obtain simulation I signal and simulation Q signal；
I/Q modulation is performed to simulation I signal and simulation Q signal, to produce the analog signal modulated.
The method of 7B. paragraphs 7, wherein, it is used as the alternative arrangement to being given above expression formula, resulting number in-phase signal
IR(n) with resulting number orthogonal signalling QR(n) converted according to following formula：
IR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
QR(n)=Q (n)
8. the method for paragraph 7, wherein digital inphase signal and digital quadrature signal represent the complex exponential sound under frequency f
Adjust.
9. the method for paragraph 7, wherein digital inphase signal and digital quadrature signal carry corresponding binary message stream.
10. a kind of transmitter, including：
Digital circuit, is configured to receive digital inphase signal I (n) and digital quadrature signal Q (n), and according to following table
Up to formula conversion digital inphase signal I (n) and digital quadrature signal Q (n), to obtain resulting number in-phase signal IRAnd result (n)
Digital quadrature signal QR(n)：
IR(n)=I (n),
QR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
Wherein HT represents that Hilbert is converted, wherein, design factor a, b, c and d receive and send device at least partly to pre-compensate for
I/Q detractions under frequency f and-f, wherein each coefficient is all based on estimation and the hair of I/Q detraction of the transmitter under frequency f
The estimation of I/Q detraction of the device under frequency-f is sent to calculate；
Digital-to-analogue conversion (DAC) unit, is configured to a resulting number in-phase signal and resulting number orthogonal signalling are converted to mould
Plan form, to obtain simulation I signal and simulation Q signal respectively；
I/Q modulators, are configured to perform I/Q modulation to simulation I signal and simulation Q signal, to produce the simulation modulated
Signal.
The transmitter of 10B. paragraphs 10, wherein, it is used as the alternative arrangement to being given above expression formula, the same phase of resulting number
Signal IR(n) with resulting number orthogonal signalling QR(n) converted according to following formula：
IR(n)=a*I (n)+HT { b*I (n) }+c*Q (n)+HT { d*Q (n) },
QR(n)=Q (n)
11. the transmitter of paragraph 10, wherein digital inphase signal and digital quadrature signal represent the complex exponential under frequency f
Tone.
12. the transmitter of paragraph 10, wherein digital inphase signal and digital quadrature signal carry corresponding binary message
Stream.
Also additional embodiment is disclosed in the paragraph of following numbering.
1. a kind of method for being used to correct the I/Q detractions in the transmission signal received, this method includes：Through transmission medium
Receive transmission signal；I/Q demodulation is performed to the transmission signal received, to produce simulation I (same to phase) and Q (orthogonal) signal；It is right
Simulate each in I signal and simulation Q signal and perform analog-to-digital conversion, to produce digital I and Q signal；And digital I and Q is believed
Number perform broadband I/Q detraction correction, wherein the broadband I/Q detraction rectification building-out numeral I and Q signal in gain imbalance and phase
The unbalanced frequency dependence change in position.
2. the method for paragraph 1, wherein the broadband I/Q detracts rectification building-out because I/Q is demodulated or simulates I signal and simulation
Uneven and unbalance in phase the frequency phase of gain in the one or more digital I caused and Q signal in the analog-to-digital conversion of Q signal
Close change.
3. the method for paragraph 1, wherein this method are realized by receiving device, wherein broadband I/Q detraction corrections exist
Bridge gain in the multiple frequency offsets compensation numeral I and Q signal of receiving unit instant bandwidth uneven and unbalance in phase
Frequency dependence changes.
4. the method for paragraph 1, wherein I/Q detraction corrections in broadband are performed to digital I and Q signal includes filtering figure I signal
One or more of or digital Q signal.
5. the method for paragraph 4, wherein I/Q detraction corrections in broadband are performed to digital I and Q signal includes filtering figure Q signal
And it is constant to leave digital iota signal.
6. the method for paragraph 4, wherein I/Q detraction corrections in broadband are performed to digital I and Q signal includes filtering figure I signal
And it is constant to leave digital Q signal.
7. the method for paragraph 4, wherein digital I and Q signal are performed broadband I/Q detraction corrections include digital Q signal with
Digital iota signal is all filtered.
8. the method for paragraph 1, wherein this method are realized by receiving device, wherein this method is also included by reception
Equipment, which provides multiple known test signals and measures the I/Q introduced by receiving device in response to known test signal, to be subtracted
Damage to determine control information, wherein broadband I/Q detraction corrections compensate gain in digital I and Q signal using control information
The frequency dependence of uneven and unbalance in phase changes.
9. the method for paragraph 8, wherein to receiving device provide multiple known test signals include providing with next or
It is multiple：Multiple sine waves in different frequency；Or multiple cosine waves in different frequency.
10. the method for paragraph 1, wherein receiving transmission signal through communication media includes transmitting through following one or more receive
Signal：Wireless communication medium；Or cable.
11. the method for paragraph 1, wherein the transmission signal received is radio frequency (RF) signal.
12. a receiving device, is configured to：Transmission signal is received through transmission medium；The transmission signal received is performed
I/Q is demodulated, to produce simulation I (same to phase) and Q (orthogonal) signal；Modulus is performed to each in simulation I signal and simulation Q signal
Conversion, to produce digital I and Q signal；And perform broadband I/Q detractions to digital I and Q signal to correct, wherein the broadband I/Q
Uneven and unbalance in phase the frequency dependence of gain in rectification building-out numeral I and Q signal is detracted to change.
13. the receiving device of paragraph 12, wherein receiving device include：One or more inputs for receiving transmission signal
Port；One or more one or more outputs in digital iota signal for output calibration or the digital Q signal corrected
Port；And it is configured to perform the programmable hardware element of broadband I/Q detraction corrections.
14. the receiving device of paragraph 13, wherein programmable hardware element include FPGA (field programmable gate array).
19. a kind of method for being used to correct I/Q detractions, this method includes：Receive the digital I to be sent (same to phase) and Q is (just
Hand over) signal；I/Q detraction precorrection in broadband is performed to digital iota signal and digital Q signal, wherein performing broadband I/Q detraction precorrection
Including one or more of filtering figure I signal and digital Q signal, to produce the data signal of one or more precorrection,
Then uneven and unbalance in phase the frequency dependence of the gain introduced in the building-up process of transmission signal is become to pre-compensate for
Change；And utilize the digital signal synthesis transmission signal of one or more precorrection.
20. the method for paragraph 19, wherein broadband I/Q detraction precorrection filtering figure Q signals are performed, to produce precorrection
Digital Q signal, and it is constant to leave digital iota signal；Wherein transmission signal is the digital Q signal and constant numeral from precorrection
I signal synthesis.
21. the method for paragraph 19, wherein broadband I/Q detraction precorrection filtering figure I signals are performed, to produce precorrection
Digital iota signal, and it is constant to leave digital Q signal；Wherein transmission signal is the digital iota signal and constant numeral from precorrection
Q signal synthesis.
22. the method for paragraph 19, wherein broadband I/Q detraction precorrection filtering figure I signals are performed, to produce precorrection
Digital iota signal, and filtering figure Q signal, to produce the digital Q signal of precorrection；Wherein transmission signal is from precorrection
What digital iota signal and the digital Q signal of precorrection were synthesized.
23. the method for paragraph 19, wherein synthesis transmission signal includes：Perform the data signal of one or more precorrection
Digital-to-analogue conversion, to produce one or more of simulation I signal or simulation Q signal；And believed using simulating I signal or simulating Q
Number one or more of perform I/Q modulation, to produce transmission signal；The data signal of wherein one or more precorrection is mended in advance
Uneven and unbalance in phase the frequency dependence of one or more gains caused in digital-to-analogue conversion or I/Q modulation is repaid to change.
24. the method for paragraph 23, wherein the digital-to-analogue conversion for performing the data signal of one or more precorrection produces simulation Q
Signal；Wherein this method also includes the digital-to-analogue conversion for performing digital iota signal, to produce simulation I signal；Wherein perform I/Q modulation
Simulation Q signal and simulation I signal are used to produce transmission signal.
25. the method for paragraph 19, wherein this method are realized by transmission equipment；Wherein described broadband I/Q detracts pre- school
Gain imbalance and unbalance in phase of the positive precompensation in multiple frequency offsets across the instant bandwidth for sending equipment.
26. the method for paragraph 19, wherein this method are realized by transmission equipment；Wherein this method is also included by hair
Equipment is sent to provide multiple known test signals and measure the I/Q introduced by transmission equipment in response to known test signal
Detract to determine control information；Wherein described broadband I/Q detractions precorrection is produced one or more pre- using the control information
The data signal of correction.
27. the method for paragraph 26, wherein providing multiple known test signals including providing with next to equipment is sent
Or it is multiple：Multiple sine waves in different frequency；Or multiple cosine waves in different frequency.
28. the method for paragraph 19, in addition to transmit signal through following one or more send：Wireless communication medium；Or electricity
Cable.
29. the method for paragraph 19, wherein transmission signal is radio frequency (RF) signal.
30. one kind sends equipment, it is configured to：Receive the digital I to be sent (same to phase) and Q (orthogonal) signal；Digital I is believed
Number and digital Q signal perform broadband I/Q detraction precorrection, wherein perform broadband I/Q detraction precorrection filtering figure I signal sum
One or more of word Q signal, to produce the data signal of one or more precorrection, will then be believed with pre-compensating in transmission
Number building-up process in uneven and unbalance in phase the frequency dependence of the gain that introduces change；And utilize one or more
The digital signal synthesis transmission signal of precorrection.
31. the transmission equipment of paragraph 30, wherein sending equipment includes：One for receiving digital iota signal and digital Q signal
Individual or multiple input ports；One or more output ports for exporting transmission signal；And be configured to digital iota signal and
Digital Q signal performs the programmable hardware element that broadband I/Q detracts precorrection.
32. the transmission equipment of paragraph 31, wherein programmable hardware element include FPGA (field programmable gate array).
34. a kind of measuring system, including：Receiving device；And equipment under test；Wherein receiving device is configured to：Reception includes
The transmission signal of the measurement data collected from equipment under test；I/Q demodulation is performed to the transmission signal received, to produce simulation
I (same to phase) and Q (orthogonal) signal；The analog-to-digital conversion of each in simulation I signal and simulation Q signal is performed, is believed with producing digital I
Number and digital Q signal, wherein gain is uneven in broadband I/Q detraction rectification building-out digital iota signals and digital Q signal and phase
The unbalanced frequency dependence change in position.
35. the measuring system of paragraph 34, in addition to：Equipment is sent, the wherein transmission device configuration is：Receive what is sent
Digital iota signal and digital Q signal, wherein digital iota signal and digital Q signal provide the information of equipment under test to be sent to；Logarithm
Word I signal and digital Q signal perform broadband I/Q detraction precorrection, wherein performing broadband I/Q detraction precorrection filtering figure I letters
Number and one or more of digital Q signal, to produce the data signal of one or more precorrection, so that pre-compensate for then will be
Uneven and unbalance in phase the frequency dependence of the gain introduced in the building-up process for transmitting signal changes；Using one or more
The digital signal synthesis transmission signal of precorrection；And transmission signal is sent to equipment under test.
36. the measuring system of paragraph 35, wherein transmission signal includes the control signal for being used to control equipment under test.
37. the measuring system of paragraph 34, in addition to：Cabinet；Wherein receiving device is embodied as installing first in the chassis
Module；Wherein transmission equipment is embodied as installing the second module in the chassis.
38. the measuring system of paragraph 37, wherein cabinet are PXI (PCI for being used for instrumentation extends) cabinets.
Figure 68 is illustrated can be for performing any means embodiment described herein or methods described herein embodiment
A kind of embodiment of the computer system 6800 of the random subset of any combination or any means embodiment described herein.
Computer system 6800 can include processing unit 6810, system storage 6812, one or more storage devices
Set 6815, communication bus 6820, the set 6825 of input equipment and display system 6830.
System storage 6812 can include partly leading for one group such as RAM device (and there may also be one group of ROM device)
Body equipment.
Storage device 6815 can include any one in various storage devices, such as one or more storage mediums
And/or memory access equipment.Driven for example, storage device 6815 can include such as CD/DVD-ROM drivers, hard disk, disk
The equipment such as dynamic device, tape drive.
Processing unit 6810 be configured to read and execute program instructions, be for example stored in system storage 6812 and/or
Programmed instruction in one or more storage devices 6815.Processing unit 6810 (or can be passed through by communication bus 6820
The system of interconnection bus, or by network) it is coupled to system storage 6812.Programmed instruction allocating computer system 6800 is real
Existing method, for example, any combination of any means embodiment described herein or methods described herein embodiment, or herein
The random subset of any means embodiment, or this subset any combination.
Processing unit 6810 can include one or more processors (for example, microprocessor).
One or more users can by input equipment 6825 to computer system 6800 provide input.Input equipment
6825 can include such as keyboard, mouse, touch sensitive mat, touching sensitive screen curtain, drawing board, trace ball, light pen, data hand
Set, eyes towards and/or head towards sensor, the equipment of microphone (or set of microphone) or its any combination.
It is any one in a variety of display devices that display system 6830 can be including representing any one in various Display
Kind.For example, display system can be computer monitor, head mounted display, projecting apparatus system, three-dimensional display, Huo Zheqi
Combination.In some embodiments, display system can include multiple display devices.In one embodiment, display system can be with
Including printer and/or plotter.
In some embodiments, computer system 6800 can include miscellaneous equipment, for example, such as one or more figures
Shape accelerator, one or more loudspeakers, sound card, video camera and video card, the equipment of data collecting system.
In some embodiments, computer system 6800 can include one or more communication equipments 6835, for example, being used for
With the NIC of computer network interface.As another example, communication equipment 6835 can include being used for through it is a variety of
It is special that any one in the communication standard or agreement (for example, USB, Firewire, PCI, PCI Express, PXI) of establishment communicates
Use interface.
Computer system, which can be utilized, to be included operating system and may also have one or more figure API (such asDirect3D、Java 3DTM) software infrastructure configure.In some embodiments, basis of software is set
The LabVIEW of National Instruments (National Instruments) can be included by applyingTMSoftware, and/or
LabVIEWTNFPGA。
In some embodiments, computer system 6800 is configurable to contact with transmitter 6840.Transmitter can match somebody with somebody
Transmission signal (onto communication channel) is set to, as described in herein in a variety of different ways.Transmitter can be in processor
The software performed on 6810 and/or the operation under the control for the software that transmitter is performed with.
In some embodiments, computer system 6800 is configurable to contact with receiver 6850.Receiver can match somebody with somebody
It is set to (from communication channel) and receives signal, as described in herein in a variety of different ways.Receiver can be in processor 6810
The software of upper execution and/or the operation under the control for the software that receiver is performed with.
In some embodiments, transmitter and/or receiver can include one or more programmable hardware elements and/or
One or more microprocessors, for performing numeral to numerical data (for example, to digital baseband signal or digital IF signal)
Processing, as described in herein in a variety of different ways.
Although embodiment has had been described in considerable detail above, once disclosure above is understanding of completely, respectively
Plant and change and modifications and those skilled in the art will be become apparent.Following claims should be construed to cover all these
Change and modifications.
Appendix A
The alternative manner of transmitter I/Q detractions is estimated using shared LO
1. pair will its measure transmitter gain imbalance gT and phase it is crookedEach band bias internal amount frequency
F, the gain imbalance gR and phase of measuring receiver is crooked(in some embodiments, this class frequency offset is on zero
Symmetrically, i.e. for each frequency offset f in set, frequency offset-f is also in the set).For each f, guide
The generation of tone transmitter is in frequency v=fLOTone under+f, wherein fLOIt is LO frequencies, the tone is applied to receiver
Input, and the output of the i/q demodulator in receiver catches complex baseband sequence z (n).Gain imbalance gR and phase are crookedIt is to be calculated based on complex baseband sequence z (n), as described in " accurate measuring technique " part.
2. configure receiver and transmitter so that they use identical LO frequencies fLO.If receiver and transmitter make
With two different LO circuits, then transmitter is tuned so that its LO is phase-locked to identical reference.Therefore, the frequency of transmitter and
The frequency of receiver is all fLO。
3. the output of a transmitter is connected to the input of receiver, for example, through cable or wireless connection.
4. estimate the I/Q modulators of transmitter by using the algorithm in part " calculating the mapping between RX and TX " and connect
The DC scaling m (0) and DC rotation θ (0) of signal path between the i/q demodulator of receipts device.For best result, except DC test to
Outside amount, also to the I/Q modulator applications tones K of transmitter.It is because leakage can be sensitive to band internal power using tone K.Sound
Some the frequency application for adjusting K different from DC in instant bandwidth.(part with the DC estimations rotated is scaled as DC, it is " accurate
The method of e measurement technology " part is applied to the complex data of sampling.If the complex data of sampling does not open a window, to tone K's
Frequency places Constrained.)
5. iteration index k ← 0
Do while (mass measurement Q is less than threshold value)
The each frequency offset f of For：
Set gT (f, 0) ← 0 and
6A.If k=0：
Precorrection is not applied in transmitter, i.e. preemphasis circuit system use value α=0 and β=1 of configuration transmitter
Else (k ＞ 0)
Based on below to frequency offset f calculating precorrection factor alphas and β：Current transmitter gain imbalance estimation gT
(f, k)；The crooked estimation of current transmitter phaseThe uneven estimation gT of current transmitter gain (- f, k)；When
The crooked estimation of preceding transmitter phase(if frequency shift (FS) duration set is asymmetric on zero, to gT (- f, k)
WithFrequency under the closest-f of selection).Alternatively, transmitter precorrection wave filter can be created.Endif
Preemphasis circuit system is configured, to use the value α and β (or precorrection wave filter) calculated.
Input application complex exponential signal u (n)=exps (j2 π fn) of the 7A. to preemphasis circuit system
Output measurement complex base band signal zs (n) of the 7B. in the i/q demodulator of receiver.
7C. utilizes the computational methods in " accurate measuring technique " to determine that original gain is uneven based on complex base band signal z (n)
Gz (f) and the original phase of weighing are crooked
8. it is crooked from original gain imbalance gz (f) and original phaseRemove the gain imbalance gR (f) of receiver
It is crooked with phaseIt is crooked to obtain pre-demodulating gain imbalance gPD (f) and pre-demodulating phase(for
This removal is performed, there are at least two methods：Direct transform method and filtering method.Direct transform method can have than filtering method
There is higher quality.Direct transform method is begged in the part of entitled " removing receiver detraction from the output detraction measured "
By.Filtering method is related to the 2x2 matrixes to complex base band signal z (n)=(I (n), Q (n)) NEURAL DISCHARGE BY DIGITAL FILTER, to obtain
The signal PCS (n) of partial correction.The 2x2 matrixes of digital filter can as above contact Fig. 2A, 2B and 3 and part " broadband
Calculated as described in I/Q detractions equilibrium ".)
9. remove the optimal current estimation of signal path between the I/Q modulators of transmitter and the i/q demodulator of receiver.
M (0) and θ (0) will provide basic estimation.Preferably estimation will improve rate of convergence.For example, step 9 can be implemented as described below.
If k=0
Utilize the conversion described in " it is uneven crooked with phase to change gain by linear system " uneven from gain
GPD and phase are crookedRemove the DC scaling m (0) estimated and DC rotates 0 (0), it is uneven with modulation gain after acquisition
GPM and rear phase modulation are crookedH (f) and H (- f) is set to be equal to H (0)=exp (- j0 (0))/m (0)
Else (k ＞ 0)
The scaling m (f) being under frequency offset f is calculated based on complex base band signal z (n).Scaling m (f) can pass through
Calculate the value of the frequency content under frequency f in complex signal z (n) under f to determine, such as in " accurate measuring technique " part
In, especially in equation 6.6, explanation.
By using the conversion described in " changed by linear system gain is uneven and phase is crooked " from gain not
Balance gPD (f) and phase is crookedThe linear signal path of estimation is removed, it is uneven with the gain modulated after acquisition
GPM (f) and the phase modulated afterwards are crookedWherein H (f)=exp (- j θ (0))/m (f) and H (- f)=exp (- j
θ(0))/m(-f)
Note：If-f is not seen also by frequency offset circulation, just using in calculating in previous inferior quality iteration k-1
M (- f).
10. it is that transmitter gain imbalance gT and transmitter phase are crooked according to following formulaGeneration updates：
GT (f, k+1) ← gT (f, k) * gPM (f) and
11. it is crooked from the gain imbalance gPM (f) of rear modulation and the phase modulated afterwards based on equation (4.15)
Calculate mirror image and suppress IR(f)。
Endfor
k←k+l
Mass measurement Q=-I is calculated to f all valuesR(f) maximum.(IR(f) more negative value corresponds to higher matter
Amount.Thus, IR(f) negative corresponds to the quality under frequency f.Q is the maximum of the quality on frequency band.)
End While
Appendix B
Iterative estimate-the optimization detracted using the transmitter for offseting LO
1. configure receiver and transmitter so that the local oscillator frequencies LO of receiverRXWith the local oscillations of transmitter
Device frequency LOTXDifference be equal to selected value Δ LO：
LORX-LOTX=Δ LO.
The selected value is the non-zero (for example, sub-fraction) of the instant bandwidth of transmitter.Two local oscillators
It is lock phase.
2. the output of a transmitter is connected to the input of receiver.
3. the I/Q modulators and receiver of transmitter are estimated using the algorithm in part " calculating the mapping between RX and TX "
I/q demodulator between signal path DC scaling m (0) and DC rotate θ (0).This estimation is involved the steps of.
3A. is applied to zero stimulus signal as input the I/Q modulators of transmitter.
3B. catches response signal z in the output of the i/q demodulator of receiverA(n)。
3C. frequency displacement response signals zA(n) Δ LO is measured, to obtain the signal FSz after frequency displacementA(n)。
DC test vectors are applied to i/q demodulator by 3D. as input.
3E. catches response signal z in the output of i/q demodulatorC(n)。
3F. frequency displacement response signals zB(n) Δ LO is measured, to obtain frequency shift signal FSzB(n)。
3G. is based on frequency shift signal FSzA(n), frequency shift signal FSzB(n) DC scaling m (0) and DC are calculated and are revolved with DC test vectors
Turn θ (0), as described in part " calculating the mapping between RX and TX ".
For best result, in addition to DC test vectors, also to the I/Q modulator applications tones K of transmitter.Using sound
It is because leakage can be sensitive to band internal power to adjust K.Some frequency application different from DC in instant bandwidth tone K.
Note：Frequency displacement operation can be performed using signal FS (n), the phase of this signal be in time it is continuous and
Advanced with speed Δ LO.For example, FS (n) can have form：
FS (n)=exp { j2 π (Δ LO/ADC_SampleRate) n }
Frequency displacement operation can be according to following relational implementation：
FSz (n)=z (n) FS (n),
Wherein z (n) is the signal for wanting frequency displacement.
In one embodiment, frequency displacement operation can be realized in the FPGA of receiver.Frequency displacement operation can be by receiver
ADC sampling rate perform, i.e. can be the new output valve FSz (n) of each new adc data vector z (n) generations.Cause
And, ADC sampling clocks can be fed as input to FPGA.Then, signal FS phase continuity is by by ADC sampling clocks
Phase continuity ensures.ADC sampling clocks are phase-locked to local oscillator.
In alternative embodiment, frequency displacement operation can be performed in software.Given alternative manner is related to signal z (n)
From the repeated acquisition of i/q demodulator.Thus, in order to realize signal FS phase continuity, provided for software on this collection
Beginning and for the first time gather beginning (or preceding beginning once gathered) between time difference.For example, can be carried for software
For the time of the time of first sample z (0) of first sample z (0) relative to first time collection of this collection.M is made to determine
Justice is the sample counting that the sample counting and n continuously run is this collection.Thus, gathered for z (n) first time, m=
0 corresponds to n=0.Then, the frequency shift signal FSz (m) of Phase Continuation can be expressed as：
FS (m)=exp { j2 π (Δ LO/ADC_SampleRate) m }
Sampled distances of the k between the current collection and first time collection for first sample z (0) is made to define.In
It is
FS (m)=FS (k+n)=exp { j2 π (Δ LO/ADC_SampleRate) (k+n) }
Now, FSz (n) can be calculated from following formula
FSz (n)=FS (k+n) z (n)=FS (n) z (n) FSOffset,
Wherein
FS (n)=exp { j2 π t (Δ LO/ADC_SampleRate) n }
FSOffset=FS (k)=exp { j2 π (Δ LO/ADC_SampleRate) k }
Note, k generals change from collection next time is once collected.
For each positive pitch frequency offset f=Δs f to N Δs f, (stepping is Δ f), receives the institute in " constraint " part
The constraint of description.
k←0
S elements in For { 1, -1 }
Do while (- the Image_Rejection being used under pitch frequency offset v=S*f is less than threshold value)：
4. at least calculated based on the optimal available estimation for the transmitter detraction in frequency v
It is as follows for the α and beta coefficient of preemphasis circuit system：
If f=Δs f
If k=0
If S=1：
Set gT (v, 0) ← 1 and
The factor alpha and β of precorrection are set, to realize identical mapping (that is, directly straight-through)：α ← 0 and β ← 1
If S=-1：
GT (v, 0) ← gT (- v, ∞)
In general, notation gT (x, ∞) andRepresent respectively from the last k times of the preceding frequency x once accessed
GT that iteration is drawn andConvergent estimation.
Based on gT (v, 0) andFor traditional single-point compensation calculation α and β Else k ＞ 0
If S=1：Based on gT (v, k) andFor traditional single-point compensation calculation α and β
If S=-1：Based on gT (v, k) andGT (- v, ∞) andFor real single point correction meter
Calculate α and β
End If
Else (f ＞ Δs f)
If k=0
GT (v, 0) ← gT (v-S* Δs f, ∞)
Real single point correction calculating α and β is estimated as based on optimal can use that the transmitter in v and-v is detracted, for example, such as
Under.
If S=1：Based on gT (v- Δs f, ∞),GT (- v+ Δs f, ∞)！、
α and β is calculated for real single point correction
If S=-1：It is based on GT (- v, ∞)！、 To be real
Single point correction calculates α and β
Else k ＞ 0
If S=1, based on gT (v, k)！、GT (- v+ Δs f, ∞),For really list
Point calibration calculates α and β
If S=-1, based on gT (v, k)！、GT (- v, ∞)！、 For real single point correction
Calculate α and β
End If
5. preemphasis circuit system is configured, to use the value α and β calculated
6. input application complex exponential signal u (n)=exp (j2 π vn) of pair preemphasis circuit system
Output measurement complex base band signal zs (n) of the 7A. in the i/q demodulator of receiver
7B. the step is optional.
The I/Q detractions of receiver are removed from complex base band signal z (n), to obtain amended complex signal.For example, this
Removal can relate to the use of the 2x2 matrix filtered complex baseband signals of digital filter, or, it is multiplied by with 2x2 scalar matrixes multiple
Base band signal, as more than described in part " detracted and determined using the transmitter I/Q for offseting LO ".
7C. is to signal z (n) application Phase Continuations, frequency shift (FS) (as described above) equal to Δ LO, to obtain frequency displacement
Signal FSz (n).If having been carried out step 7B, frequency displacement is applied to amended complex signal.
8. utilize the computational methods described in " accurate measuring technique " part to be determined based on complex base band signal FSz (n)
Original gain imbalance gFSz (v) and original phase are crooked
9. it is crooked from original gain imbalance gFSz (v) and original phaseRemove (the I/Q modulation of transmitter
Between device and the i/q demodulator of receiver) the optimal current estimation of signal path, it is uneven with the rear modulation gain for obtaining estimation
GPM (v) and rear phase modulation are crookedM (0) and θ (0) will provide the basic estimation of signal path.Preferably estimation
Rate of convergence will be improved.For example, step 9 can be implemented as described below.
If f=Δs f
Utilize the conversion described in " changed by linear system gain is uneven and phase is crooked " from original gain not
Balance gFSz (v) and original phase is crookedThe DC scaling m (0) and DC rotation θ (0) of estimation are removed, to be estimated
The rear modulation gain imbalance gPM (v) and rear phase modulation of meter are crookedSo that H
(v)=exp (- j θ (0))/m (0) and H (- v)=exp (- j θ (0))/m (0).
Else f ＞ Δs f
Signal FSz (n) based on step 7C calculates the scaling m (v) in tone v.Scaling m (v) can be by frequency v
Calculate the value of the frequency content in complex signal FSz (n) to determine, such as in " accurate measuring technique ", especially in equation 6.6
In, explained.
(note：In alternative embodiment, z (n) measurement is synchronous with tone t (n) generation, for example, by using in hair
The flop signal shared between device and receiver is sent, for example, the trigger generated by controller.In this case, except contracting
Put outside m (v), rotation θ (v) can also be measured.)
By using the conversion described in " changed by linear system gain is uneven and phase is crooked " from original
Gain imbalance gFSz (v) and original phase are crookedThe linear signal of estimation is removed, to obtain the rear modulation of estimation
Gain imbalance gPM (v) and rear phase modulation are crookedSo that H (v)=exp (- j θ (0))/m (v) and H (-
V)=exp (- j θ (0))/mBAE(- v), wherein mBAE(- v) is the optimal available estimation for being used to scale m (- v).
If S=1：mBAE(- v)=m (- v+ Δs f, ∞)
If S=-1：mBAE(- v)=m (- v, ∞)
In general, notation m (x, ∞) is represented in being calculated in the preceding frequency x once accessed last k iteration
Scale m (x).
10. it is that transmitter gain imbalance gT and transmitter phase are crooked according to following formulaGeneration updates：
GT (v, k+1) ← gT (v, k) * gPM (v) and
11. it is crooked from rear modulation gain imbalance gPM (v) and rear phase modulation based on equation 4.15Calculate mirror
As suppressing IR (v).
k←k+1
EndDo
S elements in EndFor { 1, -1 }
Appendix C
Iterative estimate-the optimization detracted using shared LO transmitter
1. it is crooked for the gain imbalance gT and phase that transmitter is measured at itEach band bias internal amount frequency
The gain imbalance gR and phase of measuring receiver are crookedFor each f, tone generator generation is in frequency v=fLO+
Tone under f, wherein fLOBe LO frequencies, tone be applied to the input of receiver, and receiver i/q demodulator it is defeated
Go out to catch complex baseband sequence z (n).Gain imbalance gR and phase are crookedAs retouched in " accurate measuring technique " part
Calculated as stating.
2. configure receiver and transmitter so that they use identical LO frequencies fLO.If receiver and transmitter make
With two different LO circuits, then transmitter is tuned so that its LO is phase-locked to identical reference.Therefore, the frequency of transmitter
Frequency with receiver is all fLO。
3. the output of a transmitter is connected to the input of receiver.
4. the I/Q modulators of transmitter are estimated by using the algorithm in part " calculating the mapping between RX and TX " and connect
The DC scaling m (0) and DC rotation θ (0) of signal path between the i/q demodulator of receipts device.For best result, except DC test to
Outside amount, also to the I/Q modulator applications tones K of transmitter.
(stepping is Δ f) to each positive frequency deviation amount f=Δs f of For to N Δs f
S elements in For { 1, -1 }
k←0
Do while (- the Image_Rejection being used under frequency offset v=S*f is less than threshold value)：
5A. at least based on the optimal available estimation for the transmitter detraction in frequency v, is calculated for preemphasis circuit system
The α and beta coefficient of system.
If f=Δs f
If k=0
If S=1：
Set gT (v, 0) ← 1 andAnd the factor alpha and β of precorrection are set, to realize identical mapping
(that is, directly leading directly to)：α ← 0 and β ← 1
If S=-1：
Setting gt (v, 0) ← gT (- v, ∞),And based on gT (v, 0) andForFor traditional single-point compensation calculation α and β
Else k ＞ 0
If S=1：Based on gT (v, k) andFor traditional single-point compensation calculation α and β
If S=-1：Based on gT (v, k) andGT (- v, ∞) andCompensate and count for real single-point
Calculate α and β
End If
Else (f ＞ Δs f)
If k=0
GT (v, 0) ← gT (v-S* Δs f, ∞)
Real single point correction calculating α and β is estimated as based on optimal can use that the transmitter in v and-v is detracted, for example,
It is as follows.
If S=1：
Based on gT (v- Δs f, ∞),gT(-v+Δf∞)！、 For really list
Point calibration calculates α and β
If S=-1：
It is based on gT(- v, ∞)！、 For real single point correction meter
Calculate α and β
Else k ＞ 0
If S=1：
Based on gT (v, k),GT (- v+ Δs f, ∞),For real single point correction meter
Calculate α and β
If S=-1
Based on gT (v, k)：GT (- v, ∞)！、 α and β is calculated for real single point correction
End If
End If
5B. configures preemphasis circuit system, to use the value α and β calculated.
6. input application complex exponential signal u (n)=exp (j2 π vn) of pair preemphasis circuit system
Output measurement complex base band signal zs (n) of the 7A. in the i/q demodulator of receiver.
7B. determines original gain using the computational methods in part " accurate measuring technique " based on complex base band signal z (n)
Uneven gz (v) and original phase are crooked
8. it is crooked from original gain imbalance gz (v) and original phaseRemove the gain imbalance gR of receiver
(v) it is crooked with phaseThe phase of gain imbalance gPD (v) and pre-demodulating to obtain pre-demodulating are crookedIt there are ways to realize this removal, including mathematic(al) manipulation method and filtering method, such as above continuation method
4400 descriptions.Mathematic(al) manipulation method from the output detraction measured described in part " removing receiver detraction ".
9. remove the optimal but money estimation of signal path between the I/Q modulators of transmitter and the i/q demodulator of receiver.
M (0) and θ (0) will provide best estimate.Preferably estimation will improve convergent speed.For example, step 9 can be implemented as described below.
If f=Δs f：
Utilize the conversion described in " it is uneven crooked with phase to change gain by linear system " uneven from gain
GPD (v) and phase are crookedThe DC scaling m (0) and DC rotation θ (0) of estimation are removed, to obtain the rear modulation of estimation
Gain imbalance gPM (v) and rear phase modulation are crookedWherein H (v) and H (- v) are set equal to exp (- j θ
(0))/m(0).
Else f ＞ Δs f
Complex base band signal z (n) based on step 7A calculates the scaling m (v) in pitch frequency v.Scaling m (v) can be with
Determined by the value of the frequency content in frequency v calculating complex signal z (n), such as in " accurate measuring technique ", especially
In equation 6.6, explained.
(note：In alternative embodiment, z (n) measurement is synchronous with tone t (n) generation, for example, by using in hair
The flop signal shared between device and receiver is sent, for example, the trigger generated by controller.In this case, except contracting
Put outside m (v), rotation θ (v) can also be measured.)
By using the conversion described in " changed by linear system gain is uneven and phase is crooked " from gain
Uneven gPD (v) and phase are crookedThe linear signal path of estimation is removed, to obtain the rear modulation gain of estimation not
Balance gPM (v) and rear phase modulation is crookedWherein H (v)=exp (- j θ (0))/m (v) and H (- v)=
exp(-jθ(0))/mBAE(- v), wherein mBAE(- v) is the optimal available estimation for being used to scale m (- v).
If S=1：mBAE(- v)=m (- v+ Δs f, ∞)
If S=-1：mBAE(- v)=m (- v, ∞)
In general, notation m (x, ∞) is represented in being calculated in the preceding frequency x once accessed last k iteration
Scale m (x).
10. it is that transmitter gain imbalance gT and transmitter phase are crooked according to following formulaGeneration updates：
GT (v, k+1) ← gT (v, k) * gPM (v) and
11. it is crooked from rear modulation gain imbalance gPM (v) and rear phase modulation based on equation 4.15Calculate
Mirror image suppresses IR (v).
k←k+1
EndDo
S elements in EndFor { 1, -1 }
EndFor
Claims (46)
1. a kind of method that I/Q for compensated receiver is detracted, methods described includes：
Receive analog input signal；
I/Q demodulation is performed to analog input signal, to produce analog in-phase (I) signal and simulate orthogonal (Q) signal；
Simulation I signal and simulation Q signal are digitized, to produce digital iota signal and digital Q signal respectively；
Digital iota signal and digital Q signal are filtered according to the 2x2 matrixes of digital filter, to produce the number filtered
Word I signal and the digital Q signal filtered, the 2x2 matrixes of wherein digital filter are compensated in a frequency model at least in part
The I/Q detractions of interior receiver are enclosed, wherein, the frequency response of at least one diagonal components of the 2x2 matrixes is based on as frequency
Function I/Q detractions measurement and calculated as the measurement of I/Q detractions of the function of the negative of frequency, the wherein 2x2
The frequency response of at least one non-diagonal component of matrix is the measurement of the I/Q detractions based on the function as frequency and is used as frequency
What the I/Q of the function of the negative of rate was detracted measures to calculate.
2. the method as described in claim 1, wherein the digital iota signal filtered and the digital Q signal filtered can be used for it is extensive
Complex information bit stream.
3. method as claimed in claim 2, in addition to pass through the digital iota signal to having filtered and the digital Q signal filtered
Symbolic solution is performed to transfer to recover information bit stream.
4. the method as described in claim 1, wherein receiver include Aristogrid, the wherein Aristogrid performs the numeral
Change and the filtering, wherein the relation between the amplitude of simulation I signal and the amplitude of the digital iota signal filtered is calibrated to
The standard known, wherein to be calibrated to this known for the relation between the amplitude of simulation Q signal and the amplitude of the digital Q signal filtered
Standard.
5. the method as described in claim 1, wherein receiver are testers, wherein analog input signal is in response in passing through
Transmitter transmits what is transmitted a signal on communication media and generate, and the measurement of the wherein I/Q detractions of receiver does not include transmitter
I/Q detraction.
6. the method as described in claim 1, wherein the filtering is in programmable hardware element or application specific integrated circuit
(ASIC) performed in.
7. the method as described in claim 1, wherein the filtering is to be performed by processor response in the execution of programmed instruction
's.
8. the method as described in claim 1, wherein the 2x2 matrixes a diagonal components are discrete time unit pulse letters
Number.
9. another non-diagonal component identically vanishing of the method as described in claim 1, wherein the 2x2 matrixes.
10. a kind of method that I/Q for compensated receiver is detracted, this method includes：
Receive analog input signal；
I/Q demodulation is performed to analog input signal, to produce analog in-phase (I) signal and simulate orthogonal (Q) signal；
Simulation I signal and simulation Q signal are digitized, to produce digital iota signal and digital Q signal respectively；
It is filtered according to the 2x2 logm word I signals and digital Q signal of digital filter, to produce the digital I filtered
Signal and the digital Q signal filtered, the 2x2 matrixes of wherein digital filter are compensated in a frequency range at least in part
The I/Q detractions of receiver, wherein, the frequency under the optional frequency f of diagonal components of the 2x2 matrixes in the frequency range
Response is only based on the I/Q measurements detracted under frequency f or the measurement for the I/Q detractions being based only upon under frequency-f to calculate
, wherein frequency response of the non-diagonal component of the 2x2 matrixes under frequency f is only based on the I/Q detractions under frequency f
Measurement or the measurements of the I/Q detractions that are based only upon under frequency-f calculate.
11. method as claimed in claim 10, wherein the I/Q detractions under frequency f and the I/Q under frequency-f are detracted by about
Beam is detracted by the I/Q under-f into the I/Q detractions caused under f and determined, or the I/Q under frequency-f is detracted by f
Under I/Q detraction determine.
12. method as claimed in claim 11, wherein I/Q detraction include, gain is uneven and phase is crooked, wherein in frequency f
Under gain is uneven and the gain imbalance under frequency-f be constrained for it is equal, wherein the phase under frequency f it is crooked and
Phase under frequency-f is crooked to be constrained for being mutual negative.
13. method as claimed in claim 11, wherein I/Q detraction include, gain is uneven and phase is crooked, wherein in frequency f
Under gain is uneven and the gain imbalance under frequency-f be constrained for it is equal, wherein the phase under frequency f it is crooked and
Phase under frequency-f is crooked be constrained for it is equal.
14. a kind of receiver, including：
I/q demodulator, is configured to receive analog input signal, and I/Q demodulation is performed to analog input signal, to produce simulation
Same phase (I) signal and orthogonal (Q) signal of simulation；
Digital unit, is configured to be digitized simulation I signal and simulation Q signal, to produce digital iota signal sum respectively
Word Q signal；
Digital circuit, is configured to be filtered according to the 2x2 logm word I signals and digital Q signal of digital filter, to produce
The raw digital iota signal filtered and the digital Q signal filtered, wherein the 2x2 matrix configurations of digital filter is at least partly
The I/Q detractions of receiver of the ground compensation in a frequency range, the wherein frequency of at least one diagonal components of the 2x2 matrixes
Response be based on the function as frequency I/Q detraction measurement and as frequency negative function I/Q detraction measurement
Come what is calculated, the frequency response of wherein at least one non-diagonal component of the 2x2 matrixes is the I/Q based on the function as frequency
The measurement of detraction and calculated as the measurement of I/Q detractions of the function of the negative of frequency.
15. receiver as claimed in claim 14, wherein the digital iota signal filtered and the digital Q signal filtered can be used
To recover information bit stream.
16. receiver as claimed in claim 15, in addition to：
Transfer to recover information bit stream for performing symbolic solution by the digital iota signal to having filtered and the digital Q signal filtered
Device.
17. receiver as claimed in claim 14, wherein receiver are testers, wherein analog input signal be in response in
Signal is sent by transmitter transmission and generated, wherein the I/Q including transmitter subtracts for the measurement of the I/Q detractions of receiver
Damage.
18. receiver as claimed in claim 14, wherein digital circuit are programmable hardware element or application specific integrated circuit
(ASIC)。
19. receiver as claimed in claim 14, wherein digital circuit are arranged to hold in response to the execution of programmed instruction
The processor of the row filtering.
20. a kind of computer implemented method for being used to receiver is configured to the I/Q detractions of compensated receiver at least in part,
This method includes：
The measurement that the I/Q of the receiver on a frequency band is detracted is received, wherein receiver includes i/q demodulator, a pair of moduluses
Converter (ADC) and digital circuit, wherein i/q demodulator are configured to according to analog input signal generation simulation I signal and mould
Intend Q signal, wherein ADC is configured to sample to simulation I signal and simulation Q signal, to obtain digital iota signal and numeral respectively
Q signal, wherein digital circuit are configured to be filtered digital iota signal and digital Q signal, to obtain the numeral filtered respectively
I signal and digital Q signal；
2x2 matrixes based on the survey calculation digital filter, wherein the 2x2 matrixes of digital filter are calculated, with the frequency
The I/Q detractions of the receiver taken, which are realized, at least partly to be compensated, and wherein the frequency of at least one diagonal components of the 2x2 matrixes is rung
Should the measurement of function of the measurement based on the function as frequency and the negative as frequency calculate, the wherein 2x2 squares
The frequency response of at least one non-diagonal component of battle array is the measurement based on the function as frequency and the negative as frequency
The measurement of function is calculated；
Digital circuit is programmed, to realize the 2x2 matrixes of digital filter, wherein, digital circuit when so be programmed when,
It is configured as compensating the I/Q detractions of the receiver on the frequency band at least in part.
21. method as claimed in claim 20, wherein digital circuit are programmable hardware element or application specific integrated circuit
(ASIC)。
22. a kind of computer system for being used to receiver is configured to the I/Q detractions of compensated receiver at least in part, the calculating
Machine system includes：
Processor；And
The memory of storage program instruction, wherein programmed instruction when being executed by a processor, makes processor：
The measurement that the I/Q of the receiver on a frequency band is detracted is received, wherein receiver includes i/q demodulator, a pair of moduluses
Converter (ADC) and digital circuit, wherein i/q demodulator are configured to according to analog input signal generation simulation I signal and mould
Intend Q signal, wherein ADC is configured to sample to simulation I signal and simulation Q signal, to obtain digital iota signal and numeral respectively
Q signal, wherein digital circuit are configured to be filtered digital iota signal and digital Q signal, to obtain the numeral filtered respectively
I signal and digital Q signal；
2x2 matrixes based on the survey calculation digital filter, wherein the 2x2 matrixes of digital filter are calculated, with the frequency
The I/Q detractions of the receiver taken, which are realized, at least partly to be compensated, the wherein frequency response of at least one diagonal components of 2x2 matrixes
Be measurement based on the function as frequency and as frequency negative function survey calculation, wherein 2x2 matrixes are extremely
The frequency response of a few non-diagonal component is the function of the measurement based on the function as frequency and the negative as frequency
Survey calculation；And
Digital circuit is programmed, to realize the 2x2 matrixes of digital filter, wherein, digital circuit when so be programmed when,
It is configured as compensating the I/Q detractions of the receiver on the frequency band at least in part.
23. computer system as claimed in claim 22, wherein digital circuit are programmable hardware element or special integrated electricity
Road (ASIC).
24. a kind of method for the I/Q detractions for being used to compensate transmitter, this method includes：
Receive digital inphase (I) signal and digital quadrature (Q) signal；
It is filtered according to the 2x2 logm word I signals and digital Q signal of digital filter, to produce the digital I filtered
Signal and the digital Q signal filtered, the 2x2 matrixes of wherein digital filter are pre-compensated in a frequency range at least in part
The I/Q detractions of interior transmitter, the frequency response of wherein at least one diagonal components of the 2x2 matrixes is based on as frequency
The measurements of the I/Q detractions of the transmitter of function and as frequency negative function survey calculation, the wherein 2x2 matrixes
The frequency response of at least one non-diagonal component is the function of the measurement based on the function as frequency and the negative as frequency
Survey calculation；
The digital iota signal and digital Q signal that have filtered are converted into analog form, to obtain corresponding simulation I signal and mould
Intend Q signal；
I/Q modulation is performed to simulation I signal and simulation Q signal, to produce the analog signal modulated.
25. method as claimed in claim 24, wherein transmitter are testers, wherein the analog signal modulated passes through letter
Road is sent to receiver, and the measurement of the wherein I/Q detractions of transmitter does not include the I/Q detractions of receiver.
26. method as claimed in claim 24, wherein the filtering is in programmable hardware element (PHE) or special integrated electricity
Performed in road (ASIC).
27. method as claimed in claim 24, wherein the filtering be in response to the execution in programmed instruction and within a processor
Perform.
28. method as claimed in claim 24, wherein digital iota signal and digital Q signal carry one or more information bit streams.
29. a kind of method for the I/Q detractions for being used to compensate transmitter, this method includes：
Receive digital inphase (I) signal and digital quadrature (Q) signal；
Digital iota signal and digital Q signal are filtered according to the 2x2 matrixes of digital filter, to produce the number filtered
Word I signal and the digital Q signal filtered, the 2x2 matrixes of wherein digital filter are pre-compensated in a frequency at least in part
The I/Q of transmitter in scope is detracted, wherein at least one of the 2x2 matrixes under the optional frequency f in the frequency range
Frequency response under the f of diagonal components is only based on the measurement of the I/Q detractions under frequency f or is based only upon under frequency-f
What I/Q was detracted measures to calculate, wherein the frequency under the f of at least one non-diagonal component of the 2x2 matrixes under frequency f
Response is only based on the I/Q measurements detracted under frequency f or the measurement for the I/Q detractions being based only upon under frequency-f to calculate
's；
The digital iota signal and digital Q signal that have filtered are converted into analog form, to obtain corresponding simulation I signal and mould
Intend Q signal；
I/Q modulation is performed to simulation I signal and simulation Q signal, to produce the analog signal modulated.
30. method as claimed in claim 29, wherein the I/Q detractions under frequency f and the I/Q under frequency-f are detracted by about
Beam is detracted by the I/Q under-f into the I/Q detractions caused under f and determined, or the I/Q under frequency-f is detracted by f
Under I/Q detraction determine.
31. method as claimed in claim 30, wherein I/Q detraction include, gain is uneven and phase is crooked, wherein in frequency f
Under gain is uneven and the gain imbalance on frequency-f be constrained for it is equal, wherein the phase under frequency f it is crooked and
Phase under frequency-f is crooked to be constrained for being mutual negative.
32. method as claimed in claim 30, wherein I/Q detraction include, gain is uneven and phase is crooked, wherein in frequency f
Under gain is uneven and the gain imbalance under frequency-f be constrained for it is equal, wherein the phase under frequency f it is crooked and
Phase under frequency-f is crooked be constrained for it is equal.
33. a kind of transmitter, including：
Digital circuit, is configured to receive digital inphase (I) signal and digital quadrature (Q) signal, and according to digital filter
2x2 logm word I signals and digital Q signal are filtered, with the digital Q for producing the digital iota signal filtered He having filtered
The I/Q that the 2x2 matrixes of signal, wherein digital filter pre-compensate for the transmitter in a frequency range at least in part subtracts
Damage, the frequency response of wherein at least one diagonal components of the 2x2 matrixes is the I/Q of the transmitter based on the function as frequency
The measurement of detraction and calculated as the measurement of the function of the negative of frequency, wherein at least one non-diagonal of the 2x2 matrixes
The frequency response of component is the measurement of the function of the measurement based on the function as frequency and the negative as frequency to calculate；
Digital-to-analogue conversion (DAC) unit, is configured to the digital iota signal and digital Q signal that have filtered to be converted into analog form, so as to
Obtain corresponding simulation I signal and simulation Q signal；
I/Q modulators, are configured to perform I/Q modulation to simulation I signal and simulation Q signal, to produce the analog signal modulated.
34. transmitter as claimed in claim 33, wherein transmitter are testers, wherein the analog signal modulated passes through
Channel is sent to receiver, and the measurement of the wherein I/Q detractions of transmitter does not include the I/Q detractions of receiver.
35. transmitter as claimed in claim 33, wherein digital circuit are programmable hardware element (PHE) or special integrated electricity
Road (ASIC).
36. transmitter as claimed in claim 33, wherein digital circuit include being configured to execution in response to programmed instruction and
Perform the processor of the filtering.
37. transmitter as claimed in claim 33, wherein digital iota signal and digital Q signal carry one or more information bits
Stream.
38. a kind of method for being used to transmitter is configured to compensate the I/Q detractions of transmitter at least in part, this method includes：
The measurement that the I/Q of the transmitter in a frequency range is detracted is received, wherein transmitter includes digital circuit, a logarithm
Weighted-voltage D/A converter (DAC) and I/Q modulators, wherein digital circuit are configured to be filtered digital iota signal and digital Q signal,
The digital iota signal filtered with obtaining respectively and the digital Q signal filtered, wherein this DAC is configured to the number filtered
Word I signal and digital Q signal are converted into analog form, to obtain simulation I signal and simulation Q signal, wherein I/Q modulators respectively
It is configured to using I signal and simulation Q signal modulation carrying signal is simulated, to obtain the carrying signal modulated；
It is used for the 2x2 matrixes of the digital filter of digital circuit based on the survey calculation, wherein calculating the 2x2 squares of digital filter
Battle array, at least part precompensation detracted with the I/Q realized to transmitter, wherein at least one diagonal components of the 2x2 matrixes
Frequency response is the measurement of the function of the measurement based on the function as frequency and the negative as frequency to calculate, wherein should
The frequency response of at least one non-diagonal component of 2x2 matrixes is the measurement based on the function as frequency and bearing as frequency
The measurement of several function is calculated；
Digital circuit is programmed, to realize the 2x2 matrixes of digital filter, wherein, digital circuit when so be programmed when,
It is configured to pre-compensate for the I/Q detractions of transmitter at least in part.
39. a kind of computer system for being used to transmitter is configured to compensate the I/Q detractions of transmitter at least in part, the calculating
Machine system includes：
Processor；And
The memory of storage program instruction, wherein programmed instruction when being executed by a processor, makes processor：
The measurement that the I/Q of the transmitter in a frequency range is detracted is received, wherein transmitter includes digital circuit, a logarithm
Weighted-voltage D/A converter (DAC) and I/Q modulators, wherein digital circuit are configured to be filtered digital iota signal and digital Q signal,
The digital iota signal filtered with obtaining respectively and the digital Q signal filtered, wherein this DAC is configured to the number filtered
Word I signal and digital Q signal are converted into analog form, to obtain simulation I signal and simulation Q signal, wherein I/Q modulators respectively
It is configured to using I signal and simulation Q signal modulation carrying signal is simulated, to obtain the carrying signal modulated；
It is used for the 2x2 matrixes of the digital filter of digital circuit based on the survey calculation, wherein calculating the 2x2 squares of digital filter
Battle array, at least part precompensation detracted with the I/Q realized to transmitter, wherein at least one diagonal components of the 2x2 matrixes
Frequency response is the measurement of the function of the measurement based on the function as frequency and the negative as frequency to calculate, wherein should
The frequency response of at least one non-diagonal component of 2x2 matrixes is the measurement based on the function as frequency and bearing as frequency
The measurement of several function is calculated；
Digital circuit is programmed, to realize the 2x2 matrixes of digital filter, wherein, digital circuit when so be programmed when,
It is configured to pre-compensate for the I/Q detractions of transmitter at least in part.
40. computer system as claimed in claim 39, wherein digital circuit are programmable hardware element or special integrated electricity
Road (ASIC).
41. a kind of method for operating transmitter, this method includes：
Receive digital inphase (I) signal and digital quadrature (Q) signal；
Digital iota signal and digital Q signal are converted according to the 2x2 matrixes of constant, to produce resulting number I signal and resulting number
Q signal；
Resulting number I signal and resulting number Q signal are converted into analog form, to obtain corresponding simulation I signal and mould
Intend Q signal；
I/Q modulation is performed to simulation I signal and simulation Q signal, to produce the analog signal modulated；
Wherein the 2x2 matrix configurations are the I/Q detractions for pre-compensating for the transmitter under frequency f at least in part, wherein corresponding to
First constant of the first diagonal element of 2x2 matrixes be based on the transmitter under frequency f I/Q detraction measurement and frequency
What the I/Q of the transmitter under rate-f was detracted measures to calculate, wherein second of the first off-diagonal element corresponding to 2x2 matrixes
Individual constant is the measurement based on the measurement under frequency f and under frequency-f to calculate.
42. method as claimed in claim 41, wherein digital inphase (I) signal and digital quadrature (Q) signal are represented in frequency f
Under complex exponential tone.
43. method as claimed in claim 41, wherein digital inphase (I) signal and digital quadrature (Q) signal carry corresponding
Binary message stream.
44. a kind of transmitter, including：
Digital circuit, is configured to receive digital inphase (I) signal and digital quadrature (Q) signal, and according to the 2x2 matrixes of constant
Digital iota signal and digital Q signal are converted, to produce resulting number I signal and resulting number Q signal；
Digital-to-analogue conversion (DAC) unit, is configured to a resulting number I signal and resulting number Q signal is converted into analog form, so as to
Obtain corresponding simulation I signal and simulation Q signal；
I/Q modulators, are configured to perform I/Q modulation to simulation I signal and simulation Q signal, to produce the analog signal modulated,
Wherein the 2x2 matrix configurations are the I/Q detractions for pre-compensating for the transmitter under frequency f at least in part, wherein corresponding to 2x2 squares
Battle array the first diagonal element first constant be based on the transmitter under frequency f I/Q detraction measurement and in frequency-f
Under the measurement of I/Q detractions of transmitter calculate, wherein corresponding to second of the first off-diagonal element of 2x2 matrixes often
Amount is the measurement based on the measurement under frequency f and under frequency-f to calculate.
45. transmitter as claimed in claim 44, wherein digital inphase (I) signal and digital quadrature (Q) signal are represented in frequency
Complex exponential tone under rate f.
46. transmitter as claimed in claim 44, wherein digital inphase (I) signal and digital quadrature (Q) signal carry corresponding
Binary message stream.
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| US13/404,851 US8638893B2 (en) | 2012-02-24 | 2012-02-24 | Mechanisms for the correction of I/Q impairments |
| US13/404,896 US8442150B1 (en) | 2012-02-24 | 2012-02-24 | Measurement of transmitter impairments using offset local oscillators |
| US13/404,851 | 2012-02-24 | ||
| US13/404,896 | 2012-02-24 | ||
| PCT/US2013/026572 WO2013126302A2 (en) | 2012-02-24 | 2013-02-18 | Mechanisms for the correction of i/q impairments, and measurement of transmitter impairments using offset local oscillators |
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| US6940916B1 (en) * | 2000-01-27 | 2005-09-06 | Pmc-Sierra, Inc. | Wideband analog quadrature modulator/demodulator with pre-compensation/post-compensation correction |
| US20030072393A1 (en) * | 2001-08-02 | 2003-04-17 | Jian Gu | Quadrature transceiver substantially free of adverse circuitry mismatch effects |
| US7269394B2 (en) * | 2002-10-02 | 2007-09-11 | Agere Systems Inc. | Frequency offset compensation for communication systems |
| EP1665693A1 (en) * | 2003-09-15 | 2006-06-07 | Intel Corporation | Adaptive iq imbalance correction for multicarrier wireless communication systems |
| US7706475B1 (en) * | 2005-04-15 | 2010-04-27 | Marvell International Ltd. | Compensation of I/Q mismatch in a communication system using I/Q modulation |
| US7962113B2 (en) * | 2005-10-31 | 2011-06-14 | Silicon Laboratories Inc. | Receiver with multi-tone wideband I/Q mismatch calibration and method therefor |
| US7881402B2 (en) * | 2006-09-07 | 2011-02-01 | Via Technologies, Inc. | Compensation for gain imbalance, phase imbalance and DC offsets in a transmitter |
| CN101188430B (en) * | 2006-11-15 | 2012-05-16 | 中兴通讯股份有限公司 | A frequency deviation pre-compensation device and method for broadband code division multiple address user device |
| US7848450B2 (en) * | 2007-03-22 | 2010-12-07 | Texas Instruments Incorporated | Methods and apparatus to pre-compensate for I/Q distortion in quadrature transmitters |
| US8462898B2 (en) * | 2008-06-30 | 2013-06-11 | Entropic Communications, Inc. | System and method for blind compensation and correction of transmitter IQ imbalance at the receiver |
| US8532237B2 (en) * | 2010-02-09 | 2013-09-10 | Provigent Ltd | Correction of alternating I/Q imbalance and frequency offset impairments |
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