CN103227768B

CN103227768B - The application in OFDM modulates of a kind of new ICI self elimination method

Info

Publication number: CN103227768B
Application number: CN201310159226.1A
Authority: CN
Inventors: 曾桂根; 季叶一
Original assignee: Nanjing Post and Telecommunication University
Current assignee: Nanjing Post and Telecommunication University
Priority date: 2013-04-28
Filing date: 2013-04-28
Publication date: 2016-07-06
Anticipated expiration: 2033-04-28
Also published as: CN103227768A

Abstract

The invention is applied in an OFDM modulation system, and performs joint signal processing in the frequency domain and time domain, and is composed of an OFDM modulator, that is, a sending end, and an OFDM demodulator, that is, a receiving end. Before performing iFFT at the sending end, transmit information on even subcarriers and fill in 0s in odd subcarriers; or transmit information in odd subcarriers and fill in 0s in odd subcarriers, or extend the above method to transmit data on subcarriers divisible by 4 , and the remaining subcarriers are filled with 0 to further eliminate the influence of ICI. The time-domain waveform after the above processing has symmetry, and the waveform can be compressed; the opposite process is performed at the receiving end, and the waveform is restored and FFT transformed according to the symmetry. Under the premise of not reducing the channel transmission efficiency, the influence of ICI can be significantly reduced, and the channel utilization rate remains basically unchanged. The core idea of the present invention is: in frequency domain, adjacent subcarrier items are compressed to eliminate the influence of ICI; in time domain, time domain waveform is compressed by symmetry to improve channel transmission efficiency.

Description

Application of a new ICI self-cancellation method in OFDM modulation

技术领域technical field

本发明提出的方法，可应用于正交频分复用（OrthogonalFrequencyDivisionMultiplexing，OFDM）调制中，属于移动通信技术领域。The method proposed by the invention can be applied to Orthogonal Frequency Division Multiplexing (OFDM) modulation, and belongs to the technical field of mobile communication.

技术背景technical background

在OFDM/QAM（QAM，QuadratureAmplitudeModulation，正交幅度调制）传输系统中，移动多径信道造成的频率选择性衰落和高速移动造成的Doppler频移是影响传输可靠性的两个主要因素。移动引起的多普勒（Doppler）频移和收发信机之间的参考频率差异，都会产生频偏，频偏会导致OFDM符号中的子载波间干扰（ICI）；降低ICI影响的一个思路是，采用各种算法估计出Doppler频移值，然后设法计算出ICI大小，并设法消除，但当信道条件恶化时，准确估算出频偏值十分困难，尤其在移动环境下，Doppler频移效应的影响是动态变化的，难以准确跟踪，多径和信道噪声的联合影响，会明显降低算法效果，尤其对高阶调制（如16QAM、64QAM），在信噪比（SNR）不高时，改进效果较差。In OFDM/QAM (QAM, Quadrature Amplitude Modulation, Quadrature Amplitude Modulation) transmission system, frequency selective fading caused by mobile multipath channels and Doppler frequency shift caused by high-speed mobile are two main factors affecting transmission reliability. The Doppler (Doppler) frequency shift caused by movement and the reference frequency difference between transceivers will generate frequency offset, which will cause inter-subcarrier interference (ICI) in OFDM symbols; one idea to reduce the impact of ICI is , use various algorithms to estimate the Doppler frequency shift value, and then try to calculate the ICI size, and try to eliminate it, but when the channel condition deteriorates, it is very difficult to accurately estimate the frequency shift value, especially in the mobile environment, the Doppler frequency shift effect The impact is dynamic and difficult to track accurately. The joint impact of multipath and channel noise will significantly reduce the algorithm effect, especially for high-order modulation (such as 16QAM, 64QAM), when the signal-to-noise ratio (SNR) is not high, the improvement effect poor.

降低ICI影响的另一思路是，对OFDM调制进行改进，从根本上消除频偏对ICI的影响，这类方法称为ICI自消除方法，其本质是对OFDM调制进行改进，消除产生ICI的根本原因。最经典的一种ICI自消除方法是，在进行iFFT之前，使相邻奇偶子载波互为相反数，即用一对奇偶子载波传输相同的信息，又称邻近子载波重复（AdjacentSub-carrierRepeat，ASR）方法。基于ASR的ICI自消除OFDM调制系统，能明显降低ICI影响，但会导致调制效率（或信道利用率）降为原来的50%，得不偿失。Another idea to reduce the impact of ICI is to improve OFDM modulation and fundamentally eliminate the impact of frequency offset on ICI. This type of method is called ICI self-cancellation method. Its essence is to improve OFDM modulation and eliminate the root cause of ICI. reason. The most classic ICI self-cancellation method is to make the adjacent odd and even subcarriers opposite to each other before performing iFFT, that is, to use a pair of odd and even subcarriers to transmit the same information, also known as Adjacent Sub-carrier Repeat (Adjacent Sub-carrier Repeat, ASR) method. The ASR-based ICI self-cancellation OFDM modulation system can significantly reduce the impact of ICI, but it will cause the modulation efficiency (or channel utilization) to drop to 50% of the original, which is not worth the candle.

发明内容Contents of the invention

本发明的发明目的：在进行OFDM调制时，通过频域和时域的联合信号处理，从根本上消除ICI对传输性能的影响，而且不降低带宽利用率。The purpose of the present invention is to fundamentally eliminate the influence of ICI on transmission performance through joint signal processing in frequency domain and time domain during OFDM modulation without reducing bandwidth utilization.

本发明的技术方案为：通过观察发现，奇数子载波填0后的OFDM符号的时域波形，其前半部分与后半部分完全相同，利用这一特性，在发送端只需要发送OFDM符号的前半部分，在接收端利用以上特性恢复被压缩的时域波形。也可以在奇数子载波传输数据，偶数子载波填0；或者扩展本文方案，在能被4整除的子载波上传输数据，其余子载波填0，使其更进一步消除ICI的影响。The technical solution of the present invention is: through observation, it is found that the first half of the time-domain waveform of the OFDM symbol after the odd subcarrier is filled with 0 is exactly the same as the second half. Using this feature, only the first half of the OFDM symbol needs to be sent at the sending end part, the compressed time-domain waveform is restored at the receiving end using the above properties. It is also possible to transmit data on odd-numbered subcarriers, and fill in even-numbered subcarriers with 0; or expand the scheme in this paper, transmit data on subcarriers divisible by 4, and fill the rest of subcarriers with 0, so that it can further eliminate the influence of ICI.

为此，本发明提出：在偶数子载波传输数据，奇数子载波填0，这样就去除了所有奇数子载波带来的影响，包括影响较大的相邻子载波。For this reason, the present invention proposes: transmit data on even subcarriers, and fill in odd subcarriers with 0, thus removing the influence brought by all odd subcarriers, including adjacent subcarriers with greater influence.

在常规OFDM通信系统中，假设由于多普勒效应或者由于晶振偏差引起的载频偏差归一化值为ε。在加性高斯信道情况下，接收端第k个子载波的解调信号可以描述为：In a conventional OFDM communication system, it is assumed that the normalized value of the carrier frequency deviation caused by the Doppler effect or the deviation of the crystal oscillator is ε. In the case of an additive Gaussian channel, the demodulated signal of the kth subcarrier at the receiving end can be described as:

${Z Z}_{k k} = = {X x}_{k k} {S S}_{00} + + {Σ Σ}_{i i = = 00,, i i &NotEqual; &NotEqual; k k}^{N N - - 11} {X x}_{i i} {S S}_{k k - - i i} + + {N N}_{k k},, k k = = 0,1 0,1,, . . . . . .,, N N - - 11 - - - - - - ((11))$

其中N是载波总数，X_k是在第k个子载波上传输的数据，X_kS₀是有用信号部分，N_k是加性高斯噪声；S_k-i是第i个子载波给第k个子载波带来的ICI影响值大小，可以表述为：Where N is the total number of carriers, X _k is the data transmitted on the kth subcarrier, X _k S ₀ is the useful signal part, N _k is the additive Gaussian noise; S _ki is the i-th subcarrier to the k-th subcarrier The magnitude of the ICI influence value can be expressed as:

${S S}_{k k - - i i} = = \frac{sin sin [[π π ((k k - - i i + + ϵ ϵ))]]}{N N sin sin [[\frac{π π}{N N} ((k k - - i i + + ϵ ϵ))]]} \cdot &Center Dot; exp exp [[jπ jπ ((11 - - 11 / / N N)) ((k k - - i i + + ϵ ϵ))]] - - - - - - ((22))$

其中ε=Δf/fs为归一化载波频偏。当没有载波频偏（ε=0）时，S₀=1，S_k-i=0，此时没有ICI的影响。当存在频偏时，S_k-i≠0，为总的ICI干扰值。观察S_k-i波形发现，对干扰项贡献最大的是S_k-i=+1,-1时对应的项，而这两项正好是当前第k个子载波的相邻项；如果在调制端不发送这两个相邻子载波项，即X_k-1=X_k+1＝0，则从根本上消除了ICI干扰值中最大的两项。在OFDM调制端，如果在k=0,2,...4,N-,子载波上发送信号，在k=1,3,5,...,N-3,N-1子载波上不发送信号（即X_k=0,k=1,3,5,...,N-3,N-1），就能达到基本ICI自消除的目的。Where ε=Δf/fs is the normalized carrier frequency offset. When there is no carrier frequency offset (ε=0), S ₀ =1, S _ki =0, and there is no influence of ICI at this time. When frequency offset exists, S _ki ≠0, is the total ICI interference value. Observing the S _ki waveform, it is found that the item that contributes the most to the interference item is the corresponding item when S _ki =+1,-1, and these two items are just the adjacent items of the current kth subcarrier; if the two items are not sent at the modulation end adjacent subcarrier items, that is, X _k-1 =X _k+1 =0, fundamentally eliminates the two largest items in the ICI interference value. At the OFDM modulation end, if the signal is sent on k=0,2,...4,N-, subcarriers, on k=1,3,5,...,N-3,N-1 subcarriers The purpose of basic ICI self-cancellation can be achieved without sending signals (ie X _k =0, k=1,3,5,...,N-3,N-1).

但是，在k=1,3,5,...,N-1子载波上不发送信号，会使传输的数据量减少了一半，即信道利用率降低了一半。但是，把上面提出的ICI自消除方法（即X_k=0,k=1,3,5,...,N-1）代入OFDM调制器的输出信号表达式中，我们发现了一个奇特现象：输出OFDM符号的时域波形前后两部分完全相同。我们称该现象为时域波形的对称性。利用对称性，可以在时域上将另一般波形压缩掉，在解调端再根据对称性恢复。经过上述处理后，信道利用率与常规OFDM系统相同。However, not sending signals on k=1, 3, 5, ..., N-1 subcarriers will reduce the amount of transmitted data by half, that is, the channel utilization rate will be reduced by half. However, substituting the ICI self-cancellation method (that is, X _k = 0, k = 1, 3, 5, ..., N-1) proposed above into the output signal expression of the OFDM modulator, we found a strange phenomenon : The two parts before and after the time domain waveform of the output OFDM symbol are exactly the same. We call this phenomenon the symmetry of the time-domain waveform. Using symmetry, another general waveform can be compressed in the time domain, and then restored according to the symmetry at the demodulation end. After the above processing, the channel utilization rate is the same as that of the conventional OFDM system.

也可以在奇数子载波传输数据，偶数子载波填0；或者扩展上述方法，在能被4整除的子载波上传输数据，其余子载波填0，使其更进一步消除ICI的影响。仿真和实验测试表明，该方案在不降低信道传输效率的前提下，能显著降低ICI的影响。Data can also be transmitted on odd-numbered subcarriers, and even-numbered subcarriers can be filled with 0; or the above method can be extended to transmit data on subcarriers divisible by 4, and the remaining subcarriers can be filled with 0, so as to further eliminate the influence of ICI. Simulation and experimental tests show that the scheme can significantly reduce the impact of ICI without reducing the channel transmission efficiency.

抑制相邻子载波的ICI自消除方法的核心思想是：在频域上将相邻子载波项压缩，以消除ICI的影响；在时域上，利用对称性压缩时域波形，以提高信道传输效率。The core idea of the ICI self-cancellation method for suppressing adjacent subcarriers is: compress the adjacent subcarrier items in the frequency domain to eliminate the influence of ICI; in the time domain, use symmetry to compress the time domain waveform to improve channel transmission. efficiency.

时域波形的对称性推导如下。The symmetry of the time domain waveform is derived as follows.

在OFDM调制中，经过N点逆离散傅里叶变换/逆快速傅里叶变换（iDFT/iFFT）模块后的输出信号可表示为：In OFDM modulation, the output signal after the N-point inverse discrete Fourier transform/inverse fast Fourier transform (iDFT/iFFT) module can be expressed as:

$x x ((n no)) = = \frac{11}{N N} {Σ Σ}_{k k = = 00}^{N N - - 11} X x ((k k)) exp exp ((j j \frac{22 π π}{N N} nk nk)),, n no = = 0,1 0,1,, . . . . . .,, N N - - 11 - - - - - - ((33))$

其中N为偶数，因奇数子载波用0填充，X(1)=X(3)=...=X(2n+1)=...=X(N-3)=X(N-1)=0，代入（3）式，得：Where N is an even number, because odd subcarriers are filled with 0, X(1)=X(3)=...=X(2n+1)=...=X(N-3)=X(N-1 )=0, substituting into formula (3), we get:

$x x ((n no)) = = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) n no)),, n no = = 00,, 11,, . . . . . .,, N N - - 11$

$= = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) n no)),, n no = = 0,1 0,1,, . . . . . .,, N N / / 22 - - 11$

$= = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) n no)) exp exp ((j j 22 πi πi)),, n no = = 0,1 0,1,, . . . . . .,, N N / / 22 - - 11$

$= = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) ((n no + + N N / / 22)))),, n no = = 0,1 0,1,, . . . . . .,, N N / / 22 - - 11 - - - - - - ((44))$

$= = x x ((n no + + N N / / 22))$

x(n)=x(n+N/2)，表示时域波形x(n)的前后两部分相同。x(n)=x(n+N/2), which means that the front and rear parts of the time-domain waveform x(n) are the same.

抑制相邻子载波的ICI自消除方法在OFDM调制中的应用原理见图1所示；它是一种改进的OFDM传输系统，由改进的OFDM调制器（发送端）和改进的OFDM解调器（接收端）构成。The application principle of the ICI self-cancellation method for suppressing adjacent subcarriers in OFDM modulation is shown in Figure 1; it is an improved OFDM transmission system, which consists of an improved OFDM modulator (transmitter) and an improved OFDM demodulator (receiving end) constitutes.

在发送端，数据流先被送入nQAM映射器（Mapper）进行星座图映射，输出N/2对数据，然后送入2倍上采样器，输出N对数据；这N对数据满足抑制相邻子载波的条件：At the sending end, the data stream is first sent to the nQAM mapper (Mapper) for constellation map mapping, and N/2 pairs of data are output, and then sent to the 2-times upsampler to output N pairs of data; these N pairs of data satisfy the requirement of suppressing adjacent Conditions for subcarriers:

X_K≠0,k=0,2,...,4N-,X_k≠0,k=0,2,...,4N-,X _K ≠0,k=0,2,...,4N-,X _k ≠0,k=0,2,...,4N-,

X_k=0,k=1,3,...,5N-,X_k=0,k=1,3,...,5N-,X _k =0,k=1,3,...,5N-,X _k =0,k=1,3,...,5N-,

这N对并行数据被输入N点逆离散傅里叶变换/逆快速傅里叶变换（iDFT/iFFT）模块，输出时长为T_S的OFDM符号，经过并串变换后，成为前后两部分相同（对称性）的时域波形；该波形经过波形压缩器压缩后，输出时长为T_S/2的时域信号；该信号经过加前缀（CP）处理和正交调制模块后，输出一个全新的OFDM已调信号。The N pairs of parallel data are input into the N-point inverse discrete Fourier transform/inverse fast Fourier transform (iDFT/iFFT) module, and the OFDM symbols with a duration of T _S are output. symmetry) time-domain waveform; after the waveform is compressed by the waveform compressor, it outputs a time-domain signal with a duration of T _S /2; after the signal is processed by prefix (CP) and quadrature modulation module, it outputs a brand new OFDM modulated signal.

解调器的处理过程基本是上述过程的逆过程。The processing process of the demodulator is basically the reverse process of the above process.

本发明的有益效果为：与常规OFDM调制系统相比，本发明能较大地降低子载波间干扰（ICI）对接收性能的影响，尤其在移动、多径场景下，而信道利用率基本与常规OFDM调制系统一致；与典型的ICI自消除方法相比，信道利用率提高了近一倍，在ICI对接收性能影响的消除效果上，两种方法不相上下。The beneficial effect of the present invention is: compared with the conventional OFDM modulation system, the present invention can greatly reduce the impact of inter-carrier interference (ICI) on receiving performance, especially in mobile and multipath scenarios, and the channel utilization rate is basically the same as the conventional The OFDM modulation system is consistent; compared with the typical ICI self-cancellation method, the channel utilization rate is nearly doubled, and the two methods are comparable in the elimination effect of ICI on the receiving performance.

附图说明Description of drawings

图1是采用本发明方案对应的OFDM调制解调系统原理框图Fig. 1 is the functional block diagram of the OFDM modulation and demodulation system corresponding to the scheme of the present invention

图2是常规OFDM系统与本发明的2阶ICI自消除方案对应的OFDM系统中ICI干扰项的对比Fig. 2 is the comparison of the ICI interference item in the OFDM system corresponding to the conventional OFDM system and the 2nd-order ICI self-cancellation scheme of the present invention

图3是不同方案下图3为OFDM信号频谱分解示意图，从图可以看出频域下有用信号与ICI干扰信号在不同频偏下的变化情况Figure 3 is a schematic diagram of OFDM signal spectrum decomposition under different schemes. From the figure, we can see the changes of useful signals and ICI interference signals in the frequency domain under different frequency offsets

图4是单径、高斯信道下4种方案BER性能对比Figure 4 is a comparison of the BER performance of the four schemes under single-path and Gaussian channels

图5是乡村多径（RAX）信道下3种方案BER性能对比Figure 5 is a comparison of the BER performance of the three schemes under the rural multipath (RAX) channel

图6是城市多径（TUX）信道下2种方案BER性能对比。Figure 6 shows the BER performance comparison of the two schemes under the urban multipath (TUX) channel.

具体实施方案specific implementation plan

1基本方案1 basic plan

抑制相邻子载波的ICI自消除方法在OFDM调制中的应用原理如图1所示，在发送端，数据流先被送入nQAM映射器（Mapper）进行星座图映射，输出N/2对数据，然后2倍上采样，并经过N点逆离散傅里叶变换/逆快速傅里叶变换（iDFT/iFFT）模块。数据在上采样后，偶数子载波对应的是数据，奇数子载波用0填充，经过iDFT/iFFT模块后，输出长度为T_S的OFDM符号，该符号时域波形的前后两部分相同（对称性）。The application principle of the ICI self-cancellation method for suppressing adjacent subcarriers in OFDM modulation is shown in Figure 1. At the sending end, the data stream is first sent to the nQAM mapper (Mapper) for constellation map mapping, and N/2 pairs of data are output , and then upsampled by 2 times, and passed through the N-point inverse discrete Fourier transform/inverse fast Fourier transform (iDFT/iFFT) module. After the data is up-sampled, the even-numbered subcarriers correspond to the data, and the odd-numbered subcarriers are filled with 0. After the iDFT/iFFT module, an OFDM symbol with a length of T _S is output. ).

IDFT后，输出信号的表达式为：After IDFT, the expression of the output signal is:

$x x ((n no)) = = \frac{11}{N N} {Σ Σ}_{k k = = 00}^{N N - - 11} X x ((k k)) exp exp ((j j \frac{22 π π}{N N} nk nk)),, n no = = 0,1 0,1,, . . . . . .,, N N - - 11 - - - - - - ((55))$

其中N为偶数，因奇数子载波用0填充，X(1)=X(3)=...=X(2n+1)=...=X(N-3)=X(N-1)=0，代入（1）式，得：Where N is an even number, because odd subcarriers are filled with 0, X(1)=X(3)=...=X(2n+1)=...=X(N-3)=X(N-1 )=0, substituting into formula (1), we get:

$= = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) ((n no + + N N / / 22)))),, n no = = 0,1 0,1,, . . . . . .,, N N / / 22 - - 11 - - - - - - ((66))$

$= = x x ((n no + + N N / / 22))$

x(n)=x(n+N/2)，表示时域波形x(n)的前后两部分相同，由此，可以将x(n)序列进行时域压缩。x(n)=x(n+N/2), which means that the front and rear parts of the time-domain waveform x(n) are the same, thus, the time-domain compression of the x(n) sequence can be performed.

也可以采用奇数子载波传输的是数据、偶数子载波被0填充的方案，这时经过iDFT/iFFT后输出的OFDM符号为前后两部分负对称。It is also possible to adopt a scheme in which odd-numbered subcarriers transmit data and even-numbered subcarriers are filled with 0s. At this time, the OFDM symbols output after iDFT/iFFT are negatively symmetrical between the front and back parts.

利用OFDM符号前后两部分相同或者相反这一特性，在数据压缩模块中可以只传输OFDM符号波形的一半，另一半被压缩。在进行iFFT前，由于相邻子载波被填0，使信道利用率下降了一半，但时域上OFDM符合被压缩了一半，使信道利用率与常规OFDM传输系统的相同。但相邻子载波被去掉后，可以消除频偏引起的ICI的影响。Utilizing the characteristic that the two parts before and after the OFDM symbol are the same or opposite, only half of the waveform of the OFDM symbol can be transmitted in the data compression module, and the other half can be compressed. Before performing iFFT, because the adjacent subcarriers are filled with 0, the channel utilization rate is reduced by half, but the OFDM coincidence in the time domain is compressed by half, so that the channel utilization rate is the same as that of the conventional OFDM transmission system. However, after the adjacent subcarriers are removed, the influence of the ICI caused by the frequency offset can be eliminated.

2高阶扩展2 Higher order extensions

上述基本方案也可以进行更高阶的扩展，例如，只在载波号能被4整除的子载波上传输数据符号，在其它点上填充0，经过iFFT后的OFDM符号波形，以Ts/4为周期相同，可以将其余3/4波形压缩。这样的高阶扩展会比之前的基本方案对于ICI自消除方面会带来更好的影响，在不考虑CP影响时，因在时域上对波形进行了压缩，不会降低频带利用率；不过，由于每个压缩后的符合都要添加CP，以消除符号间干扰（ISI），当扩展得太多时，对信道利用率将会产生明显影响。同时，在实现时，高阶扩展会消耗更多的硬件资源。不过，在本方案中，很多子载波上传输的都是0，可以利用这一特性，改进IDFT算法，以此来降低对资源的要求。在实际应用中，可以利用能被2或被4整除的子载波来传输信息，其他子载波填0，这样能兼顾ICI消除效果、资源消耗与信道利用率之间的矛盾。The above basic scheme can also be extended to a higher order. For example, data symbols are only transmitted on subcarriers whose carrier numbers can be divisible by 4, and 0s are filled in other points. The OFDM symbol waveform after iFFT is represented by Ts/4 The period is the same, and the remaining 3/4 waveform can be compressed. Such a high-order extension will have a better impact on ICI self-cancellation than the previous basic scheme. When the impact of CP is not considered, because the waveform is compressed in the time domain, the frequency band utilization will not be reduced; however , because each compressed coincidence needs to add CP to eliminate inter-symbol interference (ISI), when the expansion is too large, it will have a significant impact on channel utilization. At the same time, higher-order extensions consume more hardware resources when implemented. However, in this solution, many subcarriers are transmitted with 0, and this feature can be used to improve the IDFT algorithm to reduce the resource requirements. In practical applications, subcarriers that can be divisible by 2 or 4 can be used to transmit information, and other subcarriers can be filled with 0, which can take into account the contradiction between ICI cancellation effect, resource consumption and channel utilization.

3ICI对误比特率性能的影响3ICI's influence on bit error rate performance

在常规OFDM通信系统中，假设由于多普勒效应或者由于晶振偏差引起的载频偏差归一化值为ε。在加性高斯信道情况下，在接收端第k个子载波的解调信号可以描述为：In a conventional OFDM communication system, it is assumed that the normalized value of the carrier frequency deviation caused by the Doppler effect or the deviation of the crystal oscillator is ε. In the case of an additive Gaussian channel, the demodulated signal of the kth subcarrier at the receiving end can be described as:

${Z Z}_{k k} = = {X x}_{k k} {S S}_{00} + + {Σ Σ}_{i i = = 00,, i i &NotEqual; &NotEqual; k k}^{N N - - 11} {X x}_{i i} {S S}_{k k - - i i} + + {N N}_{k k},, k k = = 0,1 0,1,, . . . . . .,, N N - - 11 - - - - - - ((77))$

其中N是载波总数，Xk是在第k个子载波上传输的数据，N_k是所对应的加性高斯噪声，S_k-i是第i个子载波给第k个子载波带来的ICI影响值大小，可以表述为：Where N is the total number of carriers, Xk is the data transmitted on the kth subcarrier, N _k is the corresponding additive Gaussian noise, S _ki is the ICI influence value brought by the ith subcarrier to the kth subcarrier, which can be Expressed as:

${S S}_{k k - - i i} = = \frac{sin sin [[π π ((k k - - i i + + ϵ ϵ))]]}{N N sin sin [[\frac{π π}{N N} ((k k - - i i + + ϵ ϵ))]]} \cdot \cdot exp exp [[jπ jπ ((11 - - 11 / / N N)) ((k k - - i i + + ϵ ϵ))]] - - - - - - ((88))$

其中ε=Δf/fs为归一化载波频偏。式(7)右半部分首项是所需要的有用信号，当没有载波频偏（ε=0）时，S₀=1，S_i-k=0，此时没有ICI的影响。Where ε=Δf/fs is the normalized carrier frequency offset. The first item in the right half of formula (7) is the required useful signal. When there is no carrier frequency offset (ε=0), S ₀ =1, S _ik =0, and there is no influence of ICI at this time.

对于本发明提出的基本ICI自消除系统（2阶方案），接收端第k个子载波的解调信号可以描述为：For the basic ICI self-cancellation system (second-order scheme) proposed by the present invention, the demodulated signal of the kth subcarrier at the receiving end can be described as:

${Z Z}_{k k}^{' '} = = {X x}_{k k} {S S}_{00} + + {Σ Σ}_{i i = = 00,, i i &NotEqual; &NotEqual; k k / / 22}^{N N / / 22 - - 11} {X x}_{22 i i} {S S}_{k k - - 22 i i} + + {N N}_{k k},, k k = = 0,2,4 0,2,4,, . . . . . .,, N N - - 22 - - - - - - ((99))$

Z'_k＝0，k=1，3，5，...,N-3,N-1Z' _k = 0, k = 1, 3, 5, ..., N-3, N-1

其中N是载波总数，X_k是在第k个子载波上传输的数据，N_k是所对应的加性高斯噪声，S_k-2i是第2i个子载波给第k个子载波带来的ICI干扰值。Where N is the total number of carriers, X _k is the data transmitted on the kth subcarrier, N _k is the corresponding additive Gaussian noise, and S _k-2i is the ICI interference value brought by the 2ith subcarrier to the kth subcarrier .

图2为常规系统与2阶ICI自消除系统S_i-k的模值比较，图中总子载波数N=16，频偏ε取为0.15和0.3，7号子载波为有用信号S₀；2阶ICI子消除系统的S_k-2i模值与常规系统的S_k-i模值相比，数量减少了1倍，而且模值最大的2个点（7号子载波的相邻子载波）没有了。Figure 2 is a comparison of the modulus between the conventional system and the second-order ICI self-cancellation system S _ik , the total number of sub-carriers in the figure is N=16, the frequency offset ε is taken as 0.15 and 0.3, and the No. 7 sub-carrier is the useful signal S ₀ ; the second-order Compared with the S _ki modulus of the conventional system, the number of S _k-2i modulus of the ICI sub-cancellation system is reduced by 1 time, and the two points with the largest modulus (adjacent subcarriers of No. 7 subcarriers) are gone.

公式（7）中，第一项是有用信号，第二项是总的干扰项；图3为OFDM信号频谱分解示意图，实线为7号子载波的频谱，点划线为常规系统中干扰子载波的频谱，虚划线为本发明2阶ICI自消除系统干扰子载波的频谱。虚点垂直线为没有频偏时的情况，此时两种系统的干扰值都为0；垂直实线为频偏ε=0.3时的情况，此时本发明2阶ICI自消除系统的干扰值明显小于常规系统的干扰值。In formula (7), the first item is the useful signal, and the second item is the total interference item; Figure 3 is a schematic diagram of OFDM signal spectrum decomposition, the solid line is the spectrum of subcarrier No. 7, and the dotted line is the interferer in the conventional system The frequency spectrum of the carrier, the dotted line is the frequency spectrum of the interference subcarrier of the second-order ICI self-cancellation system of the present invention. The dotted vertical line is the situation when there is no frequency offset, and the interference value of the two systems is 0 at this time; the vertical solid line is the situation when the frequency offset ε=0.3, and the interference value of the second-order ICI self-cancellation system of the present invention is now Significantly less than the interference value of the conventional system.

按照图1的原理框图，对本发明方案进行了仿真。仿真参数为：N=1024，可用子载波为880个，均衡方式为基于梳状导频的线性内插方法，选择QPSK映射方式。According to the principle block diagram of Fig. 1, the scheme of the present invention is simulated. The simulation parameters are: N=1024, the available subcarriers are 880, the equalization method is the linear interpolation method based on the comb pilot, and the QPSK mapping method is selected.

仿真所采用的场景描述为：The scenario used in the simulation is described as:

方案1：常规OFDM系统，接收端未采用ICI消除方法。Solution 1: In a conventional OFDM system, the receiving end does not adopt the ICI elimination method.

方案2：本发明的2阶ICI自消除系统，接收端未采用ICI消除方法。Solution 2: In the second-order ICI self-cancellation system of the present invention, the receiving end does not use the ICI cancellation method.

方案3：本发明的4阶ICI自消除系统，接收端未采用ICI消除方法。Solution 3: In the 4th-order ICI self-cancellation system of the present invention, the receiving end does not use the ICI cancellation method.

方案4：相邻奇偶子载波互为相反数的ICI自消系统，即ASR（AdjacentSub-carrierRepeat）系统。即两个相邻载波传输一对数据符号，奇数点子载波上传输的是偶数点子载波符号的取反符号。在接收端，传输相同符号的两个相邻载波的解调值共同来判决所传输的数据符号。在相同映射方式下，其信道效率下降了50%，为了使信道效率与上面3种情况一致，必须采用16QAM映射。Scheme 4: The ICI self-cancellation system in which the adjacent odd and even subcarriers are opposite to each other, that is, the ASR (AdjacentSub-carrierRepeat) system. That is, two adjacent carriers transmit a pair of data symbols, and what is transmitted on the odd-numbered sub-carriers is the inverted symbol of the even-numbered sub-carriers. At the receiving end, the demodulated values of two adjacent carriers transmitting the same symbol are used together to determine the transmitted data symbol. In the same mapping mode, the channel efficiency drops by 50%. In order to make the channel efficiency consistent with the above three cases, 16QAM mapping must be used.

图4表示上述四种方案在单径（相对于多径）条件下、不同载波频偏的结果。分别仿真了单径ε=0（相当于高斯信道）、ε=0.3情况下，4种方案的误比特率（BER）性能；ε=0时，方案1、2、3的误比特率性能几乎相同，方案4的误比特率高于系统1、2、3，这是因为在高斯信道下，信噪比相同时，QPSK的抗噪声性能强于16QAM；ε=0.3时，采用了ICI自消除方案的方案2、3、4的BER明显低于方案1，其中，采用了本发明4阶ICI自消除方案的方案3的误比特率最低，其次是本发明的2阶ICI自消除方案，最差的是相邻奇偶子载波互为相反数的ICI自消方案。Figure 4 shows the results of the above four schemes under single path (relative to multipath) conditions and different carrier frequency deviations. The bit error rate (BER) performance of the four schemes was simulated under the conditions of single path ε=0 (equivalent to Gaussian channel) and ε=0.3; when ε=0, the BER performance of schemes 1, 2, and 3 was almost Similarly, the bit error rate of scheme 4 is higher than that of systems 1, 2, and 3. This is because the anti-noise performance of QPSK is stronger than that of 16QAM when the signal-to-noise ratio is the same under the Gaussian channel; when ε=0.3, ICI self-cancellation is adopted The BER of schemes 2, 3, and 4 of the scheme is obviously lower than scheme 1, wherein, the bit error rate of scheme 3 adopting the 4th-order ICI self-cancellation scheme of the present invention is the lowest, followed by the 2-order ICI self-cancellation scheme of the present invention, and the lowest What is worse is the ICI self-cancellation scheme in which adjacent odd and even subcarriers are opposite numbers.

仿真结果表明，在不降低频带利用率的情况下，本发明方案的误码率率性能随着阶数的提高会越来越好。The simulation results show that, without reducing the utilization rate of the frequency band, the bit error rate performance of the scheme of the present invention will become better and better with the increase of the order.

在移动多径信道环境下，仿真信道采用3GPPTR25.943协议标准中定义的经典的6抽头TUX信道（城市多径信道）和RAX信道（乡村多径信道）。为了比较不同移动信道下的多普勒频偏的大小，本文定义εd为归一化最大多普勒频偏，εd为信道的最大多普勒频偏与子载波间隔的比值。In the mobile multipath channel environment, the simulation channel adopts the classic 6-tap TUX channel (urban multipath channel) and RAX channel (rural multipath channel) defined in the 3GPP TR25.943 protocol standard. In order to compare the size of the Doppler frequency deviation under different mobile channels, this paper defines εd as the normalized maximum Doppler frequency deviation, and εd is the ratio of the maximum Doppler frequency deviation of the channel to the subcarrier spacing.

图5表示方案1、2、4在RAX信道模型下的误比特率（BER）性能，仿真环境为εd=0.01、εd=0.2，对应多普勒频偏比较小和比较大两种情况。仿真结果表明本发明方案在RAX信道模型下的BER性能优于方案1、4。Figure 5 shows the bit error rate (BER) performance of schemes 1, 2, and 4 under the RAX channel model. The simulation environment is εd=0.01 and εd=0.2, corresponding to two cases where the Doppler frequency offset is relatively small and relatively large. Simulation results show that the BER performance of the scheme of the present invention under the RAX channel model is better than schemes 1 and 4.

图6表示方案2、4在TUX信道模型下、εd=0.01和εd=0.2的BER性能，分别对应多普勒频偏比较小和比较大两种情况。结果表明，在TUX信道模型下本发明提出的ICI自消除方案的BER性能优于方案4。Figure 6 shows the BER performance of schemes 2 and 4 under the TUX channel model, εd=0.01 and εd=0.2, corresponding to the two cases of relatively small and relatively large Doppler frequency offset. The results show that the BER performance of the ICI self-cancellation scheme proposed by the present invention is better than scheme 4 under the TUX channel model.

4频带利用率分析4 Frequency Band Utilization Analysis

常规OFDM系统的频带利用率为（忽略CP的影响）：The frequency band utilization of the conventional OFDM system is (ignoring the influence of CP):

${η η}_{b b} = = \frac{N N \cdot &Center Dot; {log log}_{22} M m}{{T T}_{s the s} \cdot &Center Dot; B B} bit bit / / ((s the s \cdot &Center Dot; Hz Hz)) - - - - - - ((1010))$

相邻奇偶子载波互为相反数的L阶ICI自消除方案即L阶ASR方案的频带利用率（忽略CP的影响）：The frequency band utilization rate of the L-order ICI self-cancellation scheme, that is, the L-order ASR scheme (ignoring the influence of CP), in which the adjacent odd and even subcarriers are opposite numbers:

${η η}_{b b} = = \frac{N N / / L L \cdot &Center Dot; {log log}_{22} M m}{{T T}_{s the s} \cdot &Center Dot; B B} bit bit / / ((s the s \cdot &Center Dot; Hz Hz)) - - - - - - ((1111))$

本发明L阶ICI自消除方案的频带利用率（忽略CP的影响）：The frequency band utilization of the L-order ICI self-cancellation scheme of the present invention (ignoring the influence of CP):

${η η}_{b b} = = \frac{N N / / L L \cdot \cdot {log log}_{22} M m}{{T T}_{s the s} / / L L \cdot \cdot B B} = = \frac{N N \cdot \cdot {log log}_{22} M m}{{T T}_{s the s} \cdot \cdot B B} bit bit / / ((s the s \cdot \cdot Hz Hz)) - - - - - - ((1212))$

其中，N为载波个数，Ts为经过IDFT/IFFT后的OFDM符号长度，M为采用的QAM调制方式的进制数，B为可以用的信道带宽。其中L指高阶扩展的阶数，例如L=4，在ASR方案中是每4个子载波传输同一个数据，在本发明方案中是载波号能整除4的子载波上传输数据，其余填充0。Among them, N is the number of carriers, Ts is the OFDM symbol length after IDFT/IFFT, M is the base number of the QAM modulation mode used, and B is the available channel bandwidth. Where L refers to the order of high-order expansion, for example, L=4, in the ASR scheme, the same data is transmitted every 4 subcarriers, in the scheme of the present invention, data is transmitted on subcarriers whose carrier numbers can be divisible by 4, and the rest are filled with 0 .

比较式（10）、（11）、（12）可知，在相同映射方式（即nQAM相同）下，本发明方案的频带利用率和常规OFDM系统相同，是ASR方案的频带利用率的L倍。Comparing formulas (10), (11) and (12), it can be seen that under the same mapping mode (that is, the same nQAM), the frequency band utilization rate of the present invention scheme is the same as that of the conventional OFDM system, which is L times of the frequency band utilization rate of the ASR scheme.

当考虑CP的影响时，本发明方案的带宽利用率略有降低，影响因子β为：When considering the influence of CP, the bandwidth utilization rate of the scheme of the present invention is slightly reduced, and the influence factor β is:

$β β = = \frac{{T T}_{s the s}}{{T T}_{s the s} + + L L \cdot &Center Dot; {T T}_{CP CP}} - - - - - - ((1313))$

常规OFDM系统可看成是L=1的系统；考虑CP时长是OFDM符号时长的1/16的情况，则本发明的2阶ICI自消除方案的带宽利用率是常规OFDM系统的94.4%，4阶ICI自消除方案的带宽利用率是常规OFDM系统的85%。The conventional OFDM system can be regarded as the system of L=1; Considering that the CP duration is 1/16 of the OFDM symbol duration, the bandwidth utilization rate of the second-order ICI self-cancellation scheme of the present invention is 94.4% of the conventional OFDM system, 4 The bandwidth utilization rate of the first-order ICI self-cancellation scheme is 85% of that of the conventional OFDM system.

兼顾带宽利用率和ICI的自消除效果，本发明的2阶ICI自消除方案的综合性能最好。Considering both the bandwidth utilization rate and the ICI self-cancellation effect, the second-order ICI self-cancellation scheme of the present invention has the best comprehensive performance.

以上显示和描述了本发明的基本原理和主要特征和本发明的优点。本行业的技术人员应了解，本发明不受上述实施例的限制，上述实施例和说明书中描述的只是说明本发明的原理，在不脱离本发明精神和范围的前提下，本发明还会有各种变化和改进，这些变化和改进都落入要求保护的本发明范围内，本发明要求保护范围由所附的权利要求书其等效物界定。The basic principles and main features of the present invention and the advantages of the present invention have been shown and described above. Those skilled in the industry should understand that the present invention is not limited by the above-mentioned embodiments. What are described in the above-mentioned embodiments and the description are only to illustrate the principles of the present invention. Without departing from the spirit and scope of the present invention, the present invention will also have Various changes and improvements fall within the scope of the claimed invention, which is defined by the appended claims and their equivalents.

Claims

1. Application of a new inter-carrier interference (Inter-CarrierInterference, ICI) self-cancellation method in OFDM modulation, the OFDM modulation system applicable to the new ICI self-cancellation method is composed of the OFDM modulator that is the transmitter and the OFDM demodulator that is The structure of the receiving end is characterized in that, at the sending end, the N subcarriers are divided into even subcarriers and odd subcarriers, wherein N is the total number of carriers and N is an even number, only sending information on even subcarriers, filling 0 in odd subcarriers, The time-domain waveform is then compressed using symmetry:

The information is transmitted on the even subcarriers, and the odd subcarriers are filled with 0. The process is as follows. At the sending end, the data stream is first sent to the nQAM mapper (Mapper) for constellation map mapping, and N/2 pairs of data are output, and then sent to 2 Double upsampler, output N pairs of data; these N pairs of data satisfy the condition of suppressing adjacent subcarriers:

X _k ≠0,k=0,2,4,...,N-2

X _k = 0, k = 1, 3, 5, ..., N-1

The N pairs of parallel data are input into the N-point inverse discrete Fourier transform/inverse fast Fourier transform (iDFT/iFFT) module, and the OFDM symbols with a duration of T _S are output. Time-domain waveform; after the waveform is compressed by the waveform compressor, the same second half of the waveform is compressed, and the time-domain signal with a duration of T _S /2 is output; the signal is processed by adding a prefix (CP) and the quadrature modulation module , to output a brand new OFDM modulated signal, and the processing process of the demodulator is the reverse process of the above process.

2. Application of a new inter-carrier interference (Inter-Carrier Interference, ICI) self-cancellation method in OFDM modulation, the OFDM modulation system applicable to the new ICI self-cancellation method is composed of the OFDM modulator that is the transmitter and the OFDM demodulator that is The structure of the receiving end is characterized in that, at the sending end, the N subcarriers are divided into even subcarriers and odd subcarriers, where N is the total number of carriers and N is an even number, and information is only transmitted on odd subcarriers, and even subcarriers are filled with 0 ; Then use the waveform symmetry to compress the time-domain waveform; the specific implementation process is as follows:

The information is transmitted on the odd subcarriers, and the even subcarriers are filled with 0. The process is as follows. At the sending end, the data stream is first sent to the nQAM mapper (Mapper) for constellation map mapping, and N/2 pairs of data are output, and then sent to 2 times The upsampler outputs N pairs of data; these N pairs of data satisfy the condition of suppressing adjacent subcarriers:

X _k ≠ 0, k = 1, 3, 5, ..., N-1

X _k = 0, k = 0, 2, 4, ..., N-2

The N pairs of parallel data are input into the N-point inverse discrete Fourier transform/inverse fast Fourier transform (iDFT/iFFT) module, and the OFDM symbols with a duration of T _S are output. The same time-domain waveform with opposite signs; after the waveform is compressed by the waveform compressor, the second half of the waveform with the same absolute value and opposite signs is compressed, and the time-domain signal with a duration of T _S /2 is output; the signal is added After the prefix (CP) processing and quadrature modulation module, a brand new OFDM modulated signal is output, and the processing process of the demodulator is the reverse process of the above process.

3. according to the application of a kind of new ICI self-cancellation method described in claim 1 and 2 in OFDM modulation, it is characterized in that, after inverse discrete Fourier transform (iDFT), the expression of output signal is:

X x ((n no)) = = \frac{11}{N N} {Σ Σ}_{k k = = 00}^{N N - - 11} X x ((k k)) exp exp ((j j \frac{22 π π}{N N} n no k k)),, n no = = 00,, 11,, ... ...,, N N - - 11 - - - - - - ((11))

Where N is the total number of carriers and N is an even number, because odd subcarriers are filled with 0, X(1)=X(3)=...=X(2n+1)=...=X(N-3)=X(N- 1) = 0, substitute into (1) formula, get:

X x ((n no)) = = \frac{11}{N N} {Σ Σ}_{i i = = 00}^{N N / / 22 - - 11} X x ((22 i i)) exp exp ((j j \frac{22 π π}{N N} ((22 i i)) n no)),, n no = = 00,, 11,, ... ...,, N N - - 11

x(n)=x(n+N/2), which means that the front and rear parts of the time-domain waveform x(n) are the same, thus, the x(n) sequence can be compressed in the time domain;

It is also possible to adopt a scheme in which odd-numbered subcarriers transmit data and even-numbered subcarriers are filled with 0s. At this time, the OFDM symbols output after iDFT/iFFT are negatively symmetrical between the front and back parts.

4. according to the application of a kind of new ICI self-cancellation method described in claim 1 and 2 in OFDM modulation, it is characterized in that, after receiving end is through waveform extension, FFT module, the demodulation signal of the k subcarrier can be improved described as:

\begin{matrix} {Z Z}_{k k}^{' '} = = {X x}_{k k} {S S}_{00} + + {Σ Σ}_{i i = = 00,, i i &NotEqual; &NotEqual; k k}^{N N - - 11} {X x}_{22 i i} {S S}_{k k - - 22 i i} + + {N N}_{k k},, k k = = 00,, 22,, 44,, ... ...,, N N - - 22 \\ {Z Z}_{k k}^{' '} = = 00,, k k = = 11,, 33,, 55,, ... ...,, N N - - 33,, N N - - 11 \end{matrix} - - - - - - ((22))

Where N is the total number of carriers, and N is an even number, X _k is the data transmitted on the kth subcarrier, N _k is the corresponding additive Gaussian noise, S _k-2i is the 2ith subcarrier to the kth subcarrier band coming ICI interference value.

5. according to the application of a kind of new ICI self-cancellation method described in claim 1 and 2 in OFDM modulation, it is characterized in that, can carry out higher order extension, at sending end, in the even subcarrier that can be divisible by 4 The information is transmitted on the uplink, and the rest of the subcarriers are filled with 0; then the time-domain waveform is compressed by 4 times using symmetry.