CN102624249A - Compound control method of three-phase to two-phase orthogonal inverter power supply with reactive compensation function - Google Patents

Abstract

The invention discloses a compound control method of a three-phase to two-phase orthogonal inverter power supply with a reactive compensation function on the basis of a three-phase to two-phase orthogonal inverter power supply system. The three-phase to two-phase orthogonal inverter power supply system comprises a three-phase power supply, a three-phase PWM (Pulse Width Modulation) rectifier bridge, a three-bridge-arm inverter bridge and an electromagnetic stirrer, wherein a small inductor is connected between the midpoint of a public-phase bridge arm of the three-bridge-arm inverter bridge and a grounding point of the electromagnetic stirrer in series; the current waveform in a coil of the electromagnetic stirrer can be improved and a two-phase orthogonal inverter circuit can adopt a dead-beat current control method which has rapid tracing property and is easy to realize digital control, and thus the design of the controller is simplified and the current tracking speed is improved. The compound control method disclosed by the invention can enable the three-phase PWM rectifier bridge to work in a rectifier power supply and also can be used for carrying out reactive compensation on a load outside a network side, makes full use of the equipment volume of the three-phase PWM rectifier bridge and saves the reactive compensation equipment.

Description

Three phase transformations, the two phase quadrature inverter composite control methods of band no-power compensation function

Technical field

The present invention relates to a kind of three phase transformations, two phase quadrature inverter composite control methods that are applicable to electromagnetic stirrer, particularly a kind of three phase transformations, two phase quadrature inverter composite control methods with no-power compensation function.

Background technology

Electromagnetic agitating technology makes metal ingredients such as iron and steel, aluminium alloy even owing to have contactless characteristics, shortens smelting time, reduces the melt top and the bottom temperature difference, reduces the generation of slag, is applied at smelting industries such as continuous casting steel, aluminium foundings.Because electromagnetic stirrer uses the induced field of low speed rotation in molten steel, to produce powerful stirring action, the ability of the mixing torque of this induced field and magnetic field penetration molten steel is closely related with size with the power frequency through electromagnetic stirrer.Therefore, current waveform quality and the response speed as three phase transformations, the two phase quadrature inverters of electromagnetic stirrer power supply is most important to the quality that guarantees metallurgic products such as continuous casting steel.

Three phase transformations, two phase quadrature inverters usually by rectified three-phase circuit (prime) with two mutually quadrature inverter circuit (back level) two parts form; Two traditional phase quadrature inverters normally adopt two H bridge single-phase inversion circuit as main circuit; Need 8 jumbo device for power switching altogether, hardware cost is higher.In order to reduce cost and to reduce device loss, three phase full bridge inverter circuit and two brachium pontis inverter circuits have appearred in recent years in three phase transformations two application in the quadrature inverter mutually.Relative two brachium pontis inversion roads; The harmonic content that the three phase full bridge inverter circuit produces is little; And the inhomogeneous unequal problem of two dc capacitor voltages that causes of alternating current of two dc capacitors of need not considering to flow through; The traditional control method that is used for the three phase full bridge inverter circuit can not directly be used to control back level two phase quadrature inverter circuits, has caused the research of a lot of scholars to three phase transformations, two phase quadrature inverter control methods.At present, the control method that is applied in three phase transformations, the two phase quadrature inverter systems back level two phase inverter circuits mainly by space vector control, based on the PI control of rotating coordinate system etc.Because the voltage vector of back level two phase inverter circuits is asymmetric, its spatiality vector constitutes the length of side six distortion such as non-, confirms when therefore adopting this method that switching time is very complicated.Based on the PI control method under the rotating coordinate system, though can realize the zero steady-state error control of AC signal, this control procedure does not have response for the low-order harmonic component in the feedback signal, and the harmonic current that produces in the power supply will influence the function of electromagnetic stirrer.In addition; Electromagnetic stirrer is in order to play good mixing effect; Usually just work in the three kinds of operating states of changeing, stop transport, reverse; And the control method that is used for three phase transformations, the two phase quadrature inverter prime rectified three-phase circuits of electromagnetic stirrer at present all is to pursue the rectification circuit input side to realize unity power factor, and the control method of in rectification, also as reactive-load compensation equipment external electrical network being carried out reactive power compensation about the prime rectification circuit does not have report as yet.Simultaneously, all there are a large amount of inductive loads in the electromagnetic stirrer application scenario, generally needs to install special reactive-load compensation equipment and realizes energy-saving and cost-reducing.

Summary of the invention

Technical problem to be solved by this invention is; Not enough to prior art, three phase transformations two that provide a kind of and can improve the electromagnetic stirrer coil current, are easy to realize digital control and the fireballing band no-power compensation function of current tracking are quadrature inverter composite control method mutually.

For solving the problems of the technologies described above; The technical scheme that the present invention adopted is: three phase transformations, the two phase quadrature inverter composite control methods of band no-power compensation function; Based on three phase transformations, two phase quadrature inverter systems; Three phase transformations, two phase quadrature inverter systems comprise three phase mains, three phase transformations, two phase quadrature inverters, three phase mains and electromagnetic stirrer, and three phase transformations, two phase quadrature inverters comprise three-phase PWM rectifier bridge and three brachium pontis inverter bridge, and three phase mains is connected with the three-phase PWM rectifier bridge through inductance; The three-phase PWM rectifier bridge is connected with three brachium pontis inverter bridge; Each is connected with a single-phase load mid point of two brachium pontis through outputting inductance in the three brachium pontis inverter bridge, and the mid point of the 3rd brachium pontis is connected with the earth connection of electromagnetic stirrer through outputting inductance, is parallel with inductive load between three phase mains and the three-phase PWM rectifier bridge; Said three-phase PWM rectifier bridge comprises brachium pontis and dc bus capacitor branch road of three parallel connections, and the dc bus capacitor branch road is parallelly connected with brachium pontis, and each brachium pontis comprises the switching device of two series connection; Said three brachium pontis inverter bridge comprise the brachium pontis of three parallel connections, and each brachium pontis comprises the switching device of two series connection, and this method may further comprise the steps:

1) the outside three-phase inductive load or the capacitive load currents of detected three phase transformations, two phase quadrature inverter system net sides are passed through the dq conversion, obtain the reactive current desired value

of three-phase PWM rectifier bridge input side

2) desired value of three-phase PWM rectifier bridge dc capacitor voltage and the difference of value of feedback are obtained active current desired value

through the PI controller

2)

and

After dq transform has been phase PWM Rectifier input side of the three-phase instantaneous target current value

3)

obtains the control signal of switching device in the three-phase PWM rectifier bridge through dead beat current controller and PWM modulator, thereby realizes the stable control of dc capacitor voltage and three-phase PWM rectifier bridge net side external loading is carried out reactive power compensation;

4) a current amplitude set-point with three phase transformations, two phase quadrature inverter systems multiply by a SIN function and a cosine function respectively; Obtain target current value

and that three brachium pontis inverter bridge wherein obtain three brachium pontis inverter bridge third phases after the target current value of two phases

and and sum negate and obtain the control signal of switching device in the three brachium pontis inverter bridge, thereby realize the tracking Control of two phase quadrature currents of three brachium pontis inverter bridge through dead beat current controller and PWM modulator.

Technique effect of the present invention is: the small inductor of between the earth point of the public phase brachium pontis mid point of three brachium pontis inverter bridge (two phase quadrature inverter circuits) and electromagnetic stirrer, connecting; Can improve the current waveform in the electromagnetic stirrer coil; Two phase quadrature inverter circuits can be adopted have quick tracking performance and be easy to numerically controlled dead beat current control method, simplified the design of controller and improved current tracking speed; Simultaneously; The present invention proposes a kind of three-phase PWM rectifier bridge control strategy that has no-power compensation function; When making the three-phase PWM rectifier bridge work in rectifier power source; Can also carry out reactive power compensation to net side external loading, make full use of the place capacity of three-phase PWM rectifier bridge, reduce reactive-load compensation equipment.

Description of drawings

Fig. 1 is one embodiment of the invention three phase transformations two phase quadrature inverter system structural representations;

Fig. 2 is three phase transformations, the two phase quadrature inverter composite control method schematic diagrams of one embodiment of the invention band no-power compensation function;

Wherein:

1: the three-phase PWM rectifier bridge; 2: three brachium pontis inverter bridge; Phase transformation in 3: three two phase quadrature inverters; 4: electromagnetic stirrer.

Embodiment

As shown in Figure 1, the main circuit of three phase transformations, the two phase inverter systems that one embodiment of the invention adopted comprises three phase mains, input inductance L ₁, three-phase PWM rectifier bridge, three brachium pontis inverter bridge, outputting inductance L ₂, electromagnetic stirrer; The three-phase PWM rectifier bridge connects to form three phase transformations two quadrature inverter mutually with three brachium pontis inverter bridge; Switching device in three-phase PWM rectifier bridge and the three brachium pontis inverter circuits is IGBT or SPM IPM, and the three-phase PWM rectifier bridge is through input inductance L ₁Be connected with net side three phase mains; The mid point of two brachium pontis is connected with two single-phase loads through outputting inductance in the three brachium pontis inverter circuits, and the mid point of the 3rd brachium pontis is through small inductor L ₂Be connected with the earth connection of electromagnetic stirrer.

Fig. 2 is three phase transformations, the two phase quadrature inverter composite control method schematic diagrams of present embodiment band no-power compensation function; The forward and backward level system of two phase inverters transmits electric energy successively; The controlled target of prime three-phase PWM rectifier bridge is to realize that dc voltage is stable; Guarantee that three-phase input current is sinusoidal wave, in the residual capacity scope, realize the reactive power compensation to external equipment, the controlled target of back level two phase quadrature inverter circuits is that the output current of back level two phase quadrature inverter circuits is followed the tracks of two phase quadrature current reference values quickly and accurately; Because there is not coupling in forward and backward two-stage system in control, can be considered two relatively independent controlling object.

(1) prime three-phase PWM rectifier bridge control method

According to Fig. 1, we can obtain the circuit equation of prime three-phase PWM rectifier bridge under three-phase abc coordinate system of three phase transformations, two phase quadrature inverters suc as formula shown in (1).

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}=u_{\mathrm{sa}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{ca}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\\ u_{\mathrm{ib}}=u_{\mathrm{sb}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{cb}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cb}}\\ u_{\mathrm{ic}}=u_{\mathrm{sc}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{cc}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\end{array}\right.---\left(1\right)$

In the formula, u _Sx(x=a, b c) are power distribution network points of common connection (Point ofCommon Coupling, instantaneous voltage PCC); u _Ix(x=a, b c) are three-phase PWM rectifier bridge AC side instantaneous voltage; i _Cx(x=a, b c) are the transient current of the three-phase PWM rectifier bridge input filter inductance of flowing through.

Formula (1) is put in order and can be got:

$\left\{\begin{array}{c}{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{ca}}}{\mathrm{dt}}=u_{\mathrm{sa}}-u_{\mathrm{ia}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{cb}}}{\mathrm{dt}}=u_{\mathrm{sb}}-u_{\mathrm{ib}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{<figure-callout\; id="1"\; label="cb\; L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">cb</figure-callout>}}& \\ L_{}{}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{cc}}}{\mathrm{dt}}=u_{\mathrm{sc}}-u_{\mathrm{ic}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\end{array}\right.---\left(2\right)$

Constantly formula (2) being carried out discretization K switch periods can get

$\left\{\begin{array}{c}{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{ca}}(k+1)-i_{\mathrm{ca}}\left(k\right)}{T_S}=u_{\mathrm{sa}}\left(k\right)-u_{\mathrm{ia}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\left(k\right)\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{cb}}(k+1)-i_{\mathrm{cb}}\left(k\right)}{T_S}=u_{\mathrm{sb}}\left(k\right)-u_{\mathrm{ib}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cb}}\left(k\right)\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{cc}}(k+1)-i_{\mathrm{cc}}\left(k\right)}{T_S}=u_{\mathrm{sc}}\left(k\right)-u_{\mathrm{ic}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\left(k\right)\end{array}\right.---\left(3\right)$

In the formula, T _SBe the IGBT switch periods time.

If its reference current value is realized that dead beat is tracked as controlled target with each phase current in a switch periods, then can be with the current value of switch periods finish time (or next switch periods the zero hour) reference current value, promptly as this switch periods

$\left\{\begin{array}{c}{i}_{\mathrm{ca}}^{*}\left(k\right)=i_{\mathrm{ca}}(k+1)\\ i_{\mathrm{cb}}^{*}\left(k\right)=i_{\mathrm{cb}}(k+1)\\ i_{\mathrm{cc}}^{*}\left(k\right)=i_{\mathrm{cc}}(k+1)\end{array}\right.---\left(4\right)$

Formula (4) substitution formula (3) and arrangement can be got:

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)\end{array}\right.---\left(5\right)$

Formula (5) is the dead beat current controller under the abc coordinate system.It can find out from formula (5), and a in the three-phase PWM rectifier bridge Mathematical Modeling, b, c three-phase electric weight are separate, therefore, do not have coupling phenomenon between the Current Control amount of this dead beat current controller; U wherein _Ia(k), u _Ib(k), u _Ic(k) be k three-phase PWM rectifier bridge AC side instantaneous voltage constantly, i _Ca(k), i _Cb(k), i _Cc(k) be flow through the constantly transient current of three-phase PWM rectifier bridge input filter inductance of k, u _Sa(k), u _Sb(k), u _Sc(k) be the k instantaneous voltage of power distribution network points of common connection constantly,

Be k moment reference current value, T _SBe switching device switch periods time, L ₁Be the inductance value of input reactance device, r ₁Equivalent resistance for the input reactance device.

If the function of state of switching device is S in the rectifier bridge of three-phase PWM shown in Fig. 1 _Cx(x=a, b, c), S _Cx=1 be mutually should brachium pontis last brachium pontis switch closure, S _Cx=0 be mutually should brachium pontis following brachium pontis switch closure, and the state of upper and lower two switching devices of same brachium pontis is complementary, makes DC side P point voltage U _P=U _Dc, O point voltage U _o=0, u is arranged during the power distribution network three-phase equilibrium _N=U _o=0, k the switch periods ac output voltage constantly that then can obtain the three-phase PWM rectifier circuit is:

$\left[\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)\end{array}\right]=\left[\begin{array}{c}{S}_{\mathrm{ca}}\left(k\right)\\ S_{\mathrm{cb}}\left(k\right)\\ S_{\mathrm{cc}}\left(k\right)\end{array}\right]\&CenterDot;U_{\mathrm{dc}}---\left(6\right)$

The transient current tracking Control rule that can be got the three-phase PWM rectifier bridge by formula (5) and formula (6) is:

$\left\{\begin{array}{c}{S}_{\mathrm{ca}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)]\\ S_{\mathrm{cb}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)]\\ S_{\mathrm{cc}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)]\end{array}\right.---\left(7\right)$

Three-phase PWM rectifier bridge control system theory diagram shown in (a) part among Fig. 2, this prime control system be with dc voltage stable be controlled to be outer shroud, current tracking be controlled to be in the double loop control of ring.

(2) back level two phase quadrature inverter circuit control methods

Can be listed as voltage equation and the magnetic linkage matrix equation of writing electromagnetic stirrer in the static α β coordinate system according to Fig. 1:

$\left\{\begin{array}{c}{u}_{\mathrm{\&alpha;}}={\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&alpha;}})\&CenterDot;i_{\mathrm{\&alpha;}}+\frac{{\mathrm{d\&psi;}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}\\ u_{\mathrm{\&beta;}}={\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&beta;}})\&CenterDot;i_{\mathrm{\&beta;}}+\frac{{\mathrm{d\&psi;}}_{\mathrm{\&beta;}}}{\mathrm{dt}}\\ u_c={\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2\frac{{\mathrm{di}}_c}{\mathrm{dt}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2\&CenterDot;i_c\end{array}\right.---\left(8\right)$

$\left\{\begin{array}{c}{\mathrm{\&psi;}}_{\mathrm{\&alpha;}}=L_{\mathrm{\&alpha;}}\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}\\ {\mathrm{\&psi;}}_{\mathrm{\&beta;}}=L_{\mathrm{\&beta;}}\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}\end{array}\right.---\left(9\right)$

In the formula, u _x(x=α, β c) are the output voltage instantaneous value of back level two phase quadrature inverter circuits; ψ _x(x=α β) is the magnetic linkage of electromagnetic stirrer two phase coils; i _x(x=α, β c) are the transient current of level two phase quadrature inverter circuit output inductors after flowing through; r _x(x=2, α β) are respectively the equivalent resistance of back level two phase quadrature inverter circuit output inductors, the α phase of electromagnetic stirrer and the equivalent resistance of β phase coil.

Can find out from formula (9), as long as feed equal and opposite in direction mutually with in β two phase windings mutually at α, the alternating current that the phase phasic difference is 90 °, this moment, electromagnetic stirrer two phase windings will produce a rotating magnetic field in the space.

Formula (9) substitution formula (8) can be obtained:

$\left\{\begin{array}{c}{u}_{\mathrm{\&alpha;}}=({\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&alpha;}})\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&alpha;}})\&CenterDot;i_{\mathrm{\&alpha;}}\\ u_{\mathrm{\&beta;}}=({\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&beta;}})\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&beta;}})\&CenterDot;i_{\mathrm{\&beta;}}\\ u_c={\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2\&CenterDot;\frac{{\mathrm{di}}_c}{\mathrm{dt}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2\&CenterDot;i_c\end{array}\right.---\left(10\right)$

Constantly formula (10) is carried out discretization K switch periods, and the current tracking control law that can obtain back level two phase quadrature inverter circuits according to prime three-phase rectifier transient current control law derivation method is suc as formula shown in (11).

$\left\{\begin{array}{c}{S}_{\mathrm{\&alpha;}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&alpha;}}}{T_S}\&CenterDot;i_{\mathrm{\&alpha;}}^{*}\left(k\right)+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&alpha;}}-\frac{{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&alpha;}}}{T_S})\&CenterDot;i_{\mathrm{\&alpha;}}\left(k\right)]\\ S_{\mathrm{\&beta;}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&beta;}}}{T_S}\&CenterDot;i_{\mathrm{\&beta;}}^{*}\left(k\right)+({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_{\mathrm{\&beta;}}-\frac{{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HDA0000156255080000011.png"\; state="\{\{state\}\}">L</figure-callout>}}_2+L_{\mathrm{\&beta;}}}{T_S})\&CenterDot;i_{\mathrm{\&beta;}}\left(k\right)]\\ S_c\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{L_2}{T_S}\&CenterDot;i_c^{*}\left(k\right)+(r_2-\frac{L_2}{T_S})\&CenterDot;i_c\left(k\right)]\end{array}\right.---\left(11\right)$

The control system schematic diagram of back level two phase quadrature inverter circuits shown in (b) part among Fig. 2, this back level control system be after the tracking Control of grade two phase quadrature inverter circuit output currents be target.

(Three) the expected value signal

and

access

Owing to through instantaneous active power p and the instantaneous reactive power q that input inductance flows into the three-phase PWM rectifier bridge be from electrical network PCC:

$p=\sqrt{\frac{3}{2}}{u}_s{i}_{\mathrm{cd}}---\left(12\right)$

$q=\sqrt{\frac{3}{2}}{u}_s{i}_{\mathrm{cq}}---\left(13\right)$

Therefore, control three-phase PWM rectifier bridge AC side transient current value i _Cd, i _CqCan realize control to p, q.

In order to guarantee U _DcRemain unchanged, need correspondingly control i _CdFollow the tracks of the variation of p,, adopt the PI controller promptly can realize not having steady-state error control again because three-phase PWM rectifier bridge dc voltage is a DC quantity.Therefore, can adopt a PI controller to obtain

suc as formula shown in (14):

$i_{\mathrm{cd}}^{*}=k_p\&CenterDot;(U_{\mathrm{dc}}^{*}-U_{\mathrm{dc}})+k_i\&CenterDot;\&Integral;(U_{\mathrm{dc}}^{*}-U_{\mathrm{dc}})\mathrm{dt}---\left(14\right)$

Reactive current desired value

is obtained according to the dq conversion with the outside threephase load electric current of the net side of two phase quadrature inverter systems by electromagnetic agitation:

$\left[\begin{array}{c}{i}_d\\ i_{\mathrm{cq}}^{*}\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{ccc}\sin \mathrm{\&omega;t}& \sin (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& \sin (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\\ -\cos \mathrm{\&omega;t}& -\cos (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_{\mathrm{La}}\\ i_{\mathrm{Lb}}\\ i_{\mathrm{Lc}}\end{array}\right]=T_{\mathrm{dq}}\&CenterDot;\left[\begin{array}{c}{i}_{\mathrm{La}}\\ i_{\mathrm{Lb}}\\ i_{\mathrm{Lc}}\end{array}\right]---\left(15\right)$

Will

after dq inverse transform that can be phase PWM rectifier circuit current expectations signal

and

as formula (16) below:

$\left[\begin{array}{c}{i}_{\mathrm{ca}}^{*}\\ i_{\mathrm{cb}}^{*}\\ i^{\mathrm{cc}}\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{cc}\sin \mathrm{\&omega;t}& -\cos \mathrm{\&omega;t}\\ \sin (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)\\ \sin (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_{\mathrm{cd}}^{*}\\ i_{\mathrm{cq}}^{*}\end{array}\right]=T_{\mathrm{dq}}^{-1}\&CenterDot;\left[\begin{array}{c}{i}_{\mathrm{cd}}^{*}\\ i_{\mathrm{cq}}^{*}\end{array}\right]---\left(16\right)$

As can beappreciated from fig. 1, drive current outputs to third phase c mutually on the bridge mutually with behind the β phase winding through electromagnetic stirrer α, and c is the common current phase mutually.Therefore; α phase drive current desired value is the low frequency simple sinusoidal alternating current of a given frequency and amplitude; β phase drive current desired value be one with α mutually the drive current desired value differ the simple sinusoidal alternating current at 90 degree phase angles; The target current of c phase is by α phase and β target current summation back negate acquisition mutually, and according to above-mentioned principle, two phase quadrature inverter output current desired values can be expressed as form shown in the formula (17):

$\left\{\begin{array}{c}{i}_{\mathrm{\&alpha;}}^{*}=I^{*}\&CenterDot;\sin \mathrm{\&omega;t}\\ i_{\mathrm{\&beta;}}^{*}=I^{*}\&CenterDot;\cos \mathrm{\&omega;t}\\ i_{\mathrm{\&alpha;}}^{*}=-(I^{*}\&CenterDot;\sin \mathrm{\&omega;t}+I^{*}\&CenterDot;\cos \mathrm{\&omega;t})\end{array}\right.---\left(17\right)$

In the formula, I ^*Be the amplitude of inverter desired output electric current, frequencies omega is the inverter output current frequency.

Three phase transformations, the two phase quadrature inverter composite control methods with no-power compensation function just can be realized in the transient current tracking Control rule formula (7) that the current target value branch that obtains according to formula (16) and (17) is admitted to prime three-phase PWM rectifier bridge and the back grades two current tracking control law formula (11) of quadrature inverter circuit mutually.

Claims (4)

1. three phase transformations, two phase quadrature inverter composite control methods with no-power compensation function; Based on three phase transformations, two phase quadrature inverter systems; Three phase transformations, two phase quadrature inverter systems comprise three phase transformations, two phase quadrature inverters, three phase mains and electromagnetic stirrer; Three phase transformations, two phase quadrature inverters comprise three-phase PWM rectifier bridge and three brachium pontis inverter bridge; Three phase mains is connected with the three-phase PWM rectifier bridge through inductance, and the three-phase PWM rectifier bridge is connected with three brachium pontis inverter bridge, and each is connected with a single-phase load mid point of two brachium pontis through outputting inductance in the three brachium pontis inverter bridge; The mid point of the 3rd brachium pontis is connected with the earth connection of electromagnetic stirrer through outputting inductance, is parallel with load between three phase mains and the three-phase PWM rectifier bridge; Said three-phase PWM rectifier bridge comprises brachium pontis and dc bus capacitor branch road of three parallel connections, and the dc bus capacitor branch road is parallelly connected with brachium pontis, and each brachium pontis comprises the switching device of two series connection; Said three brachium pontis inverter bridge comprise the brachium pontis of three parallel connections, and each brachium pontis comprises the switching device of two series connection, it is characterized in that, this method may further comprise the steps:

1) the external loading electric current with detected three phase transformations, two phase quadrature inverter system net sides passes through the dq conversion, obtains the reactive current desired value

of three-phase PWM rectifier bridge input side

2) desired value of three-phase PWM rectifier bridge dc capacitor voltage and the difference of value of feedback are obtained active current desired value

through the PI controller

2)

and

After dq transform has been phase PWM Rectifier input side of the three-phase instantaneous target current value

3)

obtains the control signal of switching device in the three-phase PWM rectifier bridge through dead beat current controller and PWM modulator, thereby realizes the stable control of dc capacitor voltage and three-phase PWM rectifier bridge net side external loading is carried out reactive power compensation;

4) the desired output current amplitude with three phase transformations, two phase quadrature inverter systems multiply by a SIN function and a cosine function respectively; Obtain target current value

and

that three brachium pontis inverter bridge wherein obtain three brachium pontis inverter bridge third phases after the target current value of two phases and

and

sum negate and obtain the control signal of switching device in the three brachium pontis inverter bridge, thereby realize the tracking Control of two phase quadrature currents of three brachium pontis inverter bridge through dead beat current controller and PWM modulator.

2. three phase transformations, the two phase quadrature inverter composite control methods of band no-power compensation function according to claim 1 is characterized in that said switching device is IGBT or SPM IPM.

3. three phase transformations, the two phase quadrature inverter composite control methods of band no-power compensation function according to claim 1 is characterized in that the expression formula of said dead beat current controller is:

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)\end{array}\right.,$

U wherein _Ia(k), u _Ib(k), u _Ic(k) be k three-phase PWM rectifier bridge AC side instantaneous voltage constantly, i _Ca(k), i _Cb(k), i _Cc(k) be flow through the constantly transient current of three-phase PWM rectifier bridge input filter inductance of k, u _Sa(k), u _Sb(k), u _Sc(k) be the k instantaneous voltage of power distribution network points of common connection constantly,

Be k moment reference current value, T _SBe switching device switch periods time, L ₁Be the inductance value of input reactance device, r ₁Equivalent resistance for the input reactance device.

4. three phase transformations, the two phase quadrature inverter composite control methods of band no-power compensation function according to claim 1 is characterized in that in the said step 4), the frequency of SIN function and cosine function is the output current frequency of three brachium pontis inverter bridge.

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