CN101170295A - A magnetic levitation reaction flywheel motor control system - Google Patents

Abstract

A magnetic levitation reaction flywheel motor control system, which is used for the control of the high-precision attitude control executive mechanism of a new generation of high-stability satellites—the three-phase permanent magnet brushless DC motor for the magnetic levitation reaction flywheel. It is mainly composed of a digital signal processor ( DSP) as the core controller, power amplifier, upper bridge arm drive circuit of power amplifier, lower bridge arm drive circuit of power amplifier, resonant soft switch control circuit, three-phase permanent magnet brushless DC motor, current detection circuit, 28V Composed of DC constant power supply, +15V DC constant power supply, +5V DC constant power supply. The invention controls the on-off of each power switch tube of the resonant soft switch through an isolated drive circuit without demodulation, thereby realizing the high reliability and low power consumption operation of the drive circuit of the magnetic levitation reaction flywheel motor control system.

Description

A magnetic levitation reaction flywheel motor control system

technical field

The invention relates to a magnetic levitation reaction flywheel motor control system, which is mainly used to realize the high reliability and low power consumption of the power switch tube gate of the magnetic levitation reaction flywheel motor control system, which is a long-life and high-precision attitude control actuator of a new generation of satellites drive.

Background technique

As the actuator of the high-precision and high-stability attitude control system of the new generation satellite platform, the magnetic levitation reaction flywheel has the advantages of high angular momentum output precision, good control linearity, strong anti-interference and fast response speed, etc. It has been gradually applied in the world, and will become the preferred actuator for the new generation of high-precision and high-stability satellite platforms in my country.

The highly reliable drive of the new generation of satellite attitude control actuators is the basic requirement for space applications. Improving the reliability of the reaction flywheel motor drive system and reducing the drive power consumption are always the goals pursued by space applications.

The magnetic levitation reaction flywheel motor generally adopts a three-phase permanent magnet brushless DC motor, and the drive of the three-phase permanent magnet brushless DC motor control system mostly adopts a MOSFET drive circuit. Power MOSFET has the remarkable characteristics of fast switching speed, good high-frequency performance, high input impedance, low driving power, and no secondary breakdown problem. There are currently four commonly used MOSFET drive circuits: (1) gate-source floating power supply drive, (2) transformer isolation drive, (3) bootstrap drive, and (4) integrated circuit drive. First, the gate-source floating power supply drive (as shown in Figure 7(a)) has full gate control for an indeterminate time period, and each high-side MOSFET requires an isolated power supply that performs level shifting on ground-referenced signals. complex, and the use of optical isolators is limited in bandwidth and noise sensitivity; the transformer isolation drive (as shown in Figure 7(b)) isolates the high-frequency PWM signal added to the primary side of the isolation pulse transformer, and directly rectifies the secondary to obtain Self-sufficient power supply, this rectification method only retains the signal waveform with positive amplitude and filters out the signal waveform with negative amplitude, and needs to demodulate the high-frequency modulation signal, and the size of the transformer increases significantly when the frequency of the modulation signal decreases (The volume of the transformer is inversely proportional to the square of the signal frequency), its control is complex, the volume is large, and the power consumption is large; the duty cycle and turn-on time of the bootstrap drive (as shown in Figure 7(c)) are limited by the refresh of the bootstrap capacitor , the capacitor is charged from the high-voltage line, the power consumption may be large, and the level converter is required to be more complicated; for the integrated circuit driver, it uses an integrated driver chip to drive, because the aerospace chip is limited, and the ordinary integrated driver chip is in terms of reliability. It cannot meet the requirements of space applications. For example, the integrated driver chip will produce logic flips under the radiation of space particles, and the flipping from high level to low level will cause the motor to malfunction. Moreover, the signal modulation frequency of the integrated drive chip is low, and the control method for soft switching overcomes the switching loss of the power switch tube, and can increase the frequency of the drive signal. However, at present, the chip with the highest modulation frequency of the integrated driver chip is IR21366. Its turn-on transmission delay time is 250ns, and the turn-off transmission delay time is 180ns. The width of the high-frequency PWM signal is greater than 1us. Therefore, its highest modulation frequency is 699khz, which is not suitable for high-frequency applications.

In the drive circuit of the three-phase permanent magnet brushless DC motor control system, the gate drive of the power MOSFET driven by the upper bridge arm must be guaranteed: the gate voltage must be 10-15V higher than the drain voltage, and the gate voltage is logically Must be manageable. Therefore, the control signal must be level-shifted to the source potential of the high-side power device. Therefore, the MOSFET tube needs to be floating at high voltage, which requires electrical isolation between the input and output terminals of the MOSFET driver. The drive is generally isolated in two ways: using a photocoupler for isolation or using a pulse transformer for electrical isolation. The disadvantage of using optocoupler for isolation is that the response is relatively slow and has a large delay time (high-speed optocoupler is generally greater than 500ns) and the optocoupler needs isolated auxiliary power supply. There are three approaches to drive IG power devices with pulse transformer isolation: passive, active, and self-supplied. The passive method does not require a separate driving power supply, but the primary input signal of the transformer needs to be a high-power signal, otherwise the waveform between the gate and the source will be significantly deformed; the active method has a better driving waveform, but an isolated auxiliary power supply is also required The existing self-sufficient power supply method is to perform high-frequency modulation on the PWM drive signal, and the high-frequency modulation signal after isolation and rectification by the transformer needs to be demodulated into a low-frequency signal, which is large in size and complex in design and control, and consumes less power. higher.

At present, the power MOSFETs of each bridge arm of the magnetic levitation reaction flywheel motor control system generally work in the hard switching mode, and the large switching voltage stress, current stress, and high voltage change rate du/dt and current change rate di/dt limit its high frequency. With the development of modernization, it is difficult to reduce the size and weight of the converter, and it will generate a lot of electromagnetic interference. The soft switching technology, with its low switching stress, zero switching loss and small du/dt and di/dt, greatly increases the switching frequency of power electronic devices.

Contents of the invention

The technical problem solved by the present invention is: to overcome the disadvantages of the motor drive system existing in the prior art, such as complex control, large volume, and large power consumption, and the reliability and low modulation frequency of the integrated drive circuit, and the three-phase permanent magnet The inverter circuit of the brushless DC motor adopts the disadvantages of hard switching mode power consumption, harmonic content, and high switching stress. A magnetic levitation reaction flywheel motor drive system that drives each power switch tube of a resonant soft switch.

Technical solution of the present invention: a magnetic levitation reaction flywheel motor control system includes: an isolation drive circuit, a controller with DSP as the core, a speed detection link, a current detection link, a power amplification circuit, a resonant soft switch control circuit, a three-phase The permanent magnet brushless DC motor is characterized in that: the isolated drive circuit is composed of discrete components, mainly including: a primary side voltage doubler rectifier circuit, a push-pull circuit, a pulse transformer, a secondary side voltage doubler rectifier circuit, a high-pass filter circuit, The discharge triode, the capture unit module of the controller with DSP as the core captures the signal of the motor rotor position sensor to obtain the current rotor position, and the PWM waveform generation module generates nine PWM drive signals, and the primary voltage doubler rectifier circuit receives the PWM drive The signal rectifies the drive signal to generate a unipolar PWM drive signal; the push-pull circuit receives the PWM drive signal rectified by the primary side voltage doubler rectifier circuit, and increases its drive power; the pulse transformer receives the PWM drive signal after the push-pull circuit increases the power. The drive signal is isolated by the pulse transformer and becomes a secondary drive signal; the secondary voltage doubler rectifier circuit receives the isolated drive signal and rectifies the isolated drive signal; the high-pass filter circuit receives the rectified drive signal and filters out the additional Low-frequency noise on the rectified driving signal; the filtered driving signal is used to control the opening and closing of the discharge transistor.

The isolated drive circuit also includes: a kickback diode and a decoupling capacitor. The kickback diode is connected to the secondary voltage doubler rectifier circuit for eliminating the interference from the secondary side of the pulse transformer to the primary side. Decoupling capacitors are connected to the debounce diodes to eliminate the effects of high-frequency AC-coupled signals.

The parameters of the isolated drive circuit can be selected according to the following principles: the primary side voltage doubler rectifier circuit and the secondary side voltage doubler rectifier circuit are respectively composed of a capacitor and a diode. The value of the capacitor determines the charging and discharging time of the capacitor. Generally, a smaller capacitor value is selected, and the optional range is 10 ⁴ pF to 10 ⁵ pF. The diode is selected according to the rated voltage. When the triode of the driving circuit is turned on, the reverse voltage that the diode bears is the rated voltage of 15V, and the reverse withstand voltage that the diode can withstand should be 1.8 to 2.5 times the rated voltage. The push-pull circuit is composed of NPN transistor Q1 and PNP transistor Q2, the base and emitter of the two transistors are connected together. Transistors Q1 and Q2 are selected according to the modulation frequency of the signal. The modulation frequency is generally 1 Hz to 1 MHz. It is required that the switching frequency of the triodes Q1 and Q2 be higher than the frequency of the modulating signal. The pulse transformer is composed of LC circuit, its capacitance and inductance value is selected according to the modulation frequency, and its modulation frequency $f=\frac{1}{2\mathrm{\π}\sqrt{\mathrm{LC}}},$ The transformer turns ratio is 1:1. The high-pass filter circuit takes the -3dB frequency of the amplitude attenuation of the drive signal entering the voltage doubler rectifier circuit as the cut-off frequency, and the resistance and capacitance values of the high-pass filter circuit are determined according to the limit of the cut-off frequency.

The principle of the present invention is: the capture unit module of the controller 1 with DSP as the core captures the motor rotor position sensor signal to obtain the current rotor position, and generates nine PWM drive signals by the PWM waveform generation module. The signal is controlled by the isolation drive circuit 4 and the power amplifier circuit 5 to control the on-off of each power MOSFET switch tube in the main circuit, and the six PWM drive signals are used to control the inverter circuit of the three-phase permanent magnet brushless DC motor (as shown in Figure 2 10) The on-off of each power switch tube of the upper and lower bridge arms. In addition, the three-way PWM driving signals are used to control the on-off of each power switch tube of the resonant circuit 9. By controlling each power switch tube, each power switch tube conducts the capacitance Cr (as shown in Figure 2) through the resonant circuit before it is turned on and when it is turned off. Charging and discharging make the voltage Ucr across the capacitor become zero, so as to realize the zero-voltage turn-on and turn-off of the power switch tube. The power switch tubes of the upper and lower bridge arms of the inverter circuit 10 are turned on and off in a certain order, so as to realize the forward operation and reverse operation of the three-phase permanent magnet brushless DC motor. The rotor position of the three-phase permanent magnet brushless DC motor is detected through the speed detection link 2, and the rotor speed feedback is obtained. The speed setting is compared with the speed feedback through the controller 1 with DSP as the core, so as to realize the three-phase permanent magnet Speed loop control for brushless DC motors. The current detection link 3 detects the bus current of the three-phase permanent magnet brushless DC motor, and the controller 1 with DSP as the core converts the detected current value into a digital quantity through the analog-to-digital conversion (A/D) module, thereby performing three-phase permanent magnet brushless DC motor bus current. Current Loop Control of Magnetic Brushless DC Motors.

The isolated driving circuit 4 is mainly composed of a primary side voltage doubler rectifier circuit 12, a push-pull circuit 13, a pulse transformer 14, a secondary side voltage doubler rectifier circuit 15, a high-pass filter circuit 16, a debounce diode 17, a decoupling capacitor 18, and a discharge triode 19 compositions. The following takes one PWM drive signal as an example to illustrate the working process of the drive system.

The PWM waveform generation module of the controller 1 with DSP as the core generates the PWM drive signal, and the primary side voltage doubler rectifier circuit 12 receives the PWM drive signal and rectifies the drive signal to generate a unipolar PWM drive signal; the push-pull circuit 13 receives the PWM drive signal via the primary side The PWM drive signal rectified by the voltage doubler rectifier circuit 12 increases its drive power; the pulse transformer 14 receives the PWM drive signal after the power is increased by the push-pull circuit 13, and is isolated by the pulse transformer 14 to become a secondary drive signal; The voltage doubler rectifier circuit 15 receives the isolated drive signal and rectifies the isolated drive signal; the high-pass filter circuit 16 receives the rectified drive signal and filters out low-frequency noise added to the rectified drive signal. The drive signal filtered by the high-pass filter circuit 16 is used to control the turn-on and turn-off of the discharge transistor 19 . When the level of the drive signal was low, point A (as shown in Figure 3) was low level, the control main loop power switch tube was turned off, and the discharge transistor 19 was opened, and now the junction capacitance of the power switch tube was discharged through the discharge transistor 19; When the driving signal is at a high level, the point A is at a high level to control the opening of the power switch tube of the main circuit, and the discharge transistor 19 is turned off. The anti-kickback diode 17 is connected with the secondary side voltage doubler rectifier circuit 15, and a reverse peak will appear when the switch tube is turned on or off. The anti-kickback diode 17 is used to eliminate the interference from the secondary side of the pulse transformer to the primary side. The decoupling capacitor 18 is connected to the debounce diode 17 for eliminating the influence of the high-frequency AC coupling signal.

Among them, the primary side voltage doubler rectifier circuit 12 and the secondary side voltage doubler rectifier circuit 15 are each composed of a capacitor and a diode, which function as a charge pump to rectify the drive signal to generate a unipolar PWM drive signal and reduce the voltage of the signal doubled. The push-pull circuit 13 is composed of an NPN transistor Q1 and a PNP transistor Q2, the bases and emitters of the two transistors are connected together. Q1 turns on when the signal is in the positive half cycle and Q2 turns on when the signal is in the negative half cycle. The push-pull circuit 13 is used to increase the driving power of the driving signal. The isolation drive circuit 4 uses the pulse transformer 14 to isolate and drive the insulated gate power device. Driven by a self-sufficient power supply method without demodulation, the pulse transformer 14 is composed of an LC circuit, the ratio of the primary and secondary coils of the pulse transformer is 1:1, the PWM supplied to the primary of the pulse transformer is modulated at high frequency, and the secondary is directly rectified to obtain high-frequency self-sufficiency The power supply does not need to be demodulated, and the volume of the transformer is inversely proportional to the square of the frequency of the modulating signal. The modulation signal can directly control the on-off of each power switch tube in the main circuit without high-frequency switching loss.

The working principle of the control circuit of the resonant soft switch is: before the switching tube is turned on and when it is turned off, the capacitor Cr (as shown in Figure 2) is charged and discharged through the resonant circuit, so that the voltage Ucr at both ends of the capacitor becomes zero, thereby realizing the power switch tube zero voltage turn-on and turn-off. The working process and principle of the resonant soft switch will be described below by taking one cycle (the PWM signal changes from 1 to 0, and then from 0 to 1) as an example. Since the phase current of the inverter is a square wave current and the frequency of the resonant circuit is very high, the inverter can be analyzed as a constant current source. Its simplified circuit diagram is shown in Figure 4.

In one working cycle, the trigger signals and corresponding voltage and current waveforms of the corresponding power switch tubes (Sa, Sb, Sl) are shown in FIG. 5 . Each time period corresponds to a working state. From Figure 5, it can be seen that there are six working states of the soft switch at resonance, as shown in Figure 6. State 1 corresponds to Figure 6(a), which is the normal working state when the PWM signal is 1; State 2 corresponds to Figure 6(b), which is the process of discharging Cr when the PWM signal is about to change from 1 to 0; State 3 corresponds to the 6(c) is the normal diode freewheeling process when the PWM signal is 0; state 4 corresponds to Figure 6(d), which is the process of charging Lr and simultaneously diode freewheeling when the PWM changes from 0 to 1; state 5 corresponds to Figure 6(d) 6(e) is the charging process of the resonant circuit to Cr after the PWM signal changes from 0 to 1; state 6 corresponds to Figure 6(f) when the PWM signal is 1 and the DC power supply and the resonant circuit supply power to the inverter at the same time.

Compared with the prior art, the present invention has the following advantages: (1) The drive circuit is driven by transformer isolation, and its structure is simple without the need for isolated power supply or level conversion. (2) The driving circuit adopts a driving method that does not require demodulation of the self-sufficient power supply. The high-frequency self-sufficient power supply is obtained through voltage doubling and rectification at the secondary side of the transformer without demodulation. The volume of the transformer is inversely proportional to the square of the frequency of the modulation signal, and the higher the frequency of the modulation signal The smaller the volume of the transformer, the soft switching control method of each power switch tube in the main circuit eliminates switching loss, and its high-frequency modulation signal can directly control the on-off of each power switch tube in the main circuit without high-frequency switching loss. The pulse transformer adopts a driving method without demodulation self-sufficient power supply, has the advantages of small size, low power consumption, simple control and design. (3) The drive circuit adopts a drive method that does not require demodulation of a self-sufficient power supply. The primary side and the secondary side use a voltage doubler rectifier circuit to rectify the drive signal and double the voltage of the drive signal, which makes full use of the signal waveform and increases The power of the original driving signal is reduced. However, in the existing pulse transformer drive technology, the rectification method retains the signal waveform with positive amplitude and filters out the waveform with negative amplitude, so that the signal voltage amplitude remains unchanged and the power decreases. (4) The drive circuit adopts transformer isolation drive with high reliability, and there will be no abnormal reversal under the action of space particle radiation. (5) The drive circuit adopts transformer isolation to drive its high modulation frequency. For the soft switching control method without switching loss, the switching frequency of the power switch tube can be raised to a high level without switching loss. (6) The drive circuit adopts transformer isolation to drive fast response, so the transmission delay time is short, and its output stage does not need auxiliary power supply. (7) The present invention further merges the isolated drive method without demodulation self-sufficient power supply with the soft switch technology, and the magnetic levitation reaction flywheel three-phase permanent magnet brushless DC motor control system adopts the control method of soft switch, which is isolated by the transformer without demodulation The high-frequency drive signal provided by the driver controls the on-off of each power switch tube. The resonant circuit is used to realize the zero-voltage turn-on and turn-off of its switching tube. Therefore, the switching loss of the power switch tube is greatly reduced, and the working efficiency of the motor is improved. And reduce switching voltage stress current stress and high du/dt and di/dt.

Description of drawings

Fig. 1 three-phase permanent magnet brushless DC motor control diagram of the present invention;

Fig. 2 is a three-phase permanent magnet brushless DC motor of the present invention and a main circuit diagram of a control circuit;

Fig. 3 is a power drive circuit diagram of the present invention;

Fig. 4 is a simplified main circuit diagram of the present invention.

Fig. 5 is the voltage and current waveform diagram of the resonant circuit of the present invention;

Fig. 6 is six kinds of working state diagrams of the resonant type soft switch of the present invention; Wherein: (a) is the normal working state diagram when the PWM signal is 1, (b) is that the Cr is discharged when the PWM will change from 1 to 0 Working state diagram, (c) is the working state diagram of diode normal freewheeling when the PWM signal is 0, (d) is the working state diagram of charging Lr and diode freewheeling when the PWM signal changes from 0 to 1, (e) is After the PWM signal changes from 0 to 1, the working state diagram of the resonant circuit charging Cr, (f) is the working state diagram of the DC power supply and the resonant circuit supplying power to the inverter when the PWM signal is 1.

Fig. 7 (a) is the gate-source floating power supply driving diagram of the present invention, (b) is the transformer isolation driving diagram of the present invention, and (c) is the bootstrap driving diagram of the present invention

Detailed ways

As shown in Figure 1, the present invention mainly consists of an isolated drive circuit 4, a controller 1 with DSP as the core, a speed detection link 2, a current detection link 3, a power amplifier circuit 5, a resonant soft switch control circuit 6, a three-phase permanent magnet Composed of 7 brushed DC motors. The isolated driving circuit 4 is mainly composed of a primary side voltage doubler rectifier circuit 12, a push-pull circuit 13, a pulse transformer 14, a secondary side voltage doubler rectifier circuit 15, a high-pass filter circuit 16, a debounce diode 17, a decoupling capacitor 18 and a discharge transistor 19 composition.

The capture unit module of the controller 1 with DSP as the core captures the signal of the motor rotor position sensor to obtain the current rotor position, and the PWM waveform generation module generates nine PWM drive signals, and the primary side voltage doubler rectifier circuit 12 converts the nine channels The PWM driving signal is rectified, and the push-pull circuit 13 is used to increase the driving power of the nine PWM driving signals. The pulse transformer 14 receives the nine-way PWM driving signals after the power is increased by the push-pull circuit 13, and the nine-way PWM driving signals isolated by the pulse transformer 14 are rectified by the secondary voltage doubler rectifier circuit 15, and then passed through the high-pass filter circuit 16 The low-frequency noise is filtered out, and the filtered drive signal controls the turn-on and turn-off of the discharge transistor 19 . In this way, the grid control signal of the power switch tube is generated at the collector of Q3 (A shown in Figure 3), and six of the nine grid control signals are used to control the inverter circuit of the three-phase permanent magnet brushless DC motor The on-off of each power switch tube of the upper and lower bridge arms. The other three grid control signals are used to control the on-off of each power switch tube in the resonant circuit. By controlling each power switch tube, each power switch tube charges and discharges the capacitor Cr through the resonant circuit before it is turned on and when it is turned off. The voltage Ucr at the end becomes zero, so that the zero-voltage turn-on and turn-off of the power switch tube is realized. The power switch tubes of the upper and lower bridge arms of the inverter circuit are turned on and off in a certain order, so as to realize the forward operation and reverse operation of the three-phase permanent magnet brushless DC motor. By detecting the rotor position of the three-phase permanent magnet brushless DC motor, the rotor speed feedback is obtained, thereby performing the speed loop control of the three-phase permanent magnet brushless DC motor. The bus current of the three-phase permanent magnet brushless DC motor is detected by the current detection device, and the current loop control of the three-phase permanent magnet brushless DC motor is carried out.

The selection principle of each parameter of the isolated drive circuit: the diodes D1 and D2 of the voltage doubler rectifier circuit are selected according to the rated voltage. It is 1.8 to 2.5 times of the rated voltage, so the reverse withstand voltage value of diodes D1 and D2 is generally selected as 30V. If the conditions permit, the reverse withstand voltage value of the diode is larger, the better. In this implementation, IN4148 is selected for its higher withstand voltage value. , the reverse voltage that can withstand is 75V, and the current that can withstand is 150mA; the value of the capacitor determines the charging and discharging time of the capacitor, generally choose a smaller capacitor value, the optional range is 10 ⁴ pF ~ 10 ⁵ pF, this The implementation scheme selects 10 ⁴ pF; the push-pull circuit triode Q1 and Q2 are selected according to the modulation frequency, the modulation frequency is generally 1Hz to 1MHz, the switching frequency of TIP9013 is 100MHz, and the switching frequency of S9012 is 30MHz, which can meet the general modulation frequency requirements, so this implementation scheme TIP9013 is selected for Q1, and S9012 is selected for Q2; the capacitance and inductance value of the LC circuit of the pulse transformer is selected according to the modulation frequency, and the modulation frequency $f=\frac{1}{2\mathrm{\π}\sqrt{\mathrm{LC}}},$ In this implementation, the modulation frequency is set to 160KHz, the capacitance value is 10 ³ pF, the inductance value is 1mH, and the transformer does not need to step up and down, so its turns ratio is 1:1; the cutoff frequency of the high-pass filter circuit is its signal amplitude attenuation The frequency corresponding to the -3dB point. The resistance and capacitance values of the high-pass filter circuit are determined according to the limit of the cut-off frequency; the decoupling capacitor is generally selected as 10 ⁴ pF to 10 ⁵ pF; the switch tube Q3 is selected from S9012; the reverse voltage borne by the diode D3 is also the rated voltage value, so the reverse The withstand voltage value should also be greater than or equal to 30V. In this implementation, IN5819 is selected, which can withstand a reverse voltage of 40V and a current of 1A.

The selection principle of the control circuit parameters of the resonant soft switch:

State 1: The time period corresponding to Figure 5 is 0<t<t0. In this state, SL is on, Sa and Sb are off, and the voltage across Cr is Vs. The DC power supply directly supplies power to the inverter.

State 2: The time period corresponding to Figure 5 is to < t < t1. In this state, SL is turned off and Sa triggers conduction; Cr is discharged through the resonant circuit and the inverter circuit at the same time, and at the time when Cr discharges to zero, the PWM signal is generated by 1 becomes 0, realizing the zero-voltage turn-off of the power switch tube of the inverter. Therefore, in order to accurately determine the triggering moment of Sa, it is necessary to calculate the discharge time of Cr, that is, trigger Sa one discharge time earlier before the PWM signal changes from 1 to 0. Define time t0 as zero time, constant current source current as Io, resonant inductance and wire resistance as RL, because the resonant frequency Wr is very high, WrLr>>RL, so the voltage drop and loss caused by RL can be ignored. The equivalent equation of the circuit is obtained from Figure 6(b):

$\left\{\begin{array}{c}{\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; Cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; Lr</span>}\frac{{\mathrm{<span\; class="notranslate">\; di</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}\\ {\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>\frac{{\mathrm{<span\; class="notranslate">\; du</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\\ {\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; vs.</span>}<span\; class="notranslate">\; ,</span>\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}\end{array}\right.<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

Solving the system of equations gives:

$\left\{\begin{array}{c}{\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; K</span>}\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; )</span>\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; K</span>}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; Cr</span>}}{\mathrm{<span\; class="notranslate">\; Lr</span>}}}\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; )</span>\end{array}\right.<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; )</span>$

in:

$\mathrm{<span\; class="notranslate">\; K</span>}<span\; class="notranslate">\; =</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; vs.</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; +</span>\frac{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}\mathrm{<span\; class="notranslate">\; Lr</span>}}{\mathrm{<span\; class="notranslate">\; Cr</span>}}}<span\; class="notranslate">\; ,</span>$ $\mathrm{<span\; class="notranslate">\; Wr</span>}<span\; class="notranslate">\; =</span>\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; LrCr</span>}}}<span\; class="notranslate">\; ,</span>$ $\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; tg</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; (</span>\frac{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}}{\mathrm{<span\; class="notranslate">\; vs.</span>}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; Lr</span>}}{\mathrm{<span\; class="notranslate">\; Cr</span>}}}<span\; class="notranslate">\; )</span>$

Solve the discharge time:

${\mathrm{<span\; class="notranslate">\; t</span>}}_{\mathrm{<span\; class="notranslate">\; sc</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; t</span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; t</span>}\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; \&pi;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; a</span>}}{{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 3</span>}<span\; class="notranslate">\; )</span>$

At time t1, u _cr (t1) = 0, which is substituted into formula (2) to get i _Lr (t1) = 0, so Sa is automatically turned off at time t1.

State 3: The time period corresponding to FIG. 5 is t1<t<t2. In this state, Sa, Sb, and SL are all in the cut-off state, and the phase current continues to flow normally through the diode.

State 4: The time period corresponding to Figure 5 is t2<t<t3. In this state, Sb is turned on, SL and Sa are turned off, the power supply charges Lr to i _Lr =I ₀ , and then the PWM signal changes from 0 to 1. Therefore, in order to accurately give the trigger signal of Sb, the charging time of the resonant inductor Lr must be calculated, and the time t2 is taken as zero time, and the equivalent equation of the circuit is obtained from Figure 6(d):

$\mathrm{<span\; class="notranslate">\; Lr</span>}\frac{{\mathrm{<span\; class="notranslate">\; di</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span>$

The initial moment i _Lr (0) = 0, the charging time is obtained

${\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="t\; c"\; filenames="S2007101224067E00021.png,S2007101224067E00031.png"\; state="\{\{state\}\}">t</figure-callout></span><figure-callout\; id="1"\; label="t\; c"\; filenames="S2007101224067E00021.png,S2007101224067E00031.png"\; state="\{\{state\}\}">\; </figure-callout>}}_{\mathrm{<span\; class="notranslate">\; </span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; t</span>}\mathrm{<span\; class="notranslate">\; 3</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="t"\; filenames="S2007101224067E00021.png,S2007101224067E00031.png"\; state="\{\{state\}\}">t</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; Lr</span>}}{\mathrm{<span\; class="notranslate">\; vs.</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 5</span>}<span\; class="notranslate">\; )</span>$

State 5: The time period corresponding to Figure 5 is t3<t<t4. In this state, Sb is on, SL and Sa are still off, the diode stops freewheeling, and the resonant circuit charges Cr and supplies power to the inverter at the same time. In the working state, the absolute value of i _Lr is larger than Io, and Cr is charged until u _cr =Vs, and when t3 is zero, the equivalent equation of the circuit can be obtained from Figure 6(e):

$\left\{\begin{array}{c}{\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; Cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; Lr</span>}\frac{{\mathrm{<span\; class="notranslate">\; di</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}\\ {\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>\frac{{\mathrm{<span\; class="notranslate">\; du</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\\ {\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; vs.</span>}<span\; class="notranslate">\; ,</span>\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\end{array}\right.<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 6</span>}<span\; class="notranslate">\; )</span>$

Solutions have to:

$\left\{\begin{array}{c}{\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; cr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ]</span>\\ {\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; max</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; Cr</span>}}{\mathrm{<span\; class="notranslate">\; Lr</span>}}}\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>\end{array}\right.<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 7</span>}<span\; class="notranslate">\; )</span>$

Calculate the charging time of Lr:

${\mathrm{<span\; class="notranslate">\; t</span>}}_{\mathrm{<span\; class="notranslate">\; cc</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; t</span>}\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; t</span>}\mathrm{<span\; class="notranslate">\; 3</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; \&pi;</span>}}{{\mathrm{<span\; class="notranslate">\; w</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 8</span>}<span\; class="notranslate">\; )</span>$

When Cr is charged to Vs, SI is turned on and enters the next working state, and the voltage at both ends of SI is zero before turning on, realizing zero-voltage turn-on.

State 6: The time period corresponding to Figure 5 is t4<t<t5. In this state, SL is turned on, Sa is turned off, Sb is turned on, and the resonant inductor is discharged until the current is zero. Sb is automatically turned off and enters the next working state. The discharge time of Lr is the same as the charge time of Lr in state 4.

The inverter works cyclically in 6 working states when performing PWM modulation, among which states 1, 2, 5, and 6 work when the PWM signal is 1, and states 3 and 4 work when the PWM signal is 0 , in order to ensure the normal operation of the system, the shortest time when the PWM signal is 1 is the sum of the working hours of states 2, 5, and 6, and the shortest time when the PWM signal is 0 is the working time of 4. When designing the resonant circuit, these several times should be minimized The working time of this working state makes the inverter have a larger PWM modulation range.

Parameter selection of the resonant circuit: The main task of the resonant circuit design is to determine the parameter values of the resonant inductance Lr and the resonant capacitor Cr. When determining the parameters, try to keep the resonant current impact under the condition that the reduction state is not too large. 2, 4 , 5, 6 working hours, in order to ensure that the phase current has good square wave characteristics, the PWM carrier frequency of the inverter is selected as 5kHz, and the resonant frequency f should be much higher than the carrier frequency, which is selected as 100kHz.

Depend on $f=\frac{1}{2\mathrm{\&pi;}\sqrt{\mathrm{LrCr}}}={10}^5$

Obtained: LrCr＝2.536×10 ^-12 (9)

From the formula (5) available state 4, 6 working time decreases with the increase of Lr. The peak value of the resonant current from formula (7):

${\mathrm{<span\; class="notranslate">\; i</span>}}_{\mathrm{<span\; class="notranslate">\; Lr</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; max</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; vs.</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; Cr</span>}}{\mathrm{<span\; class="notranslate">\; Lr</span>}}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 10</span>}<span\; class="notranslate">\; )</span>$

i _Lr (max) decreases with the increase of Lr. According to the rated voltage, rated current, rated speed, phase resistance, and phase inductance of the permanent magnet brushless DC motor, the best resonant inductance and resonant capacitor values can be obtained.

Software implementation: The three-phase permanent magnet brushless DC motor generally adopts the control mode of double closed-loop adjustment of speed and current, and the control of the resonant circuit belongs to the inner loop control of the current loop. The DSP chip model of the control chip is TMS320LF2407A. The minimum clock period of this chip is 0.25us, which can accurately control the working time of the resonant circuit in each working mode. The current closed loop adopts the interruption control method, that is, the current size is controlled by continuously starting the analog-to-digital (AD) conversion of the phase current. In the AD conversion interrupt service program, the duty cycle of PWM is adjusted by the detected phase current value, and the trigger conduction time of Sa, Sb, and SL is calculated according to Figure 5. The power switch tube selected is the power MOSFET model IRF3710.

Claims (8)

1. A magnetic levitation reaction flywheel motor control system includes: an isolation drive circuit (4), a controller (1) with DSP as the core, a speed detection link (2), a current detection link (3), and a power amplifier circuit (5) , a resonant soft switching control circuit (6), a three-phase permanent magnet brushless DC motor (7), characterized in that: the isolated drive circuit (4) is composed of discrete components, mainly including: a primary side voltage doubler rectifier circuit (12), push-pull circuit (13), pulse transformer (14), secondary voltage doubler rectifier circuit (15), high-pass filter circuit (16), discharge triode (19), controller with DSP as the core (1) The capture unit module captures the motor rotor position sensor signal to obtain the current rotor position, and generates nine PWM drive signals by the PWM waveform generation module, and the primary side voltage doubler rectifier circuit (12) receives the PWM drive signal and rectifies the drive signal to generate a single polarity PWM drive signal; the push-pull circuit (13) receives the PWM drive signal rectified by the primary side voltage doubler rectifier circuit (12), and increases its drive power; the pulse transformer (14) receives the push-pull circuit (13) to increase The high-power PWM drive signal is isolated by the pulse transformer (14) and becomes a secondary drive signal; the secondary voltage doubler rectifier circuit (15) receives the isolated drive signal and rectifies the isolated drive signal; the high-pass filter circuit (16) Receive the rectified driving signal, and filter out the low-frequency noise added to the rectified driving signal; the filtered driving signal is used to control the opening and closing of the discharge transistor (19). 2. A kind of magnetic levitation reaction flywheel motor control system according to claim 1, is characterized in that: described isolation driving circuit (4) also comprises: debounce diode (17) and decoupling capacitor (18), so The described debounce diode (17) is connected with the secondary side voltage doubler rectifier circuit (15), and is used to eliminate the interference of the pulse transformer secondary side to the primary side, and the described decoupling capacitor (18) is connected with the debounce diode (17 ) to eliminate the effects of high-frequency AC-coupled signals. 3. A kind of magnetic levitation reaction flywheel motor control system according to claim 1, is characterized in that: described primary side voltage doubler rectifier circuit (12) and secondary side voltage doubler rectifier circuit (15) are respectively composed of a capacitor and a composed of diodes. 4. A magnetic levitation reaction flywheel motor control system according to claim 3, characterized in that: the value of the capacitor determines the charging and discharging time of the capacitor, and the range is 10 ⁴ pF to 10 ⁵ pF; The diode is selected according to the rated voltage. When the triode of the driving circuit is turned on, the reverse voltage that the diode bears is the rated voltage of 15V, and the reverse withstand voltage that the diode can withstand should be 1.8 to 2.5 times the rated voltage. 5. A kind of magnetic levitation reaction flywheel motor control system according to claim 1 or 2, is characterized in that: described push-pull circuit (13) is made up of NPN type triode Q1 and PNP type triode Q2, the base pole of two tubes and the emitter are connected together. 6. A magnetic levitation reaction flywheel motor control system according to claim 5, characterized in that: the triode Q1 or Q2 is selected according to the modulation frequency of the signal, and the modulation frequency is generally 1Hz-1MHz, requiring the triode Q1, Q2 The switching frequency is higher than the frequency of the modulating signal. 7. A kind of magnetic levitation reaction flywheel motor control system according to claim 1 or 2, is characterized in that: the LC circuit capacitance inductance value of described pulse transformer (14) is selected according to the modulation frequency, and its modulation frequency $f=\frac{1}{2\mathrm{\&pi;}\sqrt{\mathrm{LC}}},$ The transformer turns ratio is 1:1. 8. A kind of magnetic levitation reaction flywheel motor control system according to claim 1 or 2, is characterized in that: described high-pass filter circuit (16) enters the amplitude attenuation-3dB frequency of the drive signal of voltage doubler rectifier circuit as The cut-off frequency, the resistance and capacitance values of the high-pass filter circuit are determined according to the limit of the cut-off frequency.

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