CN100559319C

CN100559319C - Be used in the hybrid switched mode/linear power amplifier power supply in the polar transmitter

Info

Publication number: CN100559319C
Application number: CNB2004800333697A
Authority: CN
Inventors: V·G·格里戈尔
Original assignee: Nokia Oyj
Current assignee: Nokia Technologies Oy
Priority date: 2003-09-16
Filing date: 2004-09-16
Publication date: 2009-11-11
Anticipated expiration: 2024-09-16
Also published as: CN1879070A

Abstract

In one aspect, the invention provides the DC-DC converter, it has the switching mode part that is coupling between DC source and the load, and switching mode partly provides x the output power of amount; And it also have and DC source and load between the linear model part of switching mode part parallel coupled, linear model partly provides y the output power of amount.X is more preferably greater than y, and can be the ratio of special applications constrained optimization x and y.On the other hand, the invention provides the RF transmitter (TX) that is coupled to antenna.TX has polar structure, and comprises amplitude modulation (AM) path of being coupled to power amplifier (PA) power supply and be coupled to phase modulation (PM) path that PA imports.

Description

Be used in the hybrid switched mode/linear power amplifier power supply in the polar transmitter

Technical field

The present invention relates generally to the DC-DC converter power, more particularly, relate to and be adapted at the switching type power supply (SMPS) used in radio frequency (RF) transmitter, transmitter is such as the RF transmitter that envelope recovers (ER) RF transmitter that is embodied as that is used for cellular mobile station, be also referred to as polar transmitter, wherein use phase place and range weight to represent symbol, rather than complicated inphase/quadrature phase (I/Q) component.

Background technology

Figure 1A is a simplified block diagram, has shown ER transmitter (TX) 1 structure that comprises amplitude modulation (AM) chain and phase modulation (PM) chain.The bit of launching is input to bit to polarity switch 2, and it outputs to amplitude modulator (AM) 4 with range signal through propagation delay (PD) 3.AM 4 (after digital to analog conversion) is provided for controlling the signal of TX power amplifier (PA) 6 output levels by using controllable electric power 5.Bit also outputs to frequency modulator (FM) 7 through propagation delay 3 with phase signal to polarity switch 2, and frequency modulator outputs to signal through phaselocked loop (PLL) 8 input of PA 6 again.Antenna 9 transmit so by using phase place and range weight to generate simultaneously.Use the obtainable advantage of ER emitter structures to comprise the littler size and the efficient of raising.

The supply voltage that is appreciated that PA 6 should carry out amplitude modulation by high-level efficiency and wide bandwidth.

Discuss power supply 5 and PA 6 now in more detail, generally depend on efficient but nonlinear power amplifier such as high-level efficiency TX structures such as polar loop modulation TX, as switching regulator power amplifier (SMPA), E class SMPA for example, perhaps they depend on usually and are driven saturated linear power amplifier, as saturated category-B power amplifier.In these structures,, provide amplitude information, shown in more detail Figure 1B by by means of being connected the supply voltage that the DC power supply that is generally battery or the power governor between power supply and the PA 6 are modulated PA 6.

In Figure 1B, the output V of power supply 5 _PaShould be able to follow the tracks of fast-changing reference voltage V _mLike this, power supply 5 must meet some bandwidth specification.Required bandwidth depends on the system that uses transmitter 1.For example, (8PSK modulation) surpassed 1MHz (for given power level to required bandwidth for the EDGE system, dynamic range～17dB), and for WCDMA (Wideband Code Division Multiple Access (WCDMA)) system surpassed 15MHz (for given power level, dynamic range～47dB).As intelligible, these are to have challenging requirement.Fig. 2 has shown the typical waveform (the RF envelope in the EDGE system) of necessary tracking, wherein modulation voltage (V _m) be shown as between minimal value and peak value and change (also having shown typical rms and mean value).

Be noted that in gsm system modulation is the GMSK with constant RF envelope, thereby, aspect bandwidth, power supply 5 do not had special constraint for given power level.

Usually, realize that power supply 5 has two kinds of major techniques.First kind of technology shown in Figure 3 used the linear regulator of realizing by summing junction 10, driver 12 and power device 14.Though can obtain high bandwidth, because the pressure drop (V on the power device 14 _Drop) reason, efficient is very low.

Second kind of technology shown in Figure 4 will be used switch mode regulator.This technology do not allowed in the past in polarity or ER transmitter and used, and in this technology, decline type switching regulaor 16 will comprise voltage-dropping type or similarly converter 18 and voltage mode control circuit 20.PA 6 is shown as by its equivalent resistance R _PaExpression.Though the efficient of switch mode regulator 16 can be very high, required bandwidth will be difficult to maybe can not obtain.More particularly, if attempt to use switching regulaor 16, then will need very high switching frequency (for example for EDGE, about at least 5 times to required bandwidth or 5-10MHz or more, and surpass 80MHz for WCDMA).Though the switching frequency of 5-10MHz will be had challenge (general commercial DC-DC converter is with the maximum switching frequency operation of about 1-2MHz scope) technically very much, but the DC-DC converter that for example has the 100MHz switching frequency is unpractiaca current implementing, particularly in such as low cost, mass-produced devices such as cell phone and personal communications terminals.

At US 6,377,784B2, Earl McCune (Tropian, Inc.) in " high efficiency modulating RF amplifier " (High-Efficiency Modulation RF Amplifier), exist and it is said and the high-level efficiency power control of high-level efficiency (for example hard restriction or the switching regulator) power amplifier described realize required modulation thus.In one embodiment, it is said, reduced the expansion between required modulation maximum frequency and the switching DC-DC converter operating frequency by after switch mode converters, using the active linear regulator.Linear regulator be it is said the operating voltage that is designed to by enough bandwidth control power amplifiers, thereby reproduces required amplitude modulation waveform reliably.Linear regulator allegedly also is designed to suppress the variation of its input voltage, even also like this when output voltage changes in response to the control signal that applies.Even compare with the frequency that controlled output changes that the variation of input voltage has quite or even lower frequency, suppress allegedly also can take place.By directly or effectively changing the operating voltage on the power amplifier, in the conversion of amplitude modulation output signal, realize high-level efficiency in main DC source simultaneously, allegedly also can realize amplitude modulation.It is said by allowing the switching DC-DC converter also to change its output voltage, hang down and relative constant level, can strengthen high-level efficiency so that the pressure drop on the linear regulator remains on.It is said that time division multiple access (TDMA) (TDMA) burst performance can be combined with effective amplitude modulation, under the control of these functions that are combined, and change according to the mean output power level of the order of communication system and also can be combined in the same structure.

Summary of the invention

Present preferred embodiment according to these teachings of this paper has overcome above-mentioned and other problem, and has realized other advantage.

On the one hand, the invention provides the DC-DC converter, it has the switching mode part that is coupling between DC source and the load, and switching mode partly provides x the output power of amount; And it also have and DC source and load between the linear model part of switching mode part parallel coupled, linear model partly provides y the output power of amount.In a preferred embodiment, x is more preferably greater than y, and can be the ratio of special applications constrained optimization x and y.In addition, the linear model part reveals the response time faster than the switching mode part to the required change list of output voltage.In one embodiment, linear model partly comprises at least one power operational amplifier that is operating as variable voltage source, and in another embodiment, linear model partly comprises at least one the Power arithmetic trsanscondutance amplifier that is operating as variable current source.

Aspect another, the invention provides the RF transmitter (TX) that is coupled to antenna.TX has polar structure, and comprises amplitude modulation (AM) path of being coupled to power amplifier (PA) power supply and be coupled to phase modulation (PM) path that PA imports.Power supply is configured to has the switching mode part that is coupling between battery and the PA, and switching mode partly provides x the output power of amount, and also has the linear model part with the switching mode part parallel coupled between battery and the PA.Linear model partly provides the output power of y amount, and wherein x is more preferably greater than y, and can be the ratio of special applications constrained optimization x and y.Best, the linear model part reveals the response time faster than the switching mode part to the required change list of output voltage.

More on the one hand, the invention provides the method that a kind of operation has the RF TX of polar structure, this structure comprises the AM path of being coupled to the PA power supply and is coupled to the PM path of PA input, this method comprises: provide power supply (power supply) so that comprise the switching mode part that is coupling between power source (power source) and the PA, switching mode partly provides x the output power of amount; With linear model part with the switching mode part parallel coupled between power source and the PA, linear model partly provides y the output power of amount, wherein x is more preferably greater than y, and can be the ratio of special applications constrained optimization x and y, and wherein the linear model part reveals the response time faster than the switching mode part to the required change list of output voltage.

In operation, power supply provides than based on the higher power conversion efficiency of pure linear voltage regulator power supply, also provides simultaneously than based on the wideer bandwidth of operation of pure switching type power supply.

Description of drawings

When read in conjunction with the accompanying drawings, the above-mentioned and others of these teachings will become more apparent in the DETAILED DESCRIPTION OF THE PREFERRED below, wherein:

Figure 1A is the block diagram of conventional ER RF transmitter;

Figure 1B provides the block diagram of the conventional SMPA of power supply amplitude modulated voltage;

Fig. 2 is the reference voltage V that shows that Figure 1A and Figure 1B power supply must be followed the tracks of _mThe oscillogram of typical case;

Fig. 3 A and 3B show the conventional example of the linear voltage regulator that PA is provided;

Fig. 4 A and 4B show the example of the switching regulaor that PA is provided;

Fig. 5 is the PA block diagram that mixed-voltage regulator according to the present invention provides; Wherein the linear segment of mixed-voltage regulator is preferably only handled the required output power of fraction, and essential bandwidth is provided, and the switching mode part preferably provides most of output power with high-level efficiency.

Fig. 6 A-6F shows that the embodiment of mixed-voltage regulator shown in Figure 5 simplifies diagrammatic sketch;

Fig. 7 A and Fig. 7 B are generically and collectively referred to as Fig. 7, and be relevant with the circuit shown in Fig. 6 A, and wherein Fig. 7 A shows the general circuit notion, and Fig. 7 B shows switch sections in greater detail;

Fig. 8 shows the waveform corresponding to Fig. 7 circuit operation;

Fig. 9 also shows the waveform corresponding to Fig. 7 circuit operation;

Figure 10 A and Figure 10 B are generically and collectively referred to as Figure 10, and be relevant with the circuit shown in Fig. 6 B, and wherein Figure 10 A shows the general circuit notion, and Figure 10 B shows switch sections in greater detail;

Figure 11 shows the waveform corresponding to Figure 10 circuit operation;

Figure 12 also shows the waveform corresponding to Figure 10 circuit operation;

Figure 13 A and Figure 13 B are generically and collectively referred to as Figure 13, and be relevant with the circuit shown in Fig. 6 C and Fig. 6 D, and wherein Figure 13 A shows the general circuit notion, and the more detailed display switch part of Figure 13 B;

Figure 14 shows the waveform corresponding to Figure 13 circuit operation;

Figure 15 also shows the waveform corresponding to Figure 13 circuit operation;

Figure 16 A and Figure 16 B are generically and collectively referred to as Figure 16, and be relevant with the circuit shown in Fig. 6 E and Fig. 6 F, and wherein Figure 16 A shows the general circuit notion, and Figure 16 B shows switch sections in greater detail;

Figure 17 shows the waveform corresponding to Figure 16 circuit operation;

Figure 18 also shows the waveform corresponding to Figure 16 circuit operation;

Figure 19 A and Figure 19 B are generically and collectively referred to as Figure 19, respectively the equivalent circuit diagram of display voltage control voltage source (VCVS) and the VCVS circuit that is embodied as power operational amplifier (POA);

Figure 20 A and Figure 20 B are generically and collectively referred to as Figure 20, respectively the equivalent circuit diagram of display voltage Control current source (VCCS) and the VCCS circuit that is embodied as operation transconductance amplifier (OTA);

Figure 21 shows the first control configuration, and wherein switch sections and linear segment be all with close loop maneuver, and with modulation signal V _mAs a reference;

Figure 22 shows the second control configuration, and wherein switch sections and linear segment are all with close loop maneuver, and wherein linear segment is with modulation signal V _mAs a reference, and switch sections with the output of linear segment as a reference;

Figure 23 shows the 3rd control configuration, wherein has only linear segment with close loop maneuver and with modulation signal V _mAs a reference, and wherein switch sections is operated with open loop, and has only modulation signal V _mInformation is used to generate the dutycycle of switch sections;

Figure 24 also according to embodiments of the invention display switch regulator and linear regulator through secondary inductor L ₁(optional) auxiliary capacitor C ₁Parallel connection;

Figure 25 A and Figure 25 B are generically and collectively referred to as Figure 25, show that wherein in Figure 25 A, switching regulaor and linear regulator are master control, and in Figure 25, linear regulator is master control according to control block diagram embodiment illustrated in fig. 24, and switching regulaor is a subordinate;

Figure 26 A and Figure 26 B are generically and collectively referred to as Figure 26, demonstration is according to first multi-mode embodiment illustrated in fig. 24 (many PA) control block diagram, wherein all PA are connected on the same power lead of linear regulator output terminal, and wherein at Figure 26 A, switching regulaor and linear regulator are master control, and at Figure 26 B, linear regulator is master control, and switching regulaor is a subordinate;

Figure 27 A and Figure 27 B are generically and collectively referred to as Figure 27, demonstration is according to second multi-mode control block diagram embodiment illustrated in fig. 24, wherein GSM/EDGE PA is connected the output terminal of switching regulaor, and WCDMA PA is connected the output terminal of linear regulator, wherein in Figure 27 A, switching regulaor and linear regulator are master control, and in Figure 27 B, linear regulator is master control, and switching regulaor is subordinate (only in the WCDMA pattern);

Figure 28 is shown as SMPA (a) block representation, (b) by its equivalent DC resistance R _PaThe simulation, and (c) by with the capacitor C that is used to realize PA stability _PaIts equivalent DC resistance R in parallel _PaSimulation;

Figure 29 A and Figure 29 B are generically and collectively referred to as Figure 29, show according to the 3rd multi-mode control block diagram embodiment illustrated in fig. 24; Wherein GSM/EDGE PA and WCDMA PA are connected respectively to the independent current source line that is associated with two linear regulators, wherein at Figure 29 A, switching regulaor and each linear regulator are master control, and at Figure 28 B, each is master control linear regulator, and switching regulaor is subordinate (only in the WCDMA pattern); And

Figure 30 A and Figure 30 B are generically and collectively referred to as Figure 30, demonstration is according to the 4th multi-mode control block diagram embodiment illustrated in fig. 24, wherein GSM/EDGE PA is connected to the output of switching regulaor, wherein WCDMA PA and CDMA PA are connected respectively to the independent current source line that is associated with two linear regulators, wherein at Figure 30 A, switching regulaor and each linear regulator are master control, and at Figure 30 B, linear regulator is master control, and switching regulaor is subordinate (only in WCDMA and a CDMA pattern).

Embodiment

With reference to Fig. 5, the invention provides a kind of mixed-voltage regulator or power supply 30, it has made up preferably by high-level efficiency but low bandwidth is handled the switch sections 32 of most of power and preferably by poor efficiency but high bandwidth is handled the linear segment 34 of smaller portions power demand.Resultant power supply has required bandwidth and a shade below pure switch power efficiency but still be higher than the efficient of pure linear regulator efficient far away.Because linear segment 34 can be used for compensating the output voltage ripple that is associated with pure switching type power supply usually, therefore, resulting AC-battery power source 30 provides a kind of output voltage quality of improvement.Because excessive output voltage ripple can have a negative impact to the output spectrum of PA 6, therefore, this is an important advantage.

Be noted that in principle the quantity of power of being handled by switch sections 32 (x) is greater than the quantity of power of being handled by linear segment 34 (y).This is an ideal situation normally, and in fact, in many examples, x is more much bigger than y.Yet this relation between the power that switch sections 32 and linear segment 34 are handled can not be considered as the restriction of the preferred embodiment of the present invention.In principle, wish the ratio maximization with x and general power: this ratio is big more, and efficient is just high more.Yet the effective rate of realizing in given application can be one or more following factors and the function of considering item:

(a) Yu Qi application (the RF system detail is as the frequency spectrum of RF envelope, the amplitude of high-frequency AC component etc.); And

(b) realize, wherein can determine to a certain extent by the quantity of power of switch sections 32 processing and the quantity of power of handling by linear segment 34.For example, in EDGE,, can handle nearly all power by switch sections 32 by using the 6-7MHz switching frequency, or by for example use the 1MHz operation than the low switch converter, can handle less power.In some cases, for example, at low-down power, but also disabled switch part 32 is also only used linear segment 34, in this case, concerns that x＞y is not suitable.

That (c) also will consider may be efficient and realize compromise between the complicacy, and this is because realize that slow dc-dc converter is simpler usually, but subsequently because major part power need be handled by linear segment 34, thereby has reduced efficient.

What (d) also will consider may be will optimize whole efficiency compromise.For example, the switch sections 32 with very high switching frequency and high bandwidth can be handled the most of power (x is more much bigger than y) in the given application, but the processing in the switch sections 32 may be owing to the high switching frequency inefficiency that becomes.Therefore, trial is by using in switch sections 32 than low switching frequency to realize better efficient and to handle than trading off between the less energy and optimize whole efficiency that this may be more favourable in switch sections 32.

Therefore, usually the power section x of switch sections 32 processing is more preferably greater than the power section y of linear segment 34 processing, and also be preferably by given application and constrained optimization x that may be also applies by special manipulation mode (for example at above-mentioned low-power mode, wherein all power can be handled by linear segment 34) and the ratio of y.Also can consider combination,, and also can be the ratio that application constraint is optimized x and y so that x is more preferably greater than y.

In fact, the part topology (being called " switch sections " in Fig. 5) by using dc-dc converter is also in parallel with voltage or current source (being called " linear segment " in Fig. 5) with it, can realize the present invention.The output capacitor (C) of the step-down of Fig. 4 A (decline type) converter 18 is removed.In step-down controller 18, capacitor serves as voltage source, to keep output voltage constant.When the voltage of output terminal need increase, must provide big electric current through inductor (L), satisfying the increase needs of load, and capacitor (C) is charged to new more high-voltage level.This operation makes switching regulaor 16 slack-off, and has limited bandwidth.Yet the voltage level that, increases (or reduction) if capacitor (C) replaces with voltage source can provide very apace through the shunt voltage source of linear part 34, and slower switch sections readjusts its working point.

Referring again to Fig. 2, switch sections 32 provides average level V _{M_av}, and linear segment 34 provides the AC that is superimposed upon on the average level component.

Based on the voltage source in the alternative use current source instead of linear part 34 of same concept.

The voltage source of linear segment 34 can use power operational amplifier (POA) to realize, and the current source of linear segment 34 can use Power arithmetic trsanscondutance amplifier (OTA) to realize.The operational amplifier of linear segment 34 can be by cell voltage (V as shown in Figure 5 _Bat) provide.Alternative current more preferred embodiment (from the angle of efficient), the operational amplifier of linear segment 34 provides the voltage V of Fig. 2 _{M_pk}, that is, and corresponding to the voltage of reference signal amplitude, wherein V _{M_pk}All the time be lower than V _BatIn fact, preferably the operational amplifier for linear segment 34 provides voltage V _{M_pk}Add and obtain the required certain tolerance limit of linear grade proper operation (for example, 0.2V).

Fig. 6 A-6F shows the various embodiment of mixed-voltage regulator 30 shown in Figure 5, wherein Fig. 6 A, 6C and 6D show the use of variable voltage source 34A (for example above-mentioned power operational amplifier), and wherein Fig. 6 B, 6E and 6F show the use of variable current source 34B (for example above-mentioned Power arithmetic trsanscondutance amplifier).Note, used two

variable voltage source

34A and 34A ' in Fig. 6 C, and in Fig. 6 D, two

variable voltage source

34A and 34A ' are through the out-put supply rail of C1 capacitive couplings to switch sections 32.It is also noted that, in Fig. 6 E, used two variable

current source

34B and 34B ', and in Fig. 6 F, two variable

current source

34B and 34B ' are through the out-put supply rail of C1 capacitive couplings to switch sections 32.

Based on above explanation, can understand, use of the present invention allows for the TX structure and realizes effective PA power supply 30, and wherein the PA supply voltage need carry out amplitude modulation.At present, because the commercial switching regulaor that required bandwidth is provided that does not exist the inventor to know, therefore, this realizes by the low linear regulator of service efficiency (referring to Fig. 3 A and Fig. 3 B) only.

To further describe above-mentioned and other embodiment of the present invention now in detail.

Circuit shown in Figure 7 is relevant with the circuit shown in Fig. 6 A, and wherein Fig. 7 A shows the general circuit notion, and wherein Fig. 7 B shows switch sections 32 in greater detail.Switch sections 32 obtains from step-down controller, and this converter is by two switching devices and the decline type switch DC-DC converter that the L-C wave filter is formed.(the d=upper switches is at the time of conducting t with dutycycle d to be shown as the switching device of complementary MOS transistor (PMOS/NMOS) among Fig. 7 B _{On_PMOS}With switching cycle T _SRatio) alternate conduction.Control signal with dutycycle d can obtain from analog pulse width modulators (PWM) parts 32A, and these parts are by comparing V _{Ctrl_sw}And have period T _sSawtooth signal and will control voltage V _{Ctr_sw}Be converted to pwm signal with dutycycle d.The pwm signal that is fed to transistor driving level 32B also can generate by other method, such as in the digital PWM parts.

Conventional step-down controller comprises the L-C output filter generally speaking, and wherein C is enough big, is the characteristic of voltage source thereby make the characteristic of step-down controller.Yet in existing preferred embodiment of the present invention, filter capacitor is removed, and perhaps is retained, but has minimum electric capacity.Like this, parts 32 are referred to herein as " switch sections " relative with " dc-dc converter ".In fact, physical circuit will have certain filter capacitor, for example, guarantee that RFPA 6 stablizes required amount.Yet, for the present invention is described, suppose that capacitance (C) is more much smaller than the capacitance that has in the conventional step-down controller, like this, the characteristic of switch sections 32 mainly is the characteristic of current source, rather than the characteristic of voltage source.

More particularly, because the reason of inductor (L) (and nothing/minimum capacitor C), switch sections 32 has the characteristic of current source, rather than actual Voltage-controlled Current Source (VCCS).Control voltage V _{Ctrl_sw}Increase determined the increase of dutycycle d, this has determined average output voltage V _PaIncrease, and this has determined PA 6 electric current I _PaIncrease, therefore and determined inductor current I _LThe increase of DC component.Yet, PA 6 electric current I _PaAbsolute value not exclusively by control voltage V _{Ctrl_sw}Determine, and at I _Pa=V _Pa/ R _PaThe time also by R _PaDetermine.Therefore, though the operation that this technology can similar " VCCS ", its is direct Control current, and therefore is called " similar VCCS ".

Linear segment 34 is as Voltage-controlled Current Source (VCVS) 34A work, and its output voltage V _oAmplify A by difference _dBy differential voltage V _dControl.

More particularly, Fig. 7 A shows this embodiment of the present invention by the hypothesis ideal source: wherein switch sections 32 effect is just as current source, and with linear segment 34 parallel connections of effect just as two-way (that is, it can be supplied and ABSORPTION CURRENT) Voltage-controlled Current Source 34A.Linear segment 34 as voltage source is provided with PA 6 voltage V _PaThe current i of switch sections 32 _SwAdd the current i of linear segment 34 _Lin, formed PA 6 current i _Pa(R _PaThe virtual impedance of expression PA 6).Can connect optional direct-current blocking-up decoupling capacitor C _dTo guarantee that 34 of linear segments provide AC component.

Fig. 7 display switch part 32 realizes by decline type step-down controller, eliminates or greatly reduce output filter capacitor C from this converter.The current i of switch sections 32 _SwTherefore be actually inductor current i _L, cause the behavior (inductor L can liken current source to) of similar in fact current source.

It should be noted that because linear segment 34 has the characteristic of voltage source, so it can fixedly be applied to the voltage level V on the PA _Pa, and have the member of controlling this voltage level.In addition, linear segment 34 very fast (wide bandwidth) therefore might provide V _PaFast modulation.It is also noted that the VCVS 34A of linear segment 34 is two-way, expression can be supplied and ABSORPTION CURRENT.

Shown in Figure 19 B, VCVS 34A can be embodied as power operational amplifier (POA).POA comprises the operational amplifier (OPAMP) with A (B) the class level that can absorb/supply required electric current.Figure 19 B shows by transistor Q ₁And Q ₂The category-B power level of forming, but can change the output stage design to improve performance.For example, in fact power level can be embodied as AB class level, to reduce cross distortion.

As described, can introduce optional decoupling capacitor C _dTo guarantee that 34 of linear segments provide AC current component.Yet in some cases, though allow linear segment 34 also to provide the DC component can bring more complicated control, this will be favourable.For example, may be deactivated and incite somebody to action under the situation that only be provided by linear segment 34 at PA 6 electric currents at switch sections 32, expectation can provide DC component from the linear segment 34 of low-power level.And for example, at V _{Pa_peak}Approach very much for example low battery voltages level of 2.9V, when for example being 2.7V, and switch sections 32 be can't provide it the time, and expectation can provide DC component from the linear segment 34 of this low battery voltages level.Under this type of situation, will remove optional C _d

Circuit operation shown in Fig. 7 A is illustrated by analog waveform shown in Fig. 8, and the circuit operation shown in Fig. 7 B is illustrated by the analog waveform shown in Fig. 9.

Uppermost waveform shows resulting PA 6 voltage V among Fig. 8 _Pa PA 6 voltage V _PaBe provided with by linear grade 34 with voltage source characteristic.In this example, V _PaHave DC component (2V) and add the AC component that is shown as the 15MHz voltage sine wave of representing fast modulation.From the electric current composition of last several second waveform display switch part 32, steady current i _SwFrom the electric current composition of last several the 3rd waveforms demonstration linear segments, AC component i _Lin(15MHz sine wave).As described, linear segment 34 is as bi-directional voltage mode work, that is, but but its both also ABSORPTION CURRENT of supply of current.Nethermost waveform shows resulting PA 6 current i _PaHave from the DC component of switch sections 32 with from the AC component of linear segment 34.

Notice that in Fig. 8, it is zero sine wave (do not have the composition from linear segment 34, be denoted as " A ") that one group of waveform is used for amplitude, and another group is used for the sine wave (to show the composition from linear grade, being denoted as " B ") of non-zero magnitude.Used identical agreement in the oscillogram of Fig. 9, Figure 11, Figure 12, Figure 14, Figure 15, Figure 17 and Figure 18.

Fig. 9 display simulation waveform is to illustrate the circuit operation shown in Fig. 7 B.For this non-limiting example, suppose that switching stage 32 has the switching frequency of 5MHz and 0.5 dutycycle.Uppermost waveform is presented at the PWM 32A voltage that applies on the inductor L of node pwm.Show resulting PA 6 voltage V from last several second waveforms _PaPA 6 voltage V _PaBe provided with by linear grade 34 with voltage source characteristic.In this example, V _PaHave DC component (2V) and add the 15MHz AC component (voltage sine wave) of representing fast modulation.From the electric current composition of last several the 3rd waveform display switch parts 32, i.e. inductor current i _L=i _SwIn the case, electric current is not ideally a steady current as shown in Figure 8, but has the specific triangles shape that runs in dc-dc converter.Switch sections 32 provides DC component and triangle AC component (inductor current ripple).From the electric current composition of last several the 4th waveforms demonstration linear segments 34, AC component i _Lin(inductor current ripples compensation on the 15MHz sine wave).Notice that linear segment 34 not only provides the 15MHz sinusoidal component, and AC is provided component, with compensating inductance device current ripples (from being denoted as AC _RipFollowing waveform can be clear that).This is the voltage source characteristic owing to linear segment 34, and it can be supplied and ABSORPTION CURRENT as bi-directional voltage mode.Nethermost waveform shows resulting PA 6 current i _PaHave from the DC component of switch sections 32 with from the AC component of linear segment 34, wherein the AC triangle component of inductor current (the 3rd curve) is by linear grade 34 compensation.

Figure 10 has shown an embodiment, and in this embodiment, Voltage-controlled Current Source (VCCS) 34B is used to constitute linear segment 34.Usually, apply the consideration item identical with Fig. 7 embodiment, unique sizable difference is fixedly PA 6 voltage levels of VCCS itself.On the contrary, PA 6 voltages are by being injected into R _PaTotal current determine.Linear segment 34 can be embodied as the operation transconductance amplifier (OTA) shown in Figure 20 B.Simplify in the view Q in the differential pair at this ₁Collected current (Ic ₁) mirror image is I ₅, and Q ₂Collected current (Ic ₂) mirror image is I ₃, and mirror image is I subsequently ₄Output current is I _o=I ₅-I ₄, and with collected current IC ₁-IC ₂Between difference proportional, and it and differential voltage V _dProportional.As described, Figure 20 B has shown the reduced representation of OTA.In fact, the purpose that circuit is realized will be to optimize the precision of current mirror, and obtain linear characteristic I _o=gV _d

Circuit operation shown in Figure 10 A is illustrated by analog waveform shown in Figure 11, and the circuit operation shown in Figure 10 B is illustrated by the analog waveform shown in Figure 12.

In Figure 11, the PA 6 voltage V of uppermost waveform display result _PaSuppose the angle from power supply, PA 6 effects are just as resistive load.Therefore, V _Pa=R _Pa(i _Sw+ i _Lin), that is, and the electric current sum setting that PA 6 voltages are provided by switch sections 32 and linear segment 34.In this example, R _PaBe assumed to be and equal 2 ohm.Switch sections 32 provides DC component i _Sw(for example, 1 ampere), and linear segment 34 provides AC component i _Lin, the 15MHz current sinusoidal ripple of expression fast modulation.From the electric current composition of last several second waveform display switch part 32, that is, and 1 ampere steady current i _SwFrom the electric current composition of last several the 3rd waveforms demonstration linear segments 34, that is, and AC component i _Lin(15MHz current sinusoidal ripple).PA 6 current i of nethermost waveform display result _PaHave from the DC component of switch sections 32 with from the AC component of linear segment 34.

In Figure 12, suppose that switching stage 32 has switching frequency=5MHz and dutycycle=0.5.Uppermost waveform is presented at the PWM 32A voltage that applies on the inductor L of node pwm.PA 6 voltage V from last several second waveform display result _PaAforesaid the same, suppose the PA6 effect just as resistive load, and so V _Pa=R _Pa(i _Sw+ i _Lin), that is, and the electric current sum setting that PA 6 voltages are provided by switch sections 32 and linear segment 34.As mentioned above, R _PaBe assumed to be and equal 2 ohm.Switch sections provides the i of the DC component with triangle AC component _Sw(1 ampere).Linear segment provides AC component i _Lin, as the 15MHz current sinusoidal ripple of expression fast modulation.From the electric current composition of last several the 3rd waveform display switch parts 32, that is, and inductor current i _L=i _SwIn the case, electric current is not ideally a steady current as shown in figure 11, but has the triangle that runs in dc-dc converter.Switch sections 32 provides DC component and triangle AC component (inductor current ripple).From the electric current composition of last several the 4th waveforms demonstration linear segments 34, that is, and AC component i _Lin(15MHz sinusoidal component).Notice that in the case, 34 of linear segments provide 15MHz sinusoidal component, are different from the respective waveforms of Fig. 9, also can see the AC component of compensating inductance device current ripples in the figure.PA 6 current i of nethermost waveform display result _PaThe DC component and the AC triangle component that have from switch sections 32 (can be from being denoted as AC _RipFollowing waveform see) with from the AC component of linear segment 34.Notice that in this embodiment, because the current source characteristic of linear grade 34, AC triangle component be can't help linear grade 34 compensation, but it can be compensated by suitably controlling VCCS.

Figure 13 and circuit shown in Figure 16 show that respectively the linear segment 34 that is made of two

VCCS

34A and 34A ' or two

VCCS

34B and 34B ' is from V _BatSupply of current, and with current absorption to ground.In the oscillogram of Figure 14 and Figure 15 and Figure 17 and Figure 18, shown operation respectively.

The waveform that circuit among Figure 13 and Figure 16 is represented and it is corresponding shows source/Su Hangwei of VCVS34A and VCCS 34B respectively, and the behavior of difference simulated power operational amplifier and power trsanscondutance amplifier.Notice that two VCVS 34A among Figure 13 do not activate simultaneously, and be preferably in and be placed in high impedance status when not activating.

More particularly, Figure 13 A and 13B have shown this embodiment with ideal source, and the top explanation of being done for Fig. 7 circuit also is applicable to herein.Difference between the circuit is that voltage

source V CVS

34A and 34A ' are unidirectional (supply of current, another ABSORPTION CURRENT) among the embodiment of Figure 13, and voltage source 34A is two-way (supply and absorption) among Fig. 7.Can comprise decoupling capacitor C _dTo guarantee that 34 of linear segments provide AC component.

For the analog waveform figure of Figure 14 and Figure 15, as above the similar explanation that provides for Fig. 8 and Fig. 9 also is suitable for, except the component i of linear segment 34 _LinBe divided into i _Aux1(supply) and i _Aux2(absorption).It should be noted that equally two

voltage source

34A and 34A ' (Yuan Hesu) preferably are placed in high impedance status when its electric current separately is zero (that is, they do not activate).

Figure 16 A and 16B have shown this embodiment of the present invention with ideal source, and the top explanation of being done for Figure 10 circuit also is applicable to herein.Difference between the circuit is that

current source VCCS

34B and 34B ' are unidirectional (supply of current, another ABSORPTION CURRENT) among the embodiment of Figure 16, and current source 34B is two-way (supply and absorption) in Figure 10.Can comprise decoupling capacitor C _dTo guarantee that 34 of linear segments provide AC component.

For the analog waveform figure of Figure 17 and Figure 18, as above for the similar explanation that Figure 11 and Figure 12 did also is suitable for, except the component i of linear segment 34 _LinBe divided into i _Aux1(supply) and i _Aux2(absorption).

Notice that Fig. 7 and Figure 10 are the expression of power stage (switch sections 32 and linear segment 34) interconnection, do not consider control.Switch sections 32 is expressed as by control voltage V _CtrlThe parts of control.Linear segment 34 is expressed as by differential voltage V _dThe parts of control.Figure 21, Figure 22 and Figure 23 show three non-limiting examples of the control technology of closed control loop.

In Figure 21, switch sections 32 is by the voltage mode control operation.Controller is by control assembly 36A and have frequencfy-dependent behavior G _C1(s) parts 36B forms, control assembly 36A generated error signal V _E1, parts 36B is with error signal V _E1As its input and will be used for the control voltage V of switch sections 32 _{Ctrl_sw}As its output.Error voltage V _E1Be as modulation signal V _mReference voltage V _{Ref_sw}With as output voltage V _PaFeedback signal V _{Feedback_sw}Between poor.In the case, controller (

assembly

36A, 36B) can be embodied as physically have the R-C corrective network operational amplifier with acquired character G _C1(s).

Linear segment 34 uses modulation signal V _mAs reference V _{Ref_lin}Feedback voltage V _{Feedback_lin}It is output voltage V _PaFeedback voltage V _{Feeaback_lin}Also can be shown in decoupling capacitor C as the figure with dashed lines _dIf (existence) obtains before.Controller class in the case is similar to switch sections 32, by generated error signal V _E2Parts 38A and have a frequencfy-dependent behavior G _C2(s) parts 38B forms.Owing to linear segment 34A in fact preferably realizes with power operational amplifier, as among Figure 19 B, therefore, and just as the skilled person will recognize, by adding the R-C corrective network with acquired character G _C2(s), can be at closed control loop around it.Notice that comprise that VCVS34A just shows the voltage source characteristic of linear grade 34, it is different with the VCVS shown in Fig. 7.The parts that are designated as " linear segment of tool feedback " are actually the expression of the power operational amplifier with R-C corrective network.

Notice that (for example, Figure 10) and during the OTA shown in Figure 20 B, aforesaid identical points for attention are applicable to closed this ring to be configured to VCCS 34B in linear grade 34.

In Figure 22, to compare unique sizable difference with Figure 21 and be, the reference signal of switch sections 32 is the output of taking from linear segment 34 (if having decoupling capacitor then be positioned at before it).When linear grade 34 was used VCCS 34B and OTA, identical points for attention were suitable for.In this embodiment, clearly, linear segment 34 is with modulation signal V _mThe AM signal is as its reference, and switch sections 32 with the output of linear segment 34 as it with reference to (that is, its " subordinate " is in linear segment 34).

In the embodiment of Figure 23, switch sections 32 is with the open loop operation, and this expression has only modulation signal V _mBe used to generate PWM dutycycle d, and do not have error signal V _E1=V _m-V _PaIn the time may having stability problem as Figure 21 and dicyclo control system shown in Figure 22, this example embodiment may be particularly useful.As mentioned above, when linear grade 34 was used VCCS 34B and OTA, identical consideration item was suitable for.

The embodiment of the invention above-mentioned is illustrated as the fast modulation that realizes PA 6 power supplys a solution is provided, and wherein fast modulation is mainly provided by linear segment 34, and uses not or have the step-down controller of minimum filter capacitor.Yet, it should be noted, the notion that switching stage and linear grade are connected in parallel can be used, and also useful under situation about using at step-down controller with conventionally form, that is, have sizable output filter capacitor C and therefore have the voltage source characteristic.For example, being used for the RF transmitter of GSM/EDGE situation can be based on the step-down controller with voltage mode control, by the addressing of high-speed switch converter.Under this exemplary scenario, can realize essential bandwidth, yet, dynamic perfromance undesirable (that is, be referenced to the output-transfer function unevenness, and may show peaking on the contrary), and be not best with reference to following the tracks of therefore.In addition, because the output voltage ripple that the converter switch action causes has formed pseudo-RF signal.Therefore, the linear grade 34 that is connected in parallel with step-down controller can be used for by " help " dc-dc converter and improves its tracking performance compensating its unfavorable dynamic perfromance.In fact, linear segment 34 also can be used for improving (widening) bandwidth, but its mainly to act on be the correcting switch part 32 already provided output characteristics that are referenced to.In addition, linear segment 34 also can compensate output voltage switch ripple (at least to be enough to satisfy the mode of the pseudo-needs of RF) with compensating inductance device current ripples by injection current.

For the above reasons, should be understood that embodiment that Fig. 5 generally shows can expand to comprises following circuit structure: wherein switch sections 32 is " common " step-down controllers, that is, the output filter capacitor C is enough big, thereby makes the step-down controller effect just as voltage source.

Based on foregoing, be understood that, the above embodiment of the present invention comprises based on not or have a circuit structure of the step-down switching formula converter of minimum filter capacitor C, promptly, output filter capacitor C enough little (or not existing) wherein, therefore step-down controller acts in fact just as current source, wherein linear segment 34 can be determined the bandwidth of PA 6 power supplys alone, promptly, even with extremely slow switch sections 32, linear segment 34 owing to do not exist yet/minimum step-down controller filter capacitor former thereby can modulate; Wherein linear segment 34 also provides the triangle AC component of inductor current; And linear segment 34 compensating switch ripples wherein.

Based on foregoing, can understand the above embodiment of the present invention and also comprise preferably circuit structure based on step-down switching formula converter with quite big filter capacitor C, that is, wherein the output filter capacitor C is enough big, makes step-down controller act on just as voltage source in fact.Therefore, embodiments of the invention also comprise the circuit structure based on " common " step-down controller circuit topology with filter capacitor; Wherein bandwidth is mainly determined by dc-dc converter.In the case, linear segment 34 can be used for improving bandwidth, but in a kind of more limited mode, because bandwidth is limited by the filter capacitor C of switching regulaor in fact.The vital role of these embodiment neutral line parts 34 is dynamic perfromances of help and correcting switch part 32 (step-down controller).In this embodiment, but also compensating switch ripple of linear segment 34.

Each side of the present invention is based on following observation: EDGE and WCDMA envelope medium-high frequency component in example have extremely low amplitude, and most of energy is at DC and low frequency component.Low bandwidth switch sections 32 is handled most of power (DC and low frequency component) with high-level efficiency, and more the linear segment 34 of wide bandwidth is only handled sub-fraction power (corresponding to the power of high fdrequency component) with lower efficient.Therefore, realize that required bandwidth also still provides good efficiency to become possibility.Usually, obtainable efficient is lower than the efficient that realizes with pure Switching Power Supply, uses the efficient that realizes based on the power supply of pure linear regulator but still be far longer than.

Principle of the present invention can not considered the actual realization of switch sections 32 and/or linear segment 34 and use, and may be used on the emitter structures that PA 6 supply voltages need carry out amplitude modulation usually.Teachings of the present invention is not limited to GSM/EDGE and WCDMA system, but also can expand to other system (for example, to cdma system).Teachings of the present invention is not limited to use the system of E class PA 6, but also may be used on using the system of the saturated PA of other type.

Others of the present invention in greater detail below are at coupling and supply with several PA 6 in the multi-mode transmitter and the control that is used for same procedure.

Referring now to Figure 24, shown among the figure among the embodiment that wherein switching regulaor 100 and linear regulator 102 are by additional electrical sensor L ₁(that is, except that for example the 32 inductor L of ordinary tap part shown in Fig. 7 B) and (optionally) capacitor C ₁, be coupled in parallel to SMPA104 (for example, E class PA).PA 104 supply voltage V _PaBy linear regulator 102 with the high precision plan.Yet, owing to low bandwidth, switch ripple and noise reason, the instantaneous output voltage V of switching regulaor 100 ₁Can't accurately be fixed as same value.Therefore, introduced additional electrical sensor L ₁, to regulate instantaneous voltage difference V _Pa-V ₁L ₁On average voltage be necessary for zero, thereby V ₁Mean value equal V _Pa

If there is decoupling capacitor C ₁, then linear regulator 102 can only provide the AC component in certain frequency range, and the lower bandwidth of this best compensating switch regulator 100 is to obtain required total bandwidth.

If C ₁Do not exist, then linear regulator 102 also can provide DC and low frequency component.Under certain conditions, for example at PA 104 voltage V _PaShould be as far as possible near cell voltage V _BatThe time, this may advantageous particularly.This type of situation be use low battery voltages (for example, in the time of 2.9V) in the maximum RF output power GSM situation of (PA 104 needs minimum voltage, for example 2.7V).In the case, be inserted in the input voltage of the arbitrary regulator between battery and the PA 104 and the difference between the output voltage extremely low (in this example only 0.2V).This is (to suppose that a pressure drop on the power device adds two inductor L and L with the extremely unobtainable value of switching regulaor 100 ₁, dutycycle is less than 100%).At this in particular cases, the supply voltage near cell voltage is provided providing more linear regulator 102, and therefore linear regulator 102 provides all power (DC component, and capacitorless C ₁).Though under these special circumstances (GSM, peak power output, low battery voltages), because the pressure drop on the linear regulator 102 is little, efficient is unaffected, at lower GSM power level (that is, bigger pressure drop on the linear regulator 102), efficient will reduce.Therefore, in lower power level, use switching regulaor 100 to provide all power (DC component) more favourable.

In Figure 24, the supply voltage of linear regulator 102 is V _Bat, identical with the voltage that is used for switching regulaor 100.Though from the angle that realizes, this may be best, says from standpoint of efficiency, this may not be best.At lower power level, wherein V _{M_pk}Compare V _BatUnder the much lower situation, the pressure drop on the linear regulator 102 is big, and its efficient is low.Therefore, a kind of more effective technology (with high-level efficiency) at a certain level, for example is higher than envelope V with the supply voltage preconditioning of linear regulator 102 _{M_pk}Peak value 200-300mV (referring to Fig. 2).

Can see that from Fig. 3 and Fig. 4 two building blocks (switch and linearity) of hybrid regulator have its oneself control loop.Overhead control must be configured to make two parts to complement each other.Figure 25 has shown two possible controlling schemes.

In Figure 25 A, two regulators 100,102 are " master control ", and this is with modulation signal V because of each regulator _mAs a reference, and each regulator 100,102 receive its feedback signal (V from its oneself output _{Feedback_sw}, V _{Feedback_lin}).

In Figure 25 B, linear regulator 102 is " master control ", that is, it is with modulation signal V _mAs a reference, and with its oneself output as feedback signal.Switching regulaor 100 is " subordinate ", and this represents voltage that it is applied to linear regulator 102 SMPA 104 as the reference signal, and attempts as far as possible accurately following it.

As described below, the embodiments of the invention are specially adapted to the application of multi-mode transmitter.

As first non-limiting example, in GSM, the RF envelope is constant, and the voltage that therefore is provided to PA 104 is constant, and its level is according to the power demand level adjustment.The major function of SMPA 104 power supplys is power control in the case.Only use switching regulaor 100 with enough in principle.Yet switch motion generates output voltage ripple and noise, and this is regarded as false signal in the RF spectrum of SMPA 104 outputs.In this pattern, linear regulator 102 can be used for the output voltage ripple of compensating switch regulator 100 when needed.Like this, also can relax the specification of the output voltage ripple of switching regulaor 100.For example, if be the exemplary voltages ripple specification of switching regulaor 100 hypothesis 5mV, then when providing ripple compensation by linear regulator 102, this specification also can be loosened to 50mV, thereby allows the LC component littler in the switching regulaor 100 and/or the dynamic perfromance faster of switching regulaor 100.In the case, switching regulaor 100 is almost handled all SMPA power that need, and few (having only the required processing of ripple compensation) that linear regulator 102 is handled.

In the EDGE system, or generally include and have moderate high dynamic perfromance that (for example, in arbitrary system of the variable RF envelope of required BW＞1MHz), the major function of SMPA 104 power supplys is power control and envelope-tracking.Can see that the pure switching regulaor with 6-7MHz switching frequency can be followed the tracks of EDGE RF envelope with good relatively precision.Yet, when using pure switching regulaor, system is not strong, and may show to such an extent that for example switching regulaor 100 is referenced to peaking in the output-transfer function and SMPA 104 loads variation (resistance of SMPA increase when supply voltage the reduces usually) sensitivity with supply voltage.In addition, as mentioned above, also there is the problem of output voltage ripple.According to this aspect of the invention, linear regulator 102 can be used for non-optimal dynamic characteristic, SMPA 104 load variations and the switch ripple of compensating switch regulator 100 when needed.If the switching frequency of switching regulaor 100 is enough high, can realize good performance for tracking, then most of power is handled by switching regulaor 100.Yet, also might use to have more low switching frequency thereby have the more switching regulaor 100 of low bandwidth, in this case, the power ratio regular meeting of being handled by linear regulator 102 increases, the reduction of being handled by switching regulaor 100 with compensation.

In the WCDMA system, or show usually and have high dynamic perfromance that (for example, in arbitrary system of the variable RF envelope of required BW＞15MHz), the major function of SMPA 104 power supplys is power control and envelope-tracking.Yet the bandwidth that is used for the EDGE system owing to required bandwidth ratio is much higher, and it is not enough therefore only using switching regulaor 100 (in the CMOS technology), thereby uses linear regulator 100 to provide required bandwidth to become very important.As in the EDGE situation, but linear regulator 102 also compensating switch ripple and SMPA 104 load variations.

From the another effectiveness of using the embodiment of the invention to obtain is the ability that a plurality of PA are provided as multi-mode operation.A non-limiting example is E class GSM/EDGE PA104A shown in Figure 26 and E class WCDMA PA 104B.In the case, all

PA

104A, 104B are connected on the same power lead of linear regulator 102 output terminals.This embodiment is the same with the embodiment of Figure 27, Figure 29 and Figure 30, supposes to exist the parts (for example, switch) of once only enabling a

PA

104A or 104B.

Notice that it is E class PA that

PA

104A and 104B are not limited to, show that these just for convenience's sake.This is equally applicable to the embodiment shown in Figure 27, Figure 29 and Figure 30.

In Figure 26 A, regulator 100,102 all can be considered " master control ", that is, and all with modulation signal V _mAs its reference, and all has its own respective output voltages so that its feedback information to be provided.In Fig. 2 B, linear regulator 102 is " master controls ", and switching regulaor 100 is " subordinates ", and this represents that its reference signal is the output V of linear regulator _Pa

Figure 27 has shown additional multi-mode configuration, and wherein GSM/EDGE PA 104A is connected the output terminal of switching regulaor 100 (between output terminal and L ₁Between), and WCDMAPA 104B is connected the output terminal of linear regulator 102.This configuration as mentioned above, in GSM/EDGE, can realize required performance by pure switching regulaor 100 of great use.By this hypothesis, in GSM/EDGE, can only use switching regulaor 100 and forbid linear regulator 102.This has active influence to efficient, because eliminated by inductor L ₁The loss that brings.Because L ₁On pressure drop be eliminated, so it also allows to obtain more near cell voltage V _BatMaximum GSM/EDGE PA supply voltage V ₁Inductor L ₁Can be littler, because it is as long as handle less PA 104B electric current in the WCDMA operator scheme.In this embodiment, 102 of linear regulators are enabled in the WCDMA pattern.

Notice that as shown in figure 26, if all

PA

104A and 104B are connected to same power lead, then total decoupling capacitor may be too big.As shown in figure 28, PA 104 (as the E class PA of non-limiting example) can first method of approximation and from the regulator angle with its equivalent DC resistance R _PaSimulation.In fact, and because the PA stability reasons, generally speaking must be with at least one decoupling capacitor C _PaIn parallel with PA 104.If several PA 104 are connected on the same power lead, then might use one or more shared (sharing) decoupling capacitor.In this case, connection shown in Figure 26 is feasible.Yet if each PA 104 must have its oneself decoupling capacitor, for example because capacitor must place in the PA module, it is excessive that total decoupling capacitor may become, and makes for example to realize required wide bandwidth in the WCDMA operator scheme.

A solution of this problem is to use switch to disconnect being connected of inertia PA and power lead, or disconnects its decoupling capacitor at least.Another possible solution is to connect

PA

104A, 104B on the independent current source line, for example, and as shown in figure 27.

In Figure 27 A,

regulator

104A, 104B all connect as " master control ".In GSM/EDGE, and hypothesis can obtain can forbid linear regulator 102, and only use switching regulaor 100 under the situation of acceptable energy.Yet, might also use linear regulator 102, bypass L ₁Improve to realize (certain) ripple compensation and dynamic property.In this case, if linear regulator 102 also is activated, and its feedback information is the V that applies by the switch 1 (SW1) in the GSM/EDGE position ₁In the WCDMA pattern, regulator 100,102 all is activated, and the feedback information that is used for linear regulator is V _Pa(SW1 is in the WCDMA position).

In the embodiment shown in Figure 27 B, switching regulaor 100 connects (SW1 and SW2 are all in the WCDMA position) as " subordinate " that be used for the WCDMA situation, and receives its V from the output terminal of linear regulator 102 through SW2 _Ref-swSignal.When GSM/EDGE pattern (SW1 and SW2 are all in the GSM/EDGE position), configuration and operation consider that item is as above described for Figure 27 A.

Figure 29 shows additional multi-mode configuration, and wherein

PA

104A, 104B are connected to the independent current source line of independent

linear regulator

102A, 102B output terminal.This configuration is the expansion of multi-mode configuration shown in Figure 26.Have only a switching regulaor 100, and

PA

104A, 104B be connected on the independent power lead, each is auxiliary by the linear regulator 102A, the 102B that are associated respectively, and the inductor L through being associated respectively ₁And L ₂Separate.This disposes and helps overcome the front with reference to the described excessive decoupling capacitor C of Figure 28 _PaProblem.

In Figure 29 A, switching regulaor 100 is connected as " master control " with two

linear regulator

102A, 102B, and in Figure 29 B, switching regulaor 100 connects as " subordinate ", and wherein its reference voltage is according to the linear regulator 102A of current active system (GSM/EDGE or WCDMA) selection or the output of 102B as S1.

Figure 30 shows additional multi-mode configuration, and wherein GSM/EDGE PA 104A is connected the output terminal of switching regulaor 100 (between output terminal and L ₁Between), and wherein WCDMA PA 104B and CDMA PA 104C are connected on the independent current source line of independent

linear regulator

102A, 102B output terminal.This embodiment can be considered the expansion of Figure 27 and multi-mode embodiment shown in Figure 29.Can be directly connected to the output terminal of switching regulaor 100 at GSM/EDGE PA 104A, and have at least two other PA to need quick supply voltage to modulate in the time of also can placing on the independent current source line, this embodiment is particularly useful.

In Figure 30 A, switching regulaor 100 all is connected as " master control " with two

linear regulator

102A, 102B, and in Figure 30 B, switching regulaor 100 only is connected with conduct " subordinate " in the CDMA operator scheme at WCDMA, and wherein its reference voltage is according to the linear regulator 102A of current active system (WCDMA or CDMA) selection or the output of 102B as three pole switch S1.In the GSM/EDGE pattern, switching regulaor 100 through S1 from V _mInput receives its V _{Ref_sw}The input, and thereby role as among Figure 30 A.

Should be understood that Figure 21 has shown that a kind of wherein switch sections 32 and linear segment 34 are the configuration of " master control "; Figure 22 has shown that a kind of wherein linear segment 34 is that " master control " and switch sections 32 are the configuration of " subordinate "; And Figure 23 has shown that a kind of wherein switch sections 32 and linear segment 34 are the configuration with the open loop operation of " master control " and switch sections 32.In another embodiment of the present invention, switch sections 32 can play " master control ", and linear segment 34 plays " subordinate ".

Because switch sections 32 is relatively slow, therefore preferably do not use its output V yet _PaAs the reference signal linear segment is made as " subordinate ".With reference to Figure 21, signal V _{Ctrl_sw}Directly related with the dutycycle d of the PWM voltage that is applied to pulse-width modulator 32 LC of node place wave filters.At steady state (SS) (constant V _{Ref_sw}) in, V _CtrlWith output voltage V _PaProportional.Yet, at dynamical state (indefinite V _{Ref_sw}) in, the situation difference.For example, if pass through V _{Ref_sw}Order V _PaIncrease fast, then the result is error signal V _E1Increase sharply, cause V _{Ctrl_sw}Increase sharply this command duty ratio d increase again.Since the dutycycle that increases, V _PaFinal meeting (slowly) is increased to new more high level.Because the LC wave filter, the response of dc-dc converter is increasing V _PaMiddle than increasing V _Ctrl-swWith low-response among the relevant dutycycle d many.In other words, V _{Ctrl_sw}Comprise and output voltage V _PaThe relevant information that content will take place.V _{Ctrl_sw}Increase hinting the increase of dutycycle d, thereby its expression output voltage V _PaMust increase.This information can be used for sending to linear segment 34 signal of supply of current, so that help to increase V _PaRelatively, V _{Ctrl_sw}Reduction hinting the reduction of dutycycle d, and therefore its expression output voltage V _PaMust reduce.This can be used for sending to linear segment 34 signal of ABSORPTION CURRENT, so that help to reduce V _PaTherefore, V _{Ctrl_sw}Comprise valuable information, these information can be used for linear grade 34 is made as " subordinate ".

With reference to foregoing, of the present inventionly provide another controlling mechanism in this respect, wherein, as among Figure 21, not V _{Ref_lin}=V _m, but have V _{Ref_lin}=G _C3* V _{Ctrl_sw}Relation, wherein G _C3(s) expression voltage scaling amount under the simplest situation, and more also have frequencfy-dependent behavior under the complicated situation.Suppose, as non-limiting example, G _C3(s)=1.As mentioned above, at steady state (SS) (constant V _{Ref_sw}In), V _{Ctrl_sw}With output voltage V _PaProportional.Suppose that again for this non-limiting example, proportionality constant is an identity element, so that V _Pa=V _{Ctrl_sw}, and thereby also have a V _{Ref_lin}=V _{Ctrl_sw}, so V _Pa=V _{Ref_lin}=＞V _E2The composition of=0=＞no linear part 34.If V is provided _{Ref_sw}Quick increase, then as mentioned above, this causes V _{Ctrl_sw}Quick increase, and thereby V _E2Also increase fast, produce the order that linear segment 34 requires the supply extra current.Equally, if V is provided _{Ref_sw}Quick reduction, then this causes V _{Ctrl_sw}Reduce fast, thereby cause V _E2Reduce fast, and linear segment 32 thereby quilt order ABSORPTION CURRENT.Therefore, linear segment 34 just is set as " subordinate " basically in switch sections 32 like this.

With respect to the embodiment of switch sections 32 with Figure 23 of open loop operation, similarly points for attention are suitable for, and also are applicable to the embodiment of Figure 25 A and relevant Figure 26, Figure 27, Figure 29 and Figure 30.Specifically with regard to Figure 25 A, above-mentioned control configuration hint is not V _{Ref_lin}=V _m, but have V _{Ref_lin}=V _{Ctrl_sw}Relation.Note, though do not show V among Figure 25 _{Ctrl_sw}, but V _{Ctrl_sw}Being assumed to be is internal signal to switching regulaor 100 parts, and switching regulaor 100 parts have for example structure shown in Figure 21,, also has switch sections 32 except that

control

36A and 36B that is.

Should be understood that these different embodiment of the present invention allow for can realize effective PA power supply to the multi-mode emitter structures that the PA supply voltage carries out amplitude modulation.Use some advantages of these embodiment to comprise: to cause the improvement efficient that the air time is longer and heat management improves; And/or realize the ability of required bandwidth; And/or realize the ability (hypothesis is that GSM/EDGE and WCDMA situation should provide independent device at least in advance) of multi-mode transmitter by a device.

The use of the embodiment of the invention provides a plurality of advantages, comprises high power conversion efficiency.Relatively, in battery powered communicator, provide the longer air time.Compare with using pure linear DC-DC converter, heat management problems is also more effectively managed, and also existence is eliminated at least one power supply filter capacitor (for example, the capacitor C among Fig. 4 A) together or reduced its big or small possibility at least.

It is to be noted that the conversion described that carries out is the decline type in switch sections 32 or switching regulaor 100, and have voltage mode control, this is a presently preferred embodiment.Yet, it should be understood that this conversion can be a rising/decline type.Rising/decline type is favourable, but it more is difficult to realize.Rising/decline type allows to reduce such as the cut-off voltage in the transfer tables such as cell phone, and this is because cell voltage can reduce when its charge depletion, and cut-off voltage is the minimum voltage that keeps operating mobile station.When voltage was too low, PA 6 can't produce full output power, and rising/decline type has solved this problem.For example, by using rising/decline type switch sections 32, can be with V _BatBe adjusted to and be lower than V _{M_pk}(Fig. 2), and when having only the decline type, V _BatMust equal V at least _{M_pk}Add certain tolerance limit, for example V _{M_pk}+ 0.2V.Quick A M modulation by as shown in Figure 2 changes being controlled between rising and the dropping characteristic, and like this, this transformation can not cause output voltage V _PaDistortion.In addition, with rising/decline type switch sections or converter, and at V _{M_pk}＞V _BatThe time, must be from greater than V _{M_pk}And therefore greater than V _BatThe DC source be linear segment 34 power supply so that can supply of current.

What also can notice is in voltage mode control, to have only information of voltage (for example, the output voltage of converter) to be used to generate control signal.Yet, also might also use Controlled in Current Mode and Based, wherein except that voltage, also use current information (for example, inductor current).In Controlled in Current Mode and Based, two control loops are arranged, one is used for electric current, and one is used for voltage.Certainly, also can use other more control of complicated type.

Above-mentioned explanation in view of the preferred embodiment of the present invention, will be appreciated that, these teachings are not limited to only be used for GSM/EDGE, WCDMA and/or cdma system, but can be used for having in arbitrary type system of variable amplitude envelope, wherein the PA supply voltage should be modulated by high-level efficiency and high bandwidth.

In view of the above-mentioned explanation of the preferred embodiment of the present invention, it should be understood that these teachings are not limited to only be used for E class PA, but the general linear PA that may be used on multiple SMPA usually and operate in state of saturation, as saturated category-B PA.

Above-mentioned explanation in view of the preferred embodiment of the present invention, it should be understood that these teachings are not limited to only be used for the dc-dc converter topology of arbitrary particular type and (for example, step-down, decline type are arranged not only, and also have risings/decline type), and not only be used for voltage mode and control.

In view of the above-mentioned explanation of the preferred embodiment of the present invention, it should be understood that these teachings are not limited to the linear segment that only is used to provide the switch sections of DC and AC is provided.In fact, it is desirable for switch sections also provide as far as possible AC (its attempt following with reference to the time), and linear segment provides the AC part (or the bandwidth that lacks) that lacks.Like this, embodiments of the invention have strengthened total efficiency as much as possible, and are big more from the composition of switch sections or converter because in principle, and efficient is just high more.

In view of the above-mentioned explanation of the preferred embodiment of the present invention, compensated the imperfect dynamic perfromance of switching stage though it should be understood that linear grade, imperfect dynamic perfromance also causes by imperfect PA behavior (for example load variations) to a certain extent, i.e. R _PaWith V _PaChange (that is, at V _PaIncrease during reduction) and be in mismatch condition.Therefore, linear grade 34, the 102 imperfect dynamic perfromance of compensating switch converter (for example, bandwidth not enough and/or be referenced to peaking in the output characteristics) at least.In addition, in this regard, linear grade 34,102 and switching stage 32,100 complement each other, to obtain the specific required output-transfer function (specific bandwidth is not only arranged, and the given shape of transport function is arranged) that is referenced to.For example, linear grade 34,102 can have this type of and be referenced to output-transfer function, and the resulting output-transfer function that is referenced to that mixes (switch/linearity) power supply is or near planar second-order Butterworth filter type.Therefore, linear grade 34,102 can be used for always being referenced to the output-transfer function shaping to resulting, so that obtain desirable characteristics.Linear grade 34,102 also helps tracking reference signal, and can be used for obtaining reference signal V _mSpecific required tracking performance.

Linear grade 34,102 is compensating switch ripple at least also, and also can compensate nonideal PA behavior at least, such as R _PaVariation with condition of work.

Should also be understood that the secondary inductor L that in Figure 24, introduces ₁In fact have the effect that is similar to the inductor of converter shown in Fig. 6 L, consequently produce current source characteristic.A difference is, in the embodiment of Fig. 6 and those embodiment subsequently, input end at inductor L has applied the PWM rectangular voltage, and in the embodiment of Figure 24 and those embodiment subsequently, level and smooth voltage (output of dc-dc converter 100) is applied to secondary inductor L ₁Input end.

In view of the above-mentioned explanation of the preferred embodiment of the present invention, it should be understood that under the GSM/GMSK modulation case AC-battery power source is carried out " power control " function, thus by regulating power level with the power supply regulate voltage levels.Like this, be appreciated that differently with AM control, used is " substep control ".Notice that target can be to improve PA 6 efficient, particularly when using linear PA.When using linear PA, will there be another mechanism that regulates power level generally speaking, even at stabilized power source voltage V _BatSituation under, but subsequently when low-power level more efficient can reduce, and the DC level can be lowered to raise the efficiency.Yet when using SMPA, output power (mainly) controlled by supply voltage.Like this, preferably use PA power supply 30 power controlling.

In any case, for rapid mixing power supply 30 according to the preferred embodiment of the invention, and for the GSM situation: a) in the TX structure, the PA power supply is used for power controlling; B) the PA power supply needn't be very fast (though there are some requirements relevant with power ramp/oblique deascension, they are lower than the requirement of EDGE situation); And c) advantageously, by AC-battery power source 30 compensating switch ripples, as in the EDGE situation.

Above-mentioned explanation provides the best approach that the present inventor imagines for enforcement the present invention at present and the complete sum informedness explanation of equipment by demonstration and non-restrictive example.Yet when reading with appended claims in conjunction with the accompanying drawings, in view of the above description, those skilled in the relevant art can understand various changes and modification.For example, though power supply of the present invention has been described in the context of polarity or ER transmitter embodiment in the above, the present invention can use other and use, and wherein power supply must satisfy strict dynamic requirements, also shows high-level efficiency simultaneously.In addition, the quantity of unintelligible possible the embodiment that can suppose for restriction mixed-voltage regulator of the different embodiment of Fig. 6-30, or limit the RF power amplifier that the embodiment of the invention can be used for and the quantity of RF communication system types.Usually, this type of and similar modification of all of teachings of the present invention will drop in the scope of the embodiment of the invention.

In addition, characteristics more of the present invention can advantageously be used and need not other characteristic of corresponding use.Similarly, above-mentioned explanation should be considered as principle of the present invention just is described, rather than limit it.

Claims

1. DC-to-DC converter comprises:

Be used to be coupling in the switching mode part between DC source and the load, described switching mode partly provides x the output power of amount; And

And described switching mode part is coupling in the linear model part between identical or different DC source and the described load in parallel, and described linear model partly provides y the output power of amount, and wherein x is greater than y, and can be the ratio of special applications constrained optimization x and y,

Wherein said linear model part partly reveals the response time faster to the required change list of output voltage than described switching mode, and wherein said linear model partly provides AC compounent to described load.。

2. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprise at least one power operational amplifier as the variable voltage source operation.

3. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprise at least one the Power arithmetic trsanscondutance amplifier as the variable current source operation.

4. DC-to-DC converter as claimed in claim 1, wherein said linear model part provides DC component and AC compounent to described load.

5. DC-to-DC converter as claimed in claim 1, the output compensation of wherein said linear model part is from the AC ripple output of described switching mode part.

6. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprises the bi-directional voltage control voltage source.

7. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprise bi-directional voltage control voltage source (VCVS), and comprise two Voltage-controlled Current Source circuit, and wherein when operation, one is operating as the place, and one is operating as the source.

8. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprise bi-directional voltage Control current source.

9. DC-to-DC converter as claimed in claim 1, wherein said linear model partly comprise bi-directional voltage Control current source (VCCS), and comprise two Voltage-controlled Current Source circuit, and one of them is operating as the place, and one is operating as the source.

10. DC-to-DC converter as claimed in claim 1, wherein said switching mode part and described linear model part are controlled by control signal jointly with closed-loop fashion.

11. by the output control from described linear model part, and wherein said linear model part is controlled by control signal with closed-loop fashion with closed-loop fashion for DC-to-DC converter as claimed in claim 1, wherein said switching mode part.

12. DC-to-DC converter as claimed in claim 1, wherein said switching mode part is with the open loop operation, and wherein said linear model part is controlled by control signal with closed-loop fashion.

13. DC-to-DC converter as claimed in claim 1, wherein said linear model part is subordinated to the operation of described switching mode part effectively, with supply or ABSORPTION CURRENT.

14. DC-to-DC converter as claimed in claim 1, wherein said switching mode partly provide minimum or the no-output filter capacity, to play current source in fact.

15. DC-to-DC converter as claimed in claim 1, wherein said switching mode partly provides output filter electric capacity, and plays voltage source in fact.

16. DC-to-DC converter as claimed in claim 1, wherein said switching mode partly is coupled to described load, and by being inductively coupled to the output of described linear model part.

17. DC-to-DC converter as claimed in claim 1, wherein said load comprises at least one radio-frequency power amplifier.

18. DC-to-DC converter as claimed in claim 10, wherein said load comprise at least one radio frequency (RF) power amplifier, and wherein said control signal comprises the radio-frequency carrier modulation signal.

19. DC-to-DC converter as claimed in claim 11, wherein said load comprise at least one radio frequency (RF) power amplifier, and wherein said control signal comprises the radio-frequency carrier modulation signal.

20. DC-to-DC converter as claimed in claim 12, wherein said load comprise at least one radio frequency (RF) power amplifier, and wherein said control signal comprises the radio-frequency carrier modulation signal.

21. DC-to-DC converter as claimed in claim 1, wherein said linear model partly are coupled to the output of described switching mode part, and are capacitively coupled to described load.

22. DC-to-DC converter as claimed in claim 1, wherein said switching mode partly is coupled to described load, and by being inductively coupled to the output of described linear model part, and wherein said linear model part is inductively coupled to the output partly of described switching mode through described, and is capacitively coupled to described load.

23. DC-to-DC converter as claimed in claim 1, wherein said linear model part compensating load at least to a certain extent changes.

24. DC-to-DC converter as claimed in claim 1, wherein said linear model part compensates the imperfect dynamic perfromance of described switching mode part at least to a certain extent.

25. radio frequency (RF) transmitter (TX) that is used to be coupled to antenna, described transmitter has the polar structure of being made up of amplitude modulation (AM) path of being coupled to power amplifier (PA) power supply and phase modulation (PM) path of being coupled to described power amplifier input end, wherein said power supply comprises the switching mode part that is used to be coupling between power source and the described power amplifier, described switching mode partly provides x the output power of amount, described power supply also comprise and described power source and described power amplifier between the linear model part of described switching mode part parallel coupled, described linear model partly provides y the output power of amount, wherein x is greater than y, and can be the ratio of special applications constrained optimization x and y, and wherein said linear model part reveals the response time faster than described switching mode part to the required change list of output voltage, and wherein said linear model partly provides AC compounent to arrive described power amplifier.

26. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprise at least one power operational amplifier as the variable voltage source operation.

27. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprise at least one the Power arithmetic trsanscondutance amplifier as the variable current source operation.

28. radiofrequency launcher as claimed in claim 25, wherein said linear model partly provide DC component and AC compounent to described power amplifier.

29. radiofrequency launcher as claimed in claim 25, the output compensation of wherein said linear model part is from the AC ripple output of described switching mode part.

30. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprises the bi-directional voltage control voltage source.

31. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprise bi-directional voltage control voltage source (VCVS), and comprise two Voltage-controlled Current Source circuit, wherein when operation, one is operating as the place, and one is operating as the source.

32. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprise bi-directional voltage Control current source.

33. radiofrequency launcher as claimed in claim 25, wherein said linear model partly comprise bi-directional voltage Control current source (VCCS), and comprise two Voltage-controlled Current Source circuit, one of them is operating as the place, and one is operating as the source.

34. radiofrequency launcher as claimed in claim 25, jointly by control signal control, wherein said control signal comprises amplitude-modulated signal with closed-loop fashion for wherein said switching mode part and described linear model part.

35. radiofrequency launcher as claimed in claim 25, wherein said switching mode part with closed-loop fashion by output control from described linear model part, and wherein said linear model part is controlled by control signal with closed-loop fashion, and wherein said control signal comprises amplitude-modulated signal.

36. radiofrequency launcher as claimed in claim 25, wherein said switching mode part be with the open loop operation, and wherein said linear model part controlled by control signal with closed-loop fashion, wherein said control signal comprises amplitude-modulated signal.

37. radiofrequency launcher as claimed in claim 25, wherein said linear model part is subordinated to the operation of described switching mode part effectively, with supply or ABSORPTION CURRENT.

38. radiofrequency launcher as claimed in claim 25, wherein said switching mode partly provide minimum or the no-output filter capacity, so that play current source in fact.

39. radiofrequency launcher as claimed in claim 25, wherein said switching mode partly provides output filter electric capacity, and plays voltage source in fact.

40. radiofrequency launcher as claimed in claim 25, wherein said switching mode partly is coupled to described power amplifier, and by being inductively coupled to the output of described linear model part.

41. radiofrequency launcher as claimed in claim 25, wherein said switching mode partly is coupled to described power amplifier, and by being inductively coupled to the output of described linear model part, and wherein said linear model part is inductively coupled to the output partly of described switching mode through described, and is capacitively coupled to described power amplifier.

42. radiofrequency launcher as claimed in claim 25, wherein said power supply provides bigger power conversion efficiency than the power supply based on pure linear voltage regulator, also provides simultaneously than based on the wideer bandwidth of operation of pure switched-mode power supply.

43. radiofrequency launcher as claimed in claim 25, wherein said linear model partly are coupled to the output of described switching mode part, and are capacitively coupled to described load.

44. radiofrequency launcher as claimed in claim 25, wherein said linear model part compensates the described load variations that is shown by described power amplifier at least to a certain extent.

45. radiofrequency launcher as claimed in claim 25, wherein said linear model part compensates the imperfect dynamic perfromance of described switching mode part at least to a certain extent.

46. radio frequency (RF) transmitter (TX) that is used to be coupled to antenna, described transmitter has the polar structure of being made up of amplitude modulation (AM) path of being coupled to power amplifier (PA) power supply and phase modulation (PM) path of being coupled to described power amplifier input end, wherein said power supply comprises the switching mode level that is used to be coupling between power source and the described power amplifier, described switching mode level provides x the output power of amount, described power supply also comprise and described power source and described power amplifier between at least one linear model level of described switching mode level parallel coupled, described linear model level provides y the output power of amount, wherein x is greater than y, and can be the ratio of special applications constrained optimization x and y, described power supply also comprises at least one auxiliary induction between the output terminal of the output terminal that is coupling in described switching mode level and described at least one linear model level, and wherein said at least one linear model level provides AC compounent to arrive described power amplifier.

47. radiofrequency launcher as claimed in claim 46, wherein said power amplifier is coupled to the output terminal of described switching mode level before the described auxiliary induction, and comprises at least one the secondary power amplifier that is coupled to described switching mode level output terminal behind the described auxiliary induction.

48. radiofrequency launcher as claimed in claim 46, wherein said power amplifier is coupled to the output terminal of described switching mode level behind the described auxiliary induction, and comprises at least one the secondary power amplifier that also is coupled to described switching mode level output terminal behind the described auxiliary induction.

49. radiofrequency launcher as claimed in claim 46, wherein said power amplifier is coupled to the output terminal of described switching mode level before first auxiliary induction and second auxiliary induction, and the 3rd power amplifier of described switching mode level output terminal after comprising second power amplifier that is coupled to described switching mode level output terminal behind first auxiliary induction and being coupled to second auxiliary induction.

50. radiofrequency launcher as claimed in claim 49, wherein second power amplifier also is coupled to the output terminal of the first linear model level, and the 3rd power amplifier is coupled to the output terminal of the second linear model level.

51. radiofrequency launcher as claimed in claim 46, wherein said power amplifier is coupled to the output terminal of described switching mode level behind first auxiliary induction, and also comprise second power amplifier that is coupled to described switching mode level output terminal behind second auxiliary induction, wherein said power amplifier also is coupled to the output terminal of the first linear model level, and second power amplifier is coupled to the output terminal of the second linear model level.

52. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level comprise at least one power operational amplifier as the variable voltage source operation.

53. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level comprise at least one the Power arithmetic trsanscondutance amplifier as the variable current source operation.

54. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level provide DC component and AC compounent to described power amplifier.

55. radiofrequency launcher as claimed in claim 46, the output compensation of wherein said at least one linear model level is from the AC ripple output of described switching mode level.

56. radiofrequency launcher as claimed in claim 46, jointly by control signal control, wherein said control signal comprises amplitude-modulated signal with closed-loop fashion for wherein said switching mode level and described at least one linear model level.

57. radiofrequency launcher as claimed in claim 46, wherein said switching mode level is controlled by the output from described at least one linear model level with closed-loop fashion, and wherein said at least one linear model level is controlled by control signal with closed-loop fashion, and wherein said control signal comprises amplitude-modulated signal.

58. radiofrequency launcher as claimed in claim 46, wherein said switching mode level is operated with open loop, and wherein said at least one linear model level controlled by control signal with closed-loop fashion, and wherein said control signal comprises amplitude-modulated signal.

59. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level is subordinated to the operation of described switching mode level effectively, with supply or ABSORPTION CURRENT.

60. radiofrequency launcher as claimed in claim 46, wherein said switching mode level provides output filter electric capacity, and plays voltage source in fact, and provides level and smooth voltage signal to arrive described auxiliary induction.

61. radiofrequency launcher as claimed in claim 46, the grade coupled output terminal of wherein said at least one linear model to described switching mode level, and be capacitively coupled to described power amplifier.

62. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level compensate the variation of the described load that is shown by described power amplifier at least to a certain extent.

63. radiofrequency launcher as claimed in claim 46, wherein said at least one linear model level compensates the imperfect dynamic perfromance of described switching mode level at least to a certain extent.

64. radiofrequency launcher as claimed in claim 46 also comprises the switch of the reference input that is coupled to described switching mode level, optionally different reference signals is applied to described switching mode level so that become with operator scheme.

65. as the described radiofrequency launcher of claim 64, wherein a kind of operator scheme comprises the GSM pattern.

66. as the described radiofrequency launcher of claim 64, wherein a kind of operator scheme comprises the EDGE pattern.

67. as the described radiofrequency launcher of claim 64, wherein a kind of operator scheme comprises the CDMA pattern.

68. as the described radiofrequency launcher of claim 64, wherein a kind of operator scheme comprises the WCDMA pattern.

69. radiofrequency launcher as claimed in claim 46 also comprises the switch of the feedback input end that is coupled to described at least one linear model level, optionally different feedback signals is applied to described at least one linear model level so that become with operator scheme.

70. as the described radiofrequency launcher of claim 69, wherein a kind of operator scheme comprises the GSM pattern.

71. as the described radiofrequency launcher of claim 69, wherein a kind of operator scheme comprises the EDGE pattern.

72. as the described radiofrequency launcher of claim 69, wherein a kind of operator scheme comprises the CDMA pattern.

73. as the described radiofrequency launcher of claim 69, wherein a kind of operator scheme comprises the WCDMA pattern.

74. the method for operation radio frequency (RF) transmitter (TX), wherein said transmitter has the polar structure of being made up of amplitude modulation (AM) path of being coupled to power amplifier (PA) power supply and phase modulation (PM) path of being coupled to described power amplifier input end, and described method comprises:

Described power supply is provided, so that comprise the switching mode part that is coupling between power source and the described power amplifier, described switching mode partly provides x the output power of amount; And

With the described switching mode part parallel coupled between linear model part and described power source and the described power amplifier, described linear model partly provides y the output power of amount, wherein x is greater than y, and can be the ratio of special applications constrained optimization x and y, and wherein said linear model part partly reveals the response time faster to the required change list of output voltage than described switching mode, and wherein said linear model partly provides AC compounent to arrive described power amplifier.

75. as the described method of claim 74, wherein said linear model partly comprises as at least one power operational amplifier of a variable voltage source operation or at least one Power arithmetic trsanscondutance amplifier of operating as variable current source.

76. as the described method of claim 74, wherein said linear model partly provides DC component and AC compounent to described power amplifier.

77., also comprise in the imperfect dynamic perfromance of described load variations that the described linear model of operation part shows from the AC ripple output of described switching mode part, described power amplifier with compensation and described switching mode part at least one as the described method of claim 74.

78. as the described method of claim 74, wherein said linear model partly comprises bi-directional voltage control voltage source or bi-directional voltage Control current source.

79. as the described method of claim 74, comprise also with closed-loop fashion and control described switching mode part and described linear model part jointly that wherein said control signal comprises amplitude-modulated signal with control signal.

80., also comprise using from the output of described linear model part and control described switching mode part, and control described linear model part with closed-loop fashion with the control signal that comprises amplitude-modulated signal with closed-loop fashion as the described method of claim 74.

81., also comprise with open loop approach and operate described switching mode part, and control described linear model part with closed-loop fashion with the control signal that comprises amplitude-modulated signal as the described method of claim 74.

82., comprise that also the described linear model part of operation is so that be subordinated to the operation of described switching mode part effectively, with supply or ABSORPTION CURRENT as the described method of claim 74.

83., comprise that the described switching mode part of operation is to play current source in fact as the described method of claim 74.

84., comprise that the described switching mode part of operation is to play voltage source in fact as the described method of claim 74.

85., also comprise described switching mode partly is coupled to described power amplifier, and by being inductively coupled to the output terminal of described linear model part as the described method of claim 74.

86. as the described method of claim 74, also comprise described switching mode partly is coupled to described power amplifier, and by being inductively coupled to the output terminal of described linear model part, and wherein said linear model part is inductively coupled to described switching mode output terminal partly through described, and is capacitively coupled to described power amplifier.

87., also comprise the output terminal that described linear model partly is coupled to described switching mode part, and be capacitively coupled to described power amplifier as the described method of claim 74.