CA2513599A1 - High voltage to low voltage bidirectional converter - Google Patents

Abstract

The invention provides an improved method of efficiently converting a high voltage, to lower voltage with the same or changed waveform, in a bi-directional manner.

Description

FIELD OF THE INVENTION
The present invention relates generally to a power supply for DC or AC devices and in particular, to a power supply for converting any type of high voltage DC or AC to another voltage DC or AC, alternately the power supply may be restricted to just High Voltage DC to DC conversion, High Voltage AC to AC, high voltage DC to AC, High voltage AC to DC, or any other combination therein.
BACKGROUND OF THE INVENTION
Efficient conversion of high voltage to a lower voltage has become a problem with the advancement of a number of technologies. One is the continuing development of the Electro-hydrodynamic or Electro-kinetic Generator, which produces a high voltage output in the order of a few l Os of kV. Such a high voltage has few useful applications directly, for that reason it must be converted to a much lower voltage.
Advancement in solid stage microwave amplifiers has necessitated the development of replacement modules for high voltage vacuum tube based microwave devices. The requirement is for a drop in replacement for the vacuum tube, which requires the converter to change the high voltage, used by the vacuum tube amplifier, to the lower voltage, required by the solid state replacement. Another emerging market pertains to advances in energy storage in high voltage capacitors may involve the need to efficiently convert a high voltage to a lower DC voltage. The demand for higher efficiency in basic power transmission has as well started the search for alternative methods of distributing AC power throughout an AC power transmission system.
Description of PRIOR ART
Typical methods currently used are represented by the following list of documents.
U.S. PATENT DOCUMENTS
3,022,430 2/ 1962 Townsend . ...... . .... '????
4,105,939 8/1978 Culberston............318/599 4,290,108 9/1981 Woehrle et al. .......364/480 4,399,499 8/1983 Butcher et al. ........363/17 4,742,441 05/1988 Akerson..............363/97 5,119,285 6/1992 Liu et al. ............363/44 5,255,174 10/1993 Murugan.............363/17 5,666,278 9/1997 Ng et al. ..............363/71 5,815,384 9/1998 Hammond et al .....363/26 5,943,229 8/1999 Sudhoff................363/125 US Patent Application by the inventor US Patent App 11/133,189 Titled: DC HIGH VOLTAGE TO DC LOW VOLTAGE CONVERTER, Kelly, David, filed 20 May, 2005 OTHER PUBLICATIONS
Dennis A. Woodford "HVDC Transmission" Manitoba HVDC Research Centre, 18 March 1998, 27 pages.
FIG. 1 shows part of the typical technology employed by the power industry for transmitting high voltage, high power DC across large distances. The technical reference Dennis A. Woodford "HVDC
Transmission" Manitoba HVDC Research Centre, 18 March 1998, 27 pages, provides much more detail. The typical voltages are 500kV, using 200 or more high voltage solid-state switches in series. These solid-state switches are slow and designed for very high levels of power, not suitable for use at the low power levels of the current invention. In FIG. 1, switches 10I, 102, 103 are in series and pull the end of capacitor 107 to Vdc+ 150 when the SWITCH DRIVE 155 is in the first state shown by the table called SWITCH DRIVE
154. Alternately, as you progress along the clock table switches 101, 102, 103 open and then 104, 105, 106 are closed and connect the end of capacitor 107 to Vdc- 152. The resulting action of alternating the connections of capacitor 107 between Vdc+ 150 and Vdc- 151 creates a square wave on the primary of transformer 108, which is then reduced in voltage, rectified into a lower voltage DC Vout+ 152 and Vout-153. Alternately, the output of transformer 108 is filtered to make a clean AC
waveform by removing rectifiers 109, 110 and replacing them with a suitable filter. The disadvantage of this technology is that for lower power operation the switch losses are large when the frequency of operation is increased as proposed in the invention. The very high losses encountered when operating at high frequency are undesirable for a cost of operation standpoint. Further, the cost benefits of operating at high frequency, smaller size for transformer 108 and capacitors 107, 111 are not possible with currently used methods. As well the large number of switches stacked in series requires special protection circuits, not shown in FIG. 2, to ensure that all switches share the voltage equally, increasing the cost of manufacture.
FIG. 2 comes from US Patent App 11/133,189 Titled: DC HIGH VOLTAGE TO DC LOW
VOLTAGE CONVERTER, inventor Kelly, David, filed 20 May, 2005 and represents prior art. A detailed explanation is as follows. The converter is capable of producing a well regulated output as it has components such as PWM MODULE 232, SWITCH DRIVER 233 that may be PWM (Pulse Width Modulation) in a similar manner as used by commercial AC to DC switching power supplies.
Switches 200, 201; 202, 203;
204, 205 form three half bridges that are connected in series. Capacitors 206, 207, 211,212 filter the switch current pulses reducing the AC that is generated by the half bridges across the high voltage DC input Vdc+
250 and Vdc- 251. The addition of resistors 214, 215 and 216 are used to force the voltages to be equal across capacitors 206, 207 and 211 during the start-up time that the half bridges are off. Capacitor 212 is used to provide a start-up for the START MODULE 231 which has various components that store sufficient charge to run the half bridges for a specific time after which an auxiliary winding 260 from transformer 218 supplies the necessary power to run the control electronics. Alternately, an external DC or AC power source, not shown, provides the power to operate the DC to DC converter and is either common to or close to either Vdc+ 250 or Vdc- 251.
In FIG. 2 the FEEDBACK 230 supplies an error signal used by the PWM MODULE 232 to generate the appropriate width clock signals that are supplied to the SWITCH DRIVER
233, which then drives the switches 200, 201, 202, 203, 204, 205, with SWITCH DRIVE 254 a typical set of waveforms. The additional circuits function as follows. When high voltage power is first applied to Vdc+
250 and Vdc- 251, the resistors 214, 215 and 216 charge capacitor 212. The START MODULE 231 determines when it has enough charge to operate the PWM MODULE 232 and SWITCH DRIVER 233 for a predetermined time.
Alternately, the START MODULE 231 may be power by an external low voltage DC or AC source.
After initially powering the converter electronics, the START MODULE 231 receives a low voltage AC from transformer 218 through secondary 260. The powered from this secondary 260 then provides the low voltage power to sustain operation of the PWM MODULE 232 and SWITCH DRIVER 233.
Further in FIG. 2, the START MODULE 231 has started the DC-to-DC converter the FEEDBACK
230 provides to the PWM MODULE 232, a signal, which that is proportional in some way to the output voltage. The FEEDBACK 230 may use optical isolation, an isolation transformer etc. none of which are shown to provide this isolated feedback signal to the PWM MODULE 232. The method is no different than that used for traditional off the shelf power supplies except that the isolation voltage rating is substantially greater. When the SWITCH DRIVE 254 is decreased from full duty, 50% of full duty is shown as an example, then the waveform that appears on the secondary of transformer 218 is not a full duty square wave but has positive and negative pulses which are proportional in width to the SWITCH DRIVE 254 wave form.
Diodes 219, 220 rectify the secondary AC into a pulsating DC, which is then filtered by inductor 221 and capacitor 210. The output inductor 221 and capacitor 210 filters the pulsating DC into a average value equal to the duty of the waveform times it's amplitude. The circuit functions exactly in the same manner as a switching power supply commonly called a FORWARD CONVERTER, except it provides a regulated low DC voltage output from a very High voltage input.
The switches, 200, 201, 202, 203, 204, 205 are typically semi-conductor devices that have a reverse diode across them to clamp any reverse voltage that may be generated by transformer 218 during the time the SWITCH DRIVE 254 changes state. The combination of the switches 200, 201, 202, 203, 204, 200 capacitor 208, 209, 213 and primary of transformer 218 may be combined in many different ways though function in the same method as shown in FIG .2.
The following patents are converters but not all are designed specifically for high voltage to low voltage, U.S. Pat. No. 5,199,285, Jun 2, 1992, "Solid State Power Transformer Circuit"; U.S. Pat. No.
5,666,278, Sept 9, 1997, "High Voltage Inverter Utilizing Low Voltage Power Switches"; U.S. Pat. No.
5,943,229, Aug 24, 1999, "Solid State Transformer".
Other related patents are art that are either bi-directional, specifically designed for DC to DC
operation or related in some way DC to AC conversion; U.S. Pat. No. 4,105,939 Aug. 8, 1978, "Direct Digital Technique For Generating An AC Waveform"; U.S. Pat. No. 4,290,108, Sept. 15, 1981, "Control Unit For A Converter"; U.S. Pat. No. 4,399,499, Aug. 16, 1983; "Bi-Lateral Four Quadrant Power Converter";
U.S. Pat. No. 4,742,441, May 3, 1988, "High Frequency Switching Power Converter"; U.S. Pat. No.
5,255,174, Oct. 19, 1993, "Regulated Bi-Directional DC-TO-DC Voltage Converter Which Maintains A
Continuous Input Current During Step-UP Conversion"; U.S. Pat. No. 5,815,384, Sept. 29, 1998, "Transformer Which Uses Bi-directional Synchronous Rectification To Transform The Voltage Of An Output Signal Having A Different Voltage And Method For Effecting Same". Of all of these fore mentioned documents U.S. Pat. No. 4,399,499, Aug. 16, 1983; "Bi-Lateral Four Quadrant Power Converter" and U.S.
Pat. No. 4,742,441, May 3, 1988, "High Frequency Switching Power Converter"
represent a technological base for all subsequent bi-directional power supply or converter art.
SUMMARY OF THE INVENTION
The purpose of the invention is to provide an improved method of converting a high voltage DC or AC into a regulated lower voltage DC or AC. The preferred embodiment of the invention consists of a plurality of switches connected in series to a high voltage DC source. The switches are operated as an even number of pairs to form a plurality of half bridges, which are further connected to make one or a plurality of full bridges. The switches are operated using a predefined, controlled switching sequence, which may change depending on the type of input DC or AC and the desired output waveform. The SWITCH DRIVE operates using a phase shifted PWM (pulse width modulated) method of control such that the phase difference between two full duty square-wave half bridge inverters, forming a full bridge is controlled, with the combined output of the bridge being a PWM (pulse width modulated) waveform.
The SWITCH DRIVE may be generated using many different methods such as using an existing integrated circuited designed for this purpose, a lookup memory device with a microprocessor combined with additional analogue and or logic circuits or a set of analogue or logic circuits with or without the use of a lookup memory device. The SWITCH DRIVE circuit may be powered by a separate power source or alternately a special start-up run control circuit that operates from the high voltage input. The outputs of the switches are then connected to the primary of a single or plurality number of isolation transformers that have a single or multiple primaries. In the preferred embodiment each primary of the isolation transformers) will have one or more capacitor in series to block the flow of DC voltage. This preferred embodiment has at least one or a plurality of isolated secondary that have the output rectified and filtered to provide the intended regulated low voltage DC output.
Another preferred embodiment is a universal converter that provides a well-regulated low voltage DC
or AC output, from either a DC or AC input. AC operation require the input switches to be bi-directional or AC. It consists of a plurality of switches connected in series to a high voltage DC or AC source. The switches are operated as an even number of pairs to form a plurality of half bridges, which are further connected to make one or a plurality of full bridges. The switches are operated using a predefined, controlled switching sequence, which may change depending on the type of input used or desired output. The SWITCH
DRIVE operates using a phase shifted PWM (pulse width modulated) duty such that the phase difference between two full duty square-wave half bridge inverters, forming a full bridge is controlled, with the combined output of the bridge being a PWM (pulse width modulated) waveform.
The SWITCH DRIVE may be generated using many different methods such as using an existing integrated circuited designed for this purpose, a lookup memory device with a microprocessor combined with additional analogue and or logic circuits or a set of analogue or logic circuits with or without the use of a lookup memory device. The switch drive circuit may be powered by a separate power source or alternately a special start-up run control circuit that operates from the high voltage input. The outputs of the switches are then connected to the primary of either a single or plurality number of isolation transformers with single or multiple primaries. In the preferred embodiment each primary of the isolation transfonner(s) will have one or more capacitor in series to block the flow of DC voltage. This preferred embodiment has at least one or a plurality of isolated secondary that have the output rectified by switches with the outputs of these switches feeding the input of one or more inductor(s). The output of this inductors) is then connected to a capacitor to filter out any undesired ripple voltage. The resulting output waveform may be then changed or regulated using feedback and a control circuit that alters the duty of the drive signals applied to the switches.
A variation of both preferred embodiments uses a plurality of diodes or DC
switches or combination in the transformer secondary circuit, for a regulated DC output.
A further variation of both preferred embodiments, a plurality of bi-direction AC switches in the secondary circuit for regulated bi-directional DC or AC output.
Another preferred embodiment uses one or plurality of converters to take power from a battery, fuel cell, capacitor or fly wheel or combination thereof and converts it to accelerate or maintain the speed of a Drive Motor. When it is desired to slow the Drive Motor then the converter transfers the power from the Drive Motor, now operated as a generator and transfers it back to the power source.
Another preferred embodiment a plurality of half bridges instead of full bridges, connected in series to a high voltage DC input to drive the primary of a transformer combined with switches on the secondary side to rectify the transformer secondary into a DC or AC output of desired polarity.
Another preferred embodiment a plurality of bi-directional half bridges in series across a high voltage input to drive the primary of a transformer instead of bi-directional full bridges connected in series, combined with bi-directional switches on the transformer secondary side to generate a DC or AC output as determined by the duty of the PWM and polarity of the secondary switches operation with respect to the primary.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 depicts a method using series switches for high voltage to low voltage DC;
FIG. 2 is a schematic representation showing a method that is capable of providing a regulated output;
FIG. 3 a schematic representation showing a method that is capable of providing a regulated DC output;
FIG. 4 is a schematic representation showing yet another method that is capable of providing a regulated bi-directional DC or AC output;

FIG. 5A, 5B, 5C is a schematic representation showing various types of switches;
FIG. 6A, 6B is a schematic representation showing various types of filter arrangements;
FIG. 7 is a schematic representation showing a different secondary side switch arrangement.
FIG. 8 is a schematic representing a method of connecting a power source to an output device.
FIG. 9 is a schematic representing a method of connecting a power source to an output device.
FIG. 10 is a schematic representing alternate switch and power supply waveforms.
DETAILED DESCRIPTION OF THE INVENTION
The Embodiment in FIG. 3 in accordance with the present invention is intended for producing a regulated DC output or unregulated DC output by omitting FEEDBACK 340. The HIGH VOLTAGE
input is connected to INPUT 350 and 351. SWITCHES 300 through 307 are DC type if DC
HIGH VOLTAGE is applied to INPUT 350, 351 and bi-directional or AC type if connected to an AC
HIGH VOLTAGE.
SWITCHES 300, 301 form one half bridge and 302, 303 form the other side of a half bridge which is combined and operated as a full bridge with capacitor C319 blocking the DC
component from being applied to the primary of transformer 321. Both half bridges operate at full duty cycle with each switch ON for 50%
of the time. To create a PWM output the phase of the half bridge 302, 303 is shifted from that of switches 300, 301 by an amount equal to the desired pulse width. SWITCH DRIVE 354 provides and example of this, with signal A the position of the SWITCH 300, HIGH or UP representing the ON
or CLOSED state. B, C, D
corresponds to SWITCH 301, 302, 303 accordingly and J represents the difference signal that appears across the primary of TRANSFORMER 321. The duty of the primary waveform J on TRANSFORMER 321 can be changed by varying the phase of SWITCH signals A, B with respect to C, D. This type of switching method facilitates the use of isolation transformers for coupling the gate signals to the switches and can accommodate any PWM duty from 0 to 100% without worry of saturation of the switch driver transformers, not shown in the drawing. SWITCHES 304, 305 form one half bridge and 306, 307 form a second half bridge together making a full bridge and function in combination with CAPACITOR 320 and TRANSFORMER 323 in the same manner as SWITCH 300, 301, 302, 303, CAPACITOR 319, TRANSFORMER 321. An example of the switch drive waveforms is shown by SWITCH DRIVE 354 which demonstrates operation at a duty cycle of 33%. In this example separate TRANSFORMERS 321, 323 are used but the primary of a single TRANSFORMER may be shared with additional capacitor isolation or through the use of separate primaries on a common transformer. DIODES 325 and 326 rectify the output of TRANSFORMER
321 and apply the pulsating rectified DC pulses shown by waveform J on the SWITCH DRIVE table 354. These pulses are filtered by INDUCTOR 329 and CAPACITOR 331 to the desired degree. In a similar manner DIODES 327, 328 rectify the output of TRANSFORMER 323 and the pulsating DC is filtered by INDUCTOR 330 and CAPACITOR 331. If the phase of TRANSFORMER 323 is operated shifted by 67 degrees from TRANSFORMER 321 as in SWITCH DRIVE 354, then the ripple frequency across CAPACITOR 331 will be increased and the pulsating DC currents greatly reduced allowing a much smaller capacitor value to be used and this technique maybe extended to a plurality of transformers.
Alternately, by combining the switches in such a way to use a single TRANSFORMER, then only one set of diodes and inductor will be required, reducing the number of components and manufacturing cost of the design.
Capacitors 308, 309, 310, 311, 312 filter the SWITCH 300 through 307 current pulses reducing the high frequency AC that is generated by the half bridges across the INPUT 350 and 351. The addition of resistors 314, 315, 316, 317 and 318 are used to force the voltages to be equal across capacitors 308, 309, 310, 311, 312 during the start-up, the time that the half bridges are off.
Capacitor 313 is used to provide start-up power for the START MODULE 341, which has various internal components that store sufficient charge to run the halfbridges for a specific time after which an auxiliary winding 324 from transformer 323 supplies the necessary power to run the control electronics. Alternately, an external DC or AC, not shown, provides the power to operate the DC to DC converter and may be common to or close to either INPUT 350 or 351.
In FIG. 3 REFERENCE 390 provides a voltage proportional to the desired output voltage and FEEDBACK 340 supplies a feedback signal proportional to the secondary output voltage, both of which are used by the PWM MODULE 342 to generate the appropriate PWM phased clock signals that are supplied to the SWITCH DRIVER 343, which then drives the switches 300, 301, 302, 303, 304, 305, 306, 307. The additional circuits function as follows. When HIGH VOLTAGE is first applied to INPUT 350 and 351, the resistors 314, 315, 316, 317 and 318 charge capacitor 313. The START MODULE
341 takes the charge from CAPACITOR 313 and determines when it is sufficient to operate the PWM MODULE
342 and SWITCH
DRIVER 343 for a predetermined time. For operation from a HIGH VOLTAGE AC
INPUT the START
MODULE 342 takes the current normally charging CAPACITOR 313 and rectifies it and stores the charge internally until a sufficient level has built up to initiate startup of the power supply. Alternately, the START
MODULE 341 may be powered by an external low voltage DC or AC source, not shown in FIG. 3. After initially powering the converter electronics, the START MODULE 341 receives low voltage AC power from transformer 323 through secondary 324. The power from this secondary 324 then provides the low voltage power to sustain operation of the PWM MODULE 342 and SWITCH DRIVER 343.
Further in FIG. 3, the START MODULE 341 has started the DC-to-DC converter the FEEDBACK
340 provides to the PWM MODULE 342, a signal, which that is proportional to the out put voltage. The FEEDBACK 340 may use optical isolation, an isolation transformer etc. none of which are shown to provide this isolated feedback signal to the PWM MODULE 342. However, the typical design requires higher isolation voltage between the primary and secondary of TRANSFORMER 321, 323 and across the FEEDBACK 340 than that required by conventional commercial power supply designs. PWM MODULE
342 generates two or more square-wave outputs that have the phase of their outputs shifted proportional to the duty of the waveform that is to be applied to the primary of transformers 321 and 323. SWITCH
DRIVE 343 provides all necessary isolation of the drive signals with the correct phase to switches 300, 301, 302, 303, 304, 305, 306 and 307. Typical waveforms are shown in SWITCH DRIVE
354, representing an operating duty of 33%. Diodes 325, 326, 327 and 328 rectify the AC of the secondary of transformer 321, 323 into a pulsating DC, shown as J, K, which is then filtered by inductor 329, 330 and capacitor 331. The output inductor 329, 330 and capacitor 331 filters the pulsating DC into a average value equal to the duty of the waveform times it's amplitude, see equation 2 further derived in a later section . The circuit functions exactly in the same manner as a switching power supply commonly called a FORWARD CONVERTER, except it provides a regulated low DC voltage output from a HIGH VOLTAGE DC
applied to INPUT 350 and 351. HIGH VOLTAGE AC maybe applied to INPUT 350 and 351 if SWITCH 300 through 307 are BI-DIRECTIONAL.
The switches, 300, 301, 302, 303, 304, 305, 306 and 307 are typically semi-conductor devices that have a reverse diode across them to clamp any reverse voltage that may be generated by transformer 321, 323 during the time the SWITCH DRIVE 354 changes state. Should an AC output be desired from the power supply then DIODE 325, 326, 327, 328 may be omitted.
The Embodiment in FIG. 4, in accordance with the present invention represents the full capability of the HIGH VOLTAGE TO LOW VOLTAGE BI-DIRECTIONAL CONVERTER. Typically HIGH
VOLTAGE in this embodiment refers to voltages greater than 800 Volt and LOW
VOLTAGE to less than 200 Volt, though the CONVERTER may be designed to operate at any INPUT and OUTPUT voltage. With proper PWM MODULE 432 signals The CONVERTER in this embodiment, when designed with AC or BI-DIRECTIONAL SWITCHES in all SWITCH locations, 400 through 407, 423, 424, 425 is capable of the following 1. Converting a HIGH VOLTAGE DC or AC INPUT into a regulated LOWER VOLTAGE DC
or AC
output.

2. Converting a LOWER VOLTAGE DC or AC INPUT into a regulated HIGH VOLTAGE DC
or AC
output.

3. Converting a HIGH VOLTAGE DC INPUT into a regulated LOW VOLTAGE AC OUTPUT
of a frequency equal too or typically much lower than the switch frequency.

4. Converting a LOW VOLTAGE DC INPUT into a regulated HIGH VOLTAGE AC OUTPUT
of a frequency equal too or typically much lower than the switch frequency.

5. Converting a HIGH VOLTAGE AC INPUT of a frequency equal too or typically much lower than the SWITCH frequency into a regulated LOW VOLTAGE DC OUTPUT.

6. Converting a LOW VOLTAGE AC INPUT of a frequency equal too or typically much lower than the SWITCH frequency into a regulated HIGH VOLTAGE DC OUTPUT.
FIG. 4 is similar in design to FIG. 3 except that the output section has been changed by replacing each DIODE in the secondary circuit with a SWITCH. The converter is intended to produce a regulated DC
or AC output but may be modified to produce an unregulated DC or AC output by omitting FEEDBACK
430.
An explanation of the function of the various parts of the embodiment is as follows. The HIGH
VOLTAGE is connected to INPUT 450 and 451. SWITCHES 401 through 407 are bi-directional or AC type and allow the operation from a DC or AC HIGH VOLTAGE power source. SWITCHES
401 through 407 may be DC type if the HIGH VOLTAGE is always going to be DC and will allow full bi-directional operation if the DC switches have a reverse diode across them to bypass reverse current around the switch.
SWITCHES 423, 424, 426 are typically bi-directional and may be replaced with diodes if the circuit operation similar to FIG. 3 is required.
SWITCH 400, 401 form one half bridge and 402, 403 form the other side of a half bridge which is combined and operated as a full bridge with CAPACITOR 419 blocking the DC
component from being applied to the primary of transformer 421. Both half bridges operate at full duty, with each switch ON for 50% of the time. To create a PWM output the phase of the half bridge 402, 403 is shifted from that of switches 400, 401 by an amount equal to the desired pulse width. SWITCH DRIVE
454 provides an example of this, with signal A the position of the SWITCH 400, HIGH or UP representing the ON or CLOSED state, B, C, D corresponds to SWITCH 401, 402, 403 accordingly and N represents the difference signal that appears across the primary of TRANSFORMER 421. The duty of the primary wavefonn N on TRANSFORMER 421 can be changed by varying the phase of SWITCH signals A, B
with respect to C, D.
This type of switching method facilitates the use of isolation transformers for coupling the gate signals to the SWITCHES and can accommodate any PWM duty from 0 to 100% without worry of saturation of the SWITCH driver transformers, not shown in the drawing. SWITCHES 404, 405 form one half bridge and 406, 407 form the other half bridge of a full-bridge and function in combination with CAPACITOR 420 and another primary of TRANSFORMER 421 in the same manner as SWITCH 400, 401, 402, 403, CAPACITOR
419. The same primary of TRANSFORMER 421 may be used for each full bridge if at least one additional CAPACITOR is used in series with the junction of SWITCH 402 and 403 or 405,407.
An example of the switch drive waveforms is shown by SWITCH DRIVE 454 which demonstrates operation at a duty cycle of 33%. In this example a single TRANSFORMER 421 is used but the use of separate TRANSFORMERS as in FIG. 3 will not change the basic function of the converter, with the secondary combined or operated with their own output SWITCH. In FIG. 4 SWITCH
423, 424 replace the DIODES 325, 326 in FIG. 3. In FIG. 4 SWITCH 425 is new and is required for different operating modes but can improve the efficiency in all designs. SWITCH 425 is closed or ON when ever SWITCH 423 or 424 are both OFF or open, thus providing a return path for INDUCTOR 426 current to CAPACITOR 427. Failure to provide a return path for INDUCTOR 426 current would cause the build up of a very high voltage across SWITCH 423 and 424 when they are first OPENED. SWITCH 425 may be omitted in designs that use PWM
control on the primary of TRANSFORMER 421 so long as the drive for SWITCH 423 and 424 is modified such that they are both ON as indicated in SWITCH DRIVE 454 as well as both are turned ON or remain ON
when ever the SWITCH 425 was indicated as being ON. When SWITCH 425 is left out or replaced by a DIODE then SWITCH 423, 424 may be operated as synchronous rectifiers and derive their switch signals directly from extra windings on TRANSFORMER 421, not shown in FIG. 4, as that method is already in use by PRIOR ART. The inclusion of SWITCH 425 allows for a special operating mode where the primary of TRANSFORMER 421 is a full duty square wave, not PWM modulated, but instead SWITCH 423, 424 are operated in a PWM mode. The output circuit of FIG. 4 comprising of SWITCH 423, 424, 425, Inductor 426 and CAPACITOR 427 may be applied and used in a similar manner to PRIOR ART
shown in FIG. l and FIG.
2, providing bi-direction operation that currently doesn't exist in those embodiments.
SWITCH 423, 424 rectify the output of TRANSFORMER 421 and apply the pulsating rectified DC
pulses shown by waveform M on the SWITCH DRIVE table 454 to an INDUCTOR 426 and CAPACITOR
427, which are then filtered to the desired degree. IF a second output circuit and TRANSFORMER is used similar to FIG. 3 and the second TRANSFORMER is operated phase shifted by 90 degrees then the ripple frequency across CAPACITOR 427 will be doubled and the pulsating DC currents greatly reduced allowing a much smaller capacitor value to be used. This technique may be extended further using a plurality of TRANSFORMERS and plurality of secondary combined in this manner.
CAPACITORS 408, 409, 410, 411, 412 filter the switch current pulses reducing the high frequency AC that is generated by the half bridges in series across the INPUT 450 and 451. The addition of RESISTORS 414, 415, 416, 417 and 418 are used to force the voltages to be equal across CAPACITORS
408, 409, 410, 411, 412 during the start-up time that the half bridges are off. CAPACITOR 413 is used to provide start-up power for the START MODULE 431, which has various internal components that store sufficient charge to run the SWITCHES for a specific time after which an auxiliary winding 422 from TRANSFORMER 421 supplies the necessary power to run the control electronics.
Alternately, an external DC or AC power source, not shown, provides the power to operate the DC to DC
converter and may be common to or close to either INPUT 450 or 451.
In FIG. 4 INPUT VOLTAGE REFERENCE 434 provides input polarity, REFERENCE 490 provides a voltage proportional to the desired output voltage and FEEDBACK 430 supplies a feedback signal proportional to the secondary output voltage all of which are used by the PWM
MODULE 432 to generate the appropriate PWM phase clock signals that are supplied to the SWITCH DRIVER
433, which then drives the switches 400, 401,402, 403, 404, 405, 406, 407, 423, 424, 425. These circuits function as follows. When HIGH VOLTAGE is first applied to INPUT 450 and 451, the RESISTORS 414, 415, 416, 417 and 418 charge CAPACITOR 413. The START MODULE 431 takes the charge from CAPACITOR 413 and determines when it has enough charge to operate the PWM MODULE 432 and SWITCH
DRIVER 433 for a predetermined time. For operation from a HIGH VOLTAGE AC INPUT the START
MODULE 431 takes the current normally charging CAPACITOR 413 and rectifies it and stores the charge internally until a sufficient level has built up to initiate startup of the power supply.
Alternately, the START MODULE 431 may be powered by an external low voltage DC or AC source, not shown in FIG.
4. After initially powering the converter electronics, the START MODULE 431 receives a low voltage AC from through SECONDARY 422. The power from this SECONDARY 422 then provides the low voltage power to sustain operation of the PWM MODULE 432 and SWITCH DRIVER 433.
Further in FIG. 4, after the START MODULE 431 has started the converter the provides to the PWM MODULE 432, a signal, that is proportional to the output voltage. The FEEDBACK
430 may use optical isolation, an isolation transformer etc., none of which are shown, to provide this isolated feedback signal to the PWM MODULE 432. However, the typical design requires higher isolation voltage between the primary and secondary of TRANSFORMER 421 and across the FEEDBACK
430 than that required by conventional commercial power supply designs. PWM MODULE 432 generates two or more square-wave outputs that have the phase of their outputs shifted proportional to the duty of the waveform that is to be applied to the primary of TRANSFORMER 421. SWITCH DRIVE 433 provides the necessary isolation of the drive signals with the correct phase to switches 400, 401, 402, 403, 404, 405, 406, 407, 423, 424 and 425. Typical waveforms are shown in SWITCH DRIVE 454, representing an operating duty of 33%.. SWITCH 423, 424 rectify the pulsating AC waveform looking as signal N of the SWITCH DRIVE
454 of the of the secondary of TRANSFORMER 421 into a pulsating DC, shown as M, which is then filtered by INDUCTOR 426 and CAPACITOR 427. The output INDUCTOR 426 and CAPACITOR 427 filters the pulsating signal M into a average value equal to the duty of the waveform times it's peak amplitude. The circuit functions in a similar manner as a switching power supply commonly called a FORWARD
CONVERTER. In FIG.4 when HIGH VOLTAGE AC is applied to INPUT 450 and 451, acceptable as all switches are bi-directional, then the low voltage output VOUT 452 and 453 will be regulated AC and reduced in amplitude. A expression for the output voltage when the input AC or DC is being converter to a regulated lower voltage of the same type of waveform is Vp = Vin / Bn 2 Vout = Vp * D / N
Where:
Vp => TRANSFORMER PRIMARY VOLTAGE
Vin => HIGH VOLTAGE INPUT
Bn => NUMBER HALF-BRIDGES
Vout => OUTPUT VOLTAGE
N => TRANSFORMER TURNS RATIO ; Number Primary turns divided by Number Secondary turns D => DUTY
DUTY is the ratio of the time the Primary is ON divided by the sum of Primary OFF plus ON time.
Equation 2 is important as it establishes the ratio between the input and output voltage. The power supply is fully bi-directional such that should the power supply output rise to a value greater than equation 2 allows, power will flow from the output back to the input. This has numerous advantages, for example accelerating a car from a high voltage battery, then by changing the power supply duty the power supply acts as a regenerative brake returning the energy from stopping the car to recharge the high voltage battery.
In FIG. 4 the INPUT VOLTAGE REFERENCE 434 is used by the PWM MODULE 432 when it is necessary to convert from AC to DC or vice versa. The signal is used to determine whether the phase of the SWITCH 423 and 424 has to be inverted from its normal condition, thus changing equation 2 to 3 Vout=P*Vp*D / N
Where P => POLARITY is either +1 or - 1 depending on whether the phase of SWITCH 423 and 424 is inverted to the normal stated, providing a reversed OUTPUT voltage with respect to the INPUT.

The effect of the phase is to change VOUT 452, 453 polarity with respect to the INPUT 450, 451. For example using a PHASE of -1 changes a positive HIGH VOLTAGE DC INPUT to a negative LOWER
VOLTAGE DC OUTPUT. Alternately, a PHASE of -1 a positive HIGH VOLTAGE AC INPUT
would change to a negative or reversed phase LOWER VOLTAGE AC OUTPUT.
INPUT VOLTAGE REFERENCE 434 has other uses as well, especially when converting a HIGH
VOLTAGE AC INPUT to a LOWER VOLTAGE DC OUTPUT or vice versa. For example when converting a HIGH VOLTAGE AC INPUT in to a positive LOWER VOLTAGE DC OUTPUT
then the POLARITY signal +1 when the HIGH VOLTAGE AC INPUT is positive and -1 when the HIGH
VOLTAGE AC INPUT is negative. The resultant DC OUTPUT will be the same as any full wave rectified AC signal, the amplitude and ripple characteristics will be determined by the value of the filter made up of INDUCTOR 426 and CAPACITOR 427 however, the voltage at point M will be determined by equation 3.
To convert a HIGH VOLTAGE DC INPUT to a LOW VOLTAGE AC OUTPUT then the POLARITY control is used to toggle or change to the opposite state the LOW
VOLTAGE OUTPUT
every time the AC REFERENCE 490 waveform goes through a zero crossing. To synthesize an AC
waveform it is necessary for the PWM MODULE 432 to use a modified REFERENCE
490 as it requires a value which is proportional to the desired output waveform. Typically a look up table in a micro-processor memory or logic storage device is used to synthesize a suitable REFERENCE 490 signal. The FEEDBACK 430 value is compared to the REFERENCE 490 waveform and the PWM is adjusted as required to produce the correct output. The technique is well known in the Industry and is may be found in a number of the Patents listed as PRIOR ART. Conversely, a LOW VOLTAGE DC
OUTPUT may be converted to a HIGH VOLTAGE AC OUTPUT if the magnitude of the DC OUTPUT
present on VOUT 452, 453 is greater than that allowed by equation 3. The power under this circumstance will then flow from the LOW VOLTAGE VOUT side to the HIGH VOLTAGE INPUT.
FIG. 7 shows that the SWITCH 423, 424 in FIG. 4 can be substituted with a full wave bridge, using 4 SWITCHES instead of the two in FIG. 4. This is the case when the use of a center tapped secondary such as that used by TRANSFORMER 421 in FIG. 4 is not desired. The arrangement in FIG. 7 shows a typical full bridge secondary circuit where SWITCH 722, 723 are the same phase as SWITCH 423 in FIG. 4 and SWITCH 721, 724 are the same phase as SWITCH 424 in FIG. 4. Other similarities between FIG. 7 and Fig. 4 are TRANSFORMER 720 & 421; SWITCH 725 & 425; INDUCTOR 726 & 426;
CAPACITOR
727 & 427; FEEDBACK 730 & 430 etc. are all the same as well as remaining components except instead of a leading 4 there is leading 7 substituted in FIG. 7. The function of the circuit with the changed secondary circuit is exactly the same as in FIG. 4 except for the substitution of the appropriate number from FIG. 7 into the description for FIG. 4.
FIG. SB shows the definition of what is meant by a DC switch. The DC switch behaves as a switch blocking voltage in one direction but when a reverse voltage is applied it either conducts as in the case of DIODE 505 or is destroyed. That is why typically a DIODE 505 is placed across the SWITCH 504 as shown in FIG. SB. The FIG. SC shows a BI-DIRECTIONAL SWITCH 508 made up using two mosfets 506 and 507. Each mosfet is in this example have their SOURCE terminals connected together and the control signal is applied across the G and S terminals. The terminals labeled A on device 506 and B on device 507 are the equivalent SWITCH input and output terminals. The Mosfets 506 and 507 may be connected Drain to Drain instead of the way shown, however in that method each will require a separate isolated G and S drive signal. Any type of switch semiconductor or otherwise may be substituted for Mosfet 506 and 507 so long as they are combined in a way that the switch will block voltage of any polarity when turned off and pass current of any polarity when turned on. It should be pointed out that most switch designs that use semi-conductor devices use additional components not shown in any FIG. 1 through 7. These additional devices are used following manufacturer's recommendation or through good design practice for the purpose of protecting the switch from overload current, reverse voltage, voltage, power, temperature and for reducing electronic radiated noise. Mounting and cooling of the switches is selected to suit a designs mechanical and performance requirements and the preferred embodiments do not have any special design requirements other than that required to meet a specific product reliability.
FIG. 6A shows a typical filter block that may be added to the LOW VOLTAGE
OUTPUT side to improve the quality of the output. FIG. 6B shows a typical filter block that may be added to the HIGH
VOLTAGE INPUT side to reduce the radiated noise caused by the INPUT SWITCH
action. These filters typically are composed of a combination of INDUCTORS, CAPACITORS and RESITORS
in differing combinations to generate the required noise attenuation ratio required by the design. The preferred embodiments does not impose or require any special filter design other than that used by good practice.
The embodiments of FIG. 3 and FIG. 4 may be operated using PWM control or alternately using a variable frequency switching rate with a fixed or variable ON pulse width.
FIG. 8 Is a preferred embodiment that uses the converter for powering an electric motor, such as in hybrid electric car etc.. The POWER SOURCE 800 such as a battery or capacitor bank, fuel cell, or any combination of these or an AC source such as the output from a motor-generator connected to a fly-wheel, provides a source of power to operate the DRIVE MOTOR 813. The DRIVE MOTOR 813 may be DC or a poly-phase AC motor using one or plurality of AC phases provided by a plurality N of converters as shown by SUPPLY A 810, SUPPLY B 811, through SUPPLY N 812. These power supplies may be wholly independent or share various common elements from each other, such as feedback or PWM CONTROL
signals and they may even have a common primary section but multiple secondary, each providing a different output phase. In FIG. 8 DC POWER SOURCE 800, it is connected the HIGH VOLTAGE
side of the converter and the low voltage side connected to the DRIVE MOTOR 813. The DRIVE
MOTOR 813 may be replaced by any other electric device. In keeping with the preferred embodiment of the converter it can use an AC source of a differing frequency, such as that put out by the motor-generator of a fly-wheel to create a different frequency AC output to the DRIVE MOTOR 813. During acceleration or steady operation of the DRIVE MOTOR 813 the converters SUPPLY A 810, SUPPLY B 811, SUPPLY N 812 takes its power from the POWER SOURCE 800, energy may be transferred from the DRIVE MOTOR 813 back to the POWER
SOURCE 800 using regenerative-breaking, where DRIVE MOTOR 813 is changed to a generator and used to decelerate the rate that it is turning. The operation in this mode exploits the use of equation 2 or 3 from earlier in this section along with specific clocking signals that are unique to each application. The best example of this would be where the DRIVE MOTOR 813 is used in an automobile as either the whole or partial motive source though, the preferred embodiment is not limited to this application.
FIG. 9 is similar to FIG. 8 but the POWER SOURCE 904 is now located on the LOW
VOLTAGE
side and the DRIVE MOTOR 903 is on the HIGH VOLTAGE side. The operation in this mode is identical just the direction of power flow is different under the same circumstance.
Another preferred embodiment, no figure provided can be a HIGH VOLTAGE to HIGH
VOLTAGE
converter where instead of a LOW VOLTAGE secondary circuit, the secondary uses a HIGH VOLTAGE
arrangement of half bridges in series similar to the primary side.
Another preferred embodiment again no figure provided, would use the HIGH
VOLTAGE switch arrangement of FIG. 1 or FIG. 2 on the primary and the secondary arrangement of FIG. 4 or FIG. 7. A
variation of this would be where the secondary switch arrangement of FIG. 4 or FIG. 7 would be PWM
operated and the primary side would be a 100% duty square-wave and the variable DUTY of the secondary switches would provide the PWM regulation. This arrangement is shown in FIG.
10 the SWITCH DRIVE
1054A, where SWITCH signals A through G are the same as those from SWITCH
DRIVE 154 of FIG. I .
SWITCH DRIVE signals J, K, L, M, N are the same signals as that from SWITCH
DRIVE 454 of FIG. 4 and would be the switching signals of the secondary SWITCH 423 as J, SWITCH 424 as K and SWITCH 425 as L where M is the rectified output the same as FIG. 4 and N is the secondary or primary waveform. The operation is similar to FIG. 4 or FIG. 7 except that the secondary side is the only switches that have their signals PWM. Another arrangement is possible as shown by FIG. 10 SWITCH DRIVE
1054B again the same circumstance but where the primary is PWM as well as the secondary. The signals from SWITCH DRIVE
1054B relates to A through G as the switch signals of FIG. 2 SWITCH 200 through 205 respectively and FIG. 4 or 7 is the secondary SWITCH 423 as J, SWITCH 424 as K and SWITCH 425 as L where M is the rectified output the same as FIG. 4 and N is the secondary or primary waveform. FIG. 10 is bi-directional of any polarity input or output if the secondary switches are bi-directional and the primary side switches of FIG.
1 or FIG. 2 are made bi-direction.
Although the invention has been described in connection with a preferred embodiment, it should be understood that various modifications, additions and alterations may be made to the invention by one skilled in the art without departing from the spirit and scope of the invention as defined in the appended claims.

Claims

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE PROPERTY OR
PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

To be submitted later