CA2024588C - Switching regulator of a simple structure capable of reducing a switching loss - Google Patents
Switching regulator of a simple structure capable of reducing a switching lossInfo
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- CA2024588C CA2024588C CA 2024588 CA2024588A CA2024588C CA 2024588 C CA2024588 C CA 2024588C CA 2024588 CA2024588 CA 2024588 CA 2024588 A CA2024588 A CA 2024588A CA 2024588 C CA2024588 C CA 2024588C
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- 239000003990 capacitor Substances 0.000 claims abstract description 37
- 238000004804 winding Methods 0.000 claims abstract description 35
- 230000001276 controlling effect Effects 0.000 claims description 4
- 230000001105 regulatory effect Effects 0.000 claims description 3
- 230000005284 excitation Effects 0.000 description 5
- 238000009499 grossing Methods 0.000 description 5
- 230000010355 oscillation Effects 0.000 description 3
- 238000003079 width control Methods 0.000 description 3
- 238000010586 diagram Methods 0.000 description 2
- 230000003071 parasitic effect Effects 0.000 description 2
- 244000228957 Ferula foetida Species 0.000 description 1
- 230000002238 attenuated effect Effects 0.000 description 1
- 230000001939 inductive effect Effects 0.000 description 1
- 230000000087 stabilizing effect Effects 0.000 description 1
- 238000006467 substitution reaction Methods 0.000 description 1
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Abstract
In a switching regulator which is operable in response to an input a.c. voltage to produce an output d.c. voltage and which comprises a main transformer having primary and secondary windings for primary and secondary circuits, respectively, the primary circuit is formed by a series connection of the primary winding and a switching transistor selectively turned on and off at a high frequency of, for example, several hundreds of kHz while the secondary circuit comprises a series connection of a saturable reactor, a diode, and an output capacitor connected in parallel to a load. The saturable reactor exhibits high and low impedances when the diode is put into nonconductive and conductive states, respectively, and serves to reduce a switching loss resulting from interception of a voltage and a current. A frequency control circuit is connected between the output capacitor and the switching transistor to put the transistor into the on-state and the off-state at the high frequency with a time interval of the off-state kept unchanged and with a time interval of the on-state varied.
Description
292~588 SWITCHING REGULATOR OF A SIMPLE STRUCTURE
CAPABLE OF REDUCING A SWITCHING LOSS
Background of the Invention:
This invention relates to a switching regulator for use in regulating an input a.c. voltage into an output d.c. voltage.
A conventional switching regulator of the type described comprises a rectifying circuit supplied with an input d.c. voltage to produce a rectified voltage and a converter circuit for converting the rectified voltage into an output d.c. voltage. The converter circuit has 10 a pair of input terminals and a pair of output terminals and comprises a main transformer having primary and secondary windings for forming parts of primary and secondary circuits connected to the input and the output terminals, respectively. The primary circuit comprises 15 a switching transistor connected to the primary winding in series while the secondary circuit comprises the subsidiary winding, a diode connected to the secondary winding in series, and a smoothing circuit which is -,, 2024S~8 connected to the diode and which comprises an output capacitor connected to the output terminals. A load is connected in parallel to the output capacitor. In addition, a pulse width control circuit is connected 5 between one of the output terminals and the switching transistor.
The pulse width control circuit is operable to selectively put the switching transistor into an on-state and an off-state in accordance with the output 10 d.c. voltage developed across the output terminals.
More specifically, the pulse width control circuit serves to vary a time interval of the on-state with a switching period of the transistor kept unchanged.
In the meanwhile, it is a recent trend that the 15 switching transistor is turned on and of~ at a high frequency of, for example, 500 kHz higher than 100 kHz.
In this event, the switching transistor should quickly switch a large current at every switching operation.
Herein, it is to be noted that the switching transistor 20 is connected to the primary winding which is operable as an inductive load of the switching transistor. Under the circumstances, a switching noise inescapably takes place in the conventional switching regulator.
In order to prevent occurrence of such a 25 switching noise, noise reduction circuits, such as a CR
snubber, must be connected to the primary and the secondary windings of the transformer. This makes the switching regulator intricate and bulky in structure and 3 20245~8 expensive. Moreover, such a switching noise tends to become large with an increase of a current which flows through the switching transistor and the diode.
In a paper contributed by Y. Masuda et al to 5 National Conference held in 1983 by the Institute of Electronics and Communication Engineers in Japan, disclosure is made about a converter which comprises a magnetic amplifier in a secondary circuit of a main transformer so as to widen a controllable range;
10 Herein, it is to be noted that the magnetic amplifier is composed of a saturable reactor having a winding and a control circuit which is coupled to the winding and which is complicated in structure. Thus, the converter inevitably comprises the complicated control circuit and 15 is therefore expensive.
Summary of the,Invention:
It is an object of this invention to provide a switching regulator which is simple in structure and inexpensive.
It is another object of this invention to provide a switching regulator of the type described, which is capable of being switched at a high frequency without a switching noise and without a switching loss.
A switching regulator to which this invention is 25 applicable is for use in regulating an input a.c.
voltage into an output d.c. voltage. The switc~hing regulator comprises input rectifying means for rectifying the input a.c. voltage into a rectified ~ 20245~8 voltage and converting means supplied with the rectified voltage for converting the rectified voltage into the output d.c. voltage. The converting means has a pair of input terminals supplied with the rectified voltage and 5 a pair of output terminals for the output d.c. voltage.
The switching regulator comprises a transformer comprising a primary winding and a secondary winding which form parts of primary and secondary circuits, respectively. The primary circuit comprises a switching 10 transistor connected in series to the primary winding to form a primary series connection circuit and operable in response to a switching control signal to be selectively put into an on-state and an off-state. The primary series connection circuit is connected between the input 15 terminals. The secondary circuit comprises a series connection of the secondary winding, a diode, a saturable reactor, and an output capacitor with the output capacitor connected to the output terminals. The converting means further comprises a frequency control 20 circuit connected between a preselected one of the output terminals and the switching transistor for controlling the switching transistor to produce an output control signal so that a time interval of the on-state of the switching transistor is varied in 25 accordance with the output d.c. voltage with a time interval of the off-state of the switching transistor kept unchanged, and means for supplying the output 202~5~
control signal to the switching transistor as the switching control signal.
Brief Description of the Drawing:
Fig. 1 is a block diagram of a conventional 5 switçhing regulator;
Fig. 2 is a block diagram of a switching regulator according to a preferred embodiment of this invention;
Fig. 3 is an equivalent circuit for use in 10 describing operation of the switching regulator illustrated in Fig. 2;
Fig. 4 shows a waveform which appears at portions of the switching regulator illustrated in Fig.
CAPABLE OF REDUCING A SWITCHING LOSS
Background of the Invention:
This invention relates to a switching regulator for use in regulating an input a.c. voltage into an output d.c. voltage.
A conventional switching regulator of the type described comprises a rectifying circuit supplied with an input d.c. voltage to produce a rectified voltage and a converter circuit for converting the rectified voltage into an output d.c. voltage. The converter circuit has 10 a pair of input terminals and a pair of output terminals and comprises a main transformer having primary and secondary windings for forming parts of primary and secondary circuits connected to the input and the output terminals, respectively. The primary circuit comprises 15 a switching transistor connected to the primary winding in series while the secondary circuit comprises the subsidiary winding, a diode connected to the secondary winding in series, and a smoothing circuit which is -,, 2024S~8 connected to the diode and which comprises an output capacitor connected to the output terminals. A load is connected in parallel to the output capacitor. In addition, a pulse width control circuit is connected 5 between one of the output terminals and the switching transistor.
The pulse width control circuit is operable to selectively put the switching transistor into an on-state and an off-state in accordance with the output 10 d.c. voltage developed across the output terminals.
More specifically, the pulse width control circuit serves to vary a time interval of the on-state with a switching period of the transistor kept unchanged.
In the meanwhile, it is a recent trend that the 15 switching transistor is turned on and of~ at a high frequency of, for example, 500 kHz higher than 100 kHz.
In this event, the switching transistor should quickly switch a large current at every switching operation.
Herein, it is to be noted that the switching transistor 20 is connected to the primary winding which is operable as an inductive load of the switching transistor. Under the circumstances, a switching noise inescapably takes place in the conventional switching regulator.
In order to prevent occurrence of such a 25 switching noise, noise reduction circuits, such as a CR
snubber, must be connected to the primary and the secondary windings of the transformer. This makes the switching regulator intricate and bulky in structure and 3 20245~8 expensive. Moreover, such a switching noise tends to become large with an increase of a current which flows through the switching transistor and the diode.
In a paper contributed by Y. Masuda et al to 5 National Conference held in 1983 by the Institute of Electronics and Communication Engineers in Japan, disclosure is made about a converter which comprises a magnetic amplifier in a secondary circuit of a main transformer so as to widen a controllable range;
10 Herein, it is to be noted that the magnetic amplifier is composed of a saturable reactor having a winding and a control circuit which is coupled to the winding and which is complicated in structure. Thus, the converter inevitably comprises the complicated control circuit and 15 is therefore expensive.
Summary of the,Invention:
It is an object of this invention to provide a switching regulator which is simple in structure and inexpensive.
It is another object of this invention to provide a switching regulator of the type described, which is capable of being switched at a high frequency without a switching noise and without a switching loss.
A switching regulator to which this invention is 25 applicable is for use in regulating an input a.c.
voltage into an output d.c. voltage. The switc~hing regulator comprises input rectifying means for rectifying the input a.c. voltage into a rectified ~ 20245~8 voltage and converting means supplied with the rectified voltage for converting the rectified voltage into the output d.c. voltage. The converting means has a pair of input terminals supplied with the rectified voltage and 5 a pair of output terminals for the output d.c. voltage.
The switching regulator comprises a transformer comprising a primary winding and a secondary winding which form parts of primary and secondary circuits, respectively. The primary circuit comprises a switching 10 transistor connected in series to the primary winding to form a primary series connection circuit and operable in response to a switching control signal to be selectively put into an on-state and an off-state. The primary series connection circuit is connected between the input 15 terminals. The secondary circuit comprises a series connection of the secondary winding, a diode, a saturable reactor, and an output capacitor with the output capacitor connected to the output terminals. The converting means further comprises a frequency control 20 circuit connected between a preselected one of the output terminals and the switching transistor for controlling the switching transistor to produce an output control signal so that a time interval of the on-state of the switching transistor is varied in 25 accordance with the output d.c. voltage with a time interval of the off-state of the switching transistor kept unchanged, and means for supplying the output 202~5~
control signal to the switching transistor as the switching control signal.
Brief Description of the Drawing:
Fig. 1 is a block diagram of a conventional 5 switçhing regulator;
Fig. 2 is a block diagram of a switching regulator according to a preferred embodiment of this invention;
Fig. 3 is an equivalent circuit for use in 10 describing operation of the switching regulator illustrated in Fig. 2;
Fig. 4 shows a waveform which appears at portions of the switching regulator illustrated in Fig.
2; and Fig. 5 is another equivalent circuit for use in describing another operation of the switching regulator illustrated in Fig. 2.
Description of the Preferred Embodiment:
Referring to Fig. 1, a conventional switching 20 regulator will be described for a better understanding of this invention and may be known as a forward converter or an on-on type voltage resonant converter in the art. The illustrated switching regulator comprises an input rectifying circuit 11 connected to an input 25 power source 12 and a converter section 15 connected to the input rectifying circuit 11 and a load resistor 16.
More specifically, the input power source 12 supplies the input rectifying circuit 11 with an input a.c.
voltage of a commercial frequency. The input rectifying circuit 11 comprises an input line filter 17, an input rectifier circuit 18, and an input capacitor 19. At any rate, the input a.c. voltage is rectified through the 5 input line filter 17, the input rectifier circuit 18, and the input capacitor 19 and is supplied to the converter section 15 as a rectified voltage which is smoothed by the input capacitor 19.
The converter section 15 has a pair of input 10 terminals and a pair of output terminals and comprises a main transformer 21 having a primary winding and a secondary winding which form parts of primary and secondary circuits of the main transformer 21, respectively. The primary circuit is connected between 15 the input terminals while the secondary circuit is connected between the output terminals.
More particularly, the primary circuit is formed by a series connection of the primary winding of the main transformer 21 and a switching transistor, namely, 20 a main transistor 22 which is controlled in a manner to be described later. ~n the illustrated example, the switching transistor 22 is connected in parallel to a first snubber 23 which is composed of a series connection of a resistor and a capacitor and which may 25 therefore be called a CR snubber.
On the other hand, the secondary circuit comprises a first diode 26 connected to the secondary winding of the main transformer 21 in series, a second 7 20245~8 diode 27 connected to the first diode 26 and the secondary winding, a coil 28 connected to a point of connection between the first and the second diodes 26 and 27, and an output capacitor 29 connected between the 5 output terminals and connected to the coil 28 in series.
As shown in Fig. 1, the output capacitor 29 is connected in parallel to the load resistor 16.
In addition, a pulse width controller 31 is connected between one of the output terminals and the 10 switching transistor 22 and is operable in a manner to be described later. As illustrated in Fig. 1, second and third snubbers 32 and 33 are connected in parallel to the first and the second diodes 26 and 27, respectively, and are similar to the first snubber 23.
15 At any rate, the first through the third snubbers 23, 32, and 33 are effective to eliminate a switching noise resulting from a switching operation of the switching transistor 22, as known in the art.
In the illustrated converter section 15, the 20 rectified voltage is converted into a sequence of rectangular voltage pulses by switching the switching transistor 22. The rectangular voltage pulses are transformed by the main transformer 21 into transformed voltage pulses which have a transformed voltage level 25 and which appear in the secondary circuit. The transformed voltage pulses are delivered to the first and the second diodes 26 and 27 to be rectified therein and supplied to a smoothing circuit formed by the coil 8 2024S~
28 and the output capacitor 29. Thus, a smoothed voltage is sent to the load resistor 16 in the form of an average d.c. voltage.
Herein, it is to be noted here that the first 5 diode 26 is turned on when the switching transistor 22 is put into an on-state. In this connection, the illustrated switching regulator is named the forward converter, as mentioned before.
Now, it is assumed that the rectified voltage lO across the input capacitor l9, the output voltage across the output capacitor 29, a switching period of the switching transistor 22, and a time interval of the on-state of the switching transistor 22 are represented by Vi (volt), Vo (volt), T (second), and Ton (second), 15 respectively, and that turns of the primary and the secondary windings of the main transformer 21 are represented by Nl and N2, respectively. Under the circumstances, a relationship between the rectified voltage Vi and the output voltage Vo is given by:
Vo = (Ton/T)-(N2/Nl)-Vi.
From the above equation, it is apparent that the output voltage Vo can be stabilized by varying the time interval Ton of the on-state with the switching period substantially kept unchanged. To this end, the pulse 25 width controller 31 is operable in response to the output voltage to deliver a switching control signaL to the switching transistor 22. Such a switching control 9 20245~8 signal may have a high frequency of, for example, 500 kHz It is noted that the switching transistor 22 is connected to the primary winding which is operable as an 5 inductance element and that a large current should be switched by the switching transistor 22 at each time when the switching transistor 22 is turned on and off.
Accordingly, the illustrated switching regulator has disadvantages as pointed out in the preamble of the 10 instant specification. In addition, the first through the third snubbers 23, 32, and 33 must be connected so as to reduce the switching noise in addition to the input line filter 17.
Referring to Fig. 2, a switching regulator 15 according to a preferred embodiment of this invention may be an on-on type resonant regulator, namely, a forward converter like in Fig. 1 and therefore comprises similar parts designated by like reference numerals.
Like in Fig. 1, the illustrated switching regulator 20 comprises an input rectifying circuit 11' and a converter section 15'. In the example, the input rectifying circuit 11' is similar in structure to that illustrated in Fig. 1 except that the input line filter 17 is removed from Fig. 2 but is operable in a manner 25 similar to that illustrated in Fig. 1. In any event, the input a.c. voltage is produced from the input power source 12 and is supplied through the input rectifier circuit 18 and the input capacitor 19 to be impressed as 20245~8 the rectified voltage across the input terminals of the converter section 15'. In Fig. 2, the primary transformer 21 is included in the converter section 15' and has a primary winding and a secondary winding which 5 form parts of primary and secondary circuits, respectively.
In Fig. 2, the primary circuit comprises a switching transistor, namely, a main transistor 22 connected in series to the primary winding of the main 10 transformer 21 to form a primary series circuit. A
capacitor 36 is connected in parallel to the switching transistor 22. The illustrated switching transistor 22 is operable as a diode specified by an internal diode 37 when it is turned off, as shown in Fig. 2.
On the other hand, the secondary circuit comprises an inductance element 39 which may be formed either by a leakage inductance of the main transformer 21 or by an external inductance. Like in Fig. 1, the illustrated secondary circuit comprises a diode 26 20 identical with the first diode 26 of Fig. 1 and an output capacitor 29 connected across the output terminals.
It is to be noted that a saturable reactor 41 is also included in the secondary circuit and has no 25 control winding, as readily understood from Fig. 2. The saturable reactor 41 exhibits a high impedance during an off-state of the diode 26 and a low impedance during an on-state of the diode 26. The high impedance falls 11 20245~8 within a range between 100 kilohms and 500 kilohms while the low impedance is smaller than 1 ohm and may be, for example, 0.01 ohm.
The inductance element 39, the saturable reactor 5 41, the diode 26, and the output capacitor 29 are connected in series to one another in the secondary circuit of the main transformer 21, as illustrated in Fig. 2. Like in Fig. 1, the output capacitor 29 is connected in parallel to the load resistor 16.
Moreover, a frequency controller 42 is connected between one of the output terminals and the switching transistor 22. The frequency controller 42 is operable in response to the output voltage and produces a sequence of switching control pulses which are specified 15 by an on-state time interval variable in accordance with the output voltage and an invariable off-state time interval. As a result, the control pulses have variable frequencies. Thu~, the switching transistor 22 is repeatedLy turned on and off in response to the 20 switching control pulses of the variable frequency.
As readily understood from Fig. 2, the diode 26 is put into a conductive state or an on-state when a positive voltage is impressed across the primary winding of the main transformer 21 and when the switching 25 transistor 22 is turned on. On the other hand, the diode 26 is put into an off-state while the switching transistor 22 is turned off as will later become clear as the description proceeds. In this connection, the 12 20245~8 illustrated switching regulator is called the forward converter, as mentioned before.
As described above, the diode 26 is repeatedly put into the on-state and the off-state in cooperation 5 with the switching transistor 22. A diode switching period is formed by a time interval of the on-state and a time interval of the off-state. The time intervals of the on-state and the off-state will be named an on-state time interval and an off-state time interval.
Referring to Fig. 3, the secondary circuit of the switching regulator illustrated in Fig. 2 is equivalently represented by a circuit shown in Fig. 3 during the on-state time interval of the diode 26.
During the on-state time interval, the diode 26 becomes 15 conductive and a forward voltage drop of the diode 26 may be therefore néglected because the forward voltage drop is sufficiently small. In addition, the output capacitor 29 is replaced in Fig. 3 by a constant voltage source 45 on the assumption that the output capacitor 29 20 has a sufficiently large capacitance. Furthermore, another constant voltage source 46 specifies an induced voltage which appears across the secondary winding during the on-state time interval of the main transformer 21 and may be referred to as an additional 25 constant voltage source. The induced voltage is invariable and is therefore a constant voltage.
Herein, it is surmised that the rectified voltage across the input capacitor 19 (Fig. 2) and the 13 202~5a8 constant voltage of the constant voltage source 46 are represented by El and E2, respectively, and that the primary and the secondary windings have first and second turns Nl and N2, respectively. Under the circumstances, 5 the constant voltage E2 is given by:
E2 = (N2/Nl)-El.
When an inductance of the inductance element 39 and the output voltage, namely, the constant voltage of the constant voltage source 45 are represented by L and 10 V0, respectively, a current IL which ~lows through the inductance element 39 is given by:
IL = (E2 - V0)~t/L = ((N2-El/Nl) - V0))-t/L.
On the other hand, an inverse voltage takes place across the secondary winding of the main 15 transformer 21 during the off-state time interval of the diode 26. In this event, no current is caused to flow throuqh the inductance element 39 even when such an inverse voltage takes place. This is because such a current is interrupted by the diode 26.
Referring to Fig. 4, description will be made about operation of the switching regulator illustrated in Fig. 2. In Fig. 4, a capacitor voltage (depicted at Vc) across the capacitor 36 is shown along a top line of Fig. 4 while a current (depicted at ITl) flowing through 25 the primary winding of the main transformer 21 is illustrated along a second line of Fig. 4. In addition, a current (depicted at IL) flowing through the inductance element 39 is shown along a bottom line of Fig. 4 and may be called a coil current.
When the on-state time interval and the off-state time interval of the diode 26 are represented 5 by Ton and Toff, respectively, an average value of the coil current IL, namely, the output current (depicted at Io) is given by:
~Ton Io = l/(Ton + Toff ,J IL dt = (((N2-El/Nl) - Vo)/(2L))-(l/((l/Ton) + (Toff/Ton ))).
(1) Under the circumstances, when a resistance of the load resistor 16 is represented by R, the output 15 voltage Vo is given by:
Vo = R-Io. (2) Substitution of Equation (1) into Equation (2) gives:
Vo = N2-El/(Nl-(l + (2L/R).((l/Ton) + (Toff/Ton2)))).
Description of the Preferred Embodiment:
Referring to Fig. 1, a conventional switching 20 regulator will be described for a better understanding of this invention and may be known as a forward converter or an on-on type voltage resonant converter in the art. The illustrated switching regulator comprises an input rectifying circuit 11 connected to an input 25 power source 12 and a converter section 15 connected to the input rectifying circuit 11 and a load resistor 16.
More specifically, the input power source 12 supplies the input rectifying circuit 11 with an input a.c.
voltage of a commercial frequency. The input rectifying circuit 11 comprises an input line filter 17, an input rectifier circuit 18, and an input capacitor 19. At any rate, the input a.c. voltage is rectified through the 5 input line filter 17, the input rectifier circuit 18, and the input capacitor 19 and is supplied to the converter section 15 as a rectified voltage which is smoothed by the input capacitor 19.
The converter section 15 has a pair of input 10 terminals and a pair of output terminals and comprises a main transformer 21 having a primary winding and a secondary winding which form parts of primary and secondary circuits of the main transformer 21, respectively. The primary circuit is connected between 15 the input terminals while the secondary circuit is connected between the output terminals.
More particularly, the primary circuit is formed by a series connection of the primary winding of the main transformer 21 and a switching transistor, namely, 20 a main transistor 22 which is controlled in a manner to be described later. ~n the illustrated example, the switching transistor 22 is connected in parallel to a first snubber 23 which is composed of a series connection of a resistor and a capacitor and which may 25 therefore be called a CR snubber.
On the other hand, the secondary circuit comprises a first diode 26 connected to the secondary winding of the main transformer 21 in series, a second 7 20245~8 diode 27 connected to the first diode 26 and the secondary winding, a coil 28 connected to a point of connection between the first and the second diodes 26 and 27, and an output capacitor 29 connected between the 5 output terminals and connected to the coil 28 in series.
As shown in Fig. 1, the output capacitor 29 is connected in parallel to the load resistor 16.
In addition, a pulse width controller 31 is connected between one of the output terminals and the 10 switching transistor 22 and is operable in a manner to be described later. As illustrated in Fig. 1, second and third snubbers 32 and 33 are connected in parallel to the first and the second diodes 26 and 27, respectively, and are similar to the first snubber 23.
15 At any rate, the first through the third snubbers 23, 32, and 33 are effective to eliminate a switching noise resulting from a switching operation of the switching transistor 22, as known in the art.
In the illustrated converter section 15, the 20 rectified voltage is converted into a sequence of rectangular voltage pulses by switching the switching transistor 22. The rectangular voltage pulses are transformed by the main transformer 21 into transformed voltage pulses which have a transformed voltage level 25 and which appear in the secondary circuit. The transformed voltage pulses are delivered to the first and the second diodes 26 and 27 to be rectified therein and supplied to a smoothing circuit formed by the coil 8 2024S~
28 and the output capacitor 29. Thus, a smoothed voltage is sent to the load resistor 16 in the form of an average d.c. voltage.
Herein, it is to be noted here that the first 5 diode 26 is turned on when the switching transistor 22 is put into an on-state. In this connection, the illustrated switching regulator is named the forward converter, as mentioned before.
Now, it is assumed that the rectified voltage lO across the input capacitor l9, the output voltage across the output capacitor 29, a switching period of the switching transistor 22, and a time interval of the on-state of the switching transistor 22 are represented by Vi (volt), Vo (volt), T (second), and Ton (second), 15 respectively, and that turns of the primary and the secondary windings of the main transformer 21 are represented by Nl and N2, respectively. Under the circumstances, a relationship between the rectified voltage Vi and the output voltage Vo is given by:
Vo = (Ton/T)-(N2/Nl)-Vi.
From the above equation, it is apparent that the output voltage Vo can be stabilized by varying the time interval Ton of the on-state with the switching period substantially kept unchanged. To this end, the pulse 25 width controller 31 is operable in response to the output voltage to deliver a switching control signaL to the switching transistor 22. Such a switching control 9 20245~8 signal may have a high frequency of, for example, 500 kHz It is noted that the switching transistor 22 is connected to the primary winding which is operable as an 5 inductance element and that a large current should be switched by the switching transistor 22 at each time when the switching transistor 22 is turned on and off.
Accordingly, the illustrated switching regulator has disadvantages as pointed out in the preamble of the 10 instant specification. In addition, the first through the third snubbers 23, 32, and 33 must be connected so as to reduce the switching noise in addition to the input line filter 17.
Referring to Fig. 2, a switching regulator 15 according to a preferred embodiment of this invention may be an on-on type resonant regulator, namely, a forward converter like in Fig. 1 and therefore comprises similar parts designated by like reference numerals.
Like in Fig. 1, the illustrated switching regulator 20 comprises an input rectifying circuit 11' and a converter section 15'. In the example, the input rectifying circuit 11' is similar in structure to that illustrated in Fig. 1 except that the input line filter 17 is removed from Fig. 2 but is operable in a manner 25 similar to that illustrated in Fig. 1. In any event, the input a.c. voltage is produced from the input power source 12 and is supplied through the input rectifier circuit 18 and the input capacitor 19 to be impressed as 20245~8 the rectified voltage across the input terminals of the converter section 15'. In Fig. 2, the primary transformer 21 is included in the converter section 15' and has a primary winding and a secondary winding which 5 form parts of primary and secondary circuits, respectively.
In Fig. 2, the primary circuit comprises a switching transistor, namely, a main transistor 22 connected in series to the primary winding of the main 10 transformer 21 to form a primary series circuit. A
capacitor 36 is connected in parallel to the switching transistor 22. The illustrated switching transistor 22 is operable as a diode specified by an internal diode 37 when it is turned off, as shown in Fig. 2.
On the other hand, the secondary circuit comprises an inductance element 39 which may be formed either by a leakage inductance of the main transformer 21 or by an external inductance. Like in Fig. 1, the illustrated secondary circuit comprises a diode 26 20 identical with the first diode 26 of Fig. 1 and an output capacitor 29 connected across the output terminals.
It is to be noted that a saturable reactor 41 is also included in the secondary circuit and has no 25 control winding, as readily understood from Fig. 2. The saturable reactor 41 exhibits a high impedance during an off-state of the diode 26 and a low impedance during an on-state of the diode 26. The high impedance falls 11 20245~8 within a range between 100 kilohms and 500 kilohms while the low impedance is smaller than 1 ohm and may be, for example, 0.01 ohm.
The inductance element 39, the saturable reactor 5 41, the diode 26, and the output capacitor 29 are connected in series to one another in the secondary circuit of the main transformer 21, as illustrated in Fig. 2. Like in Fig. 1, the output capacitor 29 is connected in parallel to the load resistor 16.
Moreover, a frequency controller 42 is connected between one of the output terminals and the switching transistor 22. The frequency controller 42 is operable in response to the output voltage and produces a sequence of switching control pulses which are specified 15 by an on-state time interval variable in accordance with the output voltage and an invariable off-state time interval. As a result, the control pulses have variable frequencies. Thu~, the switching transistor 22 is repeatedLy turned on and off in response to the 20 switching control pulses of the variable frequency.
As readily understood from Fig. 2, the diode 26 is put into a conductive state or an on-state when a positive voltage is impressed across the primary winding of the main transformer 21 and when the switching 25 transistor 22 is turned on. On the other hand, the diode 26 is put into an off-state while the switching transistor 22 is turned off as will later become clear as the description proceeds. In this connection, the 12 20245~8 illustrated switching regulator is called the forward converter, as mentioned before.
As described above, the diode 26 is repeatedly put into the on-state and the off-state in cooperation 5 with the switching transistor 22. A diode switching period is formed by a time interval of the on-state and a time interval of the off-state. The time intervals of the on-state and the off-state will be named an on-state time interval and an off-state time interval.
Referring to Fig. 3, the secondary circuit of the switching regulator illustrated in Fig. 2 is equivalently represented by a circuit shown in Fig. 3 during the on-state time interval of the diode 26.
During the on-state time interval, the diode 26 becomes 15 conductive and a forward voltage drop of the diode 26 may be therefore néglected because the forward voltage drop is sufficiently small. In addition, the output capacitor 29 is replaced in Fig. 3 by a constant voltage source 45 on the assumption that the output capacitor 29 20 has a sufficiently large capacitance. Furthermore, another constant voltage source 46 specifies an induced voltage which appears across the secondary winding during the on-state time interval of the main transformer 21 and may be referred to as an additional 25 constant voltage source. The induced voltage is invariable and is therefore a constant voltage.
Herein, it is surmised that the rectified voltage across the input capacitor 19 (Fig. 2) and the 13 202~5a8 constant voltage of the constant voltage source 46 are represented by El and E2, respectively, and that the primary and the secondary windings have first and second turns Nl and N2, respectively. Under the circumstances, 5 the constant voltage E2 is given by:
E2 = (N2/Nl)-El.
When an inductance of the inductance element 39 and the output voltage, namely, the constant voltage of the constant voltage source 45 are represented by L and 10 V0, respectively, a current IL which ~lows through the inductance element 39 is given by:
IL = (E2 - V0)~t/L = ((N2-El/Nl) - V0))-t/L.
On the other hand, an inverse voltage takes place across the secondary winding of the main 15 transformer 21 during the off-state time interval of the diode 26. In this event, no current is caused to flow throuqh the inductance element 39 even when such an inverse voltage takes place. This is because such a current is interrupted by the diode 26.
Referring to Fig. 4, description will be made about operation of the switching regulator illustrated in Fig. 2. In Fig. 4, a capacitor voltage (depicted at Vc) across the capacitor 36 is shown along a top line of Fig. 4 while a current (depicted at ITl) flowing through 25 the primary winding of the main transformer 21 is illustrated along a second line of Fig. 4. In addition, a current (depicted at IL) flowing through the inductance element 39 is shown along a bottom line of Fig. 4 and may be called a coil current.
When the on-state time interval and the off-state time interval of the diode 26 are represented 5 by Ton and Toff, respectively, an average value of the coil current IL, namely, the output current (depicted at Io) is given by:
~Ton Io = l/(Ton + Toff ,J IL dt = (((N2-El/Nl) - Vo)/(2L))-(l/((l/Ton) + (Toff/Ton ))).
(1) Under the circumstances, when a resistance of the load resistor 16 is represented by R, the output 15 voltage Vo is given by:
Vo = R-Io. (2) Substitution of Equation (1) into Equation (2) gives:
Vo = N2-El/(Nl-(l + (2L/R).((l/Ton) + (Toff/Ton2)))).
(3) From Equation (3), it is understood that the output voltage Vo can be stabilized by controlling the on-state time interval Ton in response to a variation of the d.c. input voltage-El and the load resistor 16.
Herein, let switching operation of the switching transistor 22 be carried out within a comparatively low frequency which is lower than 100 kHz. In this case, the leakage inductance of the main transformer 21 and the like may be neglected and L may be therefore 20245~8 considered to be equal to zero. Therefore, Equation (3) is modified into:
Vo = N2-El/Nl. (4) It is apparent from Equation (4) that the output 5 voltage Vo can not be stabilized within a low frequency band because the turns Nl and N2 are invariable and the output voltage Vo is determined only by the input voltage El. This shows that the smoothing circuit and the commutation diode, such as 27, are necessary for 10 stabilizing the output voltage Vo in the low frequency band.
However, when the leakage inductance of the main transformer 21 can not be neglected or when the external inductance element is connected in addition to the 15 leakage inductance, an inductance L of such inductance elements can not be neglected from Equation (3).
Accordingly, it is possible to stabilize the output voltage Vo without any smoothing circuit by controlling the on-state time interval Ton, as is apparent from 20 Equation (3).
Referring to Fig. 5, description will be made as regards an equivalent circuit of the switching regulator which specifies a state appearing during the off-state time interval of the diode 26. In this event, the 25 switching transistor 22 is also put into the off-state, namely, a cut-off state. Therefore, the secondary circuit of the transformer 21 may be neglected from the equivalent circuit.
20245~8 Herein, an input voltage (depicted at Vin) across the input capacitor 19 is represented by a d.c.
voltage source 51. Let an excitation current Imo be caused to flow through the primary circuit of the main 5 transformer 21 when the switching regulator is seen from a primary side of the main transformer 21. In addition, let a primary inductance of the main transformer 21 be represented by Ll. Under the circumstances, the excitation current Imo is given by:
Imo = El-Ton/Ll.
The excitation current Imo may be considered as an initial current which flows through the primary inductance or coil Ll. Herein, an initial voltage across the capaci*or 36 of a capacitance C is assumed to 15 be equal to zero without loss of generality. In this case, a free oscillation is caused to occur due to the primary inductance Ll and the capacitance C in the illustrated circuit, as illustrated in Fig. 4. Under the circumstances, a voltage Vc across the capacitor 36 20 and the current Im flowing through the primary inductance Ll are given by:
Vc = ~ Imo sin(t - Ton)/ ~ , and Im = Imo cos(t - Ton)/ ~ .
In Fig. 4, a voltage appears across the 25 secondary winding of the main transformer 21 and is determined by the output voltage Vo at a time instant at which the voltage Vc across the capacitor 36 is returned 17 2~245~8 back to zero. Consequently, the diode 26 is put into the on-state.
Immediately after the diode 26 is turned on, as mentioned before, namely, for a clamp duration between 0 5 to Tcp, the excitation current of the main transformer 21 serves to feed electric power back to the input capacitor 19 and to supply electric power to the output capacitor 29. For the clamp duration, a primary current ITl is caused to flow through the internal diode 37 of 10 the switching transistor 22 in a reverse direction.
Such a primary current ITl is linearly attenuated as illustrated in Fig. 4.
If the switching transistor 22 is turned off for the clamp duration, the capacitor 36 starts to charge at 15 a time instant at which the primary current ITl of the reverse direction becomes zero. Consequently, a voltage takes place across the switching transistor 22. Taking this into consideration, it is possible to avoid occurrence of such a voltage during the on-state of the 20 diode 26, if the switching transistor 22 is kept in the on-state within the clamp duration between 0 and Tcp.
When the switching transistor 22 is turned off, a current which flows through the switching transistor 22 is immediately stopped while the excitation current 25 of the main transformer 21 is caused to flow through the capacitor 36. Therefore, the voltage across the switching transistor 22, namely, the voltage across the capacitor 36 sinusoidally increases after the switching 18 20245~8 transistor 22 is turned off, as illustrated in Fig. 4.
This shows that the voltage across the switching transistor 22 is developed after the current which flows through the switching transistor 22 is stopped.
5 Accordingly, no switching loss occurs in the illustrated switching regulator because the voltage is not crossed or intersected with the current during the off-state time interval of the switching transistor 22. In addition, when the main transformer 21 is put into the 10 clamp state, the switching transistor 22 is turned on at a time instant within the clamp duration. In this case, no switching loss occurs due to intersection of the voltage and the current because the voltage is equal to zero within the clamp state.
Furthermore, when the voltage has a sinusoidal wave, no switching noise appears due to parasitic capacitance and parasitic inductance which might bring about ringing. Therefore, it is possible to remove the input line filter 17 and the snubbers 23, 27, and 28 all 20 of which are illustrated in Fig. 1.
In Fig. 2, the capacitor 36 and the inductance element 39 may be replaced by a depletion layer capacitance of the switching transistor 22 and a leakage inductance of the main transformer 21, respectively.
In the illustrated example, the saturable reactor 41 is added to the secondary circuit of the main transformer 21 and has the high impedance and the low impedance when the saturable reactor 41 is put into an ;
19 202458.8 off-state and an on-state, respectively, as mentioned before. The saturable reactor 41 serves to suppress a steep residual current which otherwise flows due to the leakage inductance appearing when the diode 26 is turned 5 off. This enables a reduction of attenuation in a voltage oscillation wave when the switching transistor 22 is turned off. Thus, the illustrated switching regulator can favorably realize zero voltage switching (ZVS) .
As mentioned before, the on-state time interval of the switching transistor is controlled by using inductance of the leakage inductance or the external inductance. With this structure, it is possible to stabilize the output voltage without the smoothing 15 circuit and the commutation diode. Inasmuch as the inductance of the main transformer is resonant with the capacitance connected in parallel to the switching transistor, the switching transistor is supplied with the voltage of the sinusoidal wave. Consequently, low 20 loss and high efficiency can be accomplished by the illustrated switching regulator. In addition, it is possible to dispense with a noise filter for a reduction of noise and snubbers.
Inasmuch as the saturable reactor is connected 25 to the secondary circuit of the main transformer, it is possible to suppress a steep current resulting from the leakage inductance when the diode 26 is turned off.
Since the saturable reactor exhibits a high impedance = =
~ ' . . 20245~8 characteristic when the saturable reactor is put into an off state, attenuation of the voltage oscillation wave can be reduced when the switching transistor is turned off. This makes it possible to improve an average 5 efficiency due to the ZVS. Accordingly, the switching regulator is simple in structure and compact in size and has a high reliability.
Herein, let switching operation of the switching transistor 22 be carried out within a comparatively low frequency which is lower than 100 kHz. In this case, the leakage inductance of the main transformer 21 and the like may be neglected and L may be therefore 20245~8 considered to be equal to zero. Therefore, Equation (3) is modified into:
Vo = N2-El/Nl. (4) It is apparent from Equation (4) that the output 5 voltage Vo can not be stabilized within a low frequency band because the turns Nl and N2 are invariable and the output voltage Vo is determined only by the input voltage El. This shows that the smoothing circuit and the commutation diode, such as 27, are necessary for 10 stabilizing the output voltage Vo in the low frequency band.
However, when the leakage inductance of the main transformer 21 can not be neglected or when the external inductance element is connected in addition to the 15 leakage inductance, an inductance L of such inductance elements can not be neglected from Equation (3).
Accordingly, it is possible to stabilize the output voltage Vo without any smoothing circuit by controlling the on-state time interval Ton, as is apparent from 20 Equation (3).
Referring to Fig. 5, description will be made as regards an equivalent circuit of the switching regulator which specifies a state appearing during the off-state time interval of the diode 26. In this event, the 25 switching transistor 22 is also put into the off-state, namely, a cut-off state. Therefore, the secondary circuit of the transformer 21 may be neglected from the equivalent circuit.
20245~8 Herein, an input voltage (depicted at Vin) across the input capacitor 19 is represented by a d.c.
voltage source 51. Let an excitation current Imo be caused to flow through the primary circuit of the main 5 transformer 21 when the switching regulator is seen from a primary side of the main transformer 21. In addition, let a primary inductance of the main transformer 21 be represented by Ll. Under the circumstances, the excitation current Imo is given by:
Imo = El-Ton/Ll.
The excitation current Imo may be considered as an initial current which flows through the primary inductance or coil Ll. Herein, an initial voltage across the capaci*or 36 of a capacitance C is assumed to 15 be equal to zero without loss of generality. In this case, a free oscillation is caused to occur due to the primary inductance Ll and the capacitance C in the illustrated circuit, as illustrated in Fig. 4. Under the circumstances, a voltage Vc across the capacitor 36 20 and the current Im flowing through the primary inductance Ll are given by:
Vc = ~ Imo sin(t - Ton)/ ~ , and Im = Imo cos(t - Ton)/ ~ .
In Fig. 4, a voltage appears across the 25 secondary winding of the main transformer 21 and is determined by the output voltage Vo at a time instant at which the voltage Vc across the capacitor 36 is returned 17 2~245~8 back to zero. Consequently, the diode 26 is put into the on-state.
Immediately after the diode 26 is turned on, as mentioned before, namely, for a clamp duration between 0 5 to Tcp, the excitation current of the main transformer 21 serves to feed electric power back to the input capacitor 19 and to supply electric power to the output capacitor 29. For the clamp duration, a primary current ITl is caused to flow through the internal diode 37 of 10 the switching transistor 22 in a reverse direction.
Such a primary current ITl is linearly attenuated as illustrated in Fig. 4.
If the switching transistor 22 is turned off for the clamp duration, the capacitor 36 starts to charge at 15 a time instant at which the primary current ITl of the reverse direction becomes zero. Consequently, a voltage takes place across the switching transistor 22. Taking this into consideration, it is possible to avoid occurrence of such a voltage during the on-state of the 20 diode 26, if the switching transistor 22 is kept in the on-state within the clamp duration between 0 and Tcp.
When the switching transistor 22 is turned off, a current which flows through the switching transistor 22 is immediately stopped while the excitation current 25 of the main transformer 21 is caused to flow through the capacitor 36. Therefore, the voltage across the switching transistor 22, namely, the voltage across the capacitor 36 sinusoidally increases after the switching 18 20245~8 transistor 22 is turned off, as illustrated in Fig. 4.
This shows that the voltage across the switching transistor 22 is developed after the current which flows through the switching transistor 22 is stopped.
5 Accordingly, no switching loss occurs in the illustrated switching regulator because the voltage is not crossed or intersected with the current during the off-state time interval of the switching transistor 22. In addition, when the main transformer 21 is put into the 10 clamp state, the switching transistor 22 is turned on at a time instant within the clamp duration. In this case, no switching loss occurs due to intersection of the voltage and the current because the voltage is equal to zero within the clamp state.
Furthermore, when the voltage has a sinusoidal wave, no switching noise appears due to parasitic capacitance and parasitic inductance which might bring about ringing. Therefore, it is possible to remove the input line filter 17 and the snubbers 23, 27, and 28 all 20 of which are illustrated in Fig. 1.
In Fig. 2, the capacitor 36 and the inductance element 39 may be replaced by a depletion layer capacitance of the switching transistor 22 and a leakage inductance of the main transformer 21, respectively.
In the illustrated example, the saturable reactor 41 is added to the secondary circuit of the main transformer 21 and has the high impedance and the low impedance when the saturable reactor 41 is put into an ;
19 202458.8 off-state and an on-state, respectively, as mentioned before. The saturable reactor 41 serves to suppress a steep residual current which otherwise flows due to the leakage inductance appearing when the diode 26 is turned 5 off. This enables a reduction of attenuation in a voltage oscillation wave when the switching transistor 22 is turned off. Thus, the illustrated switching regulator can favorably realize zero voltage switching (ZVS) .
As mentioned before, the on-state time interval of the switching transistor is controlled by using inductance of the leakage inductance or the external inductance. With this structure, it is possible to stabilize the output voltage without the smoothing 15 circuit and the commutation diode. Inasmuch as the inductance of the main transformer is resonant with the capacitance connected in parallel to the switching transistor, the switching transistor is supplied with the voltage of the sinusoidal wave. Consequently, low 20 loss and high efficiency can be accomplished by the illustrated switching regulator. In addition, it is possible to dispense with a noise filter for a reduction of noise and snubbers.
Inasmuch as the saturable reactor is connected 25 to the secondary circuit of the main transformer, it is possible to suppress a steep current resulting from the leakage inductance when the diode 26 is turned off.
Since the saturable reactor exhibits a high impedance = =
~ ' . . 20245~8 characteristic when the saturable reactor is put into an off state, attenuation of the voltage oscillation wave can be reduced when the switching transistor is turned off. This makes it possible to improve an average 5 efficiency due to the ZVS. Accordingly, the switching regulator is simple in structure and compact in size and has a high reliability.
Claims (6)
1. A switching regulator for use in regulating an input a.c. voltage into an output d.c. voltage, said switching regulator comprising input rectifying means for rectifying said input a.c. voltage into a rectified voltage and converting means supplied with said rectified voltage for converting said rectified voltage into said output d.c. voltage, said converting means having a pair of input terminals supplied with said rectified voltage and a pair of output terminals for said output d.c. voltage and comprising:
a transformer comprising a primary winding and a secondary winding which form parts of primary and secondary circuits, respectively;
said primary circuit comprising:
a switching transistor connected in series to said primary winding to form a primary series connection circuit and operable in response to a switching control signal to be selectively put into an on-state and an off-state, said primary series connection circuit being connected between said input terminals;
said secondary circuit comprising:
a series connection of said secondary winding, a diode, a saturable reactor, and an output capacitor with said output capacitor connected to said output terminals;
(Claim 1 continued) said converting means further comprising:
a frequency control circuit connected between a preselected one of said output terminals and said switching transistor for controlling said switching transistor to produce an output control signal so that a time interval of said on-state of said switching transistor is varied in accordance with said output d.c.
voltage with a time interval of said off-state of the switching transistor kept unchanged; and means for supplying said output control signal to said switching transistor as said switching control signal.
a transformer comprising a primary winding and a secondary winding which form parts of primary and secondary circuits, respectively;
said primary circuit comprising:
a switching transistor connected in series to said primary winding to form a primary series connection circuit and operable in response to a switching control signal to be selectively put into an on-state and an off-state, said primary series connection circuit being connected between said input terminals;
said secondary circuit comprising:
a series connection of said secondary winding, a diode, a saturable reactor, and an output capacitor with said output capacitor connected to said output terminals;
(Claim 1 continued) said converting means further comprising:
a frequency control circuit connected between a preselected one of said output terminals and said switching transistor for controlling said switching transistor to produce an output control signal so that a time interval of said on-state of said switching transistor is varied in accordance with said output d.c.
voltage with a time interval of said off-state of the switching transistor kept unchanged; and means for supplying said output control signal to said switching transistor as said switching control signal.
2. A switching regulator as claimed in Claim 1, wherein said primary circuit comprises:
a capacitor connected in parallel to said switching transistor.
a capacitor connected in parallel to said switching transistor.
3. A switching regulator as claimed in Claim 1, wherein said secondary circuit comprises:
an inductance element connected in series to said saturable reactor.
an inductance element connected in series to said saturable reactor.
4. A switching regulator as claimed in Claim 1, wherein said saturable reactor has no control circuit.
5. A switching regulator as claimed in Clam 4, wherein said saturable reactor has a high impedance and a low impedance while said diode is nonconductive and conductive, respectively.
6. A switching regulator as claimed in Claim 5, wherein said high impedance falls within a range between 100 kilohms and several hundreds of kilohms and said low impedance is smaller than 1 ohm.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP227622/1989 | 1989-09-04 | ||
| JP22762289A JPH0393464A (en) | 1989-09-04 | 1989-09-04 | Switching regulator |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CA2024588A1 CA2024588A1 (en) | 1991-03-05 |
| CA2024588C true CA2024588C (en) | 1996-03-12 |
Family
ID=16863815
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA 2024588 Expired - Fee Related CA2024588C (en) | 1989-09-04 | 1990-09-04 | Switching regulator of a simple structure capable of reducing a switching loss |
Country Status (2)
| Country | Link |
|---|---|
| JP (1) | JPH0393464A (en) |
| CA (1) | CA2024588C (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2578145A (en) * | 2018-10-18 | 2020-04-22 | Owlstone Med Ltd | Waveform generator |
-
1989
- 1989-09-04 JP JP22762289A patent/JPH0393464A/en active Pending
-
1990
- 1990-09-04 CA CA 2024588 patent/CA2024588C/en not_active Expired - Fee Related
Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2578145A (en) * | 2018-10-18 | 2020-04-22 | Owlstone Med Ltd | Waveform generator |
| US11469742B2 (en) | 2018-10-18 | 2022-10-11 | Owlstone Medical Limited | Waveform generator |
| GB2578145B (en) * | 2018-10-18 | 2023-08-09 | Owlstone Med Ltd | Waveform generator |
Also Published As
| Publication number | Publication date |
|---|---|
| CA2024588A1 (en) | 1991-03-05 |
| JPH0393464A (en) | 1991-04-18 |
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| Date | Code | Title | Description |
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| EEER | Examination request | ||
| MKLA | Lapsed |