CA1218695A - Pulse-width modulated ac motor drive - Google Patents
Pulse-width modulated ac motor driveInfo
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- CA1218695A CA1218695A CA000459482A CA459482A CA1218695A CA 1218695 A CA1218695 A CA 1218695A CA 000459482 A CA000459482 A CA 000459482A CA 459482 A CA459482 A CA 459482A CA 1218695 A CA1218695 A CA 1218695A
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Abstract
ABSTRACT OF THE DISCLOSURE
A pulse-width modulated motor drive for an AC
motor includes components to sense the primary current of the motor and derive signals corresponding to it and a variable frequency inverter supplying AC to the motor. The system includes timing wave reference signals, and a pulse width modulator, which is controlled by the reference signals, for controlling the inverter. A motor speed signal generator and components to compensate any phase or amplitude difference between the time wave reference and the primary current wave form in accordance with the speed signal.
A pulse-width modulated motor drive for an AC
motor includes components to sense the primary current of the motor and derive signals corresponding to it and a variable frequency inverter supplying AC to the motor. The system includes timing wave reference signals, and a pulse width modulator, which is controlled by the reference signals, for controlling the inverter. A motor speed signal generator and components to compensate any phase or amplitude difference between the time wave reference and the primary current wave form in accordance with the speed signal.
Description
51, 865 IMPROVl~:D PULSE-WIDTH MODULATED
AC MOTOR DRIVE
_CK(~ROUND OF THE INVENTION
This invention relates to pulse-width modulation for controlling the instantaneous AC waveform of the pri-mary current of an AC motor. In the past, this kind of AC motor drive included a combination of: an AC motor, a current sensor to detec-t the primaxy current of the AC
motor, an adjustable frequency inverter unit to drive the AC motor under variable frequency, a reference timing wave generator current waveforms to control the instantaneous amount of primary AC current to be supplied to the AC motor, and a pulse-width generator circuit for generating control signals applied to the adjustable frequency inverter unit and acting as a comparator between the output of the current time wave generator and the output of the current sensor.
With such a pulse-width modulation circuit when controlling the instantaneous AC waveform of the primary current of the AC motor, as the frequency oE the primary current supplied to the AC motor increases, in other words, as the speed of the AC motor increases~ the applied timing waveform and the waveform of the primary current which is fed back from the current sensors to the AC motor tend to be displaced from one another. As a result, the errors in amplitude and phase between the two compared current waveforms will increase.
~,.,
AC MOTOR DRIVE
_CK(~ROUND OF THE INVENTION
This invention relates to pulse-width modulation for controlling the instantaneous AC waveform of the pri-mary current of an AC motor. In the past, this kind of AC motor drive included a combination of: an AC motor, a current sensor to detec-t the primaxy current of the AC
motor, an adjustable frequency inverter unit to drive the AC motor under variable frequency, a reference timing wave generator current waveforms to control the instantaneous amount of primary AC current to be supplied to the AC motor, and a pulse-width generator circuit for generating control signals applied to the adjustable frequency inverter unit and acting as a comparator between the output of the current time wave generator and the output of the current sensor.
With such a pulse-width modulation circuit when controlling the instantaneous AC waveform of the primary current of the AC motor, as the frequency oE the primary current supplied to the AC motor increases, in other words, as the speed of the AC motor increases~ the applied timing waveform and the waveform of the primary current which is fed back from the current sensors to the AC motor tend to be displaced from one another. As a result, the errors in amplitude and phase between the two compared current waveforms will increase.
~,.,
2 51,865 SUMMARY OF T~IE INVENTION
The object of the present invention is to provide an improved pulse-~width modulation technique which eliminates the drawbacks of the aforementioned prior art approach, there-by to enable the supply of a reference time wave having the same waveform as the primary current fed-back signal regard-less of the speed of -the AC motor.
This is achieved, according to the present invention, by compensating both the amplitude and the phase in -the reference time wave being compared to the instantaneous AC
primary current. This is achieved automatically in relation to the speed of the AC motor.
BRIEF DESCRIPTION OF THE DRAWINGS
Figure 1 is a block diagram of a pulse-width modulated AC motor drive of the prior art;
Fig. 2 is the basic block diagram of the adjustable frequency inverter unit of the AC motor drive of Fig. l;
Fig. 3 is a block diagram illustrating -the implemen-tation of one phase the current control system in the AC
motor drive of Fig. l;
Fig. 4 is a block diagram useful in explaining the operation of the control system of Fig. 3;
Fig. 5 is a block diagram illustrating a pulse-~idth modulation ~C motor drive according to the invention;
Fig. 6 is a block diagram showing one embodiment oE the time wave generator which is part of the apparatus of Fig. 5;
Fig. 7 shows the equivalent circuit diagram for one phase of an induction motor; and Flg. 8 is a block diagram showing another embodiment of the time wave generator which is part of the apparatus of Fig. 5 DETAILED DESCRIPTION OF THE INVENTION
The invention relates to pulse-width modulation for controllinq the instantaneous AC waveform of the primary current of an AC motor. As shown in Fig. 1, a motor drive of the prior art includes an AC motor 1, a current sensor 2 for sensing the primary currents of the AC motor 1 under variable frequency, a time wafe generator 4 for generating ~$~S
The object of the present invention is to provide an improved pulse-~width modulation technique which eliminates the drawbacks of the aforementioned prior art approach, there-by to enable the supply of a reference time wave having the same waveform as the primary current fed-back signal regard-less of the speed of -the AC motor.
This is achieved, according to the present invention, by compensating both the amplitude and the phase in -the reference time wave being compared to the instantaneous AC
primary current. This is achieved automatically in relation to the speed of the AC motor.
BRIEF DESCRIPTION OF THE DRAWINGS
Figure 1 is a block diagram of a pulse-width modulated AC motor drive of the prior art;
Fig. 2 is the basic block diagram of the adjustable frequency inverter unit of the AC motor drive of Fig. l;
Fig. 3 is a block diagram illustrating -the implemen-tation of one phase the current control system in the AC
motor drive of Fig. l;
Fig. 4 is a block diagram useful in explaining the operation of the control system of Fig. 3;
Fig. 5 is a block diagram illustrating a pulse-~idth modulation ~C motor drive according to the invention;
Fig. 6 is a block diagram showing one embodiment oE the time wave generator which is part of the apparatus of Fig. 5;
Fig. 7 shows the equivalent circuit diagram for one phase of an induction motor; and Flg. 8 is a block diagram showing another embodiment of the time wave generator which is part of the apparatus of Fig. 5 DETAILED DESCRIPTION OF THE INVENTION
The invention relates to pulse-width modulation for controllinq the instantaneous AC waveform of the primary current of an AC motor. As shown in Fig. 1, a motor drive of the prior art includes an AC motor 1, a current sensor 2 for sensing the primary currents of the AC motor 1 under variable frequency, a time wafe generator 4 for generating ~$~S
3 51,865 reference waves used to control the instantaneous value of the primary currents supplied on phase lines U, V, W to the AC motor, and a pulse~width modulation circui~ 5 for generat-ing control signals for controlling the switches of the adjust-able frequency inverter unit 3. Circuit 5 operates by compar-ing the outputs of the reference time wave generator 4 and the outputs of current sensors 2. The AC motor of Fig. 1 may be an induction motor, or a synchronous motor, for instance~
The ad~ustable frequency inverter unit 3 may consist of an inverter includin~ solid-state semiconductor devices such as power transistors, or MOSFET's as are capable of performing high-speed switching.
Referring to Fig. 2, the basic organization of the driver circuit for the AC motor is shown to include an adjustable frequency inverter unit 3 coupled to the AC motor 1 and comprising, a solid-state semiconductor devices, power transistors Trl, Tr2, Tr3, Tr4, Tr5 and Tr6 with respective diodes Dl, D2, D3, D4, D5 and D6 associated therewith. The base electrodes bl, b2, b3, b4, b5 and b6 of the respective power transistors Trl, Tr2, Tr3, Tr4, Tr5 and Tr6 are supplied via lines 10, with the control signals from the pulse width modulation circuit 5. The latter is Fig. 3.
Referring to Fig. 3, the pulse-width modulation circuit includes subtractors 501, 502, 503; adders 504 and 505; operational amplifiers 506, 507 and 508; a saw-tooth wave generator 509; and comparators 510, 511 and 512. The respective outputs are applied by lines 25-30 to the six trans-istors NOT ~ircuits (NOT gate circuits) 513, 514, 515 are provided for inverting the signals of lines 26, 28, 30 which are alternate to lines 25, 27 and 29. For the sake of simpli-city, the pulse-width modulation circuit 3 is shown con-trol-ling a three-phase AC motor. Phases U, V, W of the AC motor 1 are respectively connected between the emitter of the power transistor Trl and the collector of transistor Tr2, the emitter of transistor Tr3 and the collector of transistor Tr4, . ~.
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The ad~ustable frequency inverter unit 3 may consist of an inverter includin~ solid-state semiconductor devices such as power transistors, or MOSFET's as are capable of performing high-speed switching.
Referring to Fig. 2, the basic organization of the driver circuit for the AC motor is shown to include an adjustable frequency inverter unit 3 coupled to the AC motor 1 and comprising, a solid-state semiconductor devices, power transistors Trl, Tr2, Tr3, Tr4, Tr5 and Tr6 with respective diodes Dl, D2, D3, D4, D5 and D6 associated therewith. The base electrodes bl, b2, b3, b4, b5 and b6 of the respective power transistors Trl, Tr2, Tr3, Tr4, Tr5 and Tr6 are supplied via lines 10, with the control signals from the pulse width modulation circuit 5. The latter is Fig. 3.
Referring to Fig. 3, the pulse-width modulation circuit includes subtractors 501, 502, 503; adders 504 and 505; operational amplifiers 506, 507 and 508; a saw-tooth wave generator 509; and comparators 510, 511 and 512. The respective outputs are applied by lines 25-30 to the six trans-istors NOT ~ircuits (NOT gate circuits) 513, 514, 515 are provided for inverting the signals of lines 26, 28, 30 which are alternate to lines 25, 27 and 29. For the sake of simpli-city, the pulse-width modulation circuit 3 is shown con-trol-ling a three-phase AC motor. Phases U, V, W of the AC motor 1 are respectively connected between the emitter of the power transistor Trl and the collector of transistor Tr2, the emitter of transistor Tr3 and the collector of transistor Tr4, . ~.
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4 51,865 and the emitter of transistor Tr5 and between the collector of Tr6. When controlling the instantaneous AC waveform of the primary current flowing through phase U, an error signal (i - i ) is derived at 501 between the reference value uso us iUso for the primary current of phase U (supplied on line 11 from the reference time wave generator 4 of Fig~ 1) and the sensed value iUS of the primary current of phase U derived on line 11 from current sensor 2. An output is derived on line 15 from subtractor 501 (Fig. 3). The error signal is ampli-fied by amplifier 506, then, fed via line 1~ to comparator510. The latter compares in amplitude such error signal with the saw-tooth signal derived on line 21 from saw-tooth signal generator 509. During the periods that the error signal is larger in amplitude than the saw-tooth signal, comparator 510 applies to the base electrode bl of transistor Trl, via line 25, a signal which will be high, whereas during the periods that the error signal is smaller than the saw-tooth wave signal, the comparator output of line 25 to b'l is low, and conversely for line 26 to transistor Tx2 and its base electrode b2. A similar reasoning applies to the other lines 27-30 and the other comparators 511, 512 in relation to the other phases V, W, respectively. Thus referring to Figs. 2 and 3, comparator 510 applies its output via line 25 to the base electrode bl of power transistor Trl through a base driving circuit (not shown) so that when the logic of line 25 is high, the power transistor Trl becomes conductive, whereas when the logic of line 25 is low transistor Trl becomes non-conductive. On the other hand, the base elec-trode b2 of power transistor Tr2 to which is applied the control signal of line 26 which follows an inversion of ~.he comparator output by the NOT circuit 513, the signal being applied through a base driving circuit (not shown). The conducting periods of transistor Tr2 correspond -to the non-conducting periods of transistor Trl and conversely, so that the states of operation are opposite to one another. There-fore, a voltage corresponding -to the error signal between the primary current command signal iUso and the detected
5 51,865 primary current iUS is applied to phase U of the AC motor, whereby the waveform of the sensed primary current iUS is controlled so as -to match the assigned re~erence current signal iUso.
Similarly, the primary current of phase V is control-led ~like in the case of phase U) from lines 12 and 14 through the combination of subtractor 502, amplifier 507, saw-tooth generator 509, comparator 511, and NOT circuit 514. Control of the primary current of phase W is achieved by using the primary currents for phase U and phase ~, based on the fact tha~ the sum of the primary currents in each phase supplied to the AC motor 1 is zero~ Therefore, the primary current reference signal (iuso) for phase U and the primary current reference signal (ivso) for phase V are from lines 11' and 12' added by adder 504 to provide the primary current reference signal (-iwso) for phase W. The sensed primary current (ius) of line 13, and phase U, and the sensed primary current (ivs) of line 14, and phase V, are added by adder 505 to provide the feedback primary current the feedback signal (-iws) of phase W. Fur-ther, the current reference signal (-iwso) is subtracted from (-iws) by subtractor 503 to provide on line 17 the error signal (iWso iws) between wso ws mode of operation of circuit 5 is similar to the one described for phases U and V. When controlling the instantaneous AC
waveform of the primary current of the AC motor by the pulse-width modulation circuit 5 just described, it is observed that, as the ~requency of the primary currents supplied on lines U, V, W to the AC motor increases, in other words, as the speed of the AC motor increases, the reference time waveform for the instantaneous AC primary current and the fed-back waveform from the primary current supplied to the AC motor tend to increase. This can be explained by reference to Fig~
3 as follows:
The amplifiers 506, 507 and 508 may be considered as coefficient multipliers which produce an output by multi-plying the input supplied thereto by a certain coefficient.
Comparator 510, 511 and 512 may also be considered as ~`'`'.'i
Similarly, the primary current of phase V is control-led ~like in the case of phase U) from lines 12 and 14 through the combination of subtractor 502, amplifier 507, saw-tooth generator 509, comparator 511, and NOT circuit 514. Control of the primary current of phase W is achieved by using the primary currents for phase U and phase ~, based on the fact tha~ the sum of the primary currents in each phase supplied to the AC motor 1 is zero~ Therefore, the primary current reference signal (iuso) for phase U and the primary current reference signal (ivso) for phase V are from lines 11' and 12' added by adder 504 to provide the primary current reference signal (-iwso) for phase W. The sensed primary current (ius) of line 13, and phase U, and the sensed primary current (ivs) of line 14, and phase V, are added by adder 505 to provide the feedback primary current the feedback signal (-iws) of phase W. Fur-ther, the current reference signal (-iwso) is subtracted from (-iws) by subtractor 503 to provide on line 17 the error signal (iWso iws) between wso ws mode of operation of circuit 5 is similar to the one described for phases U and V. When controlling the instantaneous AC
waveform of the primary current of the AC motor by the pulse-width modulation circuit 5 just described, it is observed that, as the ~requency of the primary currents supplied on lines U, V, W to the AC motor increases, in other words, as the speed of the AC motor increases, the reference time waveform for the instantaneous AC primary current and the fed-back waveform from the primary current supplied to the AC motor tend to increase. This can be explained by reference to Fig~
3 as follows:
The amplifiers 506, 507 and 508 may be considered as coefficient multipliers which produce an output by multi-plying the input supplied thereto by a certain coefficient.
Comparator 510, 511 and 512 may also be considered as ~`'`'.'i
6 51,865 coe~icient multipliers for the outputs of the amplifiers 506, 507 and 508. Thus, comparators 510, 511 and 512 compare in amplitude the outputs o~ amplifiers 506, 507 and 508 wi~h the output of the saw--tooth wave genera-tor 509.
In relation to such amplitude comparison, a square wave voltage is generated having a frequency which is equal to the frequency of the saw-tooth wave signal of lines 21-23 at the output of saw tooth wave generator 509. When this square wave voltage is applied to each of the phases of the AC motor, since the duty ratio of the square wave vol-tage varies according to the output amplitudes of the amplifiers 506, 507 and 508 (provided the frequency of the saw-tooth wave is high enough compared wi-th the frequency of the out-puts of amplifiers 506, 507 and 508), the amplitude of the AC voltage component having the same frequency as that of the outputs of the amplifiers 506, 507 and 508 contained in the square wave voltage fed to each phase of the AC motor 1 will normally be proportional to the output amplitudes of these amplifiers. Moreover, this AC component will be normally in phase with the generated outputs of amplifiers 506, 507 and 508.
The operation of the current control system of Fig. 3 can be described on a single phase basis for the AC
motor in the light of the previous explanation, by reference to the block diagram of Fig. 4. In Fig. 4 a transfer function
In relation to such amplitude comparison, a square wave voltage is generated having a frequency which is equal to the frequency of the saw-tooth wave signal of lines 21-23 at the output of saw tooth wave generator 509. When this square wave voltage is applied to each of the phases of the AC motor, since the duty ratio of the square wave vol-tage varies according to the output amplitudes of the amplifiers 506, 507 and 508 (provided the frequency of the saw-tooth wave is high enough compared wi-th the frequency of the out-puts of amplifiers 506, 507 and 508), the amplitude of the AC voltage component having the same frequency as that of the outputs of the amplifiers 506, 507 and 508 contained in the square wave voltage fed to each phase of the AC motor 1 will normally be proportional to the output amplitudes of these amplifiers. Moreover, this AC component will be normally in phase with the generated outputs of amplifiers 506, 507 and 508.
The operation of the current control system of Fig. 3 can be described on a single phase basis for the AC
motor in the light of the previous explanation, by reference to the block diagram of Fig. 4. In Fig. 4 a transfer function
7 per-phase for the current limiting circuit of Fig. 3 is shown having a proportional gain as stated above. Another transfer Eunction is shown at 8, for the primary current is relative to phase voltage VS selected for illustration on a single phase basis. This function can be expressed in terms of the parameters belonging to the AC motor, such as resis-tance and inductance. In the system shown by the block diagram of Fig. 4, as the frequency of the primary current reference signal iso increases, the frequency of the phase voltage VS increases, and at the same time, the counter electromotive force (voltage) vb generated by the AC motor 1 also increases, thereby to follow the increase of the .;
-7 51,8~5 frequency of the phase voltage vs. As a result, the voltage derived by subtracting the counter electromotive force (voltage) Vb from the phase voltage VS will result in varying the pri-mary current is in such a way that the gain of the transfer function 8 be lowered. Therefore, if the transfer function 7 per-phase has only a proportional transfer characteristic~
that is, if no compensation is provided for such gain reduction due to the counter-electromotive force (voltage) vb when the frequency of the primary current reference signal iso increases, both the amplitude error and phase error, between the primary current reference signal iso and the sensed primary current is, will increase. Therefore, with the prior art pulse-width modulation approach in AC motor drive, a drawback is as follows:
when the frequency of the primary current supplied to the AC
motor will increase as the speed of the AC motor increases the reference waveform for the instantaneous AC primary current and the waveform of the primary current supplied to the AC
motor tend to displace each other and the amplitude and phase errors between the two current waveforms will increase.
The object of the present invention is to provide an improved pulse-width modulation circuit for an AC motor eli-minating the aforementioned drawback of the apparatus of the prior art so that a primary current will be supplied having a waveform which remains identical to that of -the primary current reference signal, regardless of the speed of the AC
motor. This is achieved by compensating both the amplitude and -the phase of the reference waveform compared with the instantaneous AC primary current and this is done automatically in relation to the speed of the AC motor.
Referring to Fig. 5 (in which like references are used to den~te like parts in Fig. 1), a speed detector is provided which is associated with the reference time wave generator 4 of Fig. 1, and the reference time wave generator 4 is modified by providing internal compensation as shown according to one embodiment in Fig. 6.
i`~`,.~,
-7 51,8~5 frequency of the phase voltage vs. As a result, the voltage derived by subtracting the counter electromotive force (voltage) Vb from the phase voltage VS will result in varying the pri-mary current is in such a way that the gain of the transfer function 8 be lowered. Therefore, if the transfer function 7 per-phase has only a proportional transfer characteristic~
that is, if no compensation is provided for such gain reduction due to the counter-electromotive force (voltage) vb when the frequency of the primary current reference signal iso increases, both the amplitude error and phase error, between the primary current reference signal iso and the sensed primary current is, will increase. Therefore, with the prior art pulse-width modulation approach in AC motor drive, a drawback is as follows:
when the frequency of the primary current supplied to the AC
motor will increase as the speed of the AC motor increases the reference waveform for the instantaneous AC primary current and the waveform of the primary current supplied to the AC
motor tend to displace each other and the amplitude and phase errors between the two current waveforms will increase.
The object of the present invention is to provide an improved pulse-width modulation circuit for an AC motor eli-minating the aforementioned drawback of the apparatus of the prior art so that a primary current will be supplied having a waveform which remains identical to that of -the primary current reference signal, regardless of the speed of the AC
motor. This is achieved by compensating both the amplitude and -the phase of the reference waveform compared with the instantaneous AC primary current and this is done automatically in relation to the speed of the AC motor.
Referring to Fig. 5 (in which like references are used to den~te like parts in Fig. 1), a speed detector is provided which is associated with the reference time wave generator 4 of Fig. 1, and the reference time wave generator 4 is modified by providing internal compensation as shown according to one embodiment in Fig. 6.
i`~`,.~,
8 51,865 Referring to Fig~ 6, 401 is an adder, 402 a multiplier, 403 a V/F converter, 404 a counter and 405 is a digital adder (AD~ER). 406 and 407 are read-only-memories (ROM), 408 and 409 D/A converters of the multiplier type, 410 a compensating circuit, and 411 is an A/D converter.
Within reference wave generator 4 is provided a function generator 410 working as a compensating circuit, in re-sponse to external inputs carrying an analog signal ~so on ~ines 31, 31' corresponding to the angular velocity refer-ence value (~so) and an analog signal ~r on line 32 cor-responding to the angular velocity (~r) which is detected by an analog speed detector such as a tachometer (9 in Fig.
5) providing the rotative speed of the motor. These two analog signals are added between lines 31 and 32' to the adder 401. Then -the sum signal of line 33 so derived is converted into a pulse train by a V/F converter 403. The resulting pulse train of line 34 is in turn counted by counter 404 to provide a count obtained by integration of (~r + ~so)~ that is, a digital signal representing a phase angle 9O such that 0O = (wr +~Jso)t. An analog compensating signal representing a phase angle ~9O/ to be applied as a compensating angle to phase angle 9O~ is generated on line 36 by the function gener-ator, or compensating circuit, 410. This analog compensating signal is converted into a digital signal by A/D converter 411, and the outputted digital signal is fed to digital adder 40S, thereby to digitally sum up 0O of line 35 and ~0O of line 36. On the other hand, read-only-memories 406 and 407 have stored therein data representing sinusoidal waves which are at 120 phase from one another. Therefore, digital signals representing sinusoidal waveforms sin (~O ~ ~O) and sin (0O ~ ~O - 2/3 II) are effectively obtained at the outputs of read-only-memories 406 and 407, respectively. The com-pensating circuit 410 also generates on line 45 an analog signal for the purpose of amplitude compensation by a co-efficient Ks, which is applied to multiplier 402 whereby multiplier 402 provides on line 43 the analog value of Ks ¦iso by multiplying coefficient Ks of line 45 by the analog value ¦isol of line 40 derived as an external input representing the amplitude of the primary current reference time wave
Within reference wave generator 4 is provided a function generator 410 working as a compensating circuit, in re-sponse to external inputs carrying an analog signal ~so on ~ines 31, 31' corresponding to the angular velocity refer-ence value (~so) and an analog signal ~r on line 32 cor-responding to the angular velocity (~r) which is detected by an analog speed detector such as a tachometer (9 in Fig.
5) providing the rotative speed of the motor. These two analog signals are added between lines 31 and 32' to the adder 401. Then -the sum signal of line 33 so derived is converted into a pulse train by a V/F converter 403. The resulting pulse train of line 34 is in turn counted by counter 404 to provide a count obtained by integration of (~r + ~so)~ that is, a digital signal representing a phase angle 9O such that 0O = (wr +~Jso)t. An analog compensating signal representing a phase angle ~9O/ to be applied as a compensating angle to phase angle 9O~ is generated on line 36 by the function gener-ator, or compensating circuit, 410. This analog compensating signal is converted into a digital signal by A/D converter 411, and the outputted digital signal is fed to digital adder 40S, thereby to digitally sum up 0O of line 35 and ~0O of line 36. On the other hand, read-only-memories 406 and 407 have stored therein data representing sinusoidal waves which are at 120 phase from one another. Therefore, digital signals representing sinusoidal waveforms sin (~O ~ ~O) and sin (0O ~ ~O - 2/3 II) are effectively obtained at the outputs of read-only-memories 406 and 407, respectively. The com-pensating circuit 410 also generates on line 45 an analog signal for the purpose of amplitude compensation by a co-efficient Ks, which is applied to multiplier 402 whereby multiplier 402 provides on line 43 the analog value of Ks ¦iso by multiplying coefficient Ks of line 45 by the analog value ¦isol of line 40 derived as an external input representing the amplitude of the primary current reference time wave
9 51,~65 signal. The signal of line 43 derived at the output of multi-plier 402 is applied by lines 43~ 43' to D/A conVerters 408, 409, respectively which are of the multiplier type~ Therefore, two phase signals characteristic of the primary current refer-ence time waves: Ks ¦isOl gin (~O + ~O) ¦ so¦
(~O ~ ~O - 2/3 II) where aO = (~r + ~ so) t, are derived at the outputs of D/A converters 408, 409.
Of these two time wave reference signals, the ampli-tude component and phase component are compensated as proposed in accordance with the present invention. When compensating circuit 410 is not operating, the compensation coefficient Ks = 1 and compensating phase angle ~ = O. For the two values obtained at the output of compensating circuit 410, the com-pensation coefficient Ks and compensating phase angle ~O can be calculated as explained hereinafter. Assuming first that the AC motor of Fig. 5 is an induction motor, the motor can be represented by its equivalent circuit shown in Fig. 7.
There, Rl and R2 are, respectively, the primary winding re-sistance and secondary winding resistance, whereas ~l and ~2 are, respectively, the primary winding leakage inductance and secondary winding leakage inductance, and M is the mutual inductance between primary and secondary windings. S denotes the slip of the motor for which the equations S =~Js/~o and (~0 = ~Jr + ~s) hold. Since Rl, R2, el, e2, and M are known and the frequency component of the reference signal iso applied (on a single phase basis) to the primary current is is, as shown earlier, equal to ~r + ~Jso~ it follows that ~Jr and ~so are also known. To be exact in the transient state, ~so and ~s o:E Fig. 7 do not coincide, but in the stationary state they will be the same. Therefore, if Gs is the trans-fer function of the primary current is with respect to the phase voltage VS oE an induction motor (formulated for simpli-city on a single phase basis), the following equation holds:
isvs = Gs (l) GS can be calculated si.nce ~Jr iS known when ~s is known being fed externally as a reference signal. The transfer function &c of block 7 in Fig. 4, which is the transfer function per phase of the pulse width modulation circuit 5 of Fig. 3, leads to the following equation after equation (1):
_ = GsGc iso l + GsGc (2) The fraction on the right side can be generally converted to the form GS GC = ¦ G ¦ e 1 + GSGC
where ~G ¦ = ¦ GsGc l l ¦l + GsGc ¦ ( ) From equations (2) and (3) it appears ~hat in order to have iS iso, e j~6iso I G¦
should be used as a new reference value for is. From equation (3), then:
¦G ¦ ej I I e~j~aiSO = iSO
Thus, Ks, the compensation coefficient for the amplitude component of the primary current reference time waveform and the compensation angle ~OO~ for the phase component of the primary current can both be calculated from the following equations:
~.~
11 51,865 Ks = l = ¦l ~ GsGc ¦
IGI ¦ GsGc ~0 = ~0 where ~0 is the phase angle of the -transfer function G.
It appears that since ~so is externally supplied and the parameters of the induction motor such as Rl and R2, and the transfer function per phase of block 7 of the pulse-width modulation circuits of Fig. 3 are known values, a compensating circuit such as 410 shown in Fig. 6 ean be designed whieh responds to a speed detector 9 providing the angular velocity lJr of the AC motor.
If the AC motor l of Fig. 5 is a synchronous motor, the slip angular velocity ~s is zero. Moreover when a synchro-nous motor has a field current which is controlled, or where a permanent magnet is used, the transfer function Gs of the primary eurrent is with respect to the phase voltage VS per phase ean be easily obtained from the parameters of the syn-chronous motor, that is, from its resistance and inductance.
Therefore, like in the case of an induction motor, a compen-sating eireuit is easily designed. For example, one embodi-ment of the reference time wave generator 4 provided with an integral eompensating circuit sueh as 410 can be arranged so that it does not require wsO as an input (like in Fig. 6), thus dispensing with adder 401. Here, the angular veloeity ~r is directly applied by line 31 to the ~/F eonverter 403 and the compensating circuit 410 receives only ~r on line 32.
Fig. 8 shows another embodiment of the time wave referenee generator 4. ~ere, a rotation angle detector, in the form of an encoder, is used instead of the tachometer 9 of Fig. 5. Bloeks 402 through 411 are the same as those in Fig. 6. In addition, a counter 412, a digital adder (ADDER) 413, and a F/V eonverter 414 are provided. The operation of the cireuit of Fig. 8, is as follows: An analog signal exter~
nally supplied and corresponding to the slip angular velocity reference value ~so is eonverted by the V/F eonverter ~03 into '`?~
12 51,865 the V/F converter 403 into a pulse train outputted on line 34. The pulses are counted by counter 404 providing a value resulting from the in-tegration of ~so~ that is, a digital signal representing phase angle 9so~ where so = ~sot. On the other hand, on line 32 is derived a pulse ~rain the fre-quency of which is proportional to the rotation speed~r as derived from the speed detector 9 (Fig. 5). When applied by line 47 to counter 412, the digital value of the revolving angle 9r~ where ~r =lJr~ is obtained, since counter 412 is counting the pulses of the pulse train. An adder 413 is adding the counts of ~so and 0r from lines 35 and 48 so that the digital value of the phase angle 0O = (~r + ~so)t is derived at the output on line 35'. In addition, the pulse train derived on line 32 from the speed detector 9, the frequency of which is proportional to the speed of rotation ~r' is fed to a function ~enerator, or compensating circuit 410 after being converted into an analog value by a frequency to voltage F/V converter 414. Otherwise the operation of the circuit of Fig. 3 is the same as explained for the embodiment shown in Fig. 6. The embodiment shown in Fig. 8 is applicable where the AC motor of Fig. 5 is an induction motor. When the motor is a synchronous motor, the circuit of Fig. 8 is modified in that V/F converter 403, counter 404, and digital adder 413 are eliminated, the output of counter 412 being directly supplied to digital adder 405. Moreover, the compensating circuit 410 receives only the output of F/V converter 414.
Although in the aforementioned embodiments the time wave reference generator circuit 4 is arranged so as to generate reference waveforms required for the instanta-neous AC primary current which are in number less than thenumber of phases of the AC motor l, e.g., by one, the circuit may be arranged in such a way that it generates the reference waveforms in the same number as the number of phases of the AC motor. It should be noted that when the voltage of the voltage source for the inverter (DC power supply 6) varies in amplitude, the transfer function Gc of the current limiting circuit varies accordingly. Then, it is necessary to effect a corresponding variable 13 51,865 compensation in the compensating circuit 410 in relation to the voltage of the DC power supply 6. As explained above, the pulse-width modulation circuit according to the present invention utilizes a reference time wave generator circuit which inherently automatically compensates the amplitude component and the phase component in such reference waveforms as applied for comparison with the instantaneous AC primary current fed back to the pulse-width modulation circuit through the prediction of -the operating characteristic.
Such compensation is based on anticipation made from the actual speed of rotation of the AC motor. Accordingly, the pulse-width modulation circuit will supply current for control which is precisely matching always the reference time waves, for the entire speed range of the AC motor. Special detectors, or control circuits, to effect compensation in the generated reference waveforms are no longer necessary.
(~O ~ ~O - 2/3 II) where aO = (~r + ~ so) t, are derived at the outputs of D/A converters 408, 409.
Of these two time wave reference signals, the ampli-tude component and phase component are compensated as proposed in accordance with the present invention. When compensating circuit 410 is not operating, the compensation coefficient Ks = 1 and compensating phase angle ~ = O. For the two values obtained at the output of compensating circuit 410, the com-pensation coefficient Ks and compensating phase angle ~O can be calculated as explained hereinafter. Assuming first that the AC motor of Fig. 5 is an induction motor, the motor can be represented by its equivalent circuit shown in Fig. 7.
There, Rl and R2 are, respectively, the primary winding re-sistance and secondary winding resistance, whereas ~l and ~2 are, respectively, the primary winding leakage inductance and secondary winding leakage inductance, and M is the mutual inductance between primary and secondary windings. S denotes the slip of the motor for which the equations S =~Js/~o and (~0 = ~Jr + ~s) hold. Since Rl, R2, el, e2, and M are known and the frequency component of the reference signal iso applied (on a single phase basis) to the primary current is is, as shown earlier, equal to ~r + ~Jso~ it follows that ~Jr and ~so are also known. To be exact in the transient state, ~so and ~s o:E Fig. 7 do not coincide, but in the stationary state they will be the same. Therefore, if Gs is the trans-fer function of the primary current is with respect to the phase voltage VS oE an induction motor (formulated for simpli-city on a single phase basis), the following equation holds:
isvs = Gs (l) GS can be calculated si.nce ~Jr iS known when ~s is known being fed externally as a reference signal. The transfer function &c of block 7 in Fig. 4, which is the transfer function per phase of the pulse width modulation circuit 5 of Fig. 3, leads to the following equation after equation (1):
_ = GsGc iso l + GsGc (2) The fraction on the right side can be generally converted to the form GS GC = ¦ G ¦ e 1 + GSGC
where ~G ¦ = ¦ GsGc l l ¦l + GsGc ¦ ( ) From equations (2) and (3) it appears ~hat in order to have iS iso, e j~6iso I G¦
should be used as a new reference value for is. From equation (3), then:
¦G ¦ ej I I e~j~aiSO = iSO
Thus, Ks, the compensation coefficient for the amplitude component of the primary current reference time waveform and the compensation angle ~OO~ for the phase component of the primary current can both be calculated from the following equations:
~.~
11 51,865 Ks = l = ¦l ~ GsGc ¦
IGI ¦ GsGc ~0 = ~0 where ~0 is the phase angle of the -transfer function G.
It appears that since ~so is externally supplied and the parameters of the induction motor such as Rl and R2, and the transfer function per phase of block 7 of the pulse-width modulation circuits of Fig. 3 are known values, a compensating circuit such as 410 shown in Fig. 6 ean be designed whieh responds to a speed detector 9 providing the angular velocity lJr of the AC motor.
If the AC motor l of Fig. 5 is a synchronous motor, the slip angular velocity ~s is zero. Moreover when a synchro-nous motor has a field current which is controlled, or where a permanent magnet is used, the transfer function Gs of the primary eurrent is with respect to the phase voltage VS per phase ean be easily obtained from the parameters of the syn-chronous motor, that is, from its resistance and inductance.
Therefore, like in the case of an induction motor, a compen-sating eireuit is easily designed. For example, one embodi-ment of the reference time wave generator 4 provided with an integral eompensating circuit sueh as 410 can be arranged so that it does not require wsO as an input (like in Fig. 6), thus dispensing with adder 401. Here, the angular veloeity ~r is directly applied by line 31 to the ~/F eonverter 403 and the compensating circuit 410 receives only ~r on line 32.
Fig. 8 shows another embodiment of the time wave referenee generator 4. ~ere, a rotation angle detector, in the form of an encoder, is used instead of the tachometer 9 of Fig. 5. Bloeks 402 through 411 are the same as those in Fig. 6. In addition, a counter 412, a digital adder (ADDER) 413, and a F/V eonverter 414 are provided. The operation of the cireuit of Fig. 8, is as follows: An analog signal exter~
nally supplied and corresponding to the slip angular velocity reference value ~so is eonverted by the V/F eonverter ~03 into '`?~
12 51,865 the V/F converter 403 into a pulse train outputted on line 34. The pulses are counted by counter 404 providing a value resulting from the in-tegration of ~so~ that is, a digital signal representing phase angle 9so~ where so = ~sot. On the other hand, on line 32 is derived a pulse ~rain the fre-quency of which is proportional to the rotation speed~r as derived from the speed detector 9 (Fig. 5). When applied by line 47 to counter 412, the digital value of the revolving angle 9r~ where ~r =lJr~ is obtained, since counter 412 is counting the pulses of the pulse train. An adder 413 is adding the counts of ~so and 0r from lines 35 and 48 so that the digital value of the phase angle 0O = (~r + ~so)t is derived at the output on line 35'. In addition, the pulse train derived on line 32 from the speed detector 9, the frequency of which is proportional to the speed of rotation ~r' is fed to a function ~enerator, or compensating circuit 410 after being converted into an analog value by a frequency to voltage F/V converter 414. Otherwise the operation of the circuit of Fig. 3 is the same as explained for the embodiment shown in Fig. 6. The embodiment shown in Fig. 8 is applicable where the AC motor of Fig. 5 is an induction motor. When the motor is a synchronous motor, the circuit of Fig. 8 is modified in that V/F converter 403, counter 404, and digital adder 413 are eliminated, the output of counter 412 being directly supplied to digital adder 405. Moreover, the compensating circuit 410 receives only the output of F/V converter 414.
Although in the aforementioned embodiments the time wave reference generator circuit 4 is arranged so as to generate reference waveforms required for the instanta-neous AC primary current which are in number less than thenumber of phases of the AC motor l, e.g., by one, the circuit may be arranged in such a way that it generates the reference waveforms in the same number as the number of phases of the AC motor. It should be noted that when the voltage of the voltage source for the inverter (DC power supply 6) varies in amplitude, the transfer function Gc of the current limiting circuit varies accordingly. Then, it is necessary to effect a corresponding variable 13 51,865 compensation in the compensating circuit 410 in relation to the voltage of the DC power supply 6. As explained above, the pulse-width modulation circuit according to the present invention utilizes a reference time wave generator circuit which inherently automatically compensates the amplitude component and the phase component in such reference waveforms as applied for comparison with the instantaneous AC primary current fed back to the pulse-width modulation circuit through the prediction of -the operating characteristic.
Such compensation is based on anticipation made from the actual speed of rotation of the AC motor. Accordingly, the pulse-width modulation circuit will supply current for control which is precisely matching always the reference time waves, for the entire speed range of the AC motor. Special detectors, or control circuits, to effect compensation in the generated reference waveforms are no longer necessary.
Claims (3)
1. In a pulse-width modulated motor drive includ-ing an AC motor, means for sensing the primary currents of said motor to derive corresponding current representative signals, a variable frequency inverter unit for supplying to said AC motor AC current of variable frequency, the combination of:
means for generating timing wave reference signals;
pulse-width modulation generator means responsive to said timing wave reference signals and to said current representative signals for controlling said inverter unit;
means for deriving a signal representative of motor speed; and compensating circuit means including a function generator responsive to a signal representative of the fre-quency of said timing wave reference signals and to a signal representative of motor speed for generating a first and a second corrective signal, said first corrective signal being applied to correct the amplitude of said timing wave reference signals and said second corrective signal being applied to correct the phase of said timing wave reference signals.
means for generating timing wave reference signals;
pulse-width modulation generator means responsive to said timing wave reference signals and to said current representative signals for controlling said inverter unit;
means for deriving a signal representative of motor speed; and compensating circuit means including a function generator responsive to a signal representative of the fre-quency of said timing wave reference signals and to a signal representative of motor speed for generating a first and a second corrective signal, said first corrective signal being applied to correct the amplitude of said timing wave reference signals and said second corrective signal being applied to correct the phase of said timing wave reference signals.
2. The motor drive of claim 1 with said AC motor being an induction motor, said compensating circuit means being also responsive to a signal representative of the slip frequency of said motor.
3. The motor drive of claim 1 with said timing wave generating means including ROM generator means for generating said time wave reference signals in response to a count representative of the sum of the frequency of said time wave reference signals and the speed of said motor, said second corrective signal adding a corrective count to said sum representative count, and said first corrective signal being applied to correct the magnitude of the time wave reference signals derived from said ROM generator means.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000459482A CA1218695A (en) | 1984-07-23 | 1984-07-23 | Pulse-width modulated ac motor drive |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000459482A CA1218695A (en) | 1984-07-23 | 1984-07-23 | Pulse-width modulated ac motor drive |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1218695A true CA1218695A (en) | 1987-03-03 |
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ID=4128368
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000459482A Expired CA1218695A (en) | 1984-07-23 | 1984-07-23 | Pulse-width modulated ac motor drive |
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Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110333444A (en) * | 2019-07-11 | 2019-10-15 | 中国石油大学(华东) | A kind of temporary drop critical clearing time calculation method based on name plate of motor parameter |
-
1984
- 1984-07-23 CA CA000459482A patent/CA1218695A/en not_active Expired
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110333444A (en) * | 2019-07-11 | 2019-10-15 | 中国石油大学(华东) | A kind of temporary drop critical clearing time calculation method based on name plate of motor parameter |
| CN110333444B (en) * | 2019-07-11 | 2023-10-20 | 中国石油大学(华东) | Temporary boundary excision time calculation method based on motor nameplate parameters |
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