CA1190970A - Dual frequency anti-theft system - Google Patents

Abstract

ABSTRACT OF THE DISCLOSURE

An article surveillance system employs a label or tag containing a non linear impedance element, such as a semiconductor diode, connected to a metal antenna loop configured to pick up two distinct radio frequency transmissions displaced on either side of a selected center frequency. The non-linear impedance element connects opposing sides of a closed loop section at one end of the antenna to form a tuned tank circuit having a resonant frequency twice that of the selected center frequency. A first transmitter generates a tone modulated radio frequency displaced on one side of the center frequency, and a second transmitter generates a continuous wave radio frequency displaced from the center frequency on the other side. Both transmitter signals are fed separately to respective orthogonally disposed dipole radiating antenna strips located on opposite sides of a surveillance area with the dipole strips for the different frequencies being disposed at right angles on each side while those for the same frequency are also at right angles to one another on opposite sides, thus producing cross polarized transmission of both frequencies within the surveillance area. The two different frequencies picked up by the transponder antenna are mixed by the non-linear impedance causing the tank circuit to resonate at a single higher frequency equal to their sum, which is double the center frequency; that resonant frequency is reradiated to be picked up by receiver antennas on each side to be detected by a very narrow band receiver responsive to the sum frequency. The modulating tone signal is derived from the received signal to produce a gradually increasing charge that is compared against a preselected threshhold level to trigger an alarm for a fixed interval only when the detected signal is of a sufficient strength and duration.

Description

11309~0 21 l. Field of the Invention 31 This invention relates generally to electronic 41 article surveillance systems and more particularly, to an article surveillance system that involves the trans-6 mission of two distinct radio frequency signals, one of which 7 is tone modulated, that are picked up by a transponder and 8 mixed through a non-linear impedance to be reradiated at a 9 ¦ higher frequency equal to their sum, which is detected by a narrow band receiver.

12 2. Prior Art 13 Earlier surveillance systems of this type, such as l~ that described in U.S. Patent No. 4,063,229 to Welsh et al, operate to transmit a single radio frequency to be picked up 16 by an antenna on a transponder tag or label where a non-linear 17 impedance, such as a semiconductor diode, generates a selected l~ harrnonic of the transmitted slgnal that is reradiated for 19 detection b~ a receiver circuit to the exclus,ion of the

2~ t~an~mitted frecJucnc~. Elowever ~;uch s~stems proved un-2l sa~i~factory in practice frorn the stan-3point of lacking 22 the sensitivity to reliably detect the presence of a transponder 23 within the surveillance area and of producing false alarms in 24 response to various other conditions.
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1l Signiicantly, the non-linear characteristics 2l inherent in the transrnitter circuitry and elements often

3 resulted in harmonics being transmitted along with the

4 fundamental transmission frequency causing the receiver to respond without the presence of a non-linear impedance 6 element in the transponder. If receiver sensitivity has to be 7 reduced to ignore such directly transmitted harmonics, 8 then lower energy harmonics reradiated by the transponder 9 element under some circamstances might be masked. Although this problem can be minimized by proper shielding and RF
11 filtering in both the transmitted and receiverr the filters 12 would have to be provided with extremely sharp cutoff character-13 istics so that even a small frequency drift in the transmitted 1~ signal, which is multiplied in the harmonic, could easily result in the reradiated frequency heing outside of the filter 16 pass band of the receiver. Frequency shifts may also result 17 from the Doppler effect produced in moving the transponder 18 rapidly within the surveillance area thus aggravating the 19 effect of transmitter driEt. ~i ~0 2~ ~n the other hand, such hi.gtl requenc~ signals could ~2 readil~ propagclte outside oE the intended surveillance 23 area to cause false triggering of the alarm by a remote 2~ transponder. As a result~ protected articles often could not / / /

,' 11 be located or handled anywhere in the vicinity of the sur-2 veillance area. Even then, the high frequency energy might 3 propagate by unpredictable reflections, or even along plùmbing 4 pipes or power conduits acting as wave guides, to and from remote locations within the protected structure to produce 6 ~ false iggering of the alarm system.

8 Such systems were also susceptible to false triggering 9 by metal objects such as umbrellas, baby carriages and shopping 10 ¦ carts, where a weld or contact point between dissimilar metals 11 I produces a non-linear impedance diode effect to generate and 12 ~ reradiate a harmonic of the transmitted signal. Or the 13 ~ receiver could respond to spurious radio frequency noise 1~ ¦ from other sources such as motor ignition systems and electronic 15 ¦ equipment.

17 ~ Conversely, the syste1n micJht not respond ko the 1~ ¦ actual pr-sence o~ a t~ansponder element within the survei:llance 1'3 ¦ 5Irea if ~ he ener~y pic~.ecl IJp and reradiatcd as a harmonic were 2~) ¦ in~u~fi(:ient. r~c)r exalnple, this could occur i the transponder 21 antenna were improperly oriented with respect to the polari-22 zation of the transmitted field or if the antenna were to be 23 electromagnetically shielded frorn the transmitter by the human 24 body or a metallic surface. Also, proximity of the transponder to the human body can detune the resonant tank circuit, thus 26l dissipating the harmonic energy available for reradiation 3.

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1l to the receiver. Moreover, although signal tracking circuitry 21 can be incorporated to adjust the frequency response of the 3 receiver to compensate for transmitter frequency drifts, transponder efficiency suffers badly whenever the tuned tank circuit is forced to oscillate at frequencies other than 6 its normal resonant frequency.

Later efforts to resolve the problems of such I earlier systems have resulted in several variations. In one of 10 ¦ these, which is described in U.S. Patent No. 3,631,484 to 11 Augenblick, the single radio frequency transmitted to the 1~ transponder to be reradiated as a harmonic is compared with 13 ¦ signals picked up by the receiver to detect Doppler effect la frequency shifts caused by movement of the transponder.

Although this system eliminated problems associated with 16 ¦ transmitter frequency drift and false alarms from stationary 17 transponders r.earby, an article moved slowly through the 18 surveillance area would not produce a Do~pler freq~ency shift 19 sufficient to trlyyer the alarrn~

2~ ~t~emp~:~3 ~ere also made to investigate systems 22¦ ~"herein khe non~linear impedance element in the transponder 23l operated as a signal mixer to generate sum and difference 2~ frequencies in response to two transmitted signals of different Z~ ¦ frequencies, as pointed out in the background discussion of 1 4.
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l U.S. Patent No. 3,895,368 to Gordon et al. Ho~ever, such dual 2' frequency mixer systems were considered to have many practical 3! shortcomings, which included the problem of confining higher 4 frequency transmissions to the intended surveillance area. To overcome this problem, the Gordon et al patent describes use 6 of a dual field system employing a high frequency electro-~ magnetic field in conjunction with a high power, low frequency 8 electrostatic field established between discontinuous conductors 9 disposed on opposite sides of the surveillance space. The non-linear impedance element subjected to these two fields ll operates as a mixer to produce sum and differenee frequencies 12 that are reradiated to the receiver for detection. However, 13 I the power required to establish the required electrostatic 1~ ¦ field within the surveillance area is significant, and such low freqéncy electrostatic fields can be effectively shielded 16 from the transponder by the human body or by a surrounding 17 conductor and diverted from the transponder through the l~ metallie structure o~ a shopping cart or the like. ~lso 19¦ the lo~l freclucncy electrostatic ~ield could readily be cliverted 2~1 throlJ~Jn nrarbt/ pipe-; and other metal structures to remote 21 locations t:o cause false trlggering b~ tags far outside the 2~ surveillance area, and the problem of false alarms due to 23 dissimilar metal junctions in metal carts and the like was 2~1 aggravated by concentration of the electrostatic field through 251 such metal structures.
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1 SUMMI~RY OEi' T~lE INVENTION
The present invention provides an article surveillance 3i system wherein a non-linear impedance element, such as a 41 semiconductor diode, is connected to a metal antenna within a removable label or tag attached to a garment or other item of 6 merchandise. The antenna is preferably in the form of 7 a folded dipole with the diode connected between opposite 8 sides of a closed loop section at one end to provide a tuned c3 ! tank circuit with a resonant frequency double that of a selected center frequency. The longer antenna section extendins 11 ¦ beyond the diode closely approximates a quarter wavelength at 12 i the selected center frequency, which for example may be 915 13 I megaHertz. Resonant frequency of the tank circuit, which is 1~ I determined by the capacitance of the diode and the inductance of the adjacent closed loop section of the antenna, is double 16 that of the selected middle frequency (e.g., 1830 megaHertz).
17 l 18 I T~o different radio frequency signals are hoth 19 ¦ transmitted from dipole radiatiny antennas disposed on the opposike sides o a surveiLlanccl area. One o~ the sigllals is ,?J. ¦ geneJ.,~ rls a conl:inuou3 ~/ave rOln a highJy stable crystal 22 ¦ o~-~cil:l.cltor sourc(- at a ixed frequency (e.y., 905 mec1aMert~) 231 which is displaced from the selected center frequency by 2~1 approxiJnately 1~. The other signal being transmitted is tone modulated, preferably with an audio signal in the range of 1 26 ~ / / /
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lj to 20 kiloilertz, to procluce a radio frequency deviation 2 o~ plus and minus 5 ~.iloHertz in the carrier, which is also 3 c3erived from a highly stable crystal oscillator source at a 41 frequency (e.g., 925 megaHertz) which is equally displaced

5~ from the selected center frequency on the opposite side, so 61 that the mean center frequency of the two signals equals the 7 selected center frequency. Both transmitter signals are 8 radiated across the surveillance area from dipole antenna 9 segments oriented at right angles to one another on the same lO¦¦ sides,and with the respective dipole segment for radiating the 11 same frequency from opposite sides also being oriented at 12 right angles to one another. This results in cross polarization 13 in the surveillance area of the two radio frequencies being 1~ transmitted from opposite sides, to insure that radiation of .
both frequencies in the surveillance area between the trans-16 mitters is adequate in all directions to accommodate any 17 orientation of the tag, whereas propagation of both signals 18 ¦ from the antennas on only one side to the same remote locations 19 outside the surveillance areLI is minimi.zed becau~e of their dif~c3rerlt pcJl.lrizalions~ On tile ot:her hand, aucllo modulclt.ion s21. ol- on(3 ve Ih-: ra(lio fre~{uencies avc3.i.ds creation of stanc]ing ?,~ I~/s~sVe pattetrls that can resu.lt in blind spots within the 23 ¦ surveillance area and false triggering of the system by ta~s 2~¦ outside the intended area.
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11909'~0 ll Siqnificantly, the dual frequency operation reduces 21 the effect of transmitter frequency drift and increases the 3l system bandwidth in regard to transponder efficiency in 4 reradiating the incident radio frequency signals. In particular, the frequency to which the transponder antenna is tuned may

6 fall anywhere between the two transmitted frequencies without

7 significantly reducing transponder efficiency, '~hus eliminating any need for precise antenna dimensioning and minimizing 9 problems with "body detuning" whereby the normal tuning point of the transponder is shifted downwardly in frequency due ll ¦ to the dielectric loading effect of a human body in contact 12 ¦ with or in close proximity to the tag. For example, if the 13 ¦ transponder antenna is detuned down from the selected center 14 I frequency, this merely increases the transponder efficiency l~ ~ relative to the lower transmitted frequency, and the overall 16 mixer action is not seriously affected since proper mixing 17 occurs with radio frequency power ratios of ten to one or even 18 greater. Similarl~, the ef~ects of transrnitter freq~lency 19 ¦ ~rift afe ~niniml~ed in th..lt a shift ;n one of the transrnitters ¦ L-~ not Multi~)liccl a~1 with rerac3iated harmonLcs in the single ~1 ~recluerlc~ s~stelns~ and any drift ln one can be offset by an 22 opposite shift in the other transmitter 24 ¦ The strength and frequency stability of the z51 reradiated transponder signal, and the improbability of , ~ . I

V9~0 l! triggering a false response Erorn transponders outside the 21 surveillance area permits maximum receiver sensitivity and 3i minimum receiver bandwidth. Signals received from circularly a polarized receiver antennas on either side are applied through a very narrow bandpass filter that rejects the transmitter 6 frequencies and then amplified so that the modulating tone can 7 be derived using mostly conventional demodulation techniques.
~ ~referably, the audio tone (e.g., 2 kiloElertz) is used to 9 frequency modulate the radio frequency carrier so that the 10¦ filtered and amplified signal from the receiver antenna can be 11¦ applied to a passive double balance mixer that receives a 12 lowerside injection signal (e.g., 1808.600 megaHertz) generated 13 ¦ by a stable local oscillator source to provide a suitable 1~¦ intermediate frequency (e.g., 21.4 megaHertz) at the mixer output. This intermediate frequency output from the mixer is lG ¦ amplified and applied to another precision filter with a 17 ¦ narrow passband (e;g., 30 kiloHertz) that defines the pre-18¦ detection band~idth. Detection of the modulatiny tone is then 19 accomplish(?d througtl the o~e~ation oE a narrowband (e.g., 30 2~ I kil~llqrt~) c~stal diA~;crirllination, the output of which is 2~. ¦ clarnped to gro~ d ~nti] its input i5 of suEficient strength to 22 generate an automatic gain control detector voltage that 23 exceeds a preselected reference level which is adjusted to set 241 the system sensitivity. With the clamp open, the tone is 2~1 applied to a phase locked loop tone decoder circuit whose 2~1 ///
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11909'~0 1 voltaye controlled oscillator has a free-runniny frequency 21 equal to that of the tone and is capable of acquiring any 31 steady tone within a narrow frequency range (e.g., plus or 41 minus lO percent). When the loop acquires the tone signal, a 51 quadrature detector senses the phase locked condition and 6 produces a direct current output voltage to drive an operational 7 amplifier with a capacitive feedback that sustains an output

8 signal to trigger an al-arm for some minimum time period (e.g., 91 3 seconds), no matter how brief the duration of the detected 10¦ tone. sy this means, the alarm is actuated no matter how 11¦¦ briefly the transponder remains within the surveillance area 12 ¦ once the detected signal is of sufficient strength and has the 13 ¦ proper modulated frequency content. This eliminates false 1~ alarms by weak return signals from transponders outside of the surveillance area and by signals from extraneous sources that 16 ¦ may coincidentally produce signals corresponding to the 17 ¦ reradiated frequency, but that lack the requirecl tone moclLIlatioll.

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2~. ~ G, l is a schematic block diagram of the basic 22 circuit elernents and a partial perspective showing the antenna 23 ¦ placement ~or an article surveillance system in accordance 24 ¦ ~Iith the invention;

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1~ FIG. 2 is a more detailed schematic illustratin¢,^,t the 2 ! cross polarized orientation of the transmitter antenna segments 3l with a perspective view of the operative antenna and non-linear 41 impedance elements of the transponder;

6 FIG. 3 is a more detailed block and circuit diagram 7 schematic illustrating a preferred form of the narro,~ band 8 tone modulated RF transmitter of FIG. l;

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FIG. 4 is a detailed block and circuit diagram 11 ¦ showing the preferred form of a continuous wave RF transmitter 12 ¦ of FIG. l;

1~ FIG. 5 is a block and circuit diagram illustrating 1~ I a preferred form of th¢-? linear amplifiers sho~n in FIG. l;
16¦ and, 17 l 1~ FIG. 6 i'; a detailed block and circult di.ac3ram .1~ I i.llu~t:ratirl(J Ihc l,~re~E,.Irr~.?d form c~f. the narrow banc] tone X() ¦ rnodulatc~d rec,~eiv¢-~r ol Li'IG. 1 ~hereirl the transmitted siynal. is 2~.1 ErecJuency modulated.
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1, DETI~II,E~D DESCRIPTION
21 .' 3 Referring now to FIG. 1, ~,lhich illusrates an article 4 surveillance system in accordance with the invention, appropriate transmitter and receiver antenna arrays are mounted in corre 61 sponding locations on free standing pedestals 10 and 12, or if 71 preferred on or within existing door frames on either side of 81 a surveillance area, typically at the entrance or exit 9¦ to a retail establishment, so that anyone entering or leaving 10l must traverse the space between them. Although shown slightly 11 askew in FIG. 1 for illustration purposes, the respective 12 antenna arrays on either side normally directly face one 13 another with the respective antenna elements disposed in 14 parallel vertical planes. The transmitter antenna arrays 14 and 16, as best seen in FIG. 2, both consist of orthogonally 16 ¦ disposed pairs of metal strip segrnents 18, 19, 20 and 21 17 ¦ mounted on a vertical planar backincJ on either side of the 1~ ¦ protected access or other area. rach strip extencls outwclrd 19 ~rom a cç-ntral ~lub area with irlc]ivi(1ual pair~; bein(J aligned 3idc l~ orln a convent:ional center ed dipole radiating 21¦ ant~rlna th;lt is approximately one-quarter wavelength long for 22j¦ the frequenc~ being transmitted, and may conveniently be 23 I oriented as sho"n to extend horizontally and vertically. The 24 individual strips 18-21 may be cut from conventional copper ~,5 I clad, adhesive backed tape of the type cornmonly used in 26 ~ 12.
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~ 70 11 printed circuit boards and applied to a non-conductive di-electric ~acking with suitable low loss characteristics on the 31 peqestal or door frame, or the four strip-array can simply be 4 etched out by removing the surrounding conductive surface on a printed circuit board. A conductive metal panel or a small 6 mesh grid (not shown) can be located behind and parallel to 7 the plane of the antenna strips 18 to 21 to reflect and thus 8 concentrate the transmitted signal energy and radiation 9 pattern inwardly across the protected space for greater efficiency and to inhibit radiation of the signals from 11 the opposite side to areas behind the pedestals 10 and 12. In 12 the preferred form of the system,, the copper clad tape strips 13 ¦ are applied to the surface of a G-10 fiberglass panel that is 1~¦ affixed by adhesive within a lightweight anodized aluminum 15¦ frame that covers the entire back surface of the pedestal 10 lG or 12 and structurally supports the antenna mountings and 17 associated circ~it elements.
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~r3 ~Iso rnourltr-~d on each side are receiver antenncls 22.
2(~ arld 2~ ~h,lt "~e cirularl~ r~olarizrd, such as the crossed ~1 ~olded dlpole con~ig~ration comrnonly known as a "turnstile"
22 antenna or a helical antenna. The length of each receiver 2~ dipole segment should be a quarter wavelength of the freyuency 24 reradiated signal which, as hereinafter explained, is equal to 2~ the s~m of the two transmitted frequencies.

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11~0970 1l Two distinct radio frequency siqnals fl and f2 2 ! are generated to be radiated from the respective dipole strip 3 segments 18, 19, 20 and 21 that form the transmitter antenna 4l arrays 14 and 16. The f1 signal is a narrow band modulated 5l radio frequency generated from a highly stable oscillator 6 source 26 that is coupled to the vertical dipole strip 7 segments 18 of the transmitter antenna array 14 on one side ~¦ and also through a linear amplifier 28 to the opposing 9' horizontal strip segments 21 of the transmitter array 16 on lO ¦ the other side of the surveillance area. The other transmitter 11 ! signal f2 is similarly generated at a fixed radio frequency 12 by a highly stable oscillator source 30 that is to the 13 horizontal strip segments 19 of the transmitter antenna array .~¦ 14 on one side, and on the other side throu~h a linear 15¦ amplifier 32 to the oppositely disposed vertical strip 16¦ segments 20 in the transmitter antenna array 16. Preferably lr~l both oscillator sources 26 and 30 employ respective ternperature-13 I compensatec], crystal o.sci:L:I.ators having cascaded ~recluency 1~ mu]~ipli(lr arld r~ rro~ ~?~ls-; t~and filters Eor c~enera~:irlg ~() j the contirnlous ~ave f2 ar)d the radio frec3uency carrier Eor 21 ¦ the tone modulated sicJnal fl, as rnore fully described 2211 hereinafter in connection with FIGS. 3 and 4.
~31 241 Generally, the distance between the metal strip 25 il antenna se~Jrnents 18-21 and the acljacent reflective surface of ~61~ the conducti~/e panel or grid behincd it, which depends on the 14.
1! 1 iU9~0 l thic~ness of the low loss dielectrie baeking, is selected to .
2 produce a low voltage standing wave ratio (VSWR) to match the 3 antenna input impedance with the output impedance of the a respective transmitter signal source at the transmitted 5 1 frequency so as to provide an effective radiation pattern with 6 ¦ an approximate 60 degree beam width extending outward from the 7 transmitter antenna arrays 14 and 16 on each side.

911 Both radio frequencies f1 and f2 are thus lo1 radiated from transmitter arrays 14 and 16 on opposite sides ll ¦ and with opposite polarizations to intersect and impinge from 12 ¦ both sides upon a transponder 34 located in the surveillance 13 area between the two pedestals lO and 12. The transponder 34 l~ is shown schematically in FIG. l as a circularly polarized helical antenna loop with a diode 36 connected across a short 16 ¦ elosed section of the loop. ~-]owever, as shown in more detail 17 ¦ in FIG. 2, the preferred form of the transc1ucer 34 cons.lsts oE
¦ an elonqlted 1at mekal antenrla 33 .loop with a central ~Jap on ~ On~ ~iclf that prot/i.de-; a EoLded cl:iTJole conEic.3uration. Ihe 2-) n~/era.l 1. ant-'lllla J,('nCJth i'3 i.dc-ally a quarter wavelength oE the 2~ ~nean centc~r frequency between the two transmitted radio 22 frequencies f~ and f2. The non-linear impedance element 23 36, which takes the form of a semiconductor diode, is connected 24 between opposite sides of the loop near one end about midway ~51 frorn the side cJap so that the capacitance of the diode 36 with 26 15.

l the inductance of the adjacent closed end of the conductive 2l loop form a tank circuit with a resonant frequency eq~al to or 3! approximating the sum of the two transmitter freguencies 4 and f2 or, in other words, a resonant frequency twice that 5j of the selected mean center frequency or the transmitter 61 signals. Precise placement of the diode 36 on the antenna 71 loop 38 to produce the desired resonant frequency for the 8l tank circuit is not crucial and for the most part is determined 9¦ empirically based on the capacitance of the selected diode and o¦¦ the conductive properties of the antenna loop. In operation, ~ the short straight metal segment on the diode side of the 12~, gap serves as a quarter wave dipole radiating antenna at the 13 resonant frequency of the tank circuit.
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15 . ~aximum transponder efficiency and selectivity 16 ~ is achieved where the frequency difference between the two l7¦! transmitter signals f1 and f2 i5 somewhere around two l8¦1 percent of their rnean center frequency. In the current l91l version o~ the sy5tem, the frequency of the continuou.s wave 2011 ~ignal f2 gerlerated by the source 30 is chosen at 905 2.~1¦ rnegaHertz, ~/hereas the frequency of the tone modulated 221, carrier for the other transmitted signal fl from the source 23 1 26 is at 925 megaHertz. Thus their mean center frequency is 2~ 1 915 megaHertz, and the resonant tank circuit frequency is ~5 ~
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1 l,830 mega~lertz. These particular frequencies are selected to 2' fall within the available spectrum transmission bands available 3 for such purposes in the ~nited States. On the other hand, to 4l comply with international broadcast standards, it is contem-5l plated that the system would for example be designed to have a 6 resonant tank circuit frequency of about 4,900 megaHertz with 7 transmitter frequencies of around 2,420 and 2,480 megaHertz~

9¦1 In operation, when both transmitted signals fl and lOj f2 are received by the transponder antenna loop 38, they are 11 ¦ mixed through the non-linear impedance effect of the semi-12 !- conductor diode 36 to initiate tank circuit oscillation at its 13 ¦ resonant frequency, which is equal to the sum of the f1 and 14 f2 frequencies. Increased mixing and overall transponder 1~ efficiency is enhanced through use of a planar diode exhibiting 16 high-speed switchiny, low RF threshhold and low forward bias.
17 Significantly, lower-priced germanium diodes are preferred lc~ ¦ because of their relatively low threshold oE about 0.3 volts, 19 I as compare(l to hicJher-p-riced silicon diodes with thresholds o 20 1 0.6 vol~
Zl l 22 The approximate two percent frequencv separation 231 between the transmitted signals provides important advantages 241 in maximizing transponder efficiency and in the ability of the ~51 system to avoid false alarms because the transponder return ~6 17.
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1l signal "stanclc out" ~rom that might be produced by dissimilar 2l metal objects such as umbrellas, shopping carts and the like, 3 which have tended to cause false alarms with previous systems.
4 In particular, the bandwidth of the transponder 34 relative to the incident radio frequencies is broadened without reducing 6 its efficiency because the receiver antenna 38 can be tuned to 7 fall anywhere between the two transmitter frequencies, which 8 also minimizes the effects of "body detuning" in that the 9 ~ downward shift in frequency due to such dielectric loading effects can easily be accommodated within this range. This llj results from the fact that tuning or detuning of the antenna 12¦ 38 more toward one transmitter frequency than the other only 13 ¦ serves to enhance the signal strength at that frequency 14 I without reducing mixer conversion efficiency because proper 15 I radio frequency mixing can occur with power ratios of ten to 16 ¦ one or greater between the signals~
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1~ ~oreover, because of the cross polariæation of the 1,~1 two ~requencies transmitted from each o~ the antennas 1~ and ,~0 ¦ lG, t~leir propa~3ation ~roln one translnitter location to remote 2:1 I location$ ou~si~le of the ,s~rveillance area is se.ldo~ the 22 I same or hoth .si~3nals. A freak re1ection pattérn that may 23 ¦ result in one transrnitted signal being concentrated on a 24 ¦ transponder at a rernote location will almost never result in 2~ I / / /
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1190~3'70 1¦ the other oppositely polarized transmission being reflected in 2 the same pattern to reach the same area with sufficient 3 power. Consequently, if only one signal is received, the non-linear impedance of the diode 36 can produce only a ~ I frequency-doubling effect, instead of the necessary mixing 6 ~ effect, so that the resulting return signal is at a frec~uency 7 I widely displaced from that of the desired transponder return.
8¦ For example, with the current system parameters, a transponder 9 would produce doubling frequencies of 1,810 or 1,850 megaHertz, both displaced by a full 20 me.gaHertz from the normal return 11 frequency at 1,830 meqaHertz. These displaced frequencies 12 ¦ would be subject to considerable attenuation in the tuned tank 13 I circuit and is readily distinguishable by conventional filtering 1~ techniques from a legitimate rnixed frequency response at.1,830 151 megallertz level.
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~71 In this regard, signals picked up by the receiver 1~ antenna 22 and 2~ on either side are app.l.iecl throucJh ~ con-~¦ ~Jerlt:i.oncl.l. mi~c~r ~ )n/lec~: ion ~0 to a narrow band tone modulat:ed,'O r-ceL~/er ~. 'l'he rni~i.ng ot. the~ two transMitted signals in the 2J.¦ transponder return signal permits the response of the receiver 22¦ 42 to be restricted to ver~ narrowband operation that serves 23 ¦¦ to eliminate false alarm responses due to extraneous noise and 24II transmission signals from other sources. Indeed the receiver 25 j band-,tidth needed is for the most part dependent onl~ upon the 2G i 119V9"~0 1ll frequency stability of the transmitter sources 26 and 30, thus 21¦ permitting a very narrow detection "window" corresponding to 3 the possible transmitter frequency drift. With very stable 41 transmitter oscillator sources as hereinafter described, the bandwidth of the received siynals available for detection of 61 the modulating tone (i.e., the predetection bandwidth) can be 71 extremely narrow, and the bandwidth of the receiver (post 8 il detection) can be further narrowed in precise detection of the 9l~ modulating tone. Moreover, system reliability and sensitivity

10 I is further enhanced by having the receiver 42 supply an output

11 , signal to actuate an alarm 44 only when the strength of

12 ¦ the rr.odulating tone signal detected exceeds a selected

13¦! minimum amplitude level -for a predetermined fixed interval to ~ insure the actual presence of a transponder within the 15¦' detection ~one.

1~ I ReerrincJ now to ~ J. 3, the preferrecl embodirtlent 1~, ~ now in ope~atioll (Jr?t)f;~rates l:t)e trarlr;lnitter signal fl as a .1.') I ver~ rikc-lhle, na~rol,/J~.lncl frequency rnodlllated signal to rnaximize ~li ri~/r;tcrn r;-~n-;itivity and selectivity. A stable tone generator 46 2111 0~ corlventional desiyn, which may be a simple RC type, generates ~2111 a fi~ed fre4uency tone in the audio range of one to twenty 2311 kilol3ertz. 'l'his tone, which in the current system is at 2 24 ¦ kilvl3ertz, is applied as a modulating signal to a ~oltage ~5 I controlled crystal oscillator 48 to frequency modulate its 25 ~

1 output. In the preLorrsd e~ùodimerlt the crystaI oscillator 2I 48 is of ~onventional design with precise temperature com-3 pensation capable of holding a frequency stability of 0.7 4~ cycles per million from 5 C to 45 C at a fret~uency of approxi-5~ mately 51.4 megaHertz. The amplitude of the modulating signal 6~ from the tone generator 46 applied to the voltage control 7I circuit is adjusted to produce a maximum Erequency deviation 8 1 of plus or minus only about 0.25 to 0.30 kiloHertz thus g resulting in only very narrowband modulation of the oscillator lOI¦ carrier. The modulated outpu~ of the oscillator 48 is then 11 I applied to a conventional frequency multiplier 50 which 12 I triples the oscillator frequency-that is then applied to a 13 ¦ narrowband two pole bandpass filter 52. This filtered multi-

14 plier signal is then applied to another conventional frequency multiplier 54 which again triples the available frequency to 16 be applied to another narrowband pass filter 56. The Eiltered 17 output Erom the bandpass filter 56 is then applied to yet lZ I another Erequency multiplier 58 that this t:irne only doub:le.s ~,J~¦ the :input recIu--ncy to prodl~c~? the d~sired rnodulclted output ?0I; ~;kJrIaJ (fI) at '325 rne-Ia~lertæ with a narrowband rnodulation 21¦1 deviation o~ plus or Minus 5 kilo~ertz, which is then applied 22 I to a variable gain RE arnplifier 60 and power amplifier 62.
23 I This arnplifier transmitter signal fl is passed through a 24 ¦ narrowband three pole bandpass f ilter 64 to a power divider 66 25 I that delivers the transmitter signal to the vertical antenna 21.

l1 strips 18 on the transmitter array l4 of the pedestal lO, and 21 also through a lightweight cable connector to the linear 3, amplifier 28 on the other pedestal 12.

- 5 Referring now to FIG. 4, the other transmitter 6 frequency f2 is generated in a similar fashion usin~ a q¦¦ conventional temperature compensated, crystal oscillator 68 ~11 that is capable of holding the frequency to 0.5 parts per 9 ¦ million from 5 C to 45 C with an output frequency of about lO ¦ 50.3 megaHertz. This output frequency is tripled by frequency ll 1 multiplier 70 to be filtered by a two pole bandpass fi.lter 72.
12 ¦ The narrowband output from the filter 72 is then applied to 13 ¦ another frequency multiplier 74 which again triples the l~ ¦ frequency to be applied through another two pole bandpass l51 filter 76, and the filtered output frequency is then doubled l61¦ in a final frequency multiplier 78 to produce the desired f2 17 1 signal at 905 rneya1~ertz. The ~ siynal is appl.ied to the l~, ! input oE an 1~1` variab:1.e gain ampl.ifier ~0 and the furth~r 19 1 ampli.1..i.--r ;t~ C 8~ to reach a desircd tran~mittir'cJ power 20 1 I.c~el. rhc a1npli~ied output i5 then filtered throuyh a 21 ~ narrowband, three po:Le bandpass filter 84 to rernove any ~2~1 amplified distortions or harmonics and apply it to a power 231¦ divider 86 to be applied directly to the antenna strips l9 and 24¦¦ the transmitter array 14 on the pedestal lO and through an 251¦ app~opriat~ i coupliny to the respective linear amplifier 32 2~1 / / / 22.

9()~7~

lil on the opposite pedestal 12. Because of the great eficiency 2 and sensit-ivity achieved, the transmitted power of these 3 signals is an order of magnitude below that required in 4 earlier systems, thus negating any health concerns about possible tissue damage from microwave transmissions.

t7 ReEerring to FIG. 5, the respective f1 and f2 8 signal outputs from the power divider 66 or 86 can be connected 9l to the respective linear amplifiers 28 and 32 on the opposite lol¦ antenna pedestal 12 by simple wire leads or lightweight ll ¦ cable, thus eliminating the need for the expensive and diffi-12 ¦ cult installation of heavy and bulky RF cable connections 13 ¦ requ~red in previous systems to avoid power loss. Linear l~ ¦ amplifiers 28 and 32 each simply consist of a variable radio

15 ¦ frequency amplifier stage 88, the output of which is applied

16 I through a narro~band three pole bandpass filter 90 to remove

17 ¦ any signal distortion or noise picked up on the connecting l8 I line or generated in the amplificatior1 process. rl'he yain o l-J the amr?lificr sl:a{J~ ~B is ad~uGted to rest4Ye th~ transTnitter 20 I ~ignal n~retlyth to approxirnatel~ the sarne level being supplied 2]. to the transrnitter antenna segments on the opposite side.

23 ¦ Referring now to Fig. 6, in the preferred embodiment 241 employing narrow band frequency modulation of the fl trans ?,5 mitter signal, the signals picked up by the receiver antennas 26 / / / 23.
, , I

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1~ 22 and 24 are applied through the mixer 40 to a very narro~, 2l band, four-pole band pass filter 92, the passband being 3, centered at the mean frequency of the mixed transponder return 4~ signal - for example, at 1830 megaHertz. In the particular 51 system being described, a valid return signal from the trans-6 ponder 34 is frequency modulated with a single fixed audio 7 tone, preferably at 2 kiloHertz to provide a maximum deviation 8 of only 5 kiloHertz on either side of the 1830 megaHertz 9 carrier frequency. The band pass filter is designed to reject the lower frequency transmitter signals by a minimum of 60db 11 I to prevent internal mixing due to circuit nonlinearities. A
12 filtered output from the bandpass filter 92 is applied to a 13 I double balanced mixer 94 to be mixed with lower side injection 14 frequency f3 at 1808.600 megaHertz, for example, from a stable local oscillator source to produce an intermediate 16 frequency (I~) output of 21.4 megaHertz at its oukput when a 17 ¦ valid transponder return signal is present. This lower ~¦ side injection erequency is likewise generated erom a highly .ltJ r;table, t~m~ rat:llre comperls;lt-!-l crystal oscillator 96 opercltirlcJ~ ¦ Jt at)out 5().2~ Inttla~lert.~. 'L'his oscillator erequency is 21 ¦ initially quadrupled in a frequency multiplier 98 and applied Z~ successi~ely throuyh two tripling frequency multipliers 100 23 and 102 to a four-pole narrow band pass filter 104 to supply 24 the lower side injection signal to the mixer 94.
~,5 / / /
26, / / /
1 24.

~ ~î9~19t~3 1 The intermediate frequency output of the balanced 2 mixer 94 is applied to a low noise amplifier 106 to establish 3 the overali receiver noise figure at 12db to be fed into a al four-section monolithic crystal band pass filter 108, preferably 51 the Model 1619-1622 produced by Piezo Technology, Inc. under 61 its registered trademark "CO~ILINE", wherein the response of 7, amplitude versus frequency is 30 kiloHertz at the -3 db 81 points. The crystal band pass filter 108 effectively deter-9 mines the predetection band width, and along with the 12db noise figure and modulation iridex of five, provides an 11 I overall receiver sensitivity of -113 dbm for a 20db S+~/N
12 I ratio at the output of a crystal discriminator 110 described 13 in more detail hereinafter. The output from the crystal band 14 ! pass filter 108 passes through successive RF amplifier stages 15 1 112 and 114, each of which is provided on a chip with automatic 16 I gain control capability, to provide the desired inpl]t level to 17 ~ the crystal discriminator 110. The output oE each stage 112 8¦l and 114 caused the respective autornatic ~ain control circuitc;
~ to Jerle~rate i lirect urrent: propc~rtional to the amplitude 2~ tJf ~ t~lJtp~l~. 'Ihe~e respective AGC levels from the indi~idual 21¦¦ ~ta(3t-g 112 and 11~ are sumrned together to operate as an 22 overall autornatic gain detector 116 ~hose output is a direct 23 current proportional to the combined output a~plitude 2~ of each stage ~hich is indicative of the initial transponder 25 ¦ signal strength from hand pass filter 108. This combined AGC
26 25.

1, ' 1190'3~0 1l detector output is fed to a low pass filter 118 having a 2' predetermined time constant to produce a gradually increasing 3l charge at a rate proportional to the strength of the transponder 4 return signal being detected. The output charge from the low pass filter 118 is delivered to a comparator circuit 120 to be 6 ~ compared with a preselected threshold level established by the 7 sensitivity setting on a potentiometer 122.

9l In the preferred form of the system, the crystal discriminator llO consists of a monolithic crystal filter of 11 ¦ the type available from Piezo Technology, Inc. as its Model 12 ¦ 2378F which is combined with an RCA integrated circuit Model 13 CA3089E as described in the pertinent data sheet, to produce 14 an extremely narrowband stable discriminator with a bandwidth in the order of only 30 kiloHertz. With a valid transponder 16 ¦ return signal, the output oE the discriminator 110 constitutes 17 I the modulatiny audio tone, which in the exi~ting systelil is at 1~ ¦ t~"o kiloll(-rt~. ilowever, the output of th(? discrimirlator 1:l0 ~'J l is IT~ rltair)ed at groun~ poterltia:L by a clarnp circuit 12~ until 20 ¦ a triyyeriny olltput from the comparator circuit 120 indicates 21 I that the charge built up on the low pass filter 118 exceeds 22 I the selected sensitivity setting from the potentiometer 23 ¦ 122. This perrnits the system to be set at a sensitivity ]evel 24¦ that ignores transitory or weak return signals from remote transponders or other sources.

26.

1 1909'7f~

1' Once the clamp circuit 124 i5 open, the two kiloHertz 2l¦ audio tone; is applied through a low pass filter 126 to be ¦ decoded by conventional phase locked loop techniques using a quadrature detector 128 and phase detector 130 that is 5¦ capable of acquiring any steady tone within 10% of the 61 modulating tone frequency established as the free running 71 frequency of volta~e controlled oscillator 132. In the 8~ conventional manner, the output of the phase detector 130 is 9¦ applied to a loop filter 134 to produce a signal for adjusting the frequency and phase of the voltage controlled oscillator 11 132 to achieve phase lock. The quadrature detector 128 then 12 provides its output to a conventional operational arnplifier 13 136 having feedback capacitor 138 that maintains an output 1~ signal for trig~ering a suitable alarm 44 for providing an audible or visual response for a selected time interval no 16 i matter how brief the initial response. In this manner, the 17 ¦ strong response produced by the presence of a transponder in 1~ I the surveillance area between the antenncl pec1es~c~ls ]0 arld 12 1'~ I initia(:r!~; a Cu,l.l, ç;caJe ala~rn resporl.se no matter how quickly ~1 the p~otcct:-(l itcln ic; moved through the area, but the system 21¦ is abl~ to i~;nore even continued low level response signals 22 ¦ from outside of the immediate protected area.

24 ¦ Although the system has been described in connection 2S ~"ith a preferred embodirnent employing specifically described 2~ I / / / 27.

l circuit elements and techniques ~/ith their operating parameters 2 pertinent ~o an existing preferred embodiment using audio tone 3 frequency modulation, it should be understood that the 4l invention may be implemented employing various modifications and variations of the circuit elements and techniques ~ithout 6~ departing from the spirit or scope of the invention as defined 7 in the appended claims. For example, the system might be 8 ¦ implemented to employ amplitude modulation of one of the 9 transmitted radio frequencies, rather than frequency modulation, lO I or to employ modulating tones outside the audio range without ll discarding the basic operational advantages inherent in this 12 ~ unique overall system approach.

la~ I / i /

17 ~ / / /
1~ I / / /
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23 / / / .
24 ~
25 ! / / /
ZG ' / /

Claims (13)

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. A system for detecting the presence of an article within a surveillance area comprising:
(a) transmitter means for radiating two radio frequency signals at two distinct different frequencies within the surveillance area, said radio frequencies differeing from a means center frequency by equal and opposite small amounts;
(b) transponder means removably affixed to protected articles capable of being moved with an article into said surveillance area, said transponder means having an antenna tuned to receive the radio frequency signals transmitted at both frequencies and a non-linear impedance element coupled to said antenna means, whereby said transponder means reradiates a return signal having a frequency equal to the sum of the frequencies of the two transmitted radio frequency signals;
(c) narrowband receiver means for receiving said return signal to the exclusion of the transmitted radio frequency signals and their harmonics; and (d) alarm means responsive to the detection of said return signal by said narrowband receiver means.

2. A system according to claim 1, in which said two different frequencies differ from each other by about 2%
of said mean center frequency.

3. A system according to claim 1, in which one of said two radio frequency signals is modulated.

4. A system according to claim 3, in which the modulated radio frequency signal is frequency modulated by a fixed low frequency tone.

5. A system according to claim 1, in which the means for generating each of said two radio frequency signals includes a temperature compensated crystal controlled oscillator, frequency multiplier means, and narrowband filter means.

6. A system according to claim 1, in which said transmitter means includes signal source means, antenna means located remote from said source means, linear amplifier means in proximity with said antenna means, and connector means for delivering a signal from said source means to said linear amplifier means.

7. A system according to claim 1, in which said transmitter means includes antenna means for each of two radio frequency signals arranged so that the ratio of field strengths of said two signals is substantially uniform throughout the surveillance zone.

8. A system according to claim 1, in which the antenna of said transponder means is tuned to a frequency intermediate to said two distinct different frequencies and said non-linear impedance element is connected to said antenna means so as to provide a tank circuit with a resonant frequency equal to the sum of said two distinct different frequencies for reradiating a return signal at said resonant frequency.

9. A system according to claim 3, in which receiver means comprise means including phase locked loop circuitry for decoding the modulation of the modulated radio frequency signal.

10. A system according to claim 3, in which said narrowband receiver means includes receiver antenna means for picking up said return signal, filter means for rejecting all signals picked up by the antenna except those within a narrow pass band at the frequency of said return signal, signal amplitude detection means for generating a comparison output level indicative of the amplitude of the filtered return signal, and demodulation means responsive to the comparison output level for detecting the modulation only when said comparison level exceeds a preselected level setting.

11. A system according to claim 10, in which said signal amplitude detection means includes a local oscillator, mixer means for deriving an intermediate frequency signal, and a band pass filter for said intermediate frequency signal.

12. Apparatus according to claim 1, in which one of the radio frequency signals is modulated by a fixed audio frequency tone to produce a narrowband frequency modulation and the other is transmitted as a continuous wave at a fixed radio frequency, and in which said receiver means includes a receiver antenna, filter means for rejecting signals received by said antenna outside of a narrow pass band at said resonant frequency, means for generating an intermediate frequency for demodulation of signals within said pass band, amplifier means for amplifying said intermediate frequency signal and generating a comparison level output indicative of the amplitude of said intermediate frequency, narrowband discriminator means responsive to said comparison level output for demodulating said intermediate frequency to derive said low frequency modulation only when the amplitude of said comparison level output exceeds a preselected threshold value, a phase locked loop detector tuned to the frequency of said fixed audio tone to generate an alarm output upon detection of said fixed audio tone, and operational amplifier means coupled to receive said alarm output to actuate an alarm for a fixed time period following initiation of each such alarm output.

13. A system according to claim 1, in which said receiver means decodes the return signal without reference signals derived from the transmitter means.