CA1188996A - Circuit arrangements for modifying dynamic range - Google Patents
Circuit arrangements for modifying dynamic rangeInfo
- Publication number
- CA1188996A CA1188996A CA000392068A CA392068A CA1188996A CA 1188996 A CA1188996 A CA 1188996A CA 000392068 A CA000392068 A CA 000392068A CA 392068 A CA392068 A CA 392068A CA 1188996 A CA1188996 A CA 1188996A
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- Prior art keywords
- signal
- circuit
- path
- band
- input
- Prior art date
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Classifications
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G9/00—Combinations of two or more types of control, e.g. gain control and tone control
- H03G9/02—Combinations of two or more types of control, e.g. gain control and tone control in untuned amplifiers
- H03G9/12—Combinations of two or more types of control, e.g. gain control and tone control in untuned amplifiers having semiconductor devices
- H03G9/18—Combinations of two or more types of control, e.g. gain control and tone control in untuned amplifiers having semiconductor devices for tone control and volume expansion or compression
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G7/00—Volume compression or expansion in amplifiers
- H03G7/06—Volume compression or expansion in amplifiers having semiconductor devices
- H03G7/08—Volume compression or expansion in amplifiers having semiconductor devices incorporating negative feedback
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G9/00—Combinations of two or more types of control, e.g. gain control and tone control
- H03G9/02—Combinations of two or more types of control, e.g. gain control and tone control in untuned amplifiers
- H03G9/025—Combinations of two or more types of control, e.g. gain control and tone control in untuned amplifiers frequency-dependent volume compression or expansion, e.g. multiple-band systems
Landscapes
- Tone Control, Compression And Expansion, Limiting Amplitude (AREA)
- Reduction Or Emphasis Of Bandwidth Of Signals (AREA)
- Ultra Sonic Daignosis Equipment (AREA)
Abstract
ABSTRACT
Improvements in compressors, expanders and noise reduction systems are disclosed which reduce the susceptibility of such circuit arrangements to undue control by signals outside the pass-band(s) (i.e., stop-band signals) in which the circuits are active. Level dependent circuitry operates, particularly by means of non-linear processing, to reduce the circuit response to the effect of stop-band signal components at high input signal levels.
Improvements in compressors, expanders and noise reduction systems are disclosed which reduce the susceptibility of such circuit arrangements to undue control by signals outside the pass-band(s) (i.e., stop-band signals) in which the circuits are active. Level dependent circuitry operates, particularly by means of non-linear processing, to reduce the circuit response to the effect of stop-band signal components at high input signal levels.
Description
1 Inve or: R~ M. DOLBY
2 Title: IMPROVEMENTS IN CIRCUIT ARRANGEMENTS
3 FOR MODIFYING DYNAMIC RANGE
4 The present invention is concerned in general with circuit arrangements which alter the dynamic range of audio and 6 other signals, namely compressors which compress the dynamic 7 range and expanders which expand the dynamic range. More 8 particularly, it relates to improvements in compressors and 9 expanders that reduce their susceptibility to control by undesired signals. Such improvements are designated "modulation 11 control" for reasons explained herein.
12 Compressors and complementary expanders are often used 13 together (a compander system) to effect noise reduction; the 14 signal is compressed before transmission or recording and expanded aFter reception or playback from the transmission 16 channel. However compressors may be used alone ~o reduce the 17 dynamic range, e.g., to suit the capacity of a transmission 18 ch~nnel, without subsequent expansion when the compressed signal 19 is adequate for the end purpose. In addition, compressors alone are used in certain products, especially audio products which 21 are intended only to transmit or record compressed broadcasts or 22 prerecorded signals. Expanders alone are used in certain 23 ~ products, especially audio products which are intended only to 241 receive or play back already compressed broadcasts or prerecorded 2~1 signals. In certain products, particularly audio recording and 26 playback products, a single device is often configured for 271 switchable mode operation as a compressor to record signals and 28¦ as an expander to play back compressed broadcasts or prerecorded 291 signals.
301 The amount of compression or expansion may be expressed 31¦ in dB. For example, 10 dB of compression means that an input 32 ¦ dynamic range of N dB is compressed to an output range of (N-10) I .~
1¦ dB. In a noise reduction system 10 dB of compression followed 21 by 10 dB of complementary expansion is said to provide 10 dB of 31 noise reduction.
41 The present invention relates in particular to improve-sl ments in circuit arrangements for modifying the dynamic range of 61 an input signall which circuit arrangements have a bi-linear 71 characteristic (where "linear" in this context denotes constant 31 gain) composed of:
91 1) a low level linear portion up to a threshold, 10¦ 2) an intermediate level non-linear tchanging gain) 11¦ portiont above the threshold and up to a finishing 12 ¦ point, providing a predetermined maximum compression 13 ¦ ratio or expansion ratio, and 14 ¦ 3) a high level linear portion having a gain 15 I different from the gain of the low level portion.
16 ~The characteristic is denoted a bi-linear characteristic 17 ~ because there are two portions of substantially constant gain.
18 ~ In practice, the threshold and finishing point are 19 ~ not always well defined "points". The two transition regions 20 ¦ where the intermediate level portion merges into the low 21 ¦ level and high level linear portions can each vary in shape 22 from a smooth curve to a sharp curve, depending on the control 23 characteristics of the compressor and expander.
24 It is also pointed out tha~ circuit arrangements with bi-linear characteristics are distinguished from two other known 26 classes of circuit arrangement, namelyo 27 (a) a logarithmic or non-linear circuit arrangement 28 with either a fixed or changing slope and with no linear portion:
29 the gain changes over the whole dynamic range.
lb) circuit arrangements with a characteristic 31 having two or more portions of which only one portion is linear ~"uni-linear"). The invention is also applicable to uni-linear circuits~ as is explained further below.
A circuit arrangement with a bi-linear characteristic has part-icular advantages and is widely used. The threshold can be set above *he in-put noise level or tra~smission channel noise level in order to exclucle the possibility of control of the circuit by noise. ~le high level portion of substantially constant gain avoids non-linear treatment of high level signals which would otherwise introduce distortion.
Two well known types of bi-linear circuits are referred to as slid-ing band circuits and fixed band ~or split band or multiband) circuits.
Sliding band circuits create the specified desirable character-istic for the case of high frequency audio compression or expansion by apply-ing high frequency boost ~for compression) or cut (for expansion) by way of a high pass filter with a variable lower corner frequency. As the signal level in the high frequency band increases, the filter corner frequency slides upwardly so as to narrow the boosted or cut band and exclude the use-ful signal Erom the boost or cut. Examples of such circuits are to be found in US-PS Re 28,426, US-PS 3,757,254, US-PS 4,072,914, US-PS 3,93~,190. Such circuits can also be configured to act at low frequencies in which case low ~0 frequency boost or cut is provided by way of a low pass Eilter with a vari-able upper corner frequency.
In fixed ban~ circuits the frequency spectrum is split into a plurality of bands by corresponding band-pass filters and the compression or expansion is effected in each band by a gain control device ~whether an auto-matically responsive, diode type of limiting device or a controlled limiting device) in the case of a compressor, with some form of reciprocal or comple-mentary l circuitry for an expander. Examples of such circuits are to be 2 ~ound in US PS 3,84~,719, US~PS 3,903,485 and in Journal of the 3 Audio Engineering Society, Vol. 15, NoO 4, October~ 19Ç7, pp.
4 383-388. These fixed band circuits provide independent action in the various frequency bands.
~ It is known to construct bi-linear compressors and 7 expanders, of both sliding band and fixed band type, by the use 3 of only a single signal path. However, it is generally preferred 9 to construct such devices by providing a main signal circuit which is linear with respect to dynamic range, with a combining l1 circuit in the main circuit, and a further circuit which derives 12 its input from the input or output of the further circuit and 13 has its output coupled to the combining circuitO The further 14 circuit includes a limiter (self-acting or controlled) and the limited further circuit signal boosts the main circuit signal in 16 the combining circuit for the case of compression but bucks the 17 main circuit signal for the case of expansion. The limited 18 further path signal is smaller than the main path signal in the 19 upper part of the input dynamic range. The main and further 2n circuits are preferably and most conveniently separately 21 identifiable signal paths~ More than one further circuit i5 22 usually provided in the case of fixed band devices. A bi-linear 23 device having main and further circuits is often referred to as 24 a dual path device.
Such known dual path compressors and expanders are 26 particularly advantageous because they enable the desired kind 27 of transfer characteristic to be established in a precise way 28 without problems of high level distortion. The low level 29 portion of substantially constant gain is established by giving the further path a threshold above the noise level; below this 3l ¦ threshold the further path is linear. The intermediate level 32 ¦ portion is created by the region over which the f~rther path Il -4-limiting action becomes partially efective and the high l~vel portion of substantially constant gain arises ater the limiter has become fully effec-tive so that the further path si.gnal ceases to increase and becomes negligi-ble compared to the main path signal. At the highest par-t of the input dynamic range, the output of the circuit arrangement is effectively only the signal passed by the linear main path, i.e. linear with respec~ to dynamic range.
Examples of these known circuits are to be found in IJS-PS
3,8~6,719, US-PS 3,903,~85 and US-PS Re 28,~26. lhere are also known analo-gous circui.ts which achieve like results but wherein the further path hascharacteristics inverse to limiter characteristics and the further path out-put bucks the main path signal for compression and boosts the main path signal for expansion (US-PS 3,828,280 and US-PS 3,875,537).
The invention is applicable to any of these known bi-linear cir-cuits in order to obtain the advantages inherent therein. The invention is not limited to bi-linear circuits, but also may be employed to improve the operation of the a:Eorementioned mi-linear circuits. As discussed further below, the invention may also be applied to logarithmic circuits provided that a departure from a logarithmic transfer function can be tolerated.
Ilotvever, the preferred embodiments rela-te to bi-linear circuits and except wIIere specifically noted, reference is made to bi-linear circuits throughout this specification.
As mentioned previ.ously, it is not essential to create the desired form of bi-linear characteristic by such "dual path" techniques. Alterna-tlves exist, operating with single paths, as described in US-PS 3,757725~, US-PS 3,967,219, US-PS 4,072,91~, US-PS 3,909,7337 for example. Although these alternative circuits usually are not capable of producing such good re-sults as dual path circuits, or . . .
may be less convenient and thereby less economical, -they can produce generally equivalen-t results. Accordingly, the inven-tion is also applicable to these known circuits.
The invention also pertains -to known compressors and expanders in which series connected (e.g. multistage) bi-linear circuits are employed. Such arrangements are des-cribed in selgian-PS ~89,4280 In compressors and expanders~ especially frequency selective or multi-band devices it is clearly desirable that strong signals in one frequency range should not unduly aEfect the behavior of signals in another frequency range. Filtering and equalization employed in the various circuits has been the standard method of dealing with this problem, both in logarithmic devices and in specialized devices such as the unilinear and bi-linear circuits which have been describedO
In these prior art circuits the DC control signal which controls the variable gain/loss device [e.g., a variable gain device such as~avoltage controlled amplifier (VCA) or a variolosser such as an FET attenuator] or variable Eilter is ~ormed from ~0 the linear additive combination of the pass-band signals and the stop-band signals reaching the control circuit. The pres-ent invention effectively alters this simple combination, in a level dependent way, so as to optimize the compressor or expander performance with respec-t to pass-band versus stop-band signals. Non-linear operations are performed, including rectification of the signals in various portions of the spec-trum and analyses are made of the relative and/orabsolute amplitudes. The final control can be formed by selecting one of the singals, by combining two or more, or by performing non-linear operations such as limiting a-t leas-t one of the signals.
According to one aspect, the invention contemplates a circuit arrangement for modifying the dynamic range of an input signal, comprising: frequency selective circuit means for dividing the frequency spectrum in which the input signal lies into pass-band and stop band regions, and dynamic modifi-cation means for modifying the dynamic range of signal com-ponents in the pass-band region in response to signal compon-ents lying in the pass-band and stop-band regions, the dynamic modification means being less responsive to stop-band signal components as the level of the input signal rises.
According to another aspec-t, the invention contem-plates a circuit arrangement for modifying the dynamic range of an input slgnal, comprising: frequency~selec-tive circui-t means for dividing the frequency spectrum in which the input signal lies into pass-band and s-top-band regions, the pass-band frequency region sliding in response to signal components lying ln the pass-band and stop-band regions, the frequency selec-tive circuit means becoming less responsive -to s-top-band signal componen-ts as the level of the input signal rises, ~0 and means for modifying the dynamic range of siynal components in the pass band region.
In other words, a-t low input signal levels the clrcuit arrangement performs substantially as a conventional compressor or expander. However, at high input signal levels the compressor or expander action is modifled by the modula-tion control circuitry of the invention.
A side effect is to modify the input-output level transfer characteristic of the device at any particular fre-quency or combination of frequencies. The overall effect is unimportant and may even be unnoticeable at the dominant frequency in bi-linear systems. However, in logarithmic sys--tems the effect of modulation control, which is operativeprimarily in the high level portion of the dynamic range, is to cause a departure from a purely logarithmic characteris-tic. This may or may not be important in any particular appli-cation.
The invention derives from the observation that, ideally, in compressors and expanders the compression or expan-sion is responsive only -to the levels of signals within desired frequency pass-bands and not to the levels of signals at o-ther frequencies, which frequencies can be said to be in the stop-bands. For example, in an ideal circuit, compression or expan-sion should not be affected by the levels of signals outside the pass -7a-~8B9~
1 ¦ band of the fixed band or the pass band of the sliding band 2 ¦ (whether or not in its quiescent position). In the case of a 31 sliding band circuit in accordance with the invention, the amount 41 of frequency sliding of the variable band becomes no more than 51 is necessary to assure that a dominant controlling signal is not 61 boosted (in the case of compression) above a reference level.
71 As applied to bi-linear circuits, particularly those ~¦ in the dual path configuration, the invention takes further 91 advantage of an inherent characteristic of such circuits:
10¦ at high input signal levels the main path signal is substantially 11¦ larger than the signal(s) in the further (or side) path(s).
12¦ Consequently, high-level signal manipulations in the further 13¦ path are essentially inaudible, and, except for phase shifts, 14¦ are essentially not measurable (negligible level changes)O This 15¦ property of bi-linear circuits is most easily understood in the 16¦ context of a dual path circuit. However, the principle also 17¦ applies in single-path bi-linear circuits in which there are 18¦ two or more signal components in the same path instead of in 19¦ separately identifiable paths.
20¦ The invention takes advantage of the above observations 21¦ concerninq bi-linear circuit characteristics. As compared with 22 ¦ prior art bi-linear compressors and expanders, the invention 23 ¦ provides for further manipulations of the signal (modulation 24 ¦ control) in the high input signal level region where the overall 25 ¦ compressor or expander response is linear. The comparatively 26 ¦ low level noise reduction component of the signal is manipulated 27 ¦ in this extra way only at high signal levelsj thereby assuring 28 1 that any effects important to the signal channel will be over-29 shadowed by the large main signal component~
30 1 In dual-path bi-linear circuits, an efect of the 31 invention is to modify the transfer characteristic of the side 32 (or further) path such that the side path characteristic itself _~
1¦ becomes bi-linear instead of flattening or downturning at high 21 input signal levels. This is a consequence of the proportionality 31 aspect of modulation control. That is, at high input levels 41 the side path level does not drop below a selected proportion 51 of the main path level (e.g. one-quarter or one-tenth). This 61 is acceptable because the side path signal still remains 71 substantially smaller than the main path signal at high input 81 signal levels and because the stop-band is usually substantially 91 shifted in phase with respect to the main path signal channel.
For these same reasons, the invention may be embodied 11 in uni-linear circuits which have a linear response at high 12 si~nal levels.
13 From another point of view, the action of the invention 14 is to increase the levels of stop band signal components in the output of the device at high signal levels, but not ~o such an 16 extent as to cause problems with the recordin~ or transmission 17 channel since, relatively speaking they are still small.
18 Increasing the levels of such stop band signal components is not 1~ in itself a particularly advantageous thing but is necessary in order to obtain improved dynamic action and noise reduction 21 within the pass band~ Increasing the levels of s-top band 22 ~ignals in the output oE the device at high signal levels is 23 achieved by reducing the levels of stop band signal components 24 in the control signal channel at high signal levels, or by so arranging things that the control signal is generated as if 26 there was a reduced level of stop band signal components 2~ in the signal used to produce the control signal at high signal 28 levels (e.g., by filtering and limiting in the control circuits 29 or by fre~uency dependent control signal bucking arrangements).
A further advantage o the invention is that in 31 listening tests, "pumpiny effects" of single-ended compressors 3~ ///
_g_ 1¦ and expanders are substantially reduced if not eliminated. Thus, 21 in addition to its use in complementary noise reduction systems, I s~
~; 31 the invention is particularly useful for use in ~ com-41 pressors and expanders (iOe., compressors for use in compressing 51 signals that are not subsequently expanded and expanders for use 6¦ in expanding signals that were not previously compressed)O
71 By way of background of the invention~ although various 81 practical embodiments of noise reduction circuits have proven 91 successful, in operation such circuits depart in some degree 10¦ from the ideal because of the problem of stop-band signals 11¦ unduly controlling compression and expansion. The efEect of 1~¦ such shortcomings is manifested in several interrelated ways:
13 ¦ 1) a reduction in noise reduction effect in a 14 ¦ portion of the noise reduction system pass band;
15 ¦ 2) noise modulation effects (eug., the level 16 ¦ of a signal at one frequency, modulating the noise 17 ¦ level in a different part of the frequency spectrum) 18 3) signal modulatiorl effects (e.g., the level of a 19 ¦ signal at one fre~uency modulating the level of a signal at another frequency);
21 4) cross modulation effects (e.g. spurious modulation 22 products resulting from one of more of the last two 23 enumerated modulation effects)~
24 The degree to which these shortcomings are observable 2~ depends on the type of circuits employed in the noise reduction 2~ system, the recording and playback equipment, the record/playback 27 channel or medium and the nature of the signal material. In 28 many cases, the shortcomings are essentially unobservable except 29 ¦ by test instruments. Nevertheless, it is desirable to deal with 30 ¦ these shortcomings. Because the aforementioned shortcomings of 31¦ known compressors~ expanders and noise reduction systems rela-te 32 I to modulation effects, ei-ther of signals or of noise~ -the _~o_. , ~ 6 1¦ invention described herein for reducing such shortcomings is 21 referred to as modulation control.
31 The severity of these modulation effects depends to a 41 great extent on the uniformity of the transmission channel 51 between the compressor and expander. For example, in magnetic 61 tape recording and playback systems, a frequency response 7 phenomenon known as ~Ihead bumps" exists. Even in professional 8 systems, particularly those operating at 30 ips, the playback 9 response below 100 Hz is nonuniform due to the relationship between the signal wavelength on tape and the playback head 11 dimension, which are of the same orderO If the compressor/
12 expander system is susceptible to signals in the head bump 13 region, such signals when played back may control the expander 14 in a non-complementary way such that signals or noise at higher frequencies, e.g., up to 3 kHz, may be modulated by the signals 16 in the re~ion of or below 100 Hz~
17 In prior art fixed band (single band and multi-band) 18 circuits, various filterin~ techniques have been used to minimize 19 the control of compression and expansion by undesired signals.
~0 According to these techniques, sharp filters (e.g.) with steep 21 skirts) are placed in the signal path or in the control circuit ~ ~of the limiting device).
23 However, the use of signal path filters sharper than ~ 6 dB/octave (e.g., single pole filters) in multiband compressors and expanders causes amplitude and phase efects such that when 26 the overall signal spectrum is recombined there are amplitude V and phase errors. This problem is greatly exacerbated if 28 filters sharper than 12 dB/octave are employed. However, a ~91 filter slope of only 6 or 12 dB/octave may not adequately 30 ~ discriminate against all unwanted signals. In the multi-band 311~ (fixed band) bi~linear circuit examples of US~PS 3,846,719 and 321 in'Journal of the Audio _ i eerinq Societ~, Vol. 15, No~ 4, l¦ October, 1967, pp. 383-388, filters having a 12 ds/octave slope 21 are used in the signal path of three of the four fixed bands. A
31 flat overall frequency response is obtained only by the use of a 41 complex filter characteristic in the frequency band adjacent the 51 sharp filters~ Such a solution obviously is not universally 61 applicable.
71 In the logarithmic multi-band (fixed band) compressor/
81 expander circuit described in Rundfunktechnl Mitteitun~
91 Jahr. 22 (1978) H~ 2, pp. 63-74, the input signal is divided l0¦ into four bands by single pole filters. ~oweverl th~ control ll¦ circuits for each band employ sharp 18 dB/octave fil~ers. A
12¦ sharp eontrol circuit filter (12 dB/octave~ is also employed 13¦ in a single fixed band compressor/expander circuit sold under 14¦ the trademark "dbx II." However, the use of sharp control l5¦ eircuit filters results in excessive amplification of high 16¦ level signals outside ~he control circuit filter pass band 17¦ when high amplitude signals are not present within the control l~¦ circuit filter pass band, resulting in the posslble overdriving l9¦ of the transmission channel unless sharp eutof~ filters are used 20¦ in the signal ehannel as well.
21¦ A prior art technique referred to as spectral skewing 22¦ is described in Belgian-PS 889,427; Audio, May 198l, pp~ 20-26;
~31 and paper J-6 and preprint presented at November 1981 Conven-24¦ tion~ Audio Engineering Society, New York, New York. Spectral 251 skewing is also concerned ~ith the suppression of modulation 2S¦ effects resulting from compressor/expander non-complementarity 27 ¦ due to transmission channel errors. According to the teachings 28 ¦ of spectral skewing, sharp filtering is provided at least in the 29 ¦ compressor at a frequency well within the normal bandpass of 30 ¦ the system and within the flat response region of the transmis-31 sion channel. While spectral skewing is successful in reducing 32 ! ///
~ -12-l ll 1 spurious sign~l modulation effects caused by channel irregulari-2 ties, it does not address the problem of excessive frequency 3 sliding in sliding band systems or of excessive attenuation in 4 fixed band systems~
S Thus, the present invention seeks to minimize the 6 control of expansion and compression by undesired signals without 7 the attendant side effects and/or complexity of the prior art.
8 Although measurable modulation effects are not totally 9 suppressed by the inven~ion, the effects of the invention in audio applications are supplemented by psychoacoustical masking 11 effects such that perceived effects are~ for most listeners and 12 musical material, inaudible. That is, only the modulation of a 13 signal (or signals) sufficiently spaced in frequency from the 14 modulating signal is perceived by the human ear. Such modulation is minimized by the present invention. While the modulation 16 of a signal (or signals) by another signal closely spaced in 17 frequency is less likely to be affected or improved by the 18 invention, such phenomena will likely not be perceived by the 19 ear because of two related effects:
a) a weak signal close in fre~uency to a strong 21 signal is masked by the strong signal such that the weak signal 22 is inaudible, or 23 b) if the closely spaced signal is audible before 24 compression or i9 increased in level by the compressor such that it becomes audible, then there is a psychoacoustic tolerance of 26 modulation effects because of the close frequency spacing.
27 Consequently, the human ear is not able to discern 28 modulation effects of signals at closely spaced frequencies and 29 thus the invention need not be fully effective for such signals.
The operating environment o the invention is a fixed 31 band or sliding band compressor or expander circuit in which 32 there is a variable circuit means, usually controlled by a DC
l l 1 control signal~ which is operative primarily in the lower part 2 of the overall dynamic range~ In accordance with the invention, 3 mod~lation control means are employed in the upper part of the 4 dynamic range to prevent the action of the variable circuit means from becoming any greater than is necessary to provide the 6 nominal required attenuation of dominant signals, whether such 7 signals have freauencies in the pass-band or in the stop-band.
81 In pract1ce, controlling the action of the variable circuit 91 means usually comprises operating upon the control signal 10¦ controlling the circuit means.
11¦ The modulation control may take the form of active or 12¦ passive control signal limiting means which become operative at 13¦ high signal levels or of means employing circuits which detect 14¦ the presence of high level signals and generate signals which 15¦ oppose the increase of the control signal level. Such control 16¦ signal limiting may take place in one or more frequency selective 17¦ control signal channels; if more than one, means are provided for 18¦ selecting or combining the control signals so as to provide the 19¦ variable circuit element with an optimal control signal When a 201 high level signal detection circuit~ or modulation control 21¦ generator, is usedr it may operate in various ways which will 22¦ give a measure of signal levels in at least the upper part of ~he 231 dynamic range. For example, the modulation control signal may be 24 ¦ derived from the input or output signal of ~he compressor or 25 ¦ expander. The modulation control signal in effect provides a 26 reference for the DC control signal applied to the variable cir-27 ¦ cuit element (VCA or voltage controlled filter~. The reference 28 ~ signal is combined in phase opposition with (eOgO, opposite 29 ¦ polarity or so as to buck) the DC control signal generated pri-30 1 marily in response to stop-band signal components to provide a 31 limit as to how large the control signal to the variable circuit 32 j IL181~ g6 1¦ element can become in response to signals in the stop-band, io~
21 outside the pass band of the fixed band or sliding band. In 31 practice this limit can be made relatively "hard" or relatively 41 I'soft". That is, continued increases in the control signal can be 51 rather abruptly stopped or allowed to continue at a reduced rate 61 The modulation control signal may also be derived from 7 the variable circuit (VCA or variable filter~ by measuring voltage 8 or current components of the variable circuit and, if necessary~
9 equalization in order to generate a signal usable in providing a limit as to how large the control signal to the variable circuit 11 can become in response to signals in the stop-band.
12 In terms of resul~sr the invention as applied to 13 either fixed band or sliding band devices provides a substantial 14 immunity to signals outside the pass band of the fixed band or the sliding band~ In ~he case of sllding band devices, the 16 invention provides a Eurther related advantage, i.eO~ the 17 sliding band slides only so far in response to a dominant signal 18 as is necessary to bring the gain at the signal fre~uency to 19 substantially unity, at least for levels at or above a reference level. The reference level is at or near the upper area of the 21 dynamic operating range of the device, such as within about 22 6~20dB of the maximum allowable level. Prior art sliding band 23 circuits are susceptible to excessive sliding such that the 24 variable filter corner frequency is pushed farther than is needed with hiyh level signals, causing not only potential 26 modulation effects but also resulting in a loss of noise reduc-27 tion effect in part of the spectrumO
~8 As applied to sliding band dual-path circuits, the 29 invention provides forr in the simplest embodiment, the rectifi-cation and smoothing of the input or output signal and the 31 combining of the resultant DC reference signal with the control 32 signal applied to the variable filter~ The level of the l ¦ reference signal can be set for a desired proportion limit on a 2 ¦ dominant further path signal relative to the corresponding 3 ¦ component in the main path signal~ ~or example, the modulation 41 control circuit may be made to operate such that in, say, the S ¦ upper 20dB of the dynamic range the limiter provides only that 6 ¦ attenuation required to keep the dominant signal component in 71 the further path held at a relatively constant proportion of 81 that component in the main path signal (e.g. 15 dB below).
9 ¦ As applied to fixed band dual-path circuits, the lO ¦ invention provides for, in the simplest embodimentl as in the ll¦ sliding band embodiment, the rectification and smoothing of the 12¦ input or output signal to generate a modulation control signal 13 ¦ that responds primarily to the high level signal components of 14 ¦ the input signal. However, in the case o fixed band circuits a 15 ¦ sharp filter is used in the pass-band control circuit to provide 16 a pass-band control signal. In addition a stop~band control 17 circuit is employed to provide a stop-band control signal. The l3 modulation control signal provides a reference for the stop band 19 control signal (i.e. opposes it at high signal levels).
The referenced stop-band control signal is compared with the 21 pass-band control signal and the two are combined generally to 22 favor the larger, via a maximum signal selection circu t, which 23 then controls the VCA The overall effect of the circuit is to 24 provide the required attenuation (overall compression law~ in the pass-band, while avoiding control of the pass-band attenuation 26 by larye signal components in the stop-band, and while avoiding 27 the possibility of excessive amplification of high level signals 28 in the stop-band, as seen at the output of the overall compressor 29 In these and other embodiments in which a reference bucking signal is generated, the signal may be derived from the 3l input or output because at high signal levels, where the inven-32 tion operates, the input and output levels are nearly the I
I ~ ~ame. In som mbodiments the modu1ation control signa1 may be 2 subjected to filtering or equali2ation before rectification.
3 Such equaliæation works in conjunction with the fixed or variable 4 filtering or equalization employed in the signal circuits and control circuits, to yield an overall modulation control which 6 is most effective in suppressing control by signal components in 7 the stop-band, while at the same time in~erfering as little as 8 possible with control by signal components in the pass-band.
9 Other embodiments of the invention are described hereinafter. For example, the amplified AC output of the 11 sliding band variable filter may be divided into two or more 12 band pass channels, each channel subjected to selected limiting 13 thresholds, rectified and combined to produce a control signal.
14 By selecting appropriate thresholds, the DC control circuit of ~ the sliding band compressor or expander then has a frequency 16 dependent maximum output characteristic that functions to 17 minimize the control of the compressor or expander by signals 18 outside the slidina pass band.
19 In a variation using only one control circuit channel, a low ~requency boost circuit is placed in the control amplifier~
21 This is followed by an amplitude limiter and then a high fre 22 ~uency boost circuit. The resulting AC signal is then rectified 23 and smoothed to form the control signal~
24 The invention will be described in more detail, by way of example, with reference to the accompanying drawings, in which:
26 Figure 1 is an exemplary set of curves showing com~ ¦
27 plementary bi-linear compression and expansion characteristicsO
28 Figure 2 is a schematic circuit diagram of a prior art 29 sliding band compressor.
Figure 3 is a schematic circuit diagram of a prior 31 art sliding band expander.
~ -17-l Figure 4 is a schematic circuit diagram of a modifica-21 tion to Figures 2 and 3.
3¦ Figure 5 is a block diagram of a prior art sliding 41 band compressor.
51 Figure 6 is a set of probe tone curves illustrating ~¦ the sliding band action of the circuit of Figures 2 and 4.
7 Figures 7 - 10 are a series of probe tone curves 81 illustrating the effects of modulation control according to the 91 invention embodied in a sliding band compressor.
lO¦ Figure 11 is a block diagram of a preferred embodiment ll¦ f the invention embodied in a sliding band compressor l~¦ Figures 12 - 15 are block diagrams of further embodi-13¦ ments o~ the invention embodied in slidin~ band compressors.
14¦ Figures 16 and 17 are block diagrams o a prior 15¦ art fixed band compressor and expander~
16¦ Figures 18 - 20 are response curves illustrating 17¦ the effects oE modulation control according to the invention 18¦ embodied in a sliding band compressor.
~91 Figure 21 is a block diagram of a preferred embodiment 20¦ of the invention embodied in a ixed band co~pressor.
21¦ Figure ~2 is a block diaaram of an alternative 22¦ embodiment of the invention embodied in a Eixed band compressor.
23 ¦ Exemplary bi-linear complementary compression and expan-24 ¦ sion transfer characteristics (at a particular frequency) are 25 ¦ shown in Figure 1, indicating (for the compression characteristic) 2~ ¦ the low level portion of substantially constant galn, the 27 ¦ threshold, the portion where dynamic actiorl occurs, the inishing 28 ¦ point, and the high level portion of s~bstantially constant gain~
~9 I Details of one dual path sliding band bi-linea circuit are set forth in Figures 2, 3 and 4~ The sliding band 31 I embodiments of the present invention are described with reference 32 I to this circuit, although the invention is not limited to use in j -18-l¦ such circuits. Figures 2, 3 and 4 are the same as Figure 4, 5 21 and 10 respe~tively of US-PS Re 28~426 and further details of 31 said circuits, their operation and theory are set orth therein.
41 Figure 5 is a block diagram of Figure 2 (with or without the ~¦ Figure 4 modification). The following description of Figures 2, 61 3 and 4 is taken in large part from US-PS Re 28,426.
7¦ The circuit of Figure 2 is specifically designed for 8 incorporation in the recording channel of a consumer tape 91 recorder, two such circuits being re~uired for a stereo recorder.
lO¦ The input signal is applied at ~erminal 10 to an emitter follower ll¦ stage 12 which provides a low impedance signal. This signal is 12¦ applied firstly through a main, straight-through path constituted 13¦ by a resistor 14 to an output terminal 16 and secondly through a 14¦ further path the last element of which is a resistor 18 also lS¦ connected to the terminal 16u The resistors 14 and 18 add the 16¦ outputs of the main and further paths to provide the required 17 ¦ compression law~
l8 ¦ The further path consists of a fixed filter 20, a 19 ¦ variable cut-off filter 22 including a FET 24 (these constituting the filter/limiter), and an amplifier 26 the output of which is 2~1 coupled to a double diode limiter or clipper 28 and to the ~¦ resistor 18. The non~linear limiter suppresses overshoots of 23 ¦ the output signal with abruptly increasing input signals. The 24 ¦ amplifier 26 increases the signal in the further path to a level 25 ¦ such that the knee in the characteristic of the limiter or 26 ¦ overshoot suppressor 28, comprising silicon diodes, is effective 27 ¦ at the appropriate signal level under transient conditionsO
28 ¦ The effective threshold of the overshoot suppressor is somewhat 29 above that of the syllabic filter/limiter. The resistors 14 and 18 are so proportioned that the required compensating degree of 31 attenuation is then provided for the signal in the further path~
-19 ll l The output of the amplifier 26 is also coupled to an 2 amplifier 30 the output of which is ~ectified by a germanium 3 diode 31 and integrated by a smoothing filter 32 to provide 4 the control voltage for the FET 24.
Two simple RC filters are used, though e~uivalent LC
6 or LCR filters could be used. The fixed filter 20 provides a 7 cut-off frequency of 1700 Hz (now 1500 Hz), below which diminish-8 ing compression take place. The filter 22 comprises a serles 9 capacitor 34 and shunt resistor 36 followed by a series resistor 38 and the FET 24, with i~s source-drain path connected as a ll shunt resistor. Under quiescent conditions with zero signal on 12 the gate of the FET 24, the FET is pinched off and presents 13 substantially infinite impedance; the presence of the resistor 14 38 can then be ignored. The cut-off frequency of the filter 22 is thus 800 Hz tnow 750 Hz), which it will be noted is substan~
16 tially below the cut-of frequency of the ixed filter 20.
17 When the signal on the gate increases sufficiently for l8 the resistance of the FET to fall to less than say 1 K, the 19 resistor 38 effectively shunts the resistor 36 and the cut-ofE
re~uency rises, markedly narrowing the pass band of the filter.
21 The rise in cut-off frequency is of course a progressive action.
22 The use of a FET is convenient because, within a suit-23 able restricted range of signal amplitudes, such a device acts 24 substantially as a linear resistor (for either polarity signal~, the value of which is determined by the control voltage on the 26 gate.
27 The resistor 36 and FET are returned to 2n adjustable 28 tap 46 in a potential divider which includes a temperature 29 ¦ compensating germanium diode 4~. The tap 46 enables the compres-30 ¦ sion threshold of the filter 22 to be adjusted~
311¦ The amplifier 26 comprises complementary transistors 3~11 giving high input impedance and low output impedance~ Since Il , ~ 3~
1 the amplifier drives the diode limiter 28, a finite output 2 impedance is required and is provided by a coupling resistor 3 500 The diodes 28 are, as already noted, silicon diodes 4 and have a sharp knee around 1/2 volt. I
The signal on the limiter and hence on the resistor 18 6 can be shorted to ground by a switch 53 when it is required to 7 switch the compressor out of action.
8 The amplifier 30 is an NPN ~is$~ with an emitter time 9 constant network 52 givinq increased gain at high freauenciesO
Strong high frequencies ~eag. a cymbal crash) will therefore 11 lead to rapid narrowing of the band in which compression takes 12 place~ so as to avoid signal distortionO
13 The amplifier is coupled to the smoothing filter 32 14 through the rectifying diode 310 The filter comprises a series resistor 54 and shunt capacitor 56. The resistor 54 is shunted 16 by a silicon diode 58 which allows rapid charging of the 17 capacitor 56 for fast attack, coupled wi~h good smoothing under 13 steady-state conditions. The voltage on capacitor 56 is applied 19 d.irectly t~ the gate of the FET 24.
A complete circuit diagram of the complementary expander 21 is provided in Fiyure 3, but a full description is not required as 22 substantially as the circuit is identical to Figure 2, component 23 values, are therefore not for the most part shown in Figure 3.
24 The differences between Figures 2 and 3 are as follows:
In Fi~ure 3, the further path derives its input from 26 the output terminal 16a, the amplifier 26a is inverting, 27 and the signals combined by the resistors 14 and 18 are applied 28 to the input (base) of the emitter follower 12, -the oukput 291 (emitter) of which is coupled to the terminal 16aO To ensure low driving impedance, the input terminal 1Oa i5 coupled to the 31¦ resistor 14 through an emitter follower 60. Suitable measures 32~ must be taken to prevent bias getting in the expander~
¦ -21-, 118!399~i 1¦ The ampliEier 26a is rendered inverting by taking the 21 output from the emitter, instead of the collector, of the second 31 ~PNP) transistor. This alteration involves shifting the 10 K
41 resistor 62 ~Figure 2) from the collector to the emitter ~Figure 51 2~, which automatically gives a suitable output impedance for I driving the limiter. The resistor 50 is therefore omitted in 71 Figure 3.
81 It should be noted that it is important in aligning 91 a complete noise reduction system to have equal signal levels 10¦ on the emitters of the transistors 12 in both compressor 11¦ and expander. Metering terminals M are shown connected to 12¦ these emitters.
13 Figure 4 shows a preferred circuit, for replacing the 14 circuit between points A, B and C in Figures 2 and 3. When the FET 24 is pinched off~ the second RC network 22 is inoperative, 16 and the first RC network 20 then determines the response of the 17 further path. The improved circuit combines the phase advantages 18 of having only a single RC section under quiescent conditions 19 with the 12 dB per octave attenuation characteristics of a two-section RC filter under signal conditions.
21 In the practical circuit, using MPF 104 FE~'s, the 39 K
22 resistor 36a is necessary in order to provide a finite source 23 impedance to work into the FET. In this way the compression 24 ratio at all frequencies and levels is held to a maximum of about 2u The 39 K resistor 36a serves the same compression 26 ratio limiting function in the improved circuit as the resistor 27 36 in the circuit of Figure 2 or Figure 3. In addition, this 28 resistor provides a low frequency path for the signal. I
29 Certain details of the circuit oE Figures 2, 3 and 4 have evolved over the years and more modern forms of the circuit 31 have been published and are well known in the ar-t. Reference to ~ -22--l the specific circuit in US-PS Re 28,426 is made for convenience 2 in presentation.
3 Figure 5 is a block diagram showing the major elements 4 of the compressor of Figures 2 and 4. Combining circuit 15 S represents the combining resistors 14 and 18 of Figures 2 and 3.
b The variable band action of the sliding band device 7 can be seen in Figure 6, showing an actual chart recorder probe 8 tone response obtained from the circuit of Figure 2 incorporating 9 Figure 4. The variable band action is shown by plotting the compressor frequency response by means of a low-level probe tone ll (the level of which is below the compressor threshold) in the 12 presence of a high-level signal; the probe is detected at the 13 compressor output by means of a tracking filter. The high-level 14 signal causes the compressor circuitry to operate, the graph showing the effect on the turnover frequency of the filter.
16 In a sliding band device in accordance with the 17 invention, the amplitude of the high level or dominant signal 18 that causes the sliding band action should not cause excessive 19 sliding, nor should the presence of other high level signals outside of the sliding band pass band cause excessive sliding.
21 Excessive slidinq means movement of the variable filter turnover 22 fre~uency farther than necessary to produce a sliding band 2~ compressor characteristic which avoids boosting the dominant 24 signals above a reference level. The absolute value of the reference level is chosen by the system designer, but is usually 26 some 1OdB below the highest levels normally used.
27 Figure 7 shows a further se~ of actual chart recorder 281 probe tone curves for the case of a sliding band compressor 29 circuit similar in design to that of Figure 2 (with the Figure 4 30 I modification), but with a low level gain of 8 dB and a filter 31 quiescent fre~uency of 800 Hz. The probe tone level is at -40 32 dB, below the compressor threshold. Curves are taken for a 1 100 Hz signal at -20~ -10, 0, ~10 and +20 dB9 where 0 dB is the 21 reference level. Also, a curve for no 100 Hz signal is shown.
3¦ The ~10r 0, ~10 and +20 dB chart recording curves are all started 41 at about 200 Hz. This is also the case for Figure 8~ In 51 Figures 9 and 10 there are also curves for the no signal condition ~¦ Referring again to Figure 7, ideally~ there should be 71 no sliding in response to a 100 ~z signal because it is well 81 outside the pass band of the circuit at its lowest (quiescent) 91 frequency. Nevertheless, as the 100 Hz signal increases in 10¦ level, the band slides upward. The -10, 0, ~10 and ~20 dB
11 curves need not slide any farther than the -20 dB curve in order 12¦ to avoid any substantial boosting of the 100 Hz signal. The 13¦ unnecessary sliding has two effects: a) substantial noise 14¦ reduction action is lost (during playback) because no boosting 15¦ takes place at frequencies where it otherwise can take place and 16¦ b) as the amplitude of the 100 Hz signal varies it can modulate 17¦ signals at higher frequencies as the sliding band varies under 18¦ its control~ resulting in possible incorrect restoration of the 19¦ signal by the expander if the recording or transmission channel 20¦ has an irregular frequency response in the vicinity of l00 Hz.
21¦ Figure 8 shows a set of actual chart recorder probe 22¦ tone curves for the same circuit, but with the addition of 231 modulation control circuitry as described hereinafterO Essen-?4¦ tially no sliding occurs for the same levels o~ the 100 ~æ
25¦ signal as in the Figure 7 arrangement. The sliding band 261 compressor is made essentially immune to strong signals outside 27 its pass band. The sliding band response is essentially the 28 ¦ same as its response below threshold in the presence of no 29¦ dominant signals.
30¦ The effect of modulation control for sliding band 31 ¦ compressors is further illustrated by Figures 9 and 10, which 32 are also actual chart recorder probe tone curves taken with the 1 ~2~
l l l l l ¦ same circuit and probe tone level as with Figures 7 and 8~ ln ~ ¦ this case, the effect of a dominant signal at 800 Hz, a fre-3 ¦ quency within the desired active area of the circuit, is shown 4 ¦ Ideally, slidinq is required to go only so far as not to boost
12 Compressors and complementary expanders are often used 13 together (a compander system) to effect noise reduction; the 14 signal is compressed before transmission or recording and expanded aFter reception or playback from the transmission 16 channel. However compressors may be used alone ~o reduce the 17 dynamic range, e.g., to suit the capacity of a transmission 18 ch~nnel, without subsequent expansion when the compressed signal 19 is adequate for the end purpose. In addition, compressors alone are used in certain products, especially audio products which 21 are intended only to transmit or record compressed broadcasts or 22 prerecorded signals. Expanders alone are used in certain 23 ~ products, especially audio products which are intended only to 241 receive or play back already compressed broadcasts or prerecorded 2~1 signals. In certain products, particularly audio recording and 26 playback products, a single device is often configured for 271 switchable mode operation as a compressor to record signals and 28¦ as an expander to play back compressed broadcasts or prerecorded 291 signals.
301 The amount of compression or expansion may be expressed 31¦ in dB. For example, 10 dB of compression means that an input 32 ¦ dynamic range of N dB is compressed to an output range of (N-10) I .~
1¦ dB. In a noise reduction system 10 dB of compression followed 21 by 10 dB of complementary expansion is said to provide 10 dB of 31 noise reduction.
41 The present invention relates in particular to improve-sl ments in circuit arrangements for modifying the dynamic range of 61 an input signall which circuit arrangements have a bi-linear 71 characteristic (where "linear" in this context denotes constant 31 gain) composed of:
91 1) a low level linear portion up to a threshold, 10¦ 2) an intermediate level non-linear tchanging gain) 11¦ portiont above the threshold and up to a finishing 12 ¦ point, providing a predetermined maximum compression 13 ¦ ratio or expansion ratio, and 14 ¦ 3) a high level linear portion having a gain 15 I different from the gain of the low level portion.
16 ~The characteristic is denoted a bi-linear characteristic 17 ~ because there are two portions of substantially constant gain.
18 ~ In practice, the threshold and finishing point are 19 ~ not always well defined "points". The two transition regions 20 ¦ where the intermediate level portion merges into the low 21 ¦ level and high level linear portions can each vary in shape 22 from a smooth curve to a sharp curve, depending on the control 23 characteristics of the compressor and expander.
24 It is also pointed out tha~ circuit arrangements with bi-linear characteristics are distinguished from two other known 26 classes of circuit arrangement, namelyo 27 (a) a logarithmic or non-linear circuit arrangement 28 with either a fixed or changing slope and with no linear portion:
29 the gain changes over the whole dynamic range.
lb) circuit arrangements with a characteristic 31 having two or more portions of which only one portion is linear ~"uni-linear"). The invention is also applicable to uni-linear circuits~ as is explained further below.
A circuit arrangement with a bi-linear characteristic has part-icular advantages and is widely used. The threshold can be set above *he in-put noise level or tra~smission channel noise level in order to exclucle the possibility of control of the circuit by noise. ~le high level portion of substantially constant gain avoids non-linear treatment of high level signals which would otherwise introduce distortion.
Two well known types of bi-linear circuits are referred to as slid-ing band circuits and fixed band ~or split band or multiband) circuits.
Sliding band circuits create the specified desirable character-istic for the case of high frequency audio compression or expansion by apply-ing high frequency boost ~for compression) or cut (for expansion) by way of a high pass filter with a variable lower corner frequency. As the signal level in the high frequency band increases, the filter corner frequency slides upwardly so as to narrow the boosted or cut band and exclude the use-ful signal Erom the boost or cut. Examples of such circuits are to be found in US-PS Re 28,426, US-PS 3,757,254, US-PS 4,072,914, US-PS 3,93~,190. Such circuits can also be configured to act at low frequencies in which case low ~0 frequency boost or cut is provided by way of a low pass Eilter with a vari-able upper corner frequency.
In fixed ban~ circuits the frequency spectrum is split into a plurality of bands by corresponding band-pass filters and the compression or expansion is effected in each band by a gain control device ~whether an auto-matically responsive, diode type of limiting device or a controlled limiting device) in the case of a compressor, with some form of reciprocal or comple-mentary l circuitry for an expander. Examples of such circuits are to be 2 ~ound in US PS 3,84~,719, US~PS 3,903,485 and in Journal of the 3 Audio Engineering Society, Vol. 15, NoO 4, October~ 19Ç7, pp.
4 383-388. These fixed band circuits provide independent action in the various frequency bands.
~ It is known to construct bi-linear compressors and 7 expanders, of both sliding band and fixed band type, by the use 3 of only a single signal path. However, it is generally preferred 9 to construct such devices by providing a main signal circuit which is linear with respect to dynamic range, with a combining l1 circuit in the main circuit, and a further circuit which derives 12 its input from the input or output of the further circuit and 13 has its output coupled to the combining circuitO The further 14 circuit includes a limiter (self-acting or controlled) and the limited further circuit signal boosts the main circuit signal in 16 the combining circuit for the case of compression but bucks the 17 main circuit signal for the case of expansion. The limited 18 further path signal is smaller than the main path signal in the 19 upper part of the input dynamic range. The main and further 2n circuits are preferably and most conveniently separately 21 identifiable signal paths~ More than one further circuit i5 22 usually provided in the case of fixed band devices. A bi-linear 23 device having main and further circuits is often referred to as 24 a dual path device.
Such known dual path compressors and expanders are 26 particularly advantageous because they enable the desired kind 27 of transfer characteristic to be established in a precise way 28 without problems of high level distortion. The low level 29 portion of substantially constant gain is established by giving the further path a threshold above the noise level; below this 3l ¦ threshold the further path is linear. The intermediate level 32 ¦ portion is created by the region over which the f~rther path Il -4-limiting action becomes partially efective and the high l~vel portion of substantially constant gain arises ater the limiter has become fully effec-tive so that the further path si.gnal ceases to increase and becomes negligi-ble compared to the main path signal. At the highest par-t of the input dynamic range, the output of the circuit arrangement is effectively only the signal passed by the linear main path, i.e. linear with respec~ to dynamic range.
Examples of these known circuits are to be found in IJS-PS
3,8~6,719, US-PS 3,903,~85 and US-PS Re 28,~26. lhere are also known analo-gous circui.ts which achieve like results but wherein the further path hascharacteristics inverse to limiter characteristics and the further path out-put bucks the main path signal for compression and boosts the main path signal for expansion (US-PS 3,828,280 and US-PS 3,875,537).
The invention is applicable to any of these known bi-linear cir-cuits in order to obtain the advantages inherent therein. The invention is not limited to bi-linear circuits, but also may be employed to improve the operation of the a:Eorementioned mi-linear circuits. As discussed further below, the invention may also be applied to logarithmic circuits provided that a departure from a logarithmic transfer function can be tolerated.
Ilotvever, the preferred embodiments rela-te to bi-linear circuits and except wIIere specifically noted, reference is made to bi-linear circuits throughout this specification.
As mentioned previ.ously, it is not essential to create the desired form of bi-linear characteristic by such "dual path" techniques. Alterna-tlves exist, operating with single paths, as described in US-PS 3,757725~, US-PS 3,967,219, US-PS 4,072,91~, US-PS 3,909,7337 for example. Although these alternative circuits usually are not capable of producing such good re-sults as dual path circuits, or . . .
may be less convenient and thereby less economical, -they can produce generally equivalen-t results. Accordingly, the inven-tion is also applicable to these known circuits.
The invention also pertains -to known compressors and expanders in which series connected (e.g. multistage) bi-linear circuits are employed. Such arrangements are des-cribed in selgian-PS ~89,4280 In compressors and expanders~ especially frequency selective or multi-band devices it is clearly desirable that strong signals in one frequency range should not unduly aEfect the behavior of signals in another frequency range. Filtering and equalization employed in the various circuits has been the standard method of dealing with this problem, both in logarithmic devices and in specialized devices such as the unilinear and bi-linear circuits which have been describedO
In these prior art circuits the DC control signal which controls the variable gain/loss device [e.g., a variable gain device such as~avoltage controlled amplifier (VCA) or a variolosser such as an FET attenuator] or variable Eilter is ~ormed from ~0 the linear additive combination of the pass-band signals and the stop-band signals reaching the control circuit. The pres-ent invention effectively alters this simple combination, in a level dependent way, so as to optimize the compressor or expander performance with respec-t to pass-band versus stop-band signals. Non-linear operations are performed, including rectification of the signals in various portions of the spec-trum and analyses are made of the relative and/orabsolute amplitudes. The final control can be formed by selecting one of the singals, by combining two or more, or by performing non-linear operations such as limiting a-t leas-t one of the signals.
According to one aspect, the invention contemplates a circuit arrangement for modifying the dynamic range of an input signal, comprising: frequency selective circuit means for dividing the frequency spectrum in which the input signal lies into pass-band and stop band regions, and dynamic modifi-cation means for modifying the dynamic range of signal com-ponents in the pass-band region in response to signal compon-ents lying in the pass-band and stop-band regions, the dynamic modification means being less responsive to stop-band signal components as the level of the input signal rises.
According to another aspec-t, the invention contem-plates a circuit arrangement for modifying the dynamic range of an input slgnal, comprising: frequency~selec-tive circui-t means for dividing the frequency spectrum in which the input signal lies into pass-band and s-top-band regions, the pass-band frequency region sliding in response to signal components lying ln the pass-band and stop-band regions, the frequency selec-tive circuit means becoming less responsive -to s-top-band signal componen-ts as the level of the input signal rises, ~0 and means for modifying the dynamic range of siynal components in the pass band region.
In other words, a-t low input signal levels the clrcuit arrangement performs substantially as a conventional compressor or expander. However, at high input signal levels the compressor or expander action is modifled by the modula-tion control circuitry of the invention.
A side effect is to modify the input-output level transfer characteristic of the device at any particular fre-quency or combination of frequencies. The overall effect is unimportant and may even be unnoticeable at the dominant frequency in bi-linear systems. However, in logarithmic sys--tems the effect of modulation control, which is operativeprimarily in the high level portion of the dynamic range, is to cause a departure from a purely logarithmic characteris-tic. This may or may not be important in any particular appli-cation.
The invention derives from the observation that, ideally, in compressors and expanders the compression or expan-sion is responsive only -to the levels of signals within desired frequency pass-bands and not to the levels of signals at o-ther frequencies, which frequencies can be said to be in the stop-bands. For example, in an ideal circuit, compression or expan-sion should not be affected by the levels of signals outside the pass -7a-~8B9~
1 ¦ band of the fixed band or the pass band of the sliding band 2 ¦ (whether or not in its quiescent position). In the case of a 31 sliding band circuit in accordance with the invention, the amount 41 of frequency sliding of the variable band becomes no more than 51 is necessary to assure that a dominant controlling signal is not 61 boosted (in the case of compression) above a reference level.
71 As applied to bi-linear circuits, particularly those ~¦ in the dual path configuration, the invention takes further 91 advantage of an inherent characteristic of such circuits:
10¦ at high input signal levels the main path signal is substantially 11¦ larger than the signal(s) in the further (or side) path(s).
12¦ Consequently, high-level signal manipulations in the further 13¦ path are essentially inaudible, and, except for phase shifts, 14¦ are essentially not measurable (negligible level changes)O This 15¦ property of bi-linear circuits is most easily understood in the 16¦ context of a dual path circuit. However, the principle also 17¦ applies in single-path bi-linear circuits in which there are 18¦ two or more signal components in the same path instead of in 19¦ separately identifiable paths.
20¦ The invention takes advantage of the above observations 21¦ concerninq bi-linear circuit characteristics. As compared with 22 ¦ prior art bi-linear compressors and expanders, the invention 23 ¦ provides for further manipulations of the signal (modulation 24 ¦ control) in the high input signal level region where the overall 25 ¦ compressor or expander response is linear. The comparatively 26 ¦ low level noise reduction component of the signal is manipulated 27 ¦ in this extra way only at high signal levelsj thereby assuring 28 1 that any effects important to the signal channel will be over-29 shadowed by the large main signal component~
30 1 In dual-path bi-linear circuits, an efect of the 31 invention is to modify the transfer characteristic of the side 32 (or further) path such that the side path characteristic itself _~
1¦ becomes bi-linear instead of flattening or downturning at high 21 input signal levels. This is a consequence of the proportionality 31 aspect of modulation control. That is, at high input levels 41 the side path level does not drop below a selected proportion 51 of the main path level (e.g. one-quarter or one-tenth). This 61 is acceptable because the side path signal still remains 71 substantially smaller than the main path signal at high input 81 signal levels and because the stop-band is usually substantially 91 shifted in phase with respect to the main path signal channel.
For these same reasons, the invention may be embodied 11 in uni-linear circuits which have a linear response at high 12 si~nal levels.
13 From another point of view, the action of the invention 14 is to increase the levels of stop band signal components in the output of the device at high signal levels, but not ~o such an 16 extent as to cause problems with the recordin~ or transmission 17 channel since, relatively speaking they are still small.
18 Increasing the levels of such stop band signal components is not 1~ in itself a particularly advantageous thing but is necessary in order to obtain improved dynamic action and noise reduction 21 within the pass band~ Increasing the levels of s-top band 22 ~ignals in the output oE the device at high signal levels is 23 achieved by reducing the levels of stop band signal components 24 in the control signal channel at high signal levels, or by so arranging things that the control signal is generated as if 26 there was a reduced level of stop band signal components 2~ in the signal used to produce the control signal at high signal 28 levels (e.g., by filtering and limiting in the control circuits 29 or by fre~uency dependent control signal bucking arrangements).
A further advantage o the invention is that in 31 listening tests, "pumpiny effects" of single-ended compressors 3~ ///
_g_ 1¦ and expanders are substantially reduced if not eliminated. Thus, 21 in addition to its use in complementary noise reduction systems, I s~
~; 31 the invention is particularly useful for use in ~ com-41 pressors and expanders (iOe., compressors for use in compressing 51 signals that are not subsequently expanded and expanders for use 6¦ in expanding signals that were not previously compressed)O
71 By way of background of the invention~ although various 81 practical embodiments of noise reduction circuits have proven 91 successful, in operation such circuits depart in some degree 10¦ from the ideal because of the problem of stop-band signals 11¦ unduly controlling compression and expansion. The efEect of 1~¦ such shortcomings is manifested in several interrelated ways:
13 ¦ 1) a reduction in noise reduction effect in a 14 ¦ portion of the noise reduction system pass band;
15 ¦ 2) noise modulation effects (eug., the level 16 ¦ of a signal at one frequency, modulating the noise 17 ¦ level in a different part of the frequency spectrum) 18 3) signal modulatiorl effects (e.g., the level of a 19 ¦ signal at one fre~uency modulating the level of a signal at another frequency);
21 4) cross modulation effects (e.g. spurious modulation 22 products resulting from one of more of the last two 23 enumerated modulation effects)~
24 The degree to which these shortcomings are observable 2~ depends on the type of circuits employed in the noise reduction 2~ system, the recording and playback equipment, the record/playback 27 channel or medium and the nature of the signal material. In 28 many cases, the shortcomings are essentially unobservable except 29 ¦ by test instruments. Nevertheless, it is desirable to deal with 30 ¦ these shortcomings. Because the aforementioned shortcomings of 31¦ known compressors~ expanders and noise reduction systems rela-te 32 I to modulation effects, ei-ther of signals or of noise~ -the _~o_. , ~ 6 1¦ invention described herein for reducing such shortcomings is 21 referred to as modulation control.
31 The severity of these modulation effects depends to a 41 great extent on the uniformity of the transmission channel 51 between the compressor and expander. For example, in magnetic 61 tape recording and playback systems, a frequency response 7 phenomenon known as ~Ihead bumps" exists. Even in professional 8 systems, particularly those operating at 30 ips, the playback 9 response below 100 Hz is nonuniform due to the relationship between the signal wavelength on tape and the playback head 11 dimension, which are of the same orderO If the compressor/
12 expander system is susceptible to signals in the head bump 13 region, such signals when played back may control the expander 14 in a non-complementary way such that signals or noise at higher frequencies, e.g., up to 3 kHz, may be modulated by the signals 16 in the re~ion of or below 100 Hz~
17 In prior art fixed band (single band and multi-band) 18 circuits, various filterin~ techniques have been used to minimize 19 the control of compression and expansion by undesired signals.
~0 According to these techniques, sharp filters (e.g.) with steep 21 skirts) are placed in the signal path or in the control circuit ~ ~of the limiting device).
23 However, the use of signal path filters sharper than ~ 6 dB/octave (e.g., single pole filters) in multiband compressors and expanders causes amplitude and phase efects such that when 26 the overall signal spectrum is recombined there are amplitude V and phase errors. This problem is greatly exacerbated if 28 filters sharper than 12 dB/octave are employed. However, a ~91 filter slope of only 6 or 12 dB/octave may not adequately 30 ~ discriminate against all unwanted signals. In the multi-band 311~ (fixed band) bi~linear circuit examples of US~PS 3,846,719 and 321 in'Journal of the Audio _ i eerinq Societ~, Vol. 15, No~ 4, l¦ October, 1967, pp. 383-388, filters having a 12 ds/octave slope 21 are used in the signal path of three of the four fixed bands. A
31 flat overall frequency response is obtained only by the use of a 41 complex filter characteristic in the frequency band adjacent the 51 sharp filters~ Such a solution obviously is not universally 61 applicable.
71 In the logarithmic multi-band (fixed band) compressor/
81 expander circuit described in Rundfunktechnl Mitteitun~
91 Jahr. 22 (1978) H~ 2, pp. 63-74, the input signal is divided l0¦ into four bands by single pole filters. ~oweverl th~ control ll¦ circuits for each band employ sharp 18 dB/octave fil~ers. A
12¦ sharp eontrol circuit filter (12 dB/octave~ is also employed 13¦ in a single fixed band compressor/expander circuit sold under 14¦ the trademark "dbx II." However, the use of sharp control l5¦ eircuit filters results in excessive amplification of high 16¦ level signals outside ~he control circuit filter pass band 17¦ when high amplitude signals are not present within the control l~¦ circuit filter pass band, resulting in the posslble overdriving l9¦ of the transmission channel unless sharp eutof~ filters are used 20¦ in the signal ehannel as well.
21¦ A prior art technique referred to as spectral skewing 22¦ is described in Belgian-PS 889,427; Audio, May 198l, pp~ 20-26;
~31 and paper J-6 and preprint presented at November 1981 Conven-24¦ tion~ Audio Engineering Society, New York, New York. Spectral 251 skewing is also concerned ~ith the suppression of modulation 2S¦ effects resulting from compressor/expander non-complementarity 27 ¦ due to transmission channel errors. According to the teachings 28 ¦ of spectral skewing, sharp filtering is provided at least in the 29 ¦ compressor at a frequency well within the normal bandpass of 30 ¦ the system and within the flat response region of the transmis-31 sion channel. While spectral skewing is successful in reducing 32 ! ///
~ -12-l ll 1 spurious sign~l modulation effects caused by channel irregulari-2 ties, it does not address the problem of excessive frequency 3 sliding in sliding band systems or of excessive attenuation in 4 fixed band systems~
S Thus, the present invention seeks to minimize the 6 control of expansion and compression by undesired signals without 7 the attendant side effects and/or complexity of the prior art.
8 Although measurable modulation effects are not totally 9 suppressed by the inven~ion, the effects of the invention in audio applications are supplemented by psychoacoustical masking 11 effects such that perceived effects are~ for most listeners and 12 musical material, inaudible. That is, only the modulation of a 13 signal (or signals) sufficiently spaced in frequency from the 14 modulating signal is perceived by the human ear. Such modulation is minimized by the present invention. While the modulation 16 of a signal (or signals) by another signal closely spaced in 17 frequency is less likely to be affected or improved by the 18 invention, such phenomena will likely not be perceived by the 19 ear because of two related effects:
a) a weak signal close in fre~uency to a strong 21 signal is masked by the strong signal such that the weak signal 22 is inaudible, or 23 b) if the closely spaced signal is audible before 24 compression or i9 increased in level by the compressor such that it becomes audible, then there is a psychoacoustic tolerance of 26 modulation effects because of the close frequency spacing.
27 Consequently, the human ear is not able to discern 28 modulation effects of signals at closely spaced frequencies and 29 thus the invention need not be fully effective for such signals.
The operating environment o the invention is a fixed 31 band or sliding band compressor or expander circuit in which 32 there is a variable circuit means, usually controlled by a DC
l l 1 control signal~ which is operative primarily in the lower part 2 of the overall dynamic range~ In accordance with the invention, 3 mod~lation control means are employed in the upper part of the 4 dynamic range to prevent the action of the variable circuit means from becoming any greater than is necessary to provide the 6 nominal required attenuation of dominant signals, whether such 7 signals have freauencies in the pass-band or in the stop-band.
81 In pract1ce, controlling the action of the variable circuit 91 means usually comprises operating upon the control signal 10¦ controlling the circuit means.
11¦ The modulation control may take the form of active or 12¦ passive control signal limiting means which become operative at 13¦ high signal levels or of means employing circuits which detect 14¦ the presence of high level signals and generate signals which 15¦ oppose the increase of the control signal level. Such control 16¦ signal limiting may take place in one or more frequency selective 17¦ control signal channels; if more than one, means are provided for 18¦ selecting or combining the control signals so as to provide the 19¦ variable circuit element with an optimal control signal When a 201 high level signal detection circuit~ or modulation control 21¦ generator, is usedr it may operate in various ways which will 22¦ give a measure of signal levels in at least the upper part of ~he 231 dynamic range. For example, the modulation control signal may be 24 ¦ derived from the input or output signal of ~he compressor or 25 ¦ expander. The modulation control signal in effect provides a 26 reference for the DC control signal applied to the variable cir-27 ¦ cuit element (VCA or voltage controlled filter~. The reference 28 ~ signal is combined in phase opposition with (eOgO, opposite 29 ¦ polarity or so as to buck) the DC control signal generated pri-30 1 marily in response to stop-band signal components to provide a 31 limit as to how large the control signal to the variable circuit 32 j IL181~ g6 1¦ element can become in response to signals in the stop-band, io~
21 outside the pass band of the fixed band or sliding band. In 31 practice this limit can be made relatively "hard" or relatively 41 I'soft". That is, continued increases in the control signal can be 51 rather abruptly stopped or allowed to continue at a reduced rate 61 The modulation control signal may also be derived from 7 the variable circuit (VCA or variable filter~ by measuring voltage 8 or current components of the variable circuit and, if necessary~
9 equalization in order to generate a signal usable in providing a limit as to how large the control signal to the variable circuit 11 can become in response to signals in the stop-band.
12 In terms of resul~sr the invention as applied to 13 either fixed band or sliding band devices provides a substantial 14 immunity to signals outside the pass band of the fixed band or the sliding band~ In ~he case of sllding band devices, the 16 invention provides a Eurther related advantage, i.eO~ the 17 sliding band slides only so far in response to a dominant signal 18 as is necessary to bring the gain at the signal fre~uency to 19 substantially unity, at least for levels at or above a reference level. The reference level is at or near the upper area of the 21 dynamic operating range of the device, such as within about 22 6~20dB of the maximum allowable level. Prior art sliding band 23 circuits are susceptible to excessive sliding such that the 24 variable filter corner frequency is pushed farther than is needed with hiyh level signals, causing not only potential 26 modulation effects but also resulting in a loss of noise reduc-27 tion effect in part of the spectrumO
~8 As applied to sliding band dual-path circuits, the 29 invention provides forr in the simplest embodiment, the rectifi-cation and smoothing of the input or output signal and the 31 combining of the resultant DC reference signal with the control 32 signal applied to the variable filter~ The level of the l ¦ reference signal can be set for a desired proportion limit on a 2 ¦ dominant further path signal relative to the corresponding 3 ¦ component in the main path signal~ ~or example, the modulation 41 control circuit may be made to operate such that in, say, the S ¦ upper 20dB of the dynamic range the limiter provides only that 6 ¦ attenuation required to keep the dominant signal component in 71 the further path held at a relatively constant proportion of 81 that component in the main path signal (e.g. 15 dB below).
9 ¦ As applied to fixed band dual-path circuits, the lO ¦ invention provides for, in the simplest embodimentl as in the ll¦ sliding band embodiment, the rectification and smoothing of the 12¦ input or output signal to generate a modulation control signal 13 ¦ that responds primarily to the high level signal components of 14 ¦ the input signal. However, in the case o fixed band circuits a 15 ¦ sharp filter is used in the pass-band control circuit to provide 16 a pass-band control signal. In addition a stop~band control 17 circuit is employed to provide a stop-band control signal. The l3 modulation control signal provides a reference for the stop band 19 control signal (i.e. opposes it at high signal levels).
The referenced stop-band control signal is compared with the 21 pass-band control signal and the two are combined generally to 22 favor the larger, via a maximum signal selection circu t, which 23 then controls the VCA The overall effect of the circuit is to 24 provide the required attenuation (overall compression law~ in the pass-band, while avoiding control of the pass-band attenuation 26 by larye signal components in the stop-band, and while avoiding 27 the possibility of excessive amplification of high level signals 28 in the stop-band, as seen at the output of the overall compressor 29 In these and other embodiments in which a reference bucking signal is generated, the signal may be derived from the 3l input or output because at high signal levels, where the inven-32 tion operates, the input and output levels are nearly the I
I ~ ~ame. In som mbodiments the modu1ation control signa1 may be 2 subjected to filtering or equali2ation before rectification.
3 Such equaliæation works in conjunction with the fixed or variable 4 filtering or equalization employed in the signal circuits and control circuits, to yield an overall modulation control which 6 is most effective in suppressing control by signal components in 7 the stop-band, while at the same time in~erfering as little as 8 possible with control by signal components in the pass-band.
9 Other embodiments of the invention are described hereinafter. For example, the amplified AC output of the 11 sliding band variable filter may be divided into two or more 12 band pass channels, each channel subjected to selected limiting 13 thresholds, rectified and combined to produce a control signal.
14 By selecting appropriate thresholds, the DC control circuit of ~ the sliding band compressor or expander then has a frequency 16 dependent maximum output characteristic that functions to 17 minimize the control of the compressor or expander by signals 18 outside the slidina pass band.
19 In a variation using only one control circuit channel, a low ~requency boost circuit is placed in the control amplifier~
21 This is followed by an amplitude limiter and then a high fre 22 ~uency boost circuit. The resulting AC signal is then rectified 23 and smoothed to form the control signal~
24 The invention will be described in more detail, by way of example, with reference to the accompanying drawings, in which:
26 Figure 1 is an exemplary set of curves showing com~ ¦
27 plementary bi-linear compression and expansion characteristicsO
28 Figure 2 is a schematic circuit diagram of a prior art 29 sliding band compressor.
Figure 3 is a schematic circuit diagram of a prior 31 art sliding band expander.
~ -17-l Figure 4 is a schematic circuit diagram of a modifica-21 tion to Figures 2 and 3.
3¦ Figure 5 is a block diagram of a prior art sliding 41 band compressor.
51 Figure 6 is a set of probe tone curves illustrating ~¦ the sliding band action of the circuit of Figures 2 and 4.
7 Figures 7 - 10 are a series of probe tone curves 81 illustrating the effects of modulation control according to the 91 invention embodied in a sliding band compressor.
lO¦ Figure 11 is a block diagram of a preferred embodiment ll¦ f the invention embodied in a sliding band compressor l~¦ Figures 12 - 15 are block diagrams of further embodi-13¦ ments o~ the invention embodied in slidin~ band compressors.
14¦ Figures 16 and 17 are block diagrams o a prior 15¦ art fixed band compressor and expander~
16¦ Figures 18 - 20 are response curves illustrating 17¦ the effects oE modulation control according to the invention 18¦ embodied in a sliding band compressor.
~91 Figure 21 is a block diagram of a preferred embodiment 20¦ of the invention embodied in a ixed band co~pressor.
21¦ Figure ~2 is a block diaaram of an alternative 22¦ embodiment of the invention embodied in a Eixed band compressor.
23 ¦ Exemplary bi-linear complementary compression and expan-24 ¦ sion transfer characteristics (at a particular frequency) are 25 ¦ shown in Figure 1, indicating (for the compression characteristic) 2~ ¦ the low level portion of substantially constant galn, the 27 ¦ threshold, the portion where dynamic actiorl occurs, the inishing 28 ¦ point, and the high level portion of s~bstantially constant gain~
~9 I Details of one dual path sliding band bi-linea circuit are set forth in Figures 2, 3 and 4~ The sliding band 31 I embodiments of the present invention are described with reference 32 I to this circuit, although the invention is not limited to use in j -18-l¦ such circuits. Figures 2, 3 and 4 are the same as Figure 4, 5 21 and 10 respe~tively of US-PS Re 28~426 and further details of 31 said circuits, their operation and theory are set orth therein.
41 Figure 5 is a block diagram of Figure 2 (with or without the ~¦ Figure 4 modification). The following description of Figures 2, 61 3 and 4 is taken in large part from US-PS Re 28,426.
7¦ The circuit of Figure 2 is specifically designed for 8 incorporation in the recording channel of a consumer tape 91 recorder, two such circuits being re~uired for a stereo recorder.
lO¦ The input signal is applied at ~erminal 10 to an emitter follower ll¦ stage 12 which provides a low impedance signal. This signal is 12¦ applied firstly through a main, straight-through path constituted 13¦ by a resistor 14 to an output terminal 16 and secondly through a 14¦ further path the last element of which is a resistor 18 also lS¦ connected to the terminal 16u The resistors 14 and 18 add the 16¦ outputs of the main and further paths to provide the required 17 ¦ compression law~
l8 ¦ The further path consists of a fixed filter 20, a 19 ¦ variable cut-off filter 22 including a FET 24 (these constituting the filter/limiter), and an amplifier 26 the output of which is 2~1 coupled to a double diode limiter or clipper 28 and to the ~¦ resistor 18. The non~linear limiter suppresses overshoots of 23 ¦ the output signal with abruptly increasing input signals. The 24 ¦ amplifier 26 increases the signal in the further path to a level 25 ¦ such that the knee in the characteristic of the limiter or 26 ¦ overshoot suppressor 28, comprising silicon diodes, is effective 27 ¦ at the appropriate signal level under transient conditionsO
28 ¦ The effective threshold of the overshoot suppressor is somewhat 29 above that of the syllabic filter/limiter. The resistors 14 and 18 are so proportioned that the required compensating degree of 31 attenuation is then provided for the signal in the further path~
-19 ll l The output of the amplifier 26 is also coupled to an 2 amplifier 30 the output of which is ~ectified by a germanium 3 diode 31 and integrated by a smoothing filter 32 to provide 4 the control voltage for the FET 24.
Two simple RC filters are used, though e~uivalent LC
6 or LCR filters could be used. The fixed filter 20 provides a 7 cut-off frequency of 1700 Hz (now 1500 Hz), below which diminish-8 ing compression take place. The filter 22 comprises a serles 9 capacitor 34 and shunt resistor 36 followed by a series resistor 38 and the FET 24, with i~s source-drain path connected as a ll shunt resistor. Under quiescent conditions with zero signal on 12 the gate of the FET 24, the FET is pinched off and presents 13 substantially infinite impedance; the presence of the resistor 14 38 can then be ignored. The cut-off frequency of the filter 22 is thus 800 Hz tnow 750 Hz), which it will be noted is substan~
16 tially below the cut-of frequency of the ixed filter 20.
17 When the signal on the gate increases sufficiently for l8 the resistance of the FET to fall to less than say 1 K, the 19 resistor 38 effectively shunts the resistor 36 and the cut-ofE
re~uency rises, markedly narrowing the pass band of the filter.
21 The rise in cut-off frequency is of course a progressive action.
22 The use of a FET is convenient because, within a suit-23 able restricted range of signal amplitudes, such a device acts 24 substantially as a linear resistor (for either polarity signal~, the value of which is determined by the control voltage on the 26 gate.
27 The resistor 36 and FET are returned to 2n adjustable 28 tap 46 in a potential divider which includes a temperature 29 ¦ compensating germanium diode 4~. The tap 46 enables the compres-30 ¦ sion threshold of the filter 22 to be adjusted~
311¦ The amplifier 26 comprises complementary transistors 3~11 giving high input impedance and low output impedance~ Since Il , ~ 3~
1 the amplifier drives the diode limiter 28, a finite output 2 impedance is required and is provided by a coupling resistor 3 500 The diodes 28 are, as already noted, silicon diodes 4 and have a sharp knee around 1/2 volt. I
The signal on the limiter and hence on the resistor 18 6 can be shorted to ground by a switch 53 when it is required to 7 switch the compressor out of action.
8 The amplifier 30 is an NPN ~is$~ with an emitter time 9 constant network 52 givinq increased gain at high freauenciesO
Strong high frequencies ~eag. a cymbal crash) will therefore 11 lead to rapid narrowing of the band in which compression takes 12 place~ so as to avoid signal distortionO
13 The amplifier is coupled to the smoothing filter 32 14 through the rectifying diode 310 The filter comprises a series resistor 54 and shunt capacitor 56. The resistor 54 is shunted 16 by a silicon diode 58 which allows rapid charging of the 17 capacitor 56 for fast attack, coupled wi~h good smoothing under 13 steady-state conditions. The voltage on capacitor 56 is applied 19 d.irectly t~ the gate of the FET 24.
A complete circuit diagram of the complementary expander 21 is provided in Fiyure 3, but a full description is not required as 22 substantially as the circuit is identical to Figure 2, component 23 values, are therefore not for the most part shown in Figure 3.
24 The differences between Figures 2 and 3 are as follows:
In Fi~ure 3, the further path derives its input from 26 the output terminal 16a, the amplifier 26a is inverting, 27 and the signals combined by the resistors 14 and 18 are applied 28 to the input (base) of the emitter follower 12, -the oukput 291 (emitter) of which is coupled to the terminal 16aO To ensure low driving impedance, the input terminal 1Oa i5 coupled to the 31¦ resistor 14 through an emitter follower 60. Suitable measures 32~ must be taken to prevent bias getting in the expander~
¦ -21-, 118!399~i 1¦ The ampliEier 26a is rendered inverting by taking the 21 output from the emitter, instead of the collector, of the second 31 ~PNP) transistor. This alteration involves shifting the 10 K
41 resistor 62 ~Figure 2) from the collector to the emitter ~Figure 51 2~, which automatically gives a suitable output impedance for I driving the limiter. The resistor 50 is therefore omitted in 71 Figure 3.
81 It should be noted that it is important in aligning 91 a complete noise reduction system to have equal signal levels 10¦ on the emitters of the transistors 12 in both compressor 11¦ and expander. Metering terminals M are shown connected to 12¦ these emitters.
13 Figure 4 shows a preferred circuit, for replacing the 14 circuit between points A, B and C in Figures 2 and 3. When the FET 24 is pinched off~ the second RC network 22 is inoperative, 16 and the first RC network 20 then determines the response of the 17 further path. The improved circuit combines the phase advantages 18 of having only a single RC section under quiescent conditions 19 with the 12 dB per octave attenuation characteristics of a two-section RC filter under signal conditions.
21 In the practical circuit, using MPF 104 FE~'s, the 39 K
22 resistor 36a is necessary in order to provide a finite source 23 impedance to work into the FET. In this way the compression 24 ratio at all frequencies and levels is held to a maximum of about 2u The 39 K resistor 36a serves the same compression 26 ratio limiting function in the improved circuit as the resistor 27 36 in the circuit of Figure 2 or Figure 3. In addition, this 28 resistor provides a low frequency path for the signal. I
29 Certain details of the circuit oE Figures 2, 3 and 4 have evolved over the years and more modern forms of the circuit 31 have been published and are well known in the ar-t. Reference to ~ -22--l the specific circuit in US-PS Re 28,426 is made for convenience 2 in presentation.
3 Figure 5 is a block diagram showing the major elements 4 of the compressor of Figures 2 and 4. Combining circuit 15 S represents the combining resistors 14 and 18 of Figures 2 and 3.
b The variable band action of the sliding band device 7 can be seen in Figure 6, showing an actual chart recorder probe 8 tone response obtained from the circuit of Figure 2 incorporating 9 Figure 4. The variable band action is shown by plotting the compressor frequency response by means of a low-level probe tone ll (the level of which is below the compressor threshold) in the 12 presence of a high-level signal; the probe is detected at the 13 compressor output by means of a tracking filter. The high-level 14 signal causes the compressor circuitry to operate, the graph showing the effect on the turnover frequency of the filter.
16 In a sliding band device in accordance with the 17 invention, the amplitude of the high level or dominant signal 18 that causes the sliding band action should not cause excessive 19 sliding, nor should the presence of other high level signals outside of the sliding band pass band cause excessive sliding.
21 Excessive slidinq means movement of the variable filter turnover 22 fre~uency farther than necessary to produce a sliding band 2~ compressor characteristic which avoids boosting the dominant 24 signals above a reference level. The absolute value of the reference level is chosen by the system designer, but is usually 26 some 1OdB below the highest levels normally used.
27 Figure 7 shows a further se~ of actual chart recorder 281 probe tone curves for the case of a sliding band compressor 29 circuit similar in design to that of Figure 2 (with the Figure 4 30 I modification), but with a low level gain of 8 dB and a filter 31 quiescent fre~uency of 800 Hz. The probe tone level is at -40 32 dB, below the compressor threshold. Curves are taken for a 1 100 Hz signal at -20~ -10, 0, ~10 and +20 dB9 where 0 dB is the 21 reference level. Also, a curve for no 100 Hz signal is shown.
3¦ The ~10r 0, ~10 and +20 dB chart recording curves are all started 41 at about 200 Hz. This is also the case for Figure 8~ In 51 Figures 9 and 10 there are also curves for the no signal condition ~¦ Referring again to Figure 7, ideally~ there should be 71 no sliding in response to a 100 ~z signal because it is well 81 outside the pass band of the circuit at its lowest (quiescent) 91 frequency. Nevertheless, as the 100 Hz signal increases in 10¦ level, the band slides upward. The -10, 0, ~10 and ~20 dB
11 curves need not slide any farther than the -20 dB curve in order 12¦ to avoid any substantial boosting of the 100 Hz signal. The 13¦ unnecessary sliding has two effects: a) substantial noise 14¦ reduction action is lost (during playback) because no boosting 15¦ takes place at frequencies where it otherwise can take place and 16¦ b) as the amplitude of the 100 Hz signal varies it can modulate 17¦ signals at higher frequencies as the sliding band varies under 18¦ its control~ resulting in possible incorrect restoration of the 19¦ signal by the expander if the recording or transmission channel 20¦ has an irregular frequency response in the vicinity of l00 Hz.
21¦ Figure 8 shows a set of actual chart recorder probe 22¦ tone curves for the same circuit, but with the addition of 231 modulation control circuitry as described hereinafterO Essen-?4¦ tially no sliding occurs for the same levels o~ the 100 ~æ
25¦ signal as in the Figure 7 arrangement. The sliding band 261 compressor is made essentially immune to strong signals outside 27 its pass band. The sliding band response is essentially the 28 ¦ same as its response below threshold in the presence of no 29¦ dominant signals.
30¦ The effect of modulation control for sliding band 31 ¦ compressors is further illustrated by Figures 9 and 10, which 32 are also actual chart recorder probe tone curves taken with the 1 ~2~
l l l l l ¦ same circuit and probe tone level as with Figures 7 and 8~ ln ~ ¦ this case, the effect of a dominant signal at 800 Hz, a fre-3 ¦ quency within the desired active area of the circuit, is shown 4 ¦ Ideally, slidinq is required to go only so far as not to boost
5 ¦ the 800 Hz signal above the 0 dB reference levelO Thus, in the
6 ¦ Fi~ure 9 response, without modulation control, the sliding71 produced by ~he 800 Hz signal at levels of -10, 0, ~10 and +20 81 dB are excessive. Figure 10 illustrates the response of the 91 circuit with modulation control: sliding at and above 0 dB is l~¦ greatly reduced. The effect is progressively reduced for low lll signal levels but is o~servable to some extent at the -10 dB
12¦ signal level~
13¦ Figure 11 shows generally a preferred embodiment of 14¦ the modulation control of the present invention embodied in a 15¦ dual path bi-linear sliding band device. Reference numerals 16¦ are, so far as possible, kept the same as in Figure 5 for the 17¦ same and functionally similar elements. The probe tone response l~¦ curves of Figures 7 10 are taken from a sliding band device l9¦ generally as shown in Figure 1l, with the modulation control 20¦ sub-circuitry elements within dashed block 100 taken out of 21¦ the circuit for the response curves without modulation control.
22¦ For purposes of explanation, the detailed circuitry of Figure 11 231 is essentially the same as that of Figures 2 and 4. The circuit 24 may be modified as described hereinbefore witho~t affecting the 251 basic operation of the modulation control sub-circuit.
26¦ As shown in Figure 11, the modulation control sub-~71 circuit derives a DC control signal from the circuit input (or~
28¦ optionally from the output of combining circuit 15) by means of 291 an amplifier 30'~ rectifier 31' and smoothing circuitry 32a'0 301 Potentiometer 102 is shown to indicate that the signal from 31¦ smoothing circuitry 32a' has a controlled gain. In practice 32~ ///
l -25-1181~
1¦ the gain is usually pre-set in the design. A combining circuit 21 33 subtracts the signal provided by the sub-circuit 100 from the 31 main control signal provided by way of the amplifier 30, rectifier 4¦ 31 and s~oothing circuit 32a'a 51 The smoothing circuitry of Figure 11 is broken into 61 two stages in order to minimize the cost of circuit components~
12¦ signal level~
13¦ Figure 11 shows generally a preferred embodiment of 14¦ the modulation control of the present invention embodied in a 15¦ dual path bi-linear sliding band device. Reference numerals 16¦ are, so far as possible, kept the same as in Figure 5 for the 17¦ same and functionally similar elements. The probe tone response l~¦ curves of Figures 7 10 are taken from a sliding band device l9¦ generally as shown in Figure 1l, with the modulation control 20¦ sub-circuitry elements within dashed block 100 taken out of 21¦ the circuit for the response curves without modulation control.
22¦ For purposes of explanation, the detailed circuitry of Figure 11 231 is essentially the same as that of Figures 2 and 4. The circuit 24 may be modified as described hereinbefore witho~t affecting the 251 basic operation of the modulation control sub-circuit.
26¦ As shown in Figure 11, the modulation control sub-~71 circuit derives a DC control signal from the circuit input (or~
28¦ optionally from the output of combining circuit 15) by means of 291 an amplifier 30'~ rectifier 31' and smoothing circuitry 32a'0 301 Potentiometer 102 is shown to indicate that the signal from 31¦ smoothing circuitry 32a' has a controlled gain. In practice 32~ ///
l -25-1181~
1¦ the gain is usually pre-set in the design. A combining circuit 21 33 subtracts the signal provided by the sub-circuit 100 from the 31 main control signal provided by way of the amplifier 30, rectifier 4¦ 31 and s~oothing circuit 32a'a 51 The smoothing circuitry of Figure 11 is broken into 61 two stages in order to minimize the cost of circuit components~
7 Thus, blocks 32a and 32a' may be identical and each may comprise ~¦ only a single RC filter section and block 32b which further 91 smooths the combined control signal comprises a further RC
10¦ filter sectionO
11¦ The signals are rectified to DC (by rectifiers 31 and 12¦ 31') before they are combined by the circuit 33 in order to ,31 avoid the polar-ity ambiguity that would result if AC signals 14¦ were combined and then rectified (i.e., with AC signals there 15¦ would be two possible stable states).
16¦ The arrangement of the embodiment of Figure 11 thus 17¦ provides a reference level for stabilization of the DC control 18¦ signal, a reference level that is dynamically changing with 19¦ input signal level, thereby shifting or transposing part of the 20¦ dynamic action of the variahle filter to a level region deter-21¦ mined by the reference level~ The arrangement functions to keep ~¦ the maximum amplitude of c]ominant signals in the noise reduction 231 side path at a constant proportion of the input signal at high 24 ¦ signal levels~ The relative level from the modulation control 25 ¦ sub circuit 100 is selected to minimize sliding action in 26 ¦ response to signals outside the sliding band pass band~
27 ¦ Although the embodiment of Figure 11 functions 28 ¦ effectively when the input to the modulation control sub-circuit 2~ ~ 100 is taken from the wide band inpu-t (or output), other arrange-30 I ments giving a measure of signal levels at the top end of the 31 dynamic range are possible~ For example, some modulation ~ -26-l ~, .$~
control 0fects are obtained even if the sub-circuit lOO input is taken from the output of band-pass filter 20. Ideally, equalization is employed in ampli~iers 30 and 30' to optimi~e the overall modulation control effects (control by pass-band components versus by stop-band components), taking into account the combined frequency response effects of filters 20, 22, and the equalization employed in control amplifier 26.
When the invention is embodied in series connected devices such as set forth in selgian-PS 889~428J a single modulation control sub-circuit may be used to provide a reference signal to each stage. Such a circuit advailt-ageously derives its input from the output of the last compressor stage whenthe series stages are arranged in the preferred order such that the first stage has the highest level threshold. By deriving the reference signal from the output, the low level stage(s) receive the modulation control effect at lower signal levels, thus enhancing the modulation control action.
As mentioned previously, it is also possible to achieve modulation control o:E sliding band circuits by other means than by deriving a control signal reference from the input (or output) signal. One or more control signals can be deri.ved from the variable filter output and limited so as to achieve results similar to those achieved by the bucking embod:iment o:E
~igure 11; the essential result is the same, namely to de-sensitize the dynamic modification action of the circuit to high level signals with;n the stop-b~md. Figures 12, 13, 14 cmd 15 are directed to SUC]l elllbOd:illlents employing limitlng.
In the embodiment of Figure 12, the control signal generating means (blocks 30, 31 and 33 in Figure 5) is split into three paths by ampli-iers 30, 116, and 124 cmd filters 110~ 118, and 126, namely a high fre-quency path, a mid-frequency , ~. . ;,.
l l l path and a low frequency path Each path incl~des a limiter 21 (112, 120, 128) that has a pre-set thresholdO The limiters can 3¦ be back to back diodes such as diodes 28 in Figure 2. For a 41 high frequency audio compressor having a perEormance generally 5¦ as shown in Fi~ures 7 to 10, the filter frequencies may be as 6 follows9 for example: filter 126, 200 Hz low pass, filter 118, 7l 200 to 800 Hz band pass; and filter 110, 800 Hz high pass. The 81 output of each limiter is rectified by rectifiers 114, 122, and ~¦ 130, combined (or maximum value selected) and applied to smooth l0¦ ing network 32. Alternatively, the ]imiting Eunctions can be ll¦ provided after rectificationO In operation, the low frequency 12¦ and mid-frequency band limiters are set to minimize the effect 13¦ on sliding by signals outside the pass band. Little or no 14¦ limiting may be re~ulred in the high frequency path, and the 15¦ control effected by this path may be enhanced by providing the ~6l amplifier 30 with high frequency boost, as represented by block 171 52.
l~ Figure 13 shows a further split path control circuit 19¦ embodiment. In this example~ two paths are employed, a high 20¦ frequency path and a low frequency path. The high frequency 21¦ path is essentially the ~ame as in the embodiment of Figure 12, 22¦ except that the limiter 112 is omitted. The low frequency path 231 has an amplifier 132 that has a high frequency attenuation 24¦ networ~ 134. The amplifier output is applied to a low pass 25 ¦ filter 136 and to a limiter 138. The limiter threshold is 26 I set along with the various filter and amplifier filter character- !
27 ¦ istics to achieve the best immunity from sliding band control by 28 ¦ stop-band signals~ The signals in the two paths are rectiEied 29 ¦ by rectifiers 114 and 140 and combined at the input to the 30 ¦ smoothing circuit 3~
31 ¦ A further simplified embodiment of the Figure 13 32 I embodiment is shown in Figure 14. The high pass filter 110, l -28-I
~-l¦ the low pass Eilter 136 and the amplifier hiqh frequency 21 attenuation network 134 are omitted. The high frequency pre-31 emphasis network 52' of amplifier 30 is modified from tha~ of 4¦ network 52 such that the hiqh frequency boost becomes effective 51 at a hi~her fre~uency. Consequently only the wide band path 61 containing amplifier 132 carries low fre~uency signals (along 7~ with high fre~uency signals). The threshold of limiter 138 is 81 adjusted along with the high frequency boost characteristics of 91 network 52' to minimize the effect on sliding by stop-band signals~
ll Figure 15 shows an embodiment having a single path 12 control circuit which includes a frequency dependent amplifier 13 141 having a low frequency boost network 142, followed by a 14 limiter 144 and an amplifier 146 with a hi~h frequency boost network 148. In operation the low frequency portion of the 16 spectrum which tends to cause undesirable sliding is first 17 boosted and then limited. Limiter 144 is preferably syllabic 18 with its own closed loop amplifier, rectifier, smoothing circuit 19 and controlled gain element (such as blocks 276, 280, 282 and 270 in Figure 17). Amplifier 146 having a high frequency boost 21 network 148 restores any high frequency pre-emphasis which may 22 be required. The amplifier 146 output is then rectified and 23 smoothed by blocks 114 and 32y respectively. In this single 24 path control circuit the high level stop-band signal components are significantly reduced at the rectification point 114n 26 For convenience and simplicity the sliding band 27 embodiments have been described in connection with a particular 28 configuration of sliding band compressor. The invention is 29 equally applicable to expanders, with no change in the noise reduction further path control circuits shown in the embodiments 3l of Figures 11 - *~. In noise reduction systems employing 32 compressors and expanders, it is preferred that the modulation -29~
~ ', 1 control invention be applied to both devices to assure com-2 plementarity. The invention is also equally applicable to low3 frequency sliding band circuits, in which the compression and 4 expansion action is designed to occur in the low frequency region~
6 Figure 16 shows a block diagram of a fixed band dual 7 path bi-linear compressor and expander conEiguration. The ~ Eundamental aspects of this system are disclosed in US-PS
9 ~r~467S~-, US-PS 3,903,485 and in Journal of the Audio Engineeriny Society, Vol. 15, No. 4, October, 1967, pp. 383-388.
llIn the known embodiment of Figure 16, the further path 12 networks 250 provide four bands. Bands 1, 3 and 4 have conven-13 tional 12 dB/octave input filters: an 80 Hz low pass filter 252 14 at the input of band 1, a 3 kHz high pass Eilter 254 at the 15input of band 3 and a 9 kHz high pass filter 256 at the input of 16 band 4. Each is followed by an emitter follower isolation stage 17 258. Band 2 has a frequency response which is complementary to 18 that of bands 1 and 3. Such a response is derived by adding tin 19 adder 260) the outputs of the emitter followers 258 in bands 1 and 3 and subtracting that sum from the overall input signal (in 21 subtractor 26~)~ The output of emitter follower 258 in each 22 band and the output of subtractor 262 are applied to respective 23 limiters 264 and 2641 ~ Limiters 264 and 264' are identical 24 except that limiters 2641 in bands 1 and 2 have time constants twice those in bands 3 and ~. The outputs of bands 1-4 are 26 combined with the main path signal in combiner 266r The com-27 pressor output is applied to a noisy channel for transmission to 28 the complementary expander in which the output of the identical 29 further path networks are subtracted Erom the input signal to provide the complementary expansion characteristicO
31 Figure 17 shows further details of the limiters 264 32 and 264'. Each includes an FET attenuator 270 that operates ~ ~30-I
l in response to a control signal. The attenuator output is 2 amplified by si~nal amplifier 272, the gain of which is set 3 to provide the desired low level signal gain~ The outputs 4 of all the bands are combined with the main signal in such a way as to produce a low level output from the compressor which 6 is uniformly 10 dB higher than the input signal up ~o about 7 5 kHz, above which the increase in level rises smoothly to
10¦ filter sectionO
11¦ The signals are rectified to DC (by rectifiers 31 and 12¦ 31') before they are combined by the circuit 33 in order to ,31 avoid the polar-ity ambiguity that would result if AC signals 14¦ were combined and then rectified (i.e., with AC signals there 15¦ would be two possible stable states).
16¦ The arrangement of the embodiment of Figure 11 thus 17¦ provides a reference level for stabilization of the DC control 18¦ signal, a reference level that is dynamically changing with 19¦ input signal level, thereby shifting or transposing part of the 20¦ dynamic action of the variahle filter to a level region deter-21¦ mined by the reference level~ The arrangement functions to keep ~¦ the maximum amplitude of c]ominant signals in the noise reduction 231 side path at a constant proportion of the input signal at high 24 ¦ signal levels~ The relative level from the modulation control 25 ¦ sub circuit 100 is selected to minimize sliding action in 26 ¦ response to signals outside the sliding band pass band~
27 ¦ Although the embodiment of Figure 11 functions 28 ¦ effectively when the input to the modulation control sub-circuit 2~ ~ 100 is taken from the wide band inpu-t (or output), other arrange-30 I ments giving a measure of signal levels at the top end of the 31 dynamic range are possible~ For example, some modulation ~ -26-l ~, .$~
control 0fects are obtained even if the sub-circuit lOO input is taken from the output of band-pass filter 20. Ideally, equalization is employed in ampli~iers 30 and 30' to optimi~e the overall modulation control effects (control by pass-band components versus by stop-band components), taking into account the combined frequency response effects of filters 20, 22, and the equalization employed in control amplifier 26.
When the invention is embodied in series connected devices such as set forth in selgian-PS 889~428J a single modulation control sub-circuit may be used to provide a reference signal to each stage. Such a circuit advailt-ageously derives its input from the output of the last compressor stage whenthe series stages are arranged in the preferred order such that the first stage has the highest level threshold. By deriving the reference signal from the output, the low level stage(s) receive the modulation control effect at lower signal levels, thus enhancing the modulation control action.
As mentioned previously, it is also possible to achieve modulation control o:E sliding band circuits by other means than by deriving a control signal reference from the input (or output) signal. One or more control signals can be deri.ved from the variable filter output and limited so as to achieve results similar to those achieved by the bucking embod:iment o:E
~igure 11; the essential result is the same, namely to de-sensitize the dynamic modification action of the circuit to high level signals with;n the stop-b~md. Figures 12, 13, 14 cmd 15 are directed to SUC]l elllbOd:illlents employing limitlng.
In the embodiment of Figure 12, the control signal generating means (blocks 30, 31 and 33 in Figure 5) is split into three paths by ampli-iers 30, 116, and 124 cmd filters 110~ 118, and 126, namely a high fre-quency path, a mid-frequency , ~. . ;,.
l l l path and a low frequency path Each path incl~des a limiter 21 (112, 120, 128) that has a pre-set thresholdO The limiters can 3¦ be back to back diodes such as diodes 28 in Figure 2. For a 41 high frequency audio compressor having a perEormance generally 5¦ as shown in Fi~ures 7 to 10, the filter frequencies may be as 6 follows9 for example: filter 126, 200 Hz low pass, filter 118, 7l 200 to 800 Hz band pass; and filter 110, 800 Hz high pass. The 81 output of each limiter is rectified by rectifiers 114, 122, and ~¦ 130, combined (or maximum value selected) and applied to smooth l0¦ ing network 32. Alternatively, the ]imiting Eunctions can be ll¦ provided after rectificationO In operation, the low frequency 12¦ and mid-frequency band limiters are set to minimize the effect 13¦ on sliding by signals outside the pass band. Little or no 14¦ limiting may be re~ulred in the high frequency path, and the 15¦ control effected by this path may be enhanced by providing the ~6l amplifier 30 with high frequency boost, as represented by block 171 52.
l~ Figure 13 shows a further split path control circuit 19¦ embodiment. In this example~ two paths are employed, a high 20¦ frequency path and a low frequency path. The high frequency 21¦ path is essentially the ~ame as in the embodiment of Figure 12, 22¦ except that the limiter 112 is omitted. The low frequency path 231 has an amplifier 132 that has a high frequency attenuation 24¦ networ~ 134. The amplifier output is applied to a low pass 25 ¦ filter 136 and to a limiter 138. The limiter threshold is 26 I set along with the various filter and amplifier filter character- !
27 ¦ istics to achieve the best immunity from sliding band control by 28 ¦ stop-band signals~ The signals in the two paths are rectiEied 29 ¦ by rectifiers 114 and 140 and combined at the input to the 30 ¦ smoothing circuit 3~
31 ¦ A further simplified embodiment of the Figure 13 32 I embodiment is shown in Figure 14. The high pass filter 110, l -28-I
~-l¦ the low pass Eilter 136 and the amplifier hiqh frequency 21 attenuation network 134 are omitted. The high frequency pre-31 emphasis network 52' of amplifier 30 is modified from tha~ of 4¦ network 52 such that the hiqh frequency boost becomes effective 51 at a hi~her fre~uency. Consequently only the wide band path 61 containing amplifier 132 carries low fre~uency signals (along 7~ with high fre~uency signals). The threshold of limiter 138 is 81 adjusted along with the high frequency boost characteristics of 91 network 52' to minimize the effect on sliding by stop-band signals~
ll Figure 15 shows an embodiment having a single path 12 control circuit which includes a frequency dependent amplifier 13 141 having a low frequency boost network 142, followed by a 14 limiter 144 and an amplifier 146 with a hi~h frequency boost network 148. In operation the low frequency portion of the 16 spectrum which tends to cause undesirable sliding is first 17 boosted and then limited. Limiter 144 is preferably syllabic 18 with its own closed loop amplifier, rectifier, smoothing circuit 19 and controlled gain element (such as blocks 276, 280, 282 and 270 in Figure 17). Amplifier 146 having a high frequency boost 21 network 148 restores any high frequency pre-emphasis which may 22 be required. The amplifier 146 output is then rectified and 23 smoothed by blocks 114 and 32y respectively. In this single 24 path control circuit the high level stop-band signal components are significantly reduced at the rectification point 114n 26 For convenience and simplicity the sliding band 27 embodiments have been described in connection with a particular 28 configuration of sliding band compressor. The invention is 29 equally applicable to expanders, with no change in the noise reduction further path control circuits shown in the embodiments 3l of Figures 11 - *~. In noise reduction systems employing 32 compressors and expanders, it is preferred that the modulation -29~
~ ', 1 control invention be applied to both devices to assure com-2 plementarity. The invention is also equally applicable to low3 frequency sliding band circuits, in which the compression and 4 expansion action is designed to occur in the low frequency region~
6 Figure 16 shows a block diagram of a fixed band dual 7 path bi-linear compressor and expander conEiguration. The ~ Eundamental aspects of this system are disclosed in US-PS
9 ~r~467S~-, US-PS 3,903,485 and in Journal of the Audio Engineeriny Society, Vol. 15, No. 4, October, 1967, pp. 383-388.
llIn the known embodiment of Figure 16, the further path 12 networks 250 provide four bands. Bands 1, 3 and 4 have conven-13 tional 12 dB/octave input filters: an 80 Hz low pass filter 252 14 at the input of band 1, a 3 kHz high pass Eilter 254 at the 15input of band 3 and a 9 kHz high pass filter 256 at the input of 16 band 4. Each is followed by an emitter follower isolation stage 17 258. Band 2 has a frequency response which is complementary to 18 that of bands 1 and 3. Such a response is derived by adding tin 19 adder 260) the outputs of the emitter followers 258 in bands 1 and 3 and subtracting that sum from the overall input signal (in 21 subtractor 26~)~ The output of emitter follower 258 in each 22 band and the output of subtractor 262 are applied to respective 23 limiters 264 and 2641 ~ Limiters 264 and 264' are identical 24 except that limiters 2641 in bands 1 and 2 have time constants twice those in bands 3 and ~. The outputs of bands 1-4 are 26 combined with the main path signal in combiner 266r The com-27 pressor output is applied to a noisy channel for transmission to 28 the complementary expander in which the output of the identical 29 further path networks are subtracted Erom the input signal to provide the complementary expansion characteristicO
31 Figure 17 shows further details of the limiters 264 32 and 264'. Each includes an FET attenuator 270 that operates ~ ~30-I
l in response to a control signal. The attenuator output is 2 amplified by si~nal amplifier 272, the gain of which is set 3 to provide the desired low level signal gain~ The outputs 4 of all the bands are combined with the main signal in such a way as to produce a low level output from the compressor which 6 is uniformly 10 dB higher than the input signal up ~o about 7 5 kHz, above which the increase in level rises smoothly to
8 15 dB at 15 kHz.
9 The FET attenuator is controlled by a control signal sub-circuit that provides a compression threshold of 40 dB below ll peak operating level. The control sub circuit includes control 12 signal amplifier 276 followed by a phase split~er 278 which 13 drives a full wave rectifier 280. The resulting DC is applied 14 to a smoothing network 282, the output of which is the control signal. Network 282 includes an RC pre-integrator, an emitter 16 follower and a final RC integrator that operate in con~unction 17 with diodes such that both the pre- and final integrators have l8 non-linear characteristics produced by the diodes. Fast, 19 large changes in signal amplitude are passed quickly, whereas small changes are transferred slowly. This dynamic smoothing 21 action produces optimum results with respect to modulation 22 effects, low frequency distortion, and distortion components 23 generated by the control signal. The circuit achieves both fast 24 recovery and low signal distortion.
Figure 18 shows an actual chart recording plot of 26 response below the compression threshold of a fixed band 27 compressor having a low level gain of 8 dB and a pass-band 28 filter frequency of 800 Hz high pass~ Boost is provided within 29 the active frequency area of the device (determined by the 800 Hz corner frequency) up to levels of about -10 dB (with respect 31 ¦ to a 0 dB reference level).
32 ~ ///
I , 9~6 l Figure 19 shows the effect on compression when a high 2 ¦ level signal (+10 dB) is presen~ at 100 Hz, which is well below 3 ¦ the 800 Hz filter corner frequency. The strong 100 Hz signal 4 ¦ in the stop-band effectively blocks the compressor and prevents 5 ¦ any compression within the pass-band. Consequently, desired 6 ¦ noise reduction in the pass~band is lost. In addi-tion, if the 7 ¦ 100 Hz signal is intermittent, compression in the pass-band will 8 ¦ come and go with the controlling 100 Hz signal causing noise 9 ¦ modulation and/or signal modulation.
l0¦ Figure 20 shows the effect of the addition of a ll¦ modulation control sub-circuit, described hereinafter, to a 12¦ fixed band circuit. Compression is restored to the pass-band 13¦ area even in the presence of the strong (~10 dB) signal at 100 14¦ Hz~ The modulation control sub-circuit effectively makes the 15¦ fixed band circuit immune to the strong stop-band signal.
16¦ Figure 21 shows generally the preferred embodiment 17¦ of the invention as applied ~o one band of a fixed band dual 18~ path bi~linear compressor of the type described in connection 19 with Figure 16~ Two additions are made to the circuit in order 2Q¦ to provide modulation control. A modulation control sub-circuit 21¦ 198, similar to that in the sliding band embodiment of Figure l1 22¦ is provided, which includes a rectifier 208' and a first sta~e 23 of smoothing 21Oa'. The modulation control optionally may be 241 fed from the output of the compressor. Elements 208, 208' and 25~ 210a~ ~10a' may be identical (but separate). The level of the 26¦ modulation control signal from smoothing circuit 210al is set by 271 attenuator 212 or some other suitahle means and is combined by 2~1 circuit 214 in opposite polarity with the stop~band DC control 291 si~nal from smoothing circuit 210a. In addition the output of 30l¦ VCA 204 and amplifier 206 is applied to a filter 216 which 3111 preferably has the same corner frequency as filter 202, although l -32-l ll 1 this is not essential; the comparative graphs Figures 19 and 20 21 were made ~ith a simple 6dB/octave 3kHz low-pass filter 2160 31 Nevertheless, filter 216 should ideally have a relatively steep 4 cutoff characteristic, such as 12dB or 18dB per octave (e.g. f a 51 2 or 3 pole filter) with about the same cutoff fre~uency as 61 filter 202. The filter 216 output is rectified and smoothed by blocks 218 and 220-, to form the pass band control signal. The I smoothing provided by blocks 210a, 210a' and 210a'~ may be a 9 preliminary filtering stage followed by further smoothing in circuit 210b~ The output of the pass-band filter channel is 11 applied to maximum selector 222 that receives at its other input 12 the output of combiner 214 the modulation controlled stop-band 13 control signal. In its simplest form the maximum selector 14 comprises two diodes which pass the larger of the two input signals in more sophisticated circuits, operational amplifiers 16 are employed to eliminate the diode voltage drops and to increase 17 accuracy.
18 In operation, signals in the s-top-band are subject to 19 the action of the sub circuit 198 if there are no dominant signals inside the pass band where compression action is desired.
21 Thus, although a strong signal such as that of +10 dB at 100 Hz 2~ causes a large control signal to be generated by blocks 208 and 23 210a (and 210b), that control signal is bucked by the modulation 24 control sub-circuit signal so that the VCA 204 gain is not driven down to cause a loss of compression in the pass-band.
26 If a signal of 1 OOH2 occurs in the level region of -20dB~ on the 271 other hand, the bucking action is greatly reduced, and the 28~ stop-band control signal then appropriately controls the action 291 of the compressor whenever signal conditions are such that the 30 ¦ pass-band control signal is not controlling the compressor. If 31 ¦ strong signals are present within the active area pass band, the 32 ~ ///
1 output of the sharp filter channel, the pass-band control 2 circuit will control the maximum selector and allow the VCA to 3 react accordingly~
41 The level of the modulation control sub ~ircuit ~¦ relative to the input or output is set to provide a dynamic 61 reference signal (relative to the input) of sufficient level to 71 result in substantial immunity of the compressor action to 81 strong out of pass band signals.
9 Comments made regarding equalized control and modula~
Figure 18 shows an actual chart recording plot of 26 response below the compression threshold of a fixed band 27 compressor having a low level gain of 8 dB and a pass-band 28 filter frequency of 800 Hz high pass~ Boost is provided within 29 the active frequency area of the device (determined by the 800 Hz corner frequency) up to levels of about -10 dB (with respect 31 ¦ to a 0 dB reference level).
32 ~ ///
I , 9~6 l Figure 19 shows the effect on compression when a high 2 ¦ level signal (+10 dB) is presen~ at 100 Hz, which is well below 3 ¦ the 800 Hz filter corner frequency. The strong 100 Hz signal 4 ¦ in the stop-band effectively blocks the compressor and prevents 5 ¦ any compression within the pass-band. Consequently, desired 6 ¦ noise reduction in the pass~band is lost. In addi-tion, if the 7 ¦ 100 Hz signal is intermittent, compression in the pass-band will 8 ¦ come and go with the controlling 100 Hz signal causing noise 9 ¦ modulation and/or signal modulation.
l0¦ Figure 20 shows the effect of the addition of a ll¦ modulation control sub-circuit, described hereinafter, to a 12¦ fixed band circuit. Compression is restored to the pass-band 13¦ area even in the presence of the strong (~10 dB) signal at 100 14¦ Hz~ The modulation control sub-circuit effectively makes the 15¦ fixed band circuit immune to the strong stop-band signal.
16¦ Figure 21 shows generally the preferred embodiment 17¦ of the invention as applied ~o one band of a fixed band dual 18~ path bi~linear compressor of the type described in connection 19 with Figure 16~ Two additions are made to the circuit in order 2Q¦ to provide modulation control. A modulation control sub-circuit 21¦ 198, similar to that in the sliding band embodiment of Figure l1 22¦ is provided, which includes a rectifier 208' and a first sta~e 23 of smoothing 21Oa'. The modulation control optionally may be 241 fed from the output of the compressor. Elements 208, 208' and 25~ 210a~ ~10a' may be identical (but separate). The level of the 26¦ modulation control signal from smoothing circuit 210al is set by 271 attenuator 212 or some other suitahle means and is combined by 2~1 circuit 214 in opposite polarity with the stop~band DC control 291 si~nal from smoothing circuit 210a. In addition the output of 30l¦ VCA 204 and amplifier 206 is applied to a filter 216 which 3111 preferably has the same corner frequency as filter 202, although l -32-l ll 1 this is not essential; the comparative graphs Figures 19 and 20 21 were made ~ith a simple 6dB/octave 3kHz low-pass filter 2160 31 Nevertheless, filter 216 should ideally have a relatively steep 4 cutoff characteristic, such as 12dB or 18dB per octave (e.g. f a 51 2 or 3 pole filter) with about the same cutoff fre~uency as 61 filter 202. The filter 216 output is rectified and smoothed by blocks 218 and 220-, to form the pass band control signal. The I smoothing provided by blocks 210a, 210a' and 210a'~ may be a 9 preliminary filtering stage followed by further smoothing in circuit 210b~ The output of the pass-band filter channel is 11 applied to maximum selector 222 that receives at its other input 12 the output of combiner 214 the modulation controlled stop-band 13 control signal. In its simplest form the maximum selector 14 comprises two diodes which pass the larger of the two input signals in more sophisticated circuits, operational amplifiers 16 are employed to eliminate the diode voltage drops and to increase 17 accuracy.
18 In operation, signals in the s-top-band are subject to 19 the action of the sub circuit 198 if there are no dominant signals inside the pass band where compression action is desired.
21 Thus, although a strong signal such as that of +10 dB at 100 Hz 2~ causes a large control signal to be generated by blocks 208 and 23 210a (and 210b), that control signal is bucked by the modulation 24 control sub-circuit signal so that the VCA 204 gain is not driven down to cause a loss of compression in the pass-band.
26 If a signal of 1 OOH2 occurs in the level region of -20dB~ on the 271 other hand, the bucking action is greatly reduced, and the 28~ stop-band control signal then appropriately controls the action 291 of the compressor whenever signal conditions are such that the 30 ¦ pass-band control signal is not controlling the compressor. If 31 ¦ strong signals are present within the active area pass band, the 32 ~ ///
1 output of the sharp filter channel, the pass-band control 2 circuit will control the maximum selector and allow the VCA to 3 react accordingly~
41 The level of the modulation control sub ~ircuit ~¦ relative to the input or output is set to provide a dynamic 61 reference signal (relative to the input) of sufficient level to 71 result in substantial immunity of the compressor action to 81 strong out of pass band signals.
9 Comments made regarding equalized control and modula~
10¦ tion control amplifiers in reference to slidinq band circuits
11 are also applicable to fixed band embodiments. Thus, optionally, ~ filter/equalizers 224 and 226 may be inserted in the respective 13 paths to rectifiers 208' and 208. However, the opportunities 14 for advantageously working one frequency dependent characteristic against another in the fixed band case are less than with 16 sliding bands; indeed, this is why an extra control circuit is 17 required in the fixed band case (3 circuits versus 2).
18 It is also possible to achieve modulation control of 19 fixed band circuits hy other means than deriving a control signal reference from the compressor or expander input (or 21 output) signal. One or more control signals can be derived from 22 the controllable element (attenuator or VCA) output and limited 23 so as to achieve results similar to those achieved by the 24 bucXing embodiment of Figure 210 Figure 22 is directed to such limiting embodimentsO
26 In the embodiment of Fi~ure 22, the control signal 7 generating means (blocks 276, 278, 280 and 282 in Figure 17) is c~
28 split into two paths, one having ~ amplifier 228, a sharp 29¦ cutoff filter (as in the Figure 21 embodiment) and a rectifier 30 1 218 and the other having an amplifier 230, a limiter 232 and a 31 ¦ rectifier 218~n The threshold of limiter 232 (which can be back ~ -34-I
I 1, l to back diodes, for example~ is selected such that limiting 2 action begins at relatively high levels, at about the same 3 ¦ levels at which the output from combiner 214 begins to become 4 predominant in the embodiment of Figure 210 The outputs of 51 rectifiers 218 and 218' can be combined and applied to smoothing 61 circuit 210, the output of which is applied as the control 71 signal to VCA 204 or the rectifier outputs can be applied to (or 8 serve as) a maximum selector circuit (such as block 222 in 9 Figure 21) and its output applied to smoothing network 210.
In operation, the embodiment of Figure 22 functions in ll a similar manner to the Figure 21 embodiment
18 It is also possible to achieve modulation control of 19 fixed band circuits hy other means than deriving a control signal reference from the compressor or expander input (or 21 output) signal. One or more control signals can be derived from 22 the controllable element (attenuator or VCA) output and limited 23 so as to achieve results similar to those achieved by the 24 bucXing embodiment of Figure 210 Figure 22 is directed to such limiting embodimentsO
26 In the embodiment of Fi~ure 22, the control signal 7 generating means (blocks 276, 278, 280 and 282 in Figure 17) is c~
28 split into two paths, one having ~ amplifier 228, a sharp 29¦ cutoff filter (as in the Figure 21 embodiment) and a rectifier 30 1 218 and the other having an amplifier 230, a limiter 232 and a 31 ¦ rectifier 218~n The threshold of limiter 232 (which can be back ~ -34-I
I 1, l to back diodes, for example~ is selected such that limiting 2 action begins at relatively high levels, at about the same 3 ¦ levels at which the output from combiner 214 begins to become 4 predominant in the embodiment of Figure 210 The outputs of 51 rectifiers 218 and 218' can be combined and applied to smoothing 61 circuit 210, the output of which is applied as the control 71 signal to VCA 204 or the rectifier outputs can be applied to (or 8 serve as) a maximum selector circuit (such as block 222 in 9 Figure 21) and its output applied to smoothing network 210.
In operation, the embodiment of Figure 22 functions in ll a similar manner to the Figure 21 embodiment
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Claims (42)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. A circuit arrangement for modifying the dynamic range of an input signal, comprising: frequency selective circuit means for dividing the frequency spectrum in which the input signal lies into pass-band and stop-band regions, and dynamic modification means for modifying the dynamic range of signal components in the pass-band region in response to signal com-ponents lying in the pass-band and stop-band regions, the dyn-amic modification means being less responsive to stop-band signal components as the level of the input signal rises.
2. A circuit arrangement according to claim 1 wherein the dynamic modification means includes a control circuit and a variable gain device, the variable dynamic action of which is controlled by the control circuit, the control circuit including means responsive to stop-band signal components for counteracting the dynamic range modification as the level of the input signal rises.
3. A circuit arrangement according to claim 2 wherein said means for counteracting includes means for non-linearly processing said stop-band signal components.
4. The circuit arrangement of claim 2 wherein said means for counteracting includes a sub-circuit for generating a buck-ing reference signal, the reference signal providing informa-tion as the input signal level rises for reducing the effect of stop-band signal components.
5. The circuit arrangement of claim 2 wherein the con-trol circuit includes limiting and frequency-selective circuit means for acting preferentially on signals in the stop-band to reduce the effect of stop-band signal components as the input signal level rises.
6. A circuit arrangement according to claim 1 wherein the circuit arrangement has a bi-linear characteristic composed of a low level portion of substantially constant gain up to a threshold, an intermediate level portion, above the thres-hold, of changing gain providing a maximum compression ratio, and a high level portion of substantially constant gain differ-ent from the gain of the low level portion.
7. A circuit arrangement according to claim 1 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, boosts the main path signal com-ponent, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
8. A circuit arrangement according to claim 6 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, boosts the main path signal compon-ent, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smal-ler than the main path signal component.
9. A circuit arrangement according to claim 8 wherein the further path also has a bi-linear characteristic such that as the main signal component level rises the further path signal level is substantially proportional to the main path signal level.
10. A circuit arrangement according to claim 1 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with res-pect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an up-per part of the frequency band, bucks the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
11. A circuit arrangement according to claim 6 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with res-pect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, bucks the main path signal compon-ent, but which is so limited that, in the upper part of the input dyanmic range, the further path signal component is smal-ler than the main path signal component.
12. A circuit arrangement according to claim 11 wherein the further path also has a bi-linear characteristic such that as the main signal component level rises the further path sig-nal level is substantially proportional to the main path signal level.
13. A circuit arrangement according to claim 1 wherein the circuit arrangement has a bi-linear characteristic composed of a low level portion of substantially constant gain up to a threshold, an intermediate level portion, above the threshold, of changing gain providing a maximum expansion ratio, and a high level portion of substantially constant gain different from the gain of the low level portion.
14. A circuit arrangement according to claim 13 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with respect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, boosts the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
15. A circuit arrangement according to claim 14 wherein the further path also has a bi-linear characteristic such that as the main signal component level rises the further path sig-nal level is substantially proportional to the main path signal level.
16. A circuit arrangement according to claim 15 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an up-per part of the frequency band, bucks the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
17. A circuit arrangement according to claim 16 wherein the further path also has a bi-linear characteristic such that as the main signal component level rises the further path sig-nal level is substantially proportional to the main path signal level.
18. A circuit arrangement according to claim 1, for audio signals, wherein the dynamic modification means includes a control circuit and a variable gain device, the variable dynamic action of which is controlled by the control circuit, including means providing a control signal for the variable gain device to effect the variable dynamic action, said control circuit comprising a first sub-circuit including at least one filter having frequency characteristics which more strongly favor pass-band signals than said frequency selective circuit means and rectifying means for providing a first signal, and a second sub-circuit deriving a second signal from the variable gain device output and including further rectifying means, and deri-ving a third signal from at least one of the input and the output of the circuit arrangement and including yet further rectifying means, the third signal constituting a reference signal for bucking the second signal to provide a fourth signal, the reference signal dynamically varying in level with the level of the respective one of the input and output signals, the second sub-circuit including means for setting the gain of the reference signal to reduce the effects of stop-band signal components in the third signal as the input signal level rises, and means for selecting the larger of the first and fourth signals to provide a control signal for said variable gain device.
19. A circuit arrangement according to claim 1, for audio signals, wherein the dynamic modification means includes a control. circuit and a variable gain device, the variable dyna-mic action of which is controlled by the control circuit, in-cluding means providing a control signal for the variable gain device to effect the variable dynamic action, said control circuit comprising a first sub-circuit including at least one filter having frequency characteristics which more strongly favor pass-band signals than said frequency selective circuit means and rectifying means, and a second sub-circuit including limiting and frequency-selective circuit means for acting pre-ferentially on signals in the stop-band to reduce the effect of stop-band signal components as the input signal level increa-ses.
20. A circuit arrangement for modifying the dynamic range of an input signal, comprising: frequency-selective circuit means for dividing the frequency spectrum in which the input signal lies into pass-band and stop-band regions, the pass-band frequency region sliding in response to signal components lying in the pass-band and stop-band regions, the frequency selective circuit means becoming less responsive to stop-band signal components as the level of the input signal rises, and means for modifying the dynamic range of signal components in the pass-band region.
21. A circuit arrangement according to claim 20 wherein the frequency selective circuit means includes a control circuit and a variable filter, the variable action of which is control-led by the control circuit, the control circuit including means responsive to stop-band signal components for counter-acting the sliding of the pass-band frequency region as the input signal level rises.
22. A circuit arrangement according to claim 21 wherein said means for counteracting includes means for non-linearly processing said stop-band signal components.
23. The circuit arrangement of claim 21 wherein said means for counteracting includes a sub-circuit for generating a bucking reference signal, the reference signal providing information for reducing the effect of stop-band signal compon-ents as the input signal level rises.
24. The circuit arrangement of claim 21 wherein the con-trol circuit includes limiting and frequency-selective circuit means for acting preferentially on signals in the stop-band to reduce the effect of stop-band signal components as the input signal level rises.
25. A circuit arrangement according to claim 20 wherein the circuit arrangement has a bi-linear characteristic composed of a low level portion of substantially constant gain up to a threshold, an intermediate level portion, above the threshold, of changing gain providing a maximum compression ratio, and a high level portion of substantially constant gain differ-ent from the gain of the low level portion.
26. A circuit arrangement according to claim 20 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, boosts the main path signal compon-ent, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smal-ler than the main path signal component.
27. A circuit arrangement according to claim 25 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, boosts the main path signal compon-ent, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smal-ler than the main path signal component.
28. A circuit arrangement according to claim 27 wherein the further path also has a bi-linear characteristic such that as the input signal level rises the further path signal level becomes a predetermined proportion of the main path signal level.
29. A circuit arrangement according to claim 20 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with re-spect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an upper part of the frequency band, bucks the main path signal compon-ent, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smal-ler than the main path signal component.
30. A circuit arrangement according to claim 25 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with res-pect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an up-per part of the frequency band, bucks the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
31. A circuit arrangement according to claim 30 wherein the further path also has a bi-linear characteristic such that as the input signal level rises the further path signal level becomes a predetermined proportion of the main path signal level.
32. A circuit arrangement according to claim 20 wherein the circuit arrangement has a bi-linear characteristic composed of a low level portion of substantially constant gain up to a threshold, an intermediate level portion, above the threshold, of changing gain providing a maximum expansion ratio, and a high level portion of substantially constant gain dif-ferent from the gain of the low level portion.
33. A circuit arrangement according to claim 32 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with res-pect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an up-per part of the frequency band, boosts the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
34. A circuit arrangement according to claim 33 wherein the further path also has a bi-linear characteristic such that as the input signal level rises the further path signal level becomes a predetermined proportion of the main path signal level.
35. A circuit arrangement according to claim 32 wherein the circuit arrangement is a dual-path circuit comprising a main signal component in a main path which is linear with res-pect to dynamic range and a further path which has its input signal derived from at least one point in the main path and its output signal combined with the signal in the main path, the further path providing a signal which, at least in an up-per part of the frequency band, bucks the main path signal component, but which is so limited that, in the upper part of the input dynamic range, the further path signal component is smaller than the main path signal component.
36. A circuit arrangement according to claim 35 wherein the further path also has a bi-linear characteristic such that as the input signal level rises the further path signal level becomes a predetermined proportion of the main path signal level.
37. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a boost in a high frequency region of the signal band, and responsive to dominant signals to cause the filter corner fre-quency to slide in the sense narrowing the boosted region, and wherein the control circuit includes rectifying means pro-viding a control signal, derived from at least one of the fil-ter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner frequency, said control circuit including a sub-circuit providing a refer-ence signal for bucking the control signal and derived from at least one of the input and output of the circuit arrangement, the reference signal dynamically varying in level with the level of said at least one of the input and output signals, the sub-circuit including means for setting the gain of the reference signal such that the effects of stop-band signal com-ponents are reduced as the input signal level rises.
38. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a boost in a low frequency region of the signal band, and responsive to dominant signals to cause the filter corner frequency to slide in the sense narrowing the boosted region, and wherein the control circuit includes rectifying means pro-viding a control signal, derived from at least one of the fil-ter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner frequency, said control circuit including a sub-circuit providing a refer-ence signal for bucking the control signal and derived from at least one of the input and output of the circuit arrangement, the reference signal dynamically varying in level with the level of said at least one of the input and output signals, the sub-circuit including means for setting the gain of the reference signal such that the effects of stop-band signal components are reduced as the input signal level rises.
39. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a cut in a high frequency region of the signal band, and responsive to dominant signals to cause the filter corner frequency to slide in the sense narrowing the cut region and wherein the control circuit includes rectifying means providing a control signal, derived from at least one of the filter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner frequency, said control circuit including a sub-circuit providing a reference signal for bucking the control signal and derived from at least one of the input and output of the circuit arrangement, the reference signal dynamically varying in level with the level of said at least one of the input and output signals, the sub-circuit including means for setting the gain of the reference signal such that the effects of stop-band signal components are reduced as the input signal level rises.
40. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a cut in a low frequency region of the signal band, and responsive to dominant signals to cause the filter corner frequency to slide in the sense narrowing the cut region, and wherein the control circuit includes rectifying means providing a control signal, derived from at least one of the filter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner frequency, said control circuit including a sub-circuit providing a reference signal for bucking the control signal and derived from at least one of the input and output of the circuit arrangement, the reference signal dynamically varying in level with the level of said at least one of the input and output signals, the sub-circuit including means for setting the gain of the reference signal such that the effects of stop-band signal components are reduced as the input signal level rises.
41. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a boost in a preselected frequency region of the sig-nal band, and responsive to dominant signals to cause the filter corner frequency to slide in the sense narrowing the preselected region, and wherein the control circuit includes rectifying means providing a control signal, derived from at least one of the filter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner fre-quency, said control circuit including limiting and frequency-selective circuit means for acting preferentially on signals in the stop-band to reduce the effects of stop-band signal components as the input signal level rises.
42. A circuit arrangement according to claim 20, for audio signals, wherein the frequency selective circuit means includes a control circuit and a variable filter, which filter provides a cut in a preselected frequency region of the signal band, and responsive to dominant signals to cause the filter corner frequency to slide in the sense narrowing the cut region, and wherein the control circuit includes rectifying means pro-viding a control signal, derived from at least one of the filter input and output, to a controlled impedance device of the filter to effect the sliding of the filter corner frequency, said control circuit including limiting and frequency-selective circuit means for acting preferentially on signals in the stop-band to reduce the effects of stop-band signal components as the input signal level rises.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US06/325,530 US4498055A (en) | 1981-09-10 | 1981-12-01 | Circuit arrangements for modifying dynamic range |
| US325,530 | 1981-12-01 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1188996A true CA1188996A (en) | 1985-06-18 |
Family
ID=23268273
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000392068A Expired CA1188996A (en) | 1981-12-01 | 1981-12-11 | Circuit arrangements for modifying dynamic range |
Country Status (15)
| Country | Link |
|---|---|
| KR (1) | KR900000483B1 (en) |
| BE (1) | BE901906Q (en) |
| CA (1) | CA1188996A (en) |
| CH (1) | CH656996A5 (en) |
| DE (1) | DE3151137A1 (en) |
| DK (1) | DK575681A (en) |
| ES (1) | ES8303848A1 (en) |
| FR (1) | FR2517496B1 (en) |
| GB (1) | GB2111355B (en) |
| GR (1) | GR77317B (en) |
| IE (1) | IE52495B1 (en) |
| IT (1) | IT1140402B (en) |
| LU (1) | LU83854A1 (en) |
| NL (1) | NL192860C (en) |
| SE (1) | SE449281B (en) |
Families Citing this family (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4922535A (en) * | 1986-03-03 | 1990-05-01 | Dolby Ray Milton | Transient control aspects of circuit arrangements for altering the dynamic range of audio signals |
| JP2764205B2 (en) * | 1986-03-03 | 1998-06-11 | レ−・ミルトン・ドルビ | Bootstrap signal attenuation circuit |
| CA1269138A (en) * | 1986-03-03 | 1990-05-15 | Ray Milton Dolby | Attenuator circuit employing bootstrapping |
| GB2429779A (en) * | 2005-09-01 | 2007-03-07 | Geo Kingsbury Machine Tools Lt | Testing components of drive trains |
Family Cites Families (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB1120541A (en) * | 1965-08-11 | 1968-07-17 | Dolby Ray Milton | Improvements in noise reduction systems |
| US3665345A (en) * | 1969-07-21 | 1972-05-23 | Dolby Laboratories Inc | Compressors and expanders for noise reduction systems |
| GB1367002A (en) * | 1971-04-06 | 1974-09-18 | Victor Company Of Japan | Compression and/or expansion system and circuit |
| US4101849A (en) * | 1976-11-08 | 1978-07-18 | Dbx, Inc. | Adaptive filter |
| BE889426A (en) * | 1980-06-30 | 1981-10-16 | Dolby Ray Milton | CIRCUIT FOR REDUCING THE EFFECTS OF OVERLOADING OF A SIGNAL RECORDING OR TRANSMISSION MEDIUM |
-
1981
- 1981-12-09 GB GB08137167A patent/GB2111355B/en not_active Expired
- 1981-12-11 CA CA000392068A patent/CA1188996A/en not_active Expired
- 1981-12-15 SE SE8107495A patent/SE449281B/en not_active IP Right Cessation
- 1981-12-15 IE IE2951/81A patent/IE52495B1/en not_active IP Right Cessation
- 1981-12-19 KR KR1019810005016A patent/KR900000483B1/en active
- 1981-12-22 CH CH8223/81A patent/CH656996A5/en not_active IP Right Cessation
- 1981-12-22 NL NL8105775A patent/NL192860C/en not_active IP Right Cessation
- 1981-12-23 DE DE19813151137 patent/DE3151137A1/en active Granted
- 1981-12-23 DK DK575681A patent/DK575681A/en not_active Application Discontinuation
- 1981-12-23 GR GR66897A patent/GR77317B/el unknown
- 1981-12-24 LU LU83854A patent/LU83854A1/en unknown
- 1981-12-28 FR FR8124314A patent/FR2517496B1/en not_active Expired
- 1981-12-29 IT IT25877/81A patent/IT1140402B/en active
- 1981-12-29 ES ES508394A patent/ES8303848A1/en not_active Expired
-
1985
- 1985-03-08 BE BE0/214626A patent/BE901906Q/en not_active IP Right Cessation
Also Published As
| Publication number | Publication date |
|---|---|
| DE3151137C2 (en) | 1993-04-01 |
| IE52495B1 (en) | 1987-11-25 |
| NL192860C (en) | 1998-03-04 |
| NL8105775A (en) | 1983-07-01 |
| FR2517496B1 (en) | 1985-11-15 |
| LU83854A1 (en) | 1982-05-07 |
| FR2517496A1 (en) | 1983-06-03 |
| KR830008462A (en) | 1983-11-18 |
| SE449281B (en) | 1987-04-13 |
| ES508394A0 (en) | 1983-02-01 |
| IE812951L (en) | 1983-06-01 |
| ES8303848A1 (en) | 1983-02-01 |
| SE8107495L (en) | 1983-06-02 |
| CH656996A5 (en) | 1986-07-31 |
| DK575681A (en) | 1983-06-02 |
| GB2111355A (en) | 1983-06-29 |
| GB2111355B (en) | 1985-02-13 |
| NL192860B (en) | 1997-11-03 |
| GR77317B (en) | 1984-09-11 |
| IT8125877A0 (en) | 1981-12-29 |
| BE901906Q (en) | 1985-07-01 |
| KR900000483B1 (en) | 1990-01-30 |
| DE3151137A1 (en) | 1983-07-14 |
| IT1140402B (en) | 1986-09-24 |
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| Date | Code | Title | Description |
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| MKEX | Expiry |