CA1135349A - Current sources - Google Patents
Current sourcesInfo
- Publication number
- CA1135349A CA1135349A CA000316238A CA316238A CA1135349A CA 1135349 A CA1135349 A CA 1135349A CA 000316238 A CA000316238 A CA 000316238A CA 316238 A CA316238 A CA 316238A CA 1135349 A CA1135349 A CA 1135349A
- Authority
- CA
- Canada
- Prior art keywords
- input
- current
- signal
- output
- transistor
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04M—TELEPHONIC COMMUNICATION
- H04M1/00—Substation equipment, e.g. for use by subscribers
- H04M1/60—Substation equipment, e.g. for use by subscribers including speech amplifiers
- H04M1/6025—Substation equipment, e.g. for use by subscribers including speech amplifiers implemented as integrated speech networks
-
- G—PHYSICS
- G05—CONTROLLING; REGULATING
- G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
- G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
- G05F1/10—Regulating voltage or current
- G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
- G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
- G05F1/561—Voltage to current converters
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/08—Modifications of amplifiers to reduce detrimental influences of internal impedances of amplifying elements
- H03F1/083—Modifications of amplifiers to reduce detrimental influences of internal impedances of amplifying elements in transistor amplifiers
Abstract
ABSTRACT
Current Sources.
A current source in which the output current from an output amplifier stage is controlled by an input signal, originating in a signal source, to an input amplifier stage incorporates, additionally to a negative feedback path from the output current path of the output stage to the input amplifier monitoring means in the signal input path to the output stage and control means in an input path to the input amplifier. The monitoring means monitors the current flow in the input path to the output stage and applies, via control means, a signal to the input amplifier such that the dependence of the current flow in the current output path on the gain of the output stage is reduced. A circuit is described, as well as an application to driving a telephone sub-scriber line and using two current sources working in push-pull mode.
Current Sources.
A current source in which the output current from an output amplifier stage is controlled by an input signal, originating in a signal source, to an input amplifier stage incorporates, additionally to a negative feedback path from the output current path of the output stage to the input amplifier monitoring means in the signal input path to the output stage and control means in an input path to the input amplifier. The monitoring means monitors the current flow in the input path to the output stage and applies, via control means, a signal to the input amplifier such that the dependence of the current flow in the current output path on the gain of the output stage is reduced. A circuit is described, as well as an application to driving a telephone sub-scriber line and using two current sources working in push-pull mode.
Description
~13534~
This invention relates to current sources and in partic-ular to such current sources in which the magnitude of the output current of the source is determined by the magnitude of a control voltage.
Conventional current sources which are designed to pro-duce an output current whose magnitude is a function of an input signal to the source, are frequently provided with a negative feed-back loop whose function it is to keep the output current as nearly as possible to the required magnitude.
In practice such current sources may comprise a multi-stage amplifier, with the input signal being applied to an input of a first stage and the current output being provided by an output stage. As long as the requirements of accuracy of transconductance are not too stringent, the output stage frequently consists of a single transistor, with the feedback signal being derived from the emitter circuit of the transistor. ~owever, when a high degree of accuracy is required, it has heretofore been necessary, in general, to use so-called "Darlington" or other multiple transistors in the output stage, which tend to increase the cost of the current source, particularly in application involving high voltages.
It is an object of the present invention to provide an accurate current source which does not require the use of multiple transistors.
In accordance with the present invention, there is pro-vided a current source arrangement in which the magnitude of current flowing in an output path of a transistor amplifier stage is arranged to be controlled by an input signal applied to an input of the arrangement, said input signal being applied to the
This invention relates to current sources and in partic-ular to such current sources in which the magnitude of the output current of the source is determined by the magnitude of a control voltage.
Conventional current sources which are designed to pro-duce an output current whose magnitude is a function of an input signal to the source, are frequently provided with a negative feed-back loop whose function it is to keep the output current as nearly as possible to the required magnitude.
In practice such current sources may comprise a multi-stage amplifier, with the input signal being applied to an input of a first stage and the current output being provided by an output stage. As long as the requirements of accuracy of transconductance are not too stringent, the output stage frequently consists of a single transistor, with the feedback signal being derived from the emitter circuit of the transistor. ~owever, when a high degree of accuracy is required, it has heretofore been necessary, in general, to use so-called "Darlington" or other multiple transistors in the output stage, which tend to increase the cost of the current source, particularly in application involving high voltages.
It is an object of the present invention to provide an accurate current source which does not require the use of multiple transistors.
In accordance with the present invention, there is pro-vided a current source arrangement in which the magnitude of current flowing in an output path of a transistor amplifier stage is arranged to be controlled by an input signal applied to an input of the arrangement, said input signal being applied to the
- 2 - -~
;", - : .
1~.353~9 transistor amplifier stage by way of a differential amplifier, comprising means to derive a negative feedback signal whose magni-tude is dependent upon that of said current in said output path, means to apply said negative feedback signal to an input of said differential amplifier, means to derive a further feedback signal whose magnitude is dependent upon the value of input current to said transistor amplifier stage, and means to apply said further feedback signal as a positive feedback signal to an input of said differential amplifier wh.ereby to reduce the dependence of the current flowing in said output path on the gain of the transistor amplifier stage.
In accordance with. the present invention, there is also provided a current source arrangement comprising a differential amplifier having inverting and non-inverting inputs and an output, and a transistor having an input electrode, an output electrode, and a control electrode, the input electrode being connected by way of a resistive element to one pole of power supply means and the output electrode being connected by way of a load circuit to the other pole of said power supply means, the output of the dif-ferential amplifier heing electrically coupled to the controlelectrode of the transistor, means to apply a control signal to one input of the differential amplifier, feedback means to apply to another input of the differential amplifier a first feedback signal dependent upon the voltage developed in operation across said resistive element, means to derive a second feedback signal dependent on the current flow to the control electrode of the transistor, and means to apply said second feedback signal as a positive feedback signal to said one input of the differential - 2a -:
"~
~135349 amplifier.
According to one aspect of the present invention, in acurrent source arrangement in which the magnitude of current flow in an output path of an amplifier stage is arranged to be dependent on the magnitude of an input signal applied to an amplification stage preceding said amplifier stage, there are provided also at least two feedback paths from said amplifier stage to said preced-ing amplification stage, one of which feedback paths serves to inject into the amplification stage a negative feedback signal whose magnitude is dependent on the current flow in said output path~ another one of the feedback paths being arranged to transmit a signal dependent on a current flowing in an input of said amplifier stage such as to reduce the dependence of the current flow in said output path on the - 2b -113S34~
.
~3--gain of said amplifier stageO
In accordance with another aspect of the present i~vention in a current source-arrangement in which the magni-~ude of current flowing in an output path of a transistor ampli-fier stage is arranged to be controlled by a control signal applied to an input of the arran~ement 7 said co~trol si~nal being applied to the transistor amplifier stage by way of a differential amplifier to an input of which is also applied a negative feedback signal whose magnitude is dependent upon that 10 of said current in said output path, there are provided means to provide a further feedback signal to an input of said differen-tial amplifier whose value is dependent upon the input current to said transistor amplifier stage such as to reduce the dependence of the current flowing in said output path on the gain o~ the transistor amplifier stage.
~ he control signal and the said further feedback signal are conveniently applied to a non-inverting input of the differ-ential amplifier, and the negative feedbac~ signal to an inver-ting input thereof. ~he negative feedback signal may be derived 20 from the voltage drop across a resistive element in the emitter circuit of a~ output transistor in the transistor amplifier stage. ~he said further feedback signal is conveniently derived from a first resistive element in the control signal input path of the transistor amplifier and applied to the non-inverting input via a second resisti~e element, with the control signal being applied to the same input of the differential amplifier via a third resistive element.
In accordance with a further aspect of the present invention a current source arrangement comprises a differential 30 amplifier havin~ inverting and non-inverting inputs and an out-put, and a transistor having an input electrode, an output electrode, and a control electrode; the input electrode being connected by way of a resistive element to one pole of power supply means and the output electrode being connected by way of a load circuit to the other pole of said power supply means, t~e output of the differential amplifier being electrically coupled to the control electrode of the transistor, means to apply a control signal to one input of the differential amplifier~
: 1~353~9 eedback means to apply to another input of the differential a~iplifier a signal dependent u~on the voltage developed in Gperations across said resistive element, and further means for deriving a signal dependent on the curren~ flow through the control electrode of the transistor, said further means bein~ electrically connected to ~aid on~ input o~ the di~fer-ential amplifier.
Preferably~ said differential amplifier is a high gain amplifier having substantially linear amplification characteri-stics, of the kind known as operational amplifiers.
Said ~urther means for deriving a signal dependent onthe current flow through the control electrode of the transistor conveniently comprises a first resis~ive element connected be-tween the output of the dif~erential amplifier and the control electrode of the transistor, thereby also electrically coupling said output to said control electrode, a second resistive ele-ment by means of which the output of the differential amplifier is connected to said one input thereof, and a tnird resistive element connecting said one input to a source of said control signal whereby said control signal is applied to said one input.
~ he control signal applied in operation of~the current source to said one input may be a DC signal voltage derived from a resistance divider network.
Said control signal may also include an A.C~ component superimposed on the D.C. signal in a known manner.
A current source arrange~ent in accordance with the present invention is suitable for use with electronic line units for telephone systems.
A current source arrangement in accordance with the present invention will now be described by way of example with reference to the accompanying drawing, of which:
~ igure 1 shows the current source arrangement sche-matically;
~ igure 2 shows the current source arrangement diagram-matically; and ~ igure 3 shows, in diagrammatic form, a circuit incor-porating two complimentary current source arrangements in push-pull modeO
. ~ .
~L1353 _5_ Referring first to ~igure 17 a current source inaccord~nce with ~ne present invention comprises a high gain linear am~lification stage provided by an operational amplifier 1~ and an amplifier stage in the form of a ~unction transistor 2, whose control electrode, or base, 8 is electrically coupled wi~h th~ output o~ ~he am~ icr lo ~he input el~ctrod~, o~
emitter 7, of the transistor 2 is connected to the inverting input of the amplifier 1 and also~ via a close tolerance resistor B ~ to a voltage rail 9. Connected to the non-inverting input 10 of the amplifier 1 is a voltage source 11 supplying a co~trol signal voltage to the amplifier and hence to the base 8 of the transistor 2, the value of the control voltage determining the current flow at the output electrode, or collector 6, of the transistor 2. Additionally control means 4 and 5 are conn-ected to the non-inverting input and the output respectively of the amplifier 1, arranged to derive a signal dependent on base current of the transistor 2 and to apply a signal to the non-inverting input of the amplifier 1 such that the dependence of the collector current on the gain of the transistor is reduced.
20 A load circuit (not shown) to be driven by this current source arrangement is connected in use to the collector 6 of the tran-sistor 20 A signal indicative of the base current is derived by means 5 and is fed by mea~s of the control means 4 into the non-inverting,input of the amplifier, any deviation of this signal from its design value as determined by the control voltage causing a variation'in the signal applied to the non-inverting input. A consequent change in current flow t'nrough the emitter and hence the resistor 3 results in a change of the voltage applied to the emitter and thus compensates for this signal variation thereby bringing the collector current nearer to the intended valueO
~ he operation of a particular embodiment of this con-stant current source will become apparent from the description below with reference to ~igure 2 of the drawings, in which parts easily identifiable as being equivalent to components of ~igure 1 carry the same referenceO
_ .. . .. .. ... .. . . . .
~5349 ~ eferring now to ~igure 2, the output of the operat-ional amplifier l is coupled by means of a resistor 16 to the ba~ 8 OL~ the tra~sistor 2~ ~Jhose emitter 7 is connected to the voltage rail 9 by wa~ of a resistor 3 and to the i~verting in ut of the amplifier 1 through a resistor 14 which serves as a feedback path and can also be arranged to e~ualize the source impedances to the two inputs o~ the amplifier. A resistance divider network comprising resistors 18 and 19 connected across the voltage rails 9 and 10~ from which power is also supplied 10 to the amplifier lg provides the biassing D.C. control vo~ta~e onto which may be superimposed an A.C. control voltage in the form of an A.C. signal applied to the terminal 20 of the capaci-tor 210 ~he biassing voltage derived from the resistance net-work 18, 19 and~ where applicable, the A.C. signal is applied to the non-inverting input of the amplifier 1 through a resistor 17, with a resistor 15 connecting this input to the output of the amplifiex. ~he influence of resistors 14, 18, and 19 on the A.~. performance of the current source is negligible. The accuracy of the current source can be made dependent solel~J on 20 resistor matching, thus obviating the need to adjust each indivi-dual source in dependence on the individual performance charac-teristics of the transistor used, as the following calculation shows, in which:-Vin , the voltage obtained from the resistance divider network Vref= the voltage at the non-inverting input of the amplifier V0 = the voltage at the output of the amplifier Vb = the voltage ~etween base and emitter of the transistor Ib = the base current of the transistor Ie = the emitter current of the transistor
;", - : .
1~.353~9 transistor amplifier stage by way of a differential amplifier, comprising means to derive a negative feedback signal whose magni-tude is dependent upon that of said current in said output path, means to apply said negative feedback signal to an input of said differential amplifier, means to derive a further feedback signal whose magnitude is dependent upon the value of input current to said transistor amplifier stage, and means to apply said further feedback signal as a positive feedback signal to an input of said differential amplifier wh.ereby to reduce the dependence of the current flowing in said output path on the gain of the transistor amplifier stage.
In accordance with. the present invention, there is also provided a current source arrangement comprising a differential amplifier having inverting and non-inverting inputs and an output, and a transistor having an input electrode, an output electrode, and a control electrode, the input electrode being connected by way of a resistive element to one pole of power supply means and the output electrode being connected by way of a load circuit to the other pole of said power supply means, the output of the dif-ferential amplifier heing electrically coupled to the controlelectrode of the transistor, means to apply a control signal to one input of the differential amplifier, feedback means to apply to another input of the differential amplifier a first feedback signal dependent upon the voltage developed in operation across said resistive element, means to derive a second feedback signal dependent on the current flow to the control electrode of the transistor, and means to apply said second feedback signal as a positive feedback signal to said one input of the differential - 2a -:
"~
~135349 amplifier.
According to one aspect of the present invention, in acurrent source arrangement in which the magnitude of current flow in an output path of an amplifier stage is arranged to be dependent on the magnitude of an input signal applied to an amplification stage preceding said amplifier stage, there are provided also at least two feedback paths from said amplifier stage to said preced-ing amplification stage, one of which feedback paths serves to inject into the amplification stage a negative feedback signal whose magnitude is dependent on the current flow in said output path~ another one of the feedback paths being arranged to transmit a signal dependent on a current flowing in an input of said amplifier stage such as to reduce the dependence of the current flow in said output path on the - 2b -113S34~
.
~3--gain of said amplifier stageO
In accordance with another aspect of the present i~vention in a current source-arrangement in which the magni-~ude of current flowing in an output path of a transistor ampli-fier stage is arranged to be controlled by a control signal applied to an input of the arran~ement 7 said co~trol si~nal being applied to the transistor amplifier stage by way of a differential amplifier to an input of which is also applied a negative feedback signal whose magnitude is dependent upon that 10 of said current in said output path, there are provided means to provide a further feedback signal to an input of said differen-tial amplifier whose value is dependent upon the input current to said transistor amplifier stage such as to reduce the dependence of the current flowing in said output path on the gain o~ the transistor amplifier stage.
~ he control signal and the said further feedback signal are conveniently applied to a non-inverting input of the differ-ential amplifier, and the negative feedbac~ signal to an inver-ting input thereof. ~he negative feedback signal may be derived 20 from the voltage drop across a resistive element in the emitter circuit of a~ output transistor in the transistor amplifier stage. ~he said further feedback signal is conveniently derived from a first resistive element in the control signal input path of the transistor amplifier and applied to the non-inverting input via a second resisti~e element, with the control signal being applied to the same input of the differential amplifier via a third resistive element.
In accordance with a further aspect of the present invention a current source arrangement comprises a differential 30 amplifier havin~ inverting and non-inverting inputs and an out-put, and a transistor having an input electrode, an output electrode, and a control electrode; the input electrode being connected by way of a resistive element to one pole of power supply means and the output electrode being connected by way of a load circuit to the other pole of said power supply means, t~e output of the differential amplifier being electrically coupled to the control electrode of the transistor, means to apply a control signal to one input of the differential amplifier~
: 1~353~9 eedback means to apply to another input of the differential a~iplifier a signal dependent u~on the voltage developed in Gperations across said resistive element, and further means for deriving a signal dependent on the curren~ flow through the control electrode of the transistor, said further means bein~ electrically connected to ~aid on~ input o~ the di~fer-ential amplifier.
Preferably~ said differential amplifier is a high gain amplifier having substantially linear amplification characteri-stics, of the kind known as operational amplifiers.
Said ~urther means for deriving a signal dependent onthe current flow through the control electrode of the transistor conveniently comprises a first resis~ive element connected be-tween the output of the dif~erential amplifier and the control electrode of the transistor, thereby also electrically coupling said output to said control electrode, a second resistive ele-ment by means of which the output of the differential amplifier is connected to said one input thereof, and a tnird resistive element connecting said one input to a source of said control signal whereby said control signal is applied to said one input.
~ he control signal applied in operation of~the current source to said one input may be a DC signal voltage derived from a resistance divider network.
Said control signal may also include an A.C~ component superimposed on the D.C. signal in a known manner.
A current source arrange~ent in accordance with the present invention is suitable for use with electronic line units for telephone systems.
A current source arrangement in accordance with the present invention will now be described by way of example with reference to the accompanying drawing, of which:
~ igure 1 shows the current source arrangement sche-matically;
~ igure 2 shows the current source arrangement diagram-matically; and ~ igure 3 shows, in diagrammatic form, a circuit incor-porating two complimentary current source arrangements in push-pull modeO
. ~ .
~L1353 _5_ Referring first to ~igure 17 a current source inaccord~nce with ~ne present invention comprises a high gain linear am~lification stage provided by an operational amplifier 1~ and an amplifier stage in the form of a ~unction transistor 2, whose control electrode, or base, 8 is electrically coupled wi~h th~ output o~ ~he am~ icr lo ~he input el~ctrod~, o~
emitter 7, of the transistor 2 is connected to the inverting input of the amplifier 1 and also~ via a close tolerance resistor B ~ to a voltage rail 9. Connected to the non-inverting input 10 of the amplifier 1 is a voltage source 11 supplying a co~trol signal voltage to the amplifier and hence to the base 8 of the transistor 2, the value of the control voltage determining the current flow at the output electrode, or collector 6, of the transistor 2. Additionally control means 4 and 5 are conn-ected to the non-inverting input and the output respectively of the amplifier 1, arranged to derive a signal dependent on base current of the transistor 2 and to apply a signal to the non-inverting input of the amplifier 1 such that the dependence of the collector current on the gain of the transistor is reduced.
20 A load circuit (not shown) to be driven by this current source arrangement is connected in use to the collector 6 of the tran-sistor 20 A signal indicative of the base current is derived by means 5 and is fed by mea~s of the control means 4 into the non-inverting,input of the amplifier, any deviation of this signal from its design value as determined by the control voltage causing a variation'in the signal applied to the non-inverting input. A consequent change in current flow t'nrough the emitter and hence the resistor 3 results in a change of the voltage applied to the emitter and thus compensates for this signal variation thereby bringing the collector current nearer to the intended valueO
~ he operation of a particular embodiment of this con-stant current source will become apparent from the description below with reference to ~igure 2 of the drawings, in which parts easily identifiable as being equivalent to components of ~igure 1 carry the same referenceO
_ .. . .. .. ... .. . . . .
~5349 ~ eferring now to ~igure 2, the output of the operat-ional amplifier l is coupled by means of a resistor 16 to the ba~ 8 OL~ the tra~sistor 2~ ~Jhose emitter 7 is connected to the voltage rail 9 by wa~ of a resistor 3 and to the i~verting in ut of the amplifier 1 through a resistor 14 which serves as a feedback path and can also be arranged to e~ualize the source impedances to the two inputs o~ the amplifier. A resistance divider network comprising resistors 18 and 19 connected across the voltage rails 9 and 10~ from which power is also supplied 10 to the amplifier lg provides the biassing D.C. control vo~ta~e onto which may be superimposed an A.C. control voltage in the form of an A.C. signal applied to the terminal 20 of the capaci-tor 210 ~he biassing voltage derived from the resistance net-work 18, 19 and~ where applicable, the A.C. signal is applied to the non-inverting input of the amplifier 1 through a resistor 17, with a resistor 15 connecting this input to the output of the amplifiex. ~he influence of resistors 14, 18, and 19 on the A.~. performance of the current source is negligible. The accuracy of the current source can be made dependent solel~J on 20 resistor matching, thus obviating the need to adjust each indivi-dual source in dependence on the individual performance charac-teristics of the transistor used, as the following calculation shows, in which:-Vin , the voltage obtained from the resistance divider network Vref= the voltage at the non-inverting input of the amplifier V0 = the voltage at the output of the amplifier Vb = the voltage ~etween base and emitter of the transistor Ib = the base current of the transistor Ie = the emitter current of the transistor
3 Ic ~ the collector current of the transistor re ' the dynamic emitter-base resistance of the transistor R3 = the resistance value of resistor 3, etcO
Summing currents at the non-inverting input of the operational amplifier, one obvains:
Vin ~ Vref ~VO - Vref ~ o (1) Rl~ Rlg , 1.
: : :
~135349 Rearranging this expression gives:
Vref = Vi 15 , -- ~ VO P~l~ (2) R17 ~ R15R17 + R15 3ut; since current flow th~ough the resistor R3 will tend to stabilise at a value such that the volta~e at the emitter electrode 7 of the transistor 2 is virtually equal to Vref:
V = V~ef + Vb ~ IbR16 Defining VbDc ~ Vb ~ Iere (4a) Vb ~ref R e , (4b) writing re/R = k and substituting for Vb from equation (4b) in equation (3) yields:
VO = Vref (1 ~ k) ~ VbDc + IbR16 (5) Using this equation (5) in equation (2) gives V = Vin R15 ~ VbDc R17 ~ IbR16R17 ref l5 - kR17 R15 - kR17 -R15 - kR17 (6) Xowever, I ~ Vref Ib ' (7) and by substitution from equation (6) I = VinR15 ~ VbDcR17 ~ IbR16R17 Ib (8) R3(~15 kR17) R3(R15-~R17~ 3( 15 17) In order to make the collector curre~t Ic independent of the base current Ib, and hence of the current amplification ~, where c (8a) it is necessary to ensure that R3(R-15-kR17) - 9 . (9) ,, .... . . . . . ..
-, .
1~35349 ~ . .
or given k is small~ to a first order that R~
6 ~ ~ (10) ~ 15 R17Thus under D.C. conditions the accuracy of the current source subject to th~ restrictions (9) a~d ~10) abov~ is indaed dependent only on resistor matc`ning and independen~ of the cu~rent gain ~ of the transistor. ~ny deviation of Ic from thepredicted value is caused by variations in VbDc and re-sistive tolerances which result in i~perfect cancellation of the base current Ib and hence in an error in the transconduct-ance dIc of the circuit, i~e~ the variation of collector d Vi~
current Ic in dependence on Vin.
~he error in the quiescent current under D.C. con-ditions is the same as that in the A.C. case dealt with below, except for the additional error due to VbDc variations. As this error is su~stantially independent of the absolute value of the collector current Icg only the differential of Ic with respect to VbDc needs to be calculated. ~hus, frOm equation (8) dIc _ Rl dVbDc Rl~-k~17) R~ '~
which on account of equation (10) yields dIC
dVbDc and therefore dIC = dVbDc ( 11 ) R16 .. I
i~e. the effect of variation in VbDc as defined by e~uation l~
(4a), on the collector current Ic is inversely proportional to the resistance value of resistor 16.
~ lthough being sufficient for a calculation of the accuracy of the constant current source under D.C. conditions, ': " ,`~ ~. ' :
~, :
-, ~
~ 53 _9_ the first order approxi~ation expressed in equation (9) above is not precise enough to permit an assessment of the accuracy of the current source under A.C. condi-tions~ A good approxi-mation is, however, possible by using a sensitivit~ analysis, for a given resistor R3, involving the transconductance d c m dVin and the current gain ~ of the transistor~
Rearranging equation (8), using the relationship (8a) between Ic and Ib, and defining a resistor ratio X as X = ¦ 1 - R16R17 R3(R15 - kR17) gives . gm R3 r 15 1 x r 1 (12) l R15 - kR17 ~ L ~ +
~he sensitivity factor S for a fractional change m caused by a fractional chan~e A ~ is gm defined by ~ gm = S 4 ~ , providing (13) gm a measure for the dependence between the two variables.
In the limit of infinitesimal change.s dgm and d~
expression (13) becomes dgm = S d~ . I
gm ~ (14).l iOe~ gm x dg (15 Substit~ting for gm from e~uation (12) gives gB x dgm = X . (16) m d~ -X ~
.
: ~
It follows that the dependence f gm on ~ is only slight by vir~ue o~ X being a very small number approaching 0 in the case of ideal resistor. matching. Ideally9 the ~
~epandence o~ the transconductance is removed co~pletely but for practical values of X there still is a very slight dependence on account of the inite curr0nt gain.
~ xpressions similar to (16) can be derived for the sensitivity of the current source to changes of the trans-conductance gm with respect to R3 and X, ~iz:
3 dgm ~ -1 (17) gm dX
and X x dgm = -X ~18) gm dX X ~ ~
Equation (17) shows that variations in gm are direct-ly propor~ional to variations in R3~ underlining the neeessity for a very low tolerance resistor ~.
The ef~ect of variations in the resistor ratio on gm is very low as shown by (18), the physical explanation being that the tolerance of resistor ratio X proportionally effects the base current, but the base current itself is only a small 20 proportion of the total collector current.
~ he performance of the circuit shown in ~i~ure 2 has been evaluated in practical tests, with the resistor ratio X being realized to an accuracy of 0.03%O Measurements carried out showed no discernible difference over the audio frequency range between the predicted and the actual trans-conductanee.
Using di~ferent transistors 1 with eurrent gains between 20 and 100, worst possible resistor tolerances of 0.1% and a perfeet operational amplifier, the error between 3o predicted and measured performance is still less than ~ 0.2%
under ~.C. conditions~
~ ypical VariatiOnS in VbDc for a power transistor . are I 100 mV resuiting in a change of the eolleetor ourre~t ~ ..' ' :
.
1~35345~ .
~11--o approximately ~ 5G~A~ i.e. ~ 002% of the quiescent current.
he near-independence of the output current with res-ec~ to the gain of the transistor as achieved in the fore-going current source arrangement allows easy matching of two or more sucn current source arrangements.
Althou$n a current so~rce i~ accordance ~ith the invention will give, for the same applied DC control signal voltage, the same current output over a large range of tran-sistor gains, the value at which the source output stabilizes 10 nay vary slightly between different sources, for instance on account of resistor tolerances. These variations will, in general, be small enough to be o~ no conse~uence when replacing one source with another. ~owever, if two current sources are simply connected in series to work in a push-pull mode7 e.g~ to drive a telephone subscriber's line~ even the slightest mismatch between the current outputs of the two sources may present problems in that, as each of the current sources attempts to keep the common output current to a value appropriate to its own stable state, one or the other of the sources may saturate ~0 and cease to operate in the above described manner. A circuit arrangement designed to overcome this problem is shown in ~igure 3.
~ hus, each of the two current sources making up the circuit comprises an operational amplifier 1, ll, whose output is connected, by way of a resistor 16, 161 to the base of the transistor 2,21 respectively, with transistor 2 being an n-p-n transistor, and transistor 21 being a nominall~ matched, comple-mentary p-n-p transistor. ~he use of a matched pair of comple-mentary transistors 2,21 provides for the impedance presented 3 by each current source to the respective line of the two wire line 31 to bee~ual, since thereby the collectors 6 and 61 of the transistors 2 and 21 are connected respectively to the positive line (+) and the negative line (-) of the two wire line 31. ~he otner end of the two wire line 31 is connected to a load circuit (not shown) such as e~g. a subscriber's instru-ment ,~
~ .
11;~5349 ~ he feed back paths of each of the t~o current s~u-ces~ vhat is the n~ga~ive ~eed back path via resistors 1~
ar~d i4 , and the further ~eed back path including resistors 15 and 16, and 151 and 1~1 respec~ively, are identical in arrange-rent and function to the corresponding feed back paths of the circuit sho~n in ~igure 20 ~ he power sup ly to the arrangement is by means o~
voltage rail 29, carr~ing a negative voltage of suitable magni-tude~ and grounded voltage rail 30.
~lso connected between the voltage rails 29 and 30 is a resistor divider networX com~rising resisto~s 22, 23, 2~9 25, all having the same value7 and eoual resistors 19 and 191.
~ his resistor network determines the average voltage o~ the two wire line~ i.e. the mean of the voltages on the posi-tive and the negative line of the two wire line 31.
If the two current sources of the arrangement are perfectly matched~ the average voltage of the line pair lies half way between the voltage levels at rails 29 and 30. Ifg however, the t~o current sources are imperfectly matched, that is to say that they stabilize individually at different current levels of collector current for the same applied control input voltage, the average voltage of the two wire line ~1 moves away from this half way point, in a direction so as to decrease the voltage between the collector and the corresponding current rail of that transistor which draws the higher current and increase the corresponding voltage at the other circuit. This - shi~t of the average voltage at the two wire line, also termed common mode shift causes an e~ual and opposite change, with respect to the nearest current rail~ of the input voltage levels Vin and Vinl, the change bein~ such that the current flow through that transistor~ which initially drew the higher curr-ent, is reduced and the current ~low through the other tran-sistor is increased. ~his common mode shi~t is arrested when both transistors dra~ the same current, with the average vol-tage stabilizing at the new value. ~esistors 22 to 25 thus form, in conjunction with resistors 19 and 191, a third ~eed back path which ensures that the collector currents of the two tran-;
: ~ ~
~ ' :
:1135349 sis~ors 2 and 21 a re equal~ i.e~ that lc - ICl.
A diffarential voltage change, on the other hand, which causes t~e two lines of the two wire line 31 to move indivi-dually away from the average voltage in opposite directions, leaves the average voltage unchanged~ and thus will produce no cha~e in ~he levels of Vin and ~in IL trAerefore an anti-phase AC signal is applied to the terminals 20 and 201S and the output currents Ic and ICl vary~ in antiphase, in accordance w_th that signal, no ~eed back effect will be produced. Pro-10 vided also, that the resistors 22 to 24 are approximately equalto twice the appropriate value of resistor 18 of ~igure 2 in order to leave the above calculations unchanged, the third feed back path does not~ therefore, interfere with the ~C operation of the arrangement.
In a similar way, the curren~s Ic and ICl are not affected by any differential signal produced within the two wire line~ such as may be produced in a subscriber's instrumen~.
~ s aforesaid, the collector currents Ic and ICl are largely unaffected b~ differential signals generated witnin the two wire line circuit, but respond only to either a common mode voltage shift, due to e.g. imperfect matching of the two current sources or an as~Jmmetry in the line circuit on account of lea'~age currents, or to &ntiphase signals applied to the terminals 20 and 201. ~herefore, by roviding a further circuit (not shown) which is unaffected by variations in the collector currents, but de~ects differential signals which are generated within the two wire line circuit, the present arrangement may be incorporated in, and form part of an electronic nybrid cir-cuit. ~he detection of such line generated signals may be ~0 achieved in the following way. When signals are sent to the subscriber's instrument, that is when antiphase signals are applied to the terminals 20 and 201, then the emmiter and coll-ector vo]tages generated oy a given transistor are out of phase with each other; and by suitable addition these voltages may be made to cancel each other~ &nd conse~uently no output signal is provided by said furtner circuit. Differential signals whicn are generated within the line circuit do, however, produce in-phase collector and emmiter voltages at each of the =ransistors, :: .
:
113534'13 resu ving in an outpuv signal at the said further circuit.
he arrangement including a said further circuit tnus provides ~or a separation oi incoming and outging signals as is required ~or an eiectronic h~brid circuit such as may be used in the conversion from two wire to four wire transmission and vice versa.
Resistors 26 and 27 7 and non-linear devices 28 &nd 29 on the positive and the negative line respectively form part of an overvoltage or lightning protection arrangement. ~he devices 28 and 29 may e.g. be non-linear resistors, or zener diodes. In the constant current sources described above, known means for providing the DC control voltages, other than resistor divider networks ma~ of course be employed, and &ny other modificatio~s of the current sources above, which are obvious to those skilled in the art are included in the scope of the present inventionO
., ,~ ~
. . ,
Summing currents at the non-inverting input of the operational amplifier, one obvains:
Vin ~ Vref ~VO - Vref ~ o (1) Rl~ Rlg , 1.
: : :
~135349 Rearranging this expression gives:
Vref = Vi 15 , -- ~ VO P~l~ (2) R17 ~ R15R17 + R15 3ut; since current flow th~ough the resistor R3 will tend to stabilise at a value such that the volta~e at the emitter electrode 7 of the transistor 2 is virtually equal to Vref:
V = V~ef + Vb ~ IbR16 Defining VbDc ~ Vb ~ Iere (4a) Vb ~ref R e , (4b) writing re/R = k and substituting for Vb from equation (4b) in equation (3) yields:
VO = Vref (1 ~ k) ~ VbDc + IbR16 (5) Using this equation (5) in equation (2) gives V = Vin R15 ~ VbDc R17 ~ IbR16R17 ref l5 - kR17 R15 - kR17 -R15 - kR17 (6) Xowever, I ~ Vref Ib ' (7) and by substitution from equation (6) I = VinR15 ~ VbDcR17 ~ IbR16R17 Ib (8) R3(~15 kR17) R3(R15-~R17~ 3( 15 17) In order to make the collector curre~t Ic independent of the base current Ib, and hence of the current amplification ~, where c (8a) it is necessary to ensure that R3(R-15-kR17) - 9 . (9) ,, .... . . . . . ..
-, .
1~35349 ~ . .
or given k is small~ to a first order that R~
6 ~ ~ (10) ~ 15 R17Thus under D.C. conditions the accuracy of the current source subject to th~ restrictions (9) a~d ~10) abov~ is indaed dependent only on resistor matc`ning and independen~ of the cu~rent gain ~ of the transistor. ~ny deviation of Ic from thepredicted value is caused by variations in VbDc and re-sistive tolerances which result in i~perfect cancellation of the base current Ib and hence in an error in the transconduct-ance dIc of the circuit, i~e~ the variation of collector d Vi~
current Ic in dependence on Vin.
~he error in the quiescent current under D.C. con-ditions is the same as that in the A.C. case dealt with below, except for the additional error due to VbDc variations. As this error is su~stantially independent of the absolute value of the collector current Icg only the differential of Ic with respect to VbDc needs to be calculated. ~hus, frOm equation (8) dIc _ Rl dVbDc Rl~-k~17) R~ '~
which on account of equation (10) yields dIC
dVbDc and therefore dIC = dVbDc ( 11 ) R16 .. I
i~e. the effect of variation in VbDc as defined by e~uation l~
(4a), on the collector current Ic is inversely proportional to the resistance value of resistor 16.
~ lthough being sufficient for a calculation of the accuracy of the constant current source under D.C. conditions, ': " ,`~ ~. ' :
~, :
-, ~
~ 53 _9_ the first order approxi~ation expressed in equation (9) above is not precise enough to permit an assessment of the accuracy of the current source under A.C. condi-tions~ A good approxi-mation is, however, possible by using a sensitivit~ analysis, for a given resistor R3, involving the transconductance d c m dVin and the current gain ~ of the transistor~
Rearranging equation (8), using the relationship (8a) between Ic and Ib, and defining a resistor ratio X as X = ¦ 1 - R16R17 R3(R15 - kR17) gives . gm R3 r 15 1 x r 1 (12) l R15 - kR17 ~ L ~ +
~he sensitivity factor S for a fractional change m caused by a fractional chan~e A ~ is gm defined by ~ gm = S 4 ~ , providing (13) gm a measure for the dependence between the two variables.
In the limit of infinitesimal change.s dgm and d~
expression (13) becomes dgm = S d~ . I
gm ~ (14).l iOe~ gm x dg (15 Substit~ting for gm from e~uation (12) gives gB x dgm = X . (16) m d~ -X ~
.
: ~
It follows that the dependence f gm on ~ is only slight by vir~ue o~ X being a very small number approaching 0 in the case of ideal resistor. matching. Ideally9 the ~
~epandence o~ the transconductance is removed co~pletely but for practical values of X there still is a very slight dependence on account of the inite curr0nt gain.
~ xpressions similar to (16) can be derived for the sensitivity of the current source to changes of the trans-conductance gm with respect to R3 and X, ~iz:
3 dgm ~ -1 (17) gm dX
and X x dgm = -X ~18) gm dX X ~ ~
Equation (17) shows that variations in gm are direct-ly propor~ional to variations in R3~ underlining the neeessity for a very low tolerance resistor ~.
The ef~ect of variations in the resistor ratio on gm is very low as shown by (18), the physical explanation being that the tolerance of resistor ratio X proportionally effects the base current, but the base current itself is only a small 20 proportion of the total collector current.
~ he performance of the circuit shown in ~i~ure 2 has been evaluated in practical tests, with the resistor ratio X being realized to an accuracy of 0.03%O Measurements carried out showed no discernible difference over the audio frequency range between the predicted and the actual trans-conductanee.
Using di~ferent transistors 1 with eurrent gains between 20 and 100, worst possible resistor tolerances of 0.1% and a perfeet operational amplifier, the error between 3o predicted and measured performance is still less than ~ 0.2%
under ~.C. conditions~
~ ypical VariatiOnS in VbDc for a power transistor . are I 100 mV resuiting in a change of the eolleetor ourre~t ~ ..' ' :
.
1~35345~ .
~11--o approximately ~ 5G~A~ i.e. ~ 002% of the quiescent current.
he near-independence of the output current with res-ec~ to the gain of the transistor as achieved in the fore-going current source arrangement allows easy matching of two or more sucn current source arrangements.
Althou$n a current so~rce i~ accordance ~ith the invention will give, for the same applied DC control signal voltage, the same current output over a large range of tran-sistor gains, the value at which the source output stabilizes 10 nay vary slightly between different sources, for instance on account of resistor tolerances. These variations will, in general, be small enough to be o~ no conse~uence when replacing one source with another. ~owever, if two current sources are simply connected in series to work in a push-pull mode7 e.g~ to drive a telephone subscriber's line~ even the slightest mismatch between the current outputs of the two sources may present problems in that, as each of the current sources attempts to keep the common output current to a value appropriate to its own stable state, one or the other of the sources may saturate ~0 and cease to operate in the above described manner. A circuit arrangement designed to overcome this problem is shown in ~igure 3.
~ hus, each of the two current sources making up the circuit comprises an operational amplifier 1, ll, whose output is connected, by way of a resistor 16, 161 to the base of the transistor 2,21 respectively, with transistor 2 being an n-p-n transistor, and transistor 21 being a nominall~ matched, comple-mentary p-n-p transistor. ~he use of a matched pair of comple-mentary transistors 2,21 provides for the impedance presented 3 by each current source to the respective line of the two wire line 31 to bee~ual, since thereby the collectors 6 and 61 of the transistors 2 and 21 are connected respectively to the positive line (+) and the negative line (-) of the two wire line 31. ~he otner end of the two wire line 31 is connected to a load circuit (not shown) such as e~g. a subscriber's instru-ment ,~
~ .
11;~5349 ~ he feed back paths of each of the t~o current s~u-ces~ vhat is the n~ga~ive ~eed back path via resistors 1~
ar~d i4 , and the further ~eed back path including resistors 15 and 16, and 151 and 1~1 respec~ively, are identical in arrange-rent and function to the corresponding feed back paths of the circuit sho~n in ~igure 20 ~ he power sup ly to the arrangement is by means o~
voltage rail 29, carr~ing a negative voltage of suitable magni-tude~ and grounded voltage rail 30.
~lso connected between the voltage rails 29 and 30 is a resistor divider networX com~rising resisto~s 22, 23, 2~9 25, all having the same value7 and eoual resistors 19 and 191.
~ his resistor network determines the average voltage o~ the two wire line~ i.e. the mean of the voltages on the posi-tive and the negative line of the two wire line 31.
If the two current sources of the arrangement are perfectly matched~ the average voltage of the line pair lies half way between the voltage levels at rails 29 and 30. Ifg however, the t~o current sources are imperfectly matched, that is to say that they stabilize individually at different current levels of collector current for the same applied control input voltage, the average voltage of the two wire line ~1 moves away from this half way point, in a direction so as to decrease the voltage between the collector and the corresponding current rail of that transistor which draws the higher current and increase the corresponding voltage at the other circuit. This - shi~t of the average voltage at the two wire line, also termed common mode shift causes an e~ual and opposite change, with respect to the nearest current rail~ of the input voltage levels Vin and Vinl, the change bein~ such that the current flow through that transistor~ which initially drew the higher curr-ent, is reduced and the current ~low through the other tran-sistor is increased. ~his common mode shi~t is arrested when both transistors dra~ the same current, with the average vol-tage stabilizing at the new value. ~esistors 22 to 25 thus form, in conjunction with resistors 19 and 191, a third ~eed back path which ensures that the collector currents of the two tran-;
: ~ ~
~ ' :
:1135349 sis~ors 2 and 21 a re equal~ i.e~ that lc - ICl.
A diffarential voltage change, on the other hand, which causes t~e two lines of the two wire line 31 to move indivi-dually away from the average voltage in opposite directions, leaves the average voltage unchanged~ and thus will produce no cha~e in ~he levels of Vin and ~in IL trAerefore an anti-phase AC signal is applied to the terminals 20 and 201S and the output currents Ic and ICl vary~ in antiphase, in accordance w_th that signal, no ~eed back effect will be produced. Pro-10 vided also, that the resistors 22 to 24 are approximately equalto twice the appropriate value of resistor 18 of ~igure 2 in order to leave the above calculations unchanged, the third feed back path does not~ therefore, interfere with the ~C operation of the arrangement.
In a similar way, the curren~s Ic and ICl are not affected by any differential signal produced within the two wire line~ such as may be produced in a subscriber's instrumen~.
~ s aforesaid, the collector currents Ic and ICl are largely unaffected b~ differential signals generated witnin the two wire line circuit, but respond only to either a common mode voltage shift, due to e.g. imperfect matching of the two current sources or an as~Jmmetry in the line circuit on account of lea'~age currents, or to &ntiphase signals applied to the terminals 20 and 201. ~herefore, by roviding a further circuit (not shown) which is unaffected by variations in the collector currents, but de~ects differential signals which are generated within the two wire line circuit, the present arrangement may be incorporated in, and form part of an electronic nybrid cir-cuit. ~he detection of such line generated signals may be ~0 achieved in the following way. When signals are sent to the subscriber's instrument, that is when antiphase signals are applied to the terminals 20 and 201, then the emmiter and coll-ector vo]tages generated oy a given transistor are out of phase with each other; and by suitable addition these voltages may be made to cancel each other~ &nd conse~uently no output signal is provided by said furtner circuit. Differential signals whicn are generated within the line circuit do, however, produce in-phase collector and emmiter voltages at each of the =ransistors, :: .
:
113534'13 resu ving in an outpuv signal at the said further circuit.
he arrangement including a said further circuit tnus provides ~or a separation oi incoming and outging signals as is required ~or an eiectronic h~brid circuit such as may be used in the conversion from two wire to four wire transmission and vice versa.
Resistors 26 and 27 7 and non-linear devices 28 &nd 29 on the positive and the negative line respectively form part of an overvoltage or lightning protection arrangement. ~he devices 28 and 29 may e.g. be non-linear resistors, or zener diodes. In the constant current sources described above, known means for providing the DC control voltages, other than resistor divider networks ma~ of course be employed, and &ny other modificatio~s of the current sources above, which are obvious to those skilled in the art are included in the scope of the present inventionO
., ,~ ~
. . ,
Claims (8)
1. A current source arrangement in which the magni tude of current flowing in an output path of a transistor amplifier stage is arranged to be controlled by an input signal applied to an input of the arrangement, said input signal being applied to the transistor amplifier stage by way of a differential amplifier, comprising means to derive a negative feedback signal whose magnitude is dependent upon that of said current in said output path, means to apply said negative feedback signal to an input of said differential amplifier, means to derive a further feedback signal whose magnitude is dependent upon the value of input current to said transistor amplifier stage, and means to apply said further feedback signal as a positive feedback signal to an input of said differential amplifier whereby to reduce the dependence of the current flowing in said output path on the gain of the transistor amplifier stage.
2. A current source arrangement as claimed in claim 1, wherein said input signal is applied to a non-inverting input of said differential amplifier, to which is also applied said further feedback signal.
3. A current source arrangement as claimed in claim 1, wherein the said further feedback signal is derived from at least one first resistive element in the input signal input path of the transistor amplifier and applied to the non-inverting input of the differential-amplifier via at least one second resistive element, with the input signal being applied to the same input of the differential amplifier via at least one third resistive element.
4. A current source arrangement comprising a differential amplifier having inverting and non-inverting inputs and an output, and a transistor having an input electrode, an output electrode, and a control electrode, the input electrode being connected by way of a resistive element to one pole of power supply means and the output electrode being connected by way of a load circuit to the other pole of said power supply means, the output of the differential amplifier being electrically coupled to the control electrode of the transistor, means to apply a control signal to one input of the differential amplifier, feedback means to apply to another input of the differential amplifier a first feedback signal dependent upon the voltage developed in operation across said resistive element, means to derive a second feedback signal dependent on the current flow to the control electrode of the transistor, and means to apply said second feedback signal as a positive feed-back signal to said one input of the differential amplifier.
5. A current source arrangement as claimed in claim 4, wherein said means to derive the second feedback signal comprises a first resistive element connected between the output of the differential amplifier and the control electrode of the transistor, a second resistive element connecting the output of the differential amplifier to said one input thereof, and a third resistive element connecting said one input to a source of said control signal whereby said control signal is applied to said one input.
6. A current source as claimed in claim 4 wherein the control signal applied in operation of the current source to said one input has a DC signal voltage component derived from a resistor divider network.
7. A circuit arrangement for driving a two wire tele-phone subscriber line incorporating two current source arrangements in accordance with claim 1 operating in push-pull mode and having outputs of the respective transistor amplifier stages connected to respective lines of the two wire line, and means to apply said input signal to one of said two current source arrangements inverted with respect to the other of said current sources.
8. A circuit arrangement as claimed in claim 7, wherein a third feedback path comprising a resistor divider network interconnecting said respective lines and the respective inputs of said current source arrangements is arranged to hold the mean voltage of both lines of said two wire line at a level intermediate the voltage provided by power supply means to the circuit arrangement.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB47475/77 | 1977-11-15 | ||
| GB4747577A GB1602296A (en) | 1977-11-15 | 1977-11-15 | Current sources |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1135349A true CA1135349A (en) | 1982-11-09 |
Family
ID=10445118
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000316238A Expired CA1135349A (en) | 1977-11-15 | 1978-11-14 | Current sources |
Country Status (6)
| Country | Link |
|---|---|
| BE (1) | BE872025A (en) |
| CA (1) | CA1135349A (en) |
| FR (1) | FR2408948A1 (en) |
| GB (1) | GB1602296A (en) |
| NL (1) | NL7811241A (en) |
| WO (1) | WO1979000295A1 (en) |
Families Citing this family (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2135846B (en) * | 1983-02-04 | 1986-03-12 | Standard Telephones Cables Ltd | Current splitter |
| NL8400508A (en) * | 1984-02-17 | 1985-09-16 | Philips Nv | AUDIO TRANSMISSION CIRCUIT WITH TRANSMITTER AMPLIFIED AS LINE VOLTAGE STABILIZER. |
| NL8500674A (en) * | 1985-03-11 | 1986-10-01 | Philips Nv | TRANSMITTER AMPLIFIED AS LINE VOLTAGE STABILIZER WITH CHANGEABLE REFERENCE VOLTAGE. |
| JPH0826911B2 (en) * | 1990-02-07 | 1996-03-21 | 三菱電機株式会社 | Electromagnetic clutch current controller for automobile |
Family Cites Families (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| BE568224A (en) * | 1957-06-07 | |||
| NL7505506A (en) * | 1974-05-15 | 1975-11-18 | Analog Devices Inc | TRANSISTOR AMPLIFIER OF THE DARLINGTON TYPE. |
| DE2626570C2 (en) * | 1976-06-14 | 1986-07-03 | Sachs Systemtechnik Gmbh, 8720 Schweinfurt | Regulated power supply device for a water sterilization device |
-
1977
- 1977-11-15 GB GB4747577A patent/GB1602296A/en not_active Expired
-
1978
- 1978-11-14 FR FR7832136A patent/FR2408948A1/en active Granted
- 1978-11-14 WO PCT/GB1978/000041 patent/WO1979000295A1/en unknown
- 1978-11-14 CA CA000316238A patent/CA1135349A/en not_active Expired
- 1978-11-14 NL NL7811241A patent/NL7811241A/en not_active Application Discontinuation
- 1978-11-14 BE BE191745A patent/BE872025A/en not_active IP Right Cessation
Also Published As
| Publication number | Publication date |
|---|---|
| FR2408948A1 (en) | 1979-06-08 |
| FR2408948B1 (en) | 1984-05-04 |
| NL7811241A (en) | 1979-05-17 |
| BE872025A (en) | 1979-03-01 |
| GB1602296A (en) | 1981-11-11 |
| WO1979000295A1 (en) | 1979-05-31 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US4302726A (en) | Current sources | |
| US5182526A (en) | Differential input amplifier stage with frequency compensation | |
| CA1192274A (en) | Current amplifier | |
| US6028482A (en) | Wide dynamic range transimpedance amplifier circuit | |
| US4207475A (en) | Efficient bipolar regulated power supply | |
| CA1135349A (en) | Current sources | |
| JPH0710082B2 (en) | Telephone circuit | |
| US4217555A (en) | Amplifier circuit arrangement with stabilized power-supply current | |
| EP0502056A1 (en) | Difference amplifier apparatus employing an input attenuator network. | |
| US4585987A (en) | Regulated voltage supply with sense current cancellation circuit | |
| JP2862296B2 (en) | Voltage applied current measuring device and current applied voltage measuring device | |
| US4612513A (en) | Differential amplifier | |
| US4107618A (en) | Amplifier system having signal gain independent of a reference voltage | |
| US3243719A (en) | A. g. c. circuit including a constant impedance variable-attenuation network utilizing current-sensitive impedances | |
| US4634986A (en) | Log amplifier with pole-zero compensation | |
| JP3003174B2 (en) | Amplifier circuit | |
| US2762005A (en) | Motor speed control | |
| US3950708A (en) | Gain-controlled amplifier | |
| GB2073524A (en) | Class ab amplifying circuit | |
| WO1985000684A1 (en) | Two wire circuit having an adjustable span | |
| CA2144420C (en) | Circuit arrangement for an integrated output amplifier | |
| JPS6228606B2 (en) | ||
| KR850000359B1 (en) | In-phase voltage elimination circuit for hall element | |
| US3499169A (en) | Variable frequency ac regulator | |
| KR850000358B1 (en) | In-phase voltage elimination circuit for hall element |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| MKEX | Expiry |