CA1109552A - Power supply apparatus - Google Patents

Abstract

ABSTRACT OF THE DISCLOSURE
A series connection between a pair of dividing capa-citors and a series connection between a PNP transistor and an NPN transistor are connected in parallel with a direct current voltage source. A primary winding of a saturable transformer is coupled between the junctions of the respective series connec-tions. The saturable transformer comprises a pair of feedback windings coupled to the primary winding. The pair of feedback windings are connected between the base and emitted electrodes of the respective corresponding transistors through the respective base resistors. The emitter electrodes of the respective transistors are connected to the direct current voltage source sides. A series connection of two capacitors is connected between the junctions of the respective feedback windings and base resistors, and the junction of the two capacitors is connected to the junction of the series connection of the dividing capacitors. A secondary winding is coupled to the primary winding of the saturable transformer and a direct current output voltage is obtained through rectification of the output from the secondary winding.

Description

i2 .~ ~
The pres~nt invention relates to a power supply apparatus. More specifically, the present invention relates to a direct current power supply apparatus employing a direct current/
alternate current inverter, par~icularly suited or small-sized and light-weight electrical equipment.
A sPlf-excited inver~er employing a Royer oscillator has been proposed and put into practical use as an inverter for use in a power supply apparatus. Since such a Royer-type inverter is well known, it is not believed necessary ~o describe the same in detail. Briefly described, a Royer-type inverter comprises two power transistors which are each utilized as a switching device such that a direct curren~ is on/off controlled or switched to provide an alternate current voltage of a rectangular wave formO
The switching operation of the transistors is achieved by a feedback coil coupled to the primary wlnding of a saturable transformer. In operation, one transistox is rendered conductive while the other transistor is rendered non-conductive for a given time period. Then the operation state is reversed and thereafter this sequence is repeated. In a Royer-type inverter, the excited voltage in th~ primary winding of the saturable transformer which serves as a common load of the respective transistors causes a voltage to be applied between the collector and emitter electrodes in the non-conduction stater This applied voltage may equal approximately two times the power supply voltage.
Accordingly, transistors of a higher withstand voltag are requlred for such an inverter.
- ~ A so-called half-bridge type inverter has been proposed as an improvement to the ~oyer-type inverter. In a typical example of a half-bridge-type inverter the voltage from a direct current power supply is halved by a pair of dividing capacitors.

~1~5~

Accordinyly, only half the power supply voltage is applied between -the collector and emitter electrodes of the transistor in a conductive state. It follows that transistors havir.g a lower withstand voltage can be used as compared to -those used in a Royer-type inverter.
However, even a half-bridge-type inverter still has - several problems to be solved. Specifically, because a starting resistor need be employed across one of two power transistors, the biasing circuits of the two otherwise matched power transistors becomes asymmetrical. In the result, the power transistor across which the starting resistor is connected is never rendered fully non~conductive but remains somewhat conductive even during its non-conductive interval. The output of the inverter i8 therefore asy~metrical, resulting in poor efficiency and perhaps causing a short circuiting of the two power transistors.
Accordingly, a principal object of the present invention is to provide a power supply apparatus which seeks to ~ overcome the above disadvantages of the prior art.
`~ 20 According to the present invcntion, then, there is provided a power supply apparatus comprising an inverter for converting a direct current output from a direct current voltage source .into an alternating current voltage, the inverter comprising: first and second dividing capacitors connected in ser1es, the capaci-tors being arranged to be connected in parallel wlth the direct current voltage source ~irst and second trans-istors having their main current-carrying paths connected in series, the series-connected paths also being arranged to be connected in parallel with the direct current voltage source; a saturable transformer having a primary winding connected between .~ .

.
.: , . .. . . . .

the junction of the dividing capacitors and the junc-tion of the main current-carrying paths of the transistors; biasing means for each of the transistors, each biasing means comprising a respective feedback winding magnetically coupled to the primary winding and connected in such a manner that the voltage induced thereacross by current flowing through the primary winding when the respective transistor is conductive is operable to maintain the -transistor in its conductive state until satu-ration occurs, the arrangement being such that in use the transistors are alternating conductive and thus cause an alternating current to flow through the primary winding; and the biasing means of the transistors being arranged to cause substantially symmetrical operation of the transistors.
According to a further aspect of the present invention, - there is also provided an electric shaver comprising: a casing ; having an openiny; a shaver cutt r assembly provided in -the shaver casing so as to be exposed through the opening, the :
; shaver cutter assembly comprising a stationary cutter and a movable cutter; a power supply apparatus provided within the shaver casing, the power supply apparatus comprising an inverter for converting a direct current output from a direct current voltage source into an alternating current voltage, the inverter comprising: first and second dividing capacitors connected in seri~s, the capacitors being arranged for connection in parallel ~-with the direct current voltage source; first and Second trans-istors~having their main current-carrying paths connected in ;
series, *he series-connected paths also being arranged ~or connection in parallel with the direct current voltage source;
, .
a saturable transformer having a primary winding connected between -the junction of the dividiny capacitors and the junc~ion .
~ ~ 3a -;~

:

of the main current-carrying paths of the transistors; and biasing means for each of the transistoxs, each biasing means comprising a respective feedback winding magnetically coupled to the primary winding and connected in such a manner that the voltage induced thereacross by current ~lowing through the primary winding when the respective transistor is conductive is operable to maintain the transistor in its conductive state : until ~turation occurs, the arrangement being such that in use the transistors are alternatively conductive and -thus cause an alternating current to flow through the primary winding; the biasing ~eans of the transistors being arranged to cause substantially symmetrical operation of the transistors; and the saturable transformer further including a secondary winding and the shaver further comprising motor means housed ~:
in the shaver casing and coupled to the secondary winding of : the saturable transformer for driving the movable cutter of the shaver cutter assembly.
Embodiments of the present invention will now be : described in greater detail and will be bet-ter understood when read in conjunction with the following drawings in which:
FIGURE 1 is a schematic diagram of a conventional halfbridge-type inverter;
FIGURE 2 is a schematic diagram of one embodimen-t of the present inven-tion;

FIGURE 3 ia a schematic d.iagram of a preferred embodi-ment of ~he prèsent invention;
. FIGURES 4A and 4B are load characteristic curves of : the transistors employed in the Figu~e 3 emhodiment~
` 30 ~
: :
- 3b -.~

:

FIGURE 5 is a schematic diagram of another preferred embodiment of the present invention;
FIGURES 6A and 6B are wave ~orms of the outpu~ ~rom the inverter shown in Figure 5, FIGURES 7A th.rough 7J show wave forms of th~ electrical signals at various portions in ~he Figure 5 embodiment, FIGURE 8 is a schematic diagram of a ~urther preferred embodiment of the present invention;
FIGURE5 9 and 10 are wave forms of the collector current in the Figure 8 embodiment;
FIGUR~ 11 is a w~ve form of the voltage across the time constant capacitor in the Figure 8 embodiment;
FIGURE 12 is a schematic diagram of still another preferred embodiment of the present invention;
FIGURE 13 is a schematic diagram of still a further prefèrred embodiment of the presen~ invention;
FIGURE 14 shows a fluctuation characteristic of a charging curren~ with respect to the source voltage fluctuation ~in the Figure 13 embodiment;
FIGURE 15 is a schematic diagram of still a further preferred embodiment of the present invention;
FIGURE 16 is a schematic diagram of still a ~urther preferred embodiment of the present invention;
FIGU~E 17 is a wave ~orm of a charging voltage of the secondary battery in the Figure 16 embodiment;
: FIGURE 18 is a schematic d1agram of still another preferred embodiment o~ the present invention; and FIGURE 19 ~hows a noi~e characteristic.
~ Re~erring to Figure 1, in a known half-bridge-type in~erter, when a switch 3 is turned on, a direct current ` ~ .

, voltage from a direct current voltage source 2 causes a current to flow through base resistors 7 and 11 of a pair of PNP
transistors 6 and 10. The voltage drop across the respective base resistors 7 and 11 will rénder the pair of ~NP transistors 6 and 10 conduc~ive. A starti~g resistor 13 is connected between the base and collector electrodes of the transistor 10. Hence, the current flowing through the base re~istor 11 is larger than the current flowing through the other base resistor 7 and the transistor 10 is rendered conductive earlier than the transistor 6. When` the transistor 10 is rendered conduc~ive, the electric charge in a dividing capacitor 9 is discharged through the transistor 10 and a primary winding 5 wound on a saturable magnetic core 14. On th~ other hand, a dividing capacitor 4 starts being charged by a current ~lowing thro~gh the transistor 10 and the primary winding 5 from the direct current voltage source 2. At that time, the current flowing from the point a to the point b of the primary winding 5 induces a voltage across one feedback winding 12 in the direction ~or forward biasing the transistor 10 and al50 induces a reverse bias voltage for the transistor 6 in the other feedback winding 8. Accor-dingly, the transistor 10 is rendered co~ductive, while ~he transistor 6 is rendered non-conductive, with the result that a positive half-wave output is generated in a secondary winding 15 wound on the magnetic core 14 and coupled to the primary winding 5.
Thusg the primary winding 5 is excited. When the primary winding~5 is thus excited and the magnetic core 14 of the saturable trans~ormer i~ ~aturated, the magnetomotive.force of the magnetic core 14 disappears and a voltage is induced in the - 30 respective feedback windings 8 and 12 in the reverse direction~

.

d - , . : ' ',,~ ' , ' ' :

i52 Therefore, one transistor lO is rendered non~conductive and the other transistor 6 is rendered conductive. Accordingly, the electric charge in the dividing capacitor 4 is discharged through the primary winding 5 and the transistor 6 and the dividing capacitor 9 is charged by a current flowing through the primary winding 5 and the transistor 6 from the direct current voltage source 2. The above-described charging current causes the primary winding 5 to be excited in the direction from the point b to the point a, with the result that a negative half-wave output is obtained in the secondary winding 15.
The output from the inverter l is obtained throu~h the secondary winding 15 which is wound on the saturable magnetic core 14. Winding 15 is coupled to primary winding 5O The output is applied to a load circuit 200, which may comprise a direct current motor, for example.
Since in such a half-bridge-type inverter as shown in Figure l, the voltage E of the direct current voltage source 2 is divided by two by means of a pair of dividing capacitors 4 and 9, the voltage E/2 is applied between the collector and emitter electrodes of the transistor in conduction. As a result, according to a hal~-bridge-type inverter, transistors of a lower withstand v~ltage can be employed as compared with a case of a Royer-type inverter.
As mentioned aboYe, problems remain however.
Specifically, transistox lO need be provided with a startiny res~stor 13 for preferentially rendering transistor lO conductive.
Because of the ~tarting resistor 130 the bias circuits for the two transistors 6 and lO selected as;,a pair in terms of their electrical characteristics become asymmetrical. This causes a situation wherein the transistor 10 is not rend~red fully , . . .

.

~1~5~

non-conductive or cut off, i.e., a somewhat conductive tendency exists even in the non-conduction period. In order to avoid such a situation, a reverse bias could be applied between the base and emitter electrodes of the transistor 10, although there remains some difficulty. Ano her problem i 5 that the above-described conductive tendency of the transistor 10 makes the output wave of the inverter 1 asymmetrical with respect to the positive and negative polarities, resulting in poor efficiency. This fact is also liable to cause short-circuiting of transistors 6 and 10.
Referring now to Figure 2, circuit configuration of a direct current/alternate current inverter for use according to an en~odiment of the present invention is shown. The embodiment shown comprises a series connection of a paix of dividing capacitors 104 and 109 and a series connec~ion of a PNP
transistor 106 and an NPN transistor 110, both series connections being coupled in parallel with a direct current voltage source

2 through a switch 103. A primary winding 105 wound on a saturable ~agnetic core 114 for constituting a saturable trans-former is interposed between the junctions a and b of the res~ective series connections. The saturable transformer ~urther comprises a secondary winding 115 and a pair of feedback windings 108 and 112 wound on saturable magnetic core 114 so as to be magnetically coupled to primary winding 105. Secondary winding 115 is connected to a load circuit 200.
One end of each o~ feedback windings 108 and 112 is ` connected to the base electrodes o~ transistors 106 and 110 re6pectively. ~he othex ends of each of feedback windings 108 and 112 are connected to each of the emitter electrodes o~
transistors 106 and 110 respectively through each of respective .. ....

:.

base resistors 107 and 111 so as to constitute a respectivebase biasing circuit. A capacitor 116 is interposed between the junction c of feedback winding 108 and base resistor 107 and the junction d between feedback winding 112 and base resistor lllo The capacitor 116 is shunted by a resistor 117 of a large resistance value.
When switch 103 is closed, a current flows through base resistors 107 and 111 and capacitor 116. The voltage drop across respective base resistors 107 and 111 will render xespective transistors 106 and 110 conductive. However, because of the differin~ characteristics of the components such as transistors 106 and 110, ~eedback windings 108 and 112 and base resistors 107 and 111, one transistor is likely to ke conductive earlier than the other. Assume that transistor 106 is likely to be conductive earlier than transisto~ 110 in the embodiment shown~
When PNP transistor 106 becomes conduc~ive, the charge in `~ dividing capacitor 104 is discharged through transistor 106 and primary winding 105. At the same time, dividing capacitor 109 starts being charged by a current ~lowiny through transistor 106 and primary winding 105 from direct current voltage source 2.
At tha~ time, the current flowing from the node b of primary windlng 105 to the node a induces a voltage in one feedback winding 108 in the direction for forward biasing transistor 106 and a voltage in the other feedhack winding 112 for revexse biaslng transistor 110. Accordingly, transistor 106 is rendered conductive and transistor 110 ls rendered non-conductive, with the result that a positive half~wave output is obtained in second winding 115.
The primary winding 105 is thus ex~ited and the 30 ~magnetic core 114 o~ the saturable transformer is saturated.

~: .
. ~ .. .~ . , . . .. . :

:. . . ,: ~ . :

d~2 When magnetic core 114 is saturated, the magnetomotive force of magnetic core 114 disappears and a reverse directional voltage is induced in feedback windings 108 and 112. There-fore, one transistor 106 is rendered non-conductive and the other transistor 110 is rendered conductive. Accordingly, the electric charge in dividing capacitor 109 is discharged through primary winding 105 and transistor 110 and dividing capacitor 104 is charged by a current flowing through primary winding lOS and transistor 110 from the direct current voltage source 2. The above-described charging current causes primary winding 105 to be excited in the direction from node a to node b, with the result that a negative half-wave output is obtained in secondary winding 115.
Thereafter, the conduction state of ~he transîs~ors is reversed each time magnetic core 114 of the saturable transformer is saturated as described above and a positive half-wave output -and a negative half-wave output are obtained alternately in secondary winding 115.
One ~eature to be noted in the Figure 2 embodiment is 20 the connectlon of capacitor 116 and resistor 117. The said connection of capacitor 116 and resistor 117 causes an ample current to flow through the base biasing circuit at the initial " stage of the operation, eliminating the necessity for starting resistor 13 as illustrated in Figure 1. Additionally, transistors 106 and 110 are implemented by compl~mentary txansistors. The emitter electrodes of each of these transistors are coupled to the direct current voltagç source side. As a result, the base biasing circuits of transistors 1~6 and 110 can be provlded on the voltage source side with respect to primary winding 105 of the transformer, making it possible to , _ g _ , '" .. "' '' ' ' ' ' '" ' ' ' ' "' ' ' ' ' " ;
': ' ' ' ' ' ' ' . ., ' S5~:

symmetrically construct the above-described pair of base biasing circui-ts. Accordingly, the output from the present inverter is symmetrical. The above-described symmetrical circuit configuration of the base biasing circuits facilitates the selection of impedance values for the various impedance components such as resistors and capacitors used in the base biasing circuits. Further, the continuous conductivity of one of the two power transistors described in conjunction with the prior art can be avoided. The short-circuiting o~ transistors 106 and 110 can therefore also be avoided.
Referring to Figure 3, a further preferred emhodiment of a power supply apparatus emp7oying the above-described inverter will be described. It is pointed out that the embodiment shown is particularly suited as a power supply for electrical equipment using relatively small direct current motors such as electric razors, motor-driven tooth brushes and the like. Thus, the embodiment is shown comprising a storage battery 203 and a direct current motor 205 in the load circuit 200. Both terminals of secondary winding 115 of the saturable transformer are coupled to rectifying diodes 201 and 202 constituting a rectifying circuit. The output of the rectifying circuit is connected to one end of storage battery 203. The other end o~ storage battery 203 is connected to a center tap 115a of secondary winding 115 through a limiting resistor 204~
In addition, both terminals of storage battery 203 are connected to direct current motor 205 thxough a switch contact 206.
Accordinyly, when switch contact 206 is closed, the limuting resistor 204 i9 short-circuited and ~torage battery 203 is charged with a full-wave rectified output obtained by rectifying an alternate current output in a rectangular wave form ';~
',. -' ,," ~,',', ,,, ~ . ' . ' at secondary winding 115 of inverter 100 by means of diodes 201 and 202. It is pointed out that direct current motor 205 is also energized with the above-described full-wave rectiEied output.
The direct current voltage source 2 of inverter 100 may comprise an alternate current voltage source 21 such as a commercial power supply and a bridge circuit 22 for full wave rectification of the alternate current output from alternate current voltage source 21. Both terminals of bridge circuit 22 are connected to the input termunals of inverter 100 at the output terminals of direct current voltage source 2.
Inverter 100 shown in Figure 3 is different from the Figure 2 inverter in the following respects, Speci~ically, in the Figure 3 embodiment, capacitor 116 and resistor 117 are each implemented by series connections o~ the two capacitors 116a and 116b and two resistors 117a and 117b, swch that the junctions of the respective series connections are each connected to the junction a of dividing capacitors 104 and 109.
It has been observed that the connec*ion of capacitors 116a and 116b to the point a reduces a spike voltage occurring across primary winding 105. More specifically, a spike voltage occurring across primary winding 105 cau~es a current to flow through capacitors 116a and 116b and feedback windings 108 and 112 to the base and collector junctions of transistors 106 and 110 respectively. This spike voltage also causes a current to flow through capacitors 104 and 109 to the emitter and collector junctions of transistors 106 and 110 in the reverse direction.
Accordingly, the spike voltage is absorbed in the inverter.
In the Figure 3, a capacitor 118 is connected across primary winding 105. The said capacitor 118 also serves to :

.. . . . . . .
: ' ' . , . '' : . . ' : .

absorb the spike voltage occurring in primary windiny 105. More specifically, without capacitor 1~8, it is possible that the voltage to transistors 106 and 110 will fall outside the region of safe operation enclosed by dotted line A in Figure 4A of the load characteristic curve. However, it has been observed that the addition of capacitor 118 as described above serves to confine the operation of transistors 106 and 110 within the region of safe operation A as shown in ~igure 4B. In Figures 4A
and 4B, the abscissa indicates the collector/emltter voltage (VcE) of the transistors, while the ordinate indicates the collector current IC of the transistors. Since the operation of inverter 100 in Figure 3 is the same as that of the funda-mentàl circuit configuration shown in Figure 2, the details of the inverter will not be repeated here.
; Figure 5 is a schematic diagram of another preferred ; embodiment of the present invention, wherein an improvement in the inverter i9 adopted. In comparison with the Figure 3 embodiment, the Figuxe 5 embodiment comprises diodes 119 and 120 coupled between the collector and emitter electrodes of transistors 106 and 110 respectively in the reverse direction.
As the remaining portions of the Figure 5 embodiment are substantially the same as those illustrated in Figure 3, only the salient features of the Flgure 5 embodiment will be described in greater detail below. Without diodes 119 and 120, secondary winding 115 provides a positive half-wave output o~
pulsive form as seen from Figure 6B. The no~mal wave form should be as shown in Figure 6A. Such pulsive portions P cause an over w ltage to be applied between the base and emitter electrodes of transistors 106 and 110 in the reverse direction~
whereby transistors 105 and 110 create heat and the wave form of the output from inverter 100 is distorted. Further, a low '. ' .'' .'' ' , : - .
. . . " , ~ ' ' ~, :

~` :
r~
frequency vibratory noise emanates from the saturable transformer.
;r` Actual measurement show that the wave form of the base/emltter voltage VBE of -transistor 110, ~or example, is as shown in Figure 7A. Similarly, the wave forms of the base current IB, the collector current IC and the emitter current IE f transistor 110 are shown in Figures 7s, 7C and 7D, respectively.
Thus, a conduction period of transistor 110 is shown as W in Figure 7A. Referring to Figure 7A, it is supposed that the base/emitter voltage VBE should be positive in response to the rise of the base current IB and should remain positive as shown by the dotted line. However, in actuality, a negative valley portion appears, represent~d by the solid line in Figure 7A. It is presumed that the negative voltage portion cau~es the above-described pulsive portions P.
When transistor 110 is shifted from the non-conduction state to the conduction state, a current flows through primary winding 105 in a direction from point a to point b~ However, before that, a current has been flowing through primary winding 105 in a direction from point b to point a by virtue of the conductivity of transistor 106. As a result, a counter electro-motive force is generated across primary winding 105. The said counter electromotive force causes a current I105 to flow ~hrough primary winding 105 in the negative direction, i.e., in the `
direction from point b to point a, as shown in Figure 7~, irrespective of the conductivity of transistor 110. The wave forms of the current I116b of capacitor 116b, the current I117b of resistor 117b and the current Illl of base resistor 111 are shown in Figures 7E, 7F and 7G, respectively. Taking into consideration the above-described wave forms, it is presumed that the above-described negative current in primary winding 105 flows . ` - . .

, .

.
, from the base electrode to the collector electrode of transistor 110 through capacitor 116b and resistor 117b of the parall 1 circuit and the feedback winding 112, whereby transistor 110 is reverse-biased and the negativé current flows through dividing capacitor 109 and through the emitter/collector junction o~ tran-5istor 110. In other words,it is presumed tha the ne~ative current causes a current to instantaneously flow through transistor 110 in the reverse direction, whereby transistor 110 instantaneously serves as a so-called backward transistor. ~ence, the negative voltage portion shown in Figure 7~ in base/emitter voltage VBE.
It is presumed that pulsive portions P are generated to cut off transistor 110 which is about to become conductive due to the counter electromotive forceO
In the embodiment shown, the counter electromotive force occurring in primary winding 105 bypasses transistor 110 : by means of diode 120. Hence, a current is prevented from flowin~ through the emitter/collector junction. ~s a result, the output of the wave form as shown in Figure 6A is obtained from secondary winding 115, wherein the pulsive portions P shown in Figure 6B have disappeared. Thus, the wave forM of the output from inverter 100 is not distorted and the low-frequency vibratory noise from the transformer lS eliminated. The wave form of the base/ mitter voltage VBE of transistor 110 in such a situation is shown in Figure 7I and the wave form of the current I120 of diode 120 is shown in ~igure 7J. Although in the foregoing only one transistor 110 and diode 120 were described, the same applies to the tran~istor 106 and the diode 119.
As described above 9 wave distortion of the output o~ inverter 100 and transform~r vibratory noise due to spike - i4 - .
~ ' .
~ ., , , . . .
.

voltage are eliminated by the use of diodes 119 and 120 connected in reverse direction between the emitter and collector electrodes of transistors 106 and 110.
As will also be appreciated, if transistors 106 and 110 are selected to have reverse characteristics, then diodes 119 and 120 can be dispensed with. Specifically, since diodes 119 and 120 are connected to bypass a reverse directional current between the collector and emitter electrodes of transistors 106 and 110, diodes 119 and 120 can be dispensed with, if transistors 106 and 110 each function satisfactorily as backward transistors.
Figure 8 shows a schematic diagram of a further preferred embodiment of the present invention, wherein it is contemplated that diversified current amplification factors of transistor 106 and 110 are absorbed by connecting a pair of RC time constant circuits 121 and 122 to the base biasing circuits of transistors 106 and 110. The RC time constant -` circuits 121 and 122 comprise a series connection of a resistor 121a and a capacitor 121b and a series connection of a resi~tor 122a and a capacitor 122h, respectively. These resistors 121a and 122a each function as a charging/discharging resis-tor of the corrésponding capacitors 121b and 122b, res-pectively. More specifically, the charging/discharging time constant of each of the circuits 121 and 122 is determined by the resistor 121a and the capacitor 121b, and the resistor 122a and the capacitor 122b, respectively. A further modification illustrated in Figure 8 is the noise filter circuit 23 which includes a choke coil and a capacitor inter-posed between an alternate current voltage source 21, such as a commercail voltage source, and bridge circuit 22. The : :

above-described time constant circuits 121 and 122 are inter-posed in the base biasing circuits of inverter 100.
In opera-tion, when switch 103 iS turned on~ a current flows through the pair of time constant circuits 121 and 122 and capacitors 116a and 116b. The voltage drops in respective time constant circuits 121 and 122 cause transistors 106 and 110 to be turned on. In such a situation, because of the diversified characteristics of circu.it components SUCh as transistors 106 and 110, feedback windings 108 and 112 and time constant circuits 121 and 122, either transistor may become ~.

`'`
': ' ~ .

:
:

:

3~ .

- 15a ~

... . .

conductive before the other. Assuming thak transistor 106 becomes conductive ~irst, then substantially the same operation as is described in conjunction with the embodiment : of Figure 2 follows thereafter. When the other transistor 110 becomes conductive, then collector current IC flow8 as shown in Figure ~ and the saturable magnetic core 114 of the trans-former becomes magnetically saturated immediately before the collector current Ic disappears. As a result, a peak current Ic, as shown by the dotted line in Figure 9 flows. The voltage - 10 V109 of dividing capacitor 109 abruptly decreases at such peak point to become voltage V109,. Assuming that capacitor 116b is not connected, the fluctuation ~V of the base voltage of transistor 110 in such voltage transition can be expressed by . the following equation:

~V = 122a -- (V
R122a R117b 109 109' :~ where R122a and R117b are the resistance values of resistors 122a and 117b.
Since the abo~e-described fluctuation V is small, : the base current of transistor 110 does not abruptly decrease and the collector current gives rise to a peak current Ic, as shown by the dotted line in Figure 9. By contrast, with capacitor 116b connected, since capacitor 116b is charged to the voltage V116b, the fluctuation ~V' o~ the base voltage of transistor 110 when dividing capacitor 109 is discharged -through primary winding 105 may be expressed by the followi~g e~uation:

( log V116b) - (v~og 1 ~ Vll~b) ; V109 Vlos I

' ?.

.

, The fluctuation av~ is larger than the fluctuation ~V
described previously and can decrease the base current of transistor 110~ As a result, the collector current IC exhibits : certain characteristics as shown by the solid line in Figure 9, wherein no peak characteristic is seen. The same applies to the collector current of transistor 106. Further, the current load and heat in respective transistors 106 and 110 can be mitigated.
According to the embodiment shown, the collector current IC of each of transistors 106 and 110 exhibits characteristics without a peak as shown in Figure 9. However, the cut-off time point t is differen~ depending on the value of the current amplification factor of the respective transistors 106 and 110. More specifically, and referring to Figure 10, the cut-off time point t2 of the ~ransistor having a smaller current amplification factor comes later than the cut-off time - point tl of the transistor having a larg~r current amplification factor. As a result, a difference ~t in the conductlon period of the transistors of different current amplification factors occurs. This entails asymmetry in the positive- and negative~
going directions in t~e inverter output. For example, refe.rring to Figure 8, assuming that transistor 106 has a smaller current ampliication factor compared with transistor 110, then the base switching voltage or the base threshold value of transistor : 106 ls largex than that of tran~istor 110. Therefore, the voltage V104 o~ dividing capacitor 104 corre~ponding to translstor 106 is higher than the voltage Vlog of the other dividing capacitor 109. It follows that supply voltages V10~
andV1O9 will differ ~rom each other even when the time constants of time constant circuits 121 ana 122 provided in ~he base 30 biasing circuits of transistors 106 and 110 are the sameO

- 17 ~

Accordingly~ the voltages V121b and V122b 121b and 122b exhibit the characteristics shown in Figure 11~
One transistor 106 is rendered non~conductive by virtue of the base voltage corresponding to the v~ltage vl21b, in Figure 11 : and similarly transistor 110 is rendered non-conductive by virtue of the base voltage corresponding to the voltage V122b, in Figure 11. Accordingly, the conduction periods of transistors 106 and 110 become substantially the same even when their curre~t amplification factors differ from each other~ A diversified difference in temperature increase of transistors 106 and 110 is also reduced.
Figure 12 is a schematic diagram of still another preferred embodiment of the present invention, wherein capacitors 116a' and 116b' are connected in parallel with the r spective base resistors 107 and lllr In previously described embodiments, capacitors 116a and 116b were described as being connected in ~.
parallel with resistors 117a and 117b. The modification of ~: the circuit of Figure 12 brings abouk the following advantages.
In the case where capacitors 116a and 116b are connected in parallel with resistors 117a and 117b as shown in Figure 3, the capacitors 116a and 116b mus~ possess an increased withstand voltage, inasmuch as the resistance values of resistors 117a ana 117b are larger than those of base resistors 107 and 111. Such capacitors are large and costly. By contrast, according to the FLgure 12 embodiment, since capacitors 116a' and 116b' are coupled in parallel with base resistors of decreassd resistance .
; value, capacitors 116a and 116b may possess a smaller withstand chara~teristic and may therefore be smaller-sized and less . .
co0t1y as well. This allows for the provision of a power supply apparatus employing an inexpensive inverter.

ss~

Figure 13 illustrates load circuit 200 coupled to secondary winding 115 of the saturable transformer. The direct current voltage source 2 and inverter 100 may be of the type already described above. Therefore, only load circuit 200 will be described in detail below.
Each terminal of secondary winding 115 of the saturable transformer is coupled to one terminal of bat~ery 203 and direct current motor 205 through the base/collectox junction$ of planar transistors 207 and 208, the base and emitter electrodes of which are short-circuited in the forward direction.
The other ends of battery 203 and ~irect current motor 205 are connected to first and second switch contacts 211 and 213.
Central tap 115a of secondary winding 115 is connec~ed to third contact 212 and one end of secondary winding 115 is connected to a fourth switch contact 210 through a limiting resistor 209.
A switch short-circuiting piece 206' is switched to the solid line position when direct current motor 205 is ~o be utilized When switch 206' i5 turned on, the output of inverter 100 is full-wave rectified by the base/collector junctions of transistors 207 and 208 through central tap llSa and direct current motor 205 is energized with a half of the full~wave rectified output from inverter l00. Battery 203 is charged simultaneously. On the other hand, if direct current motor 205 is not to be utilized, switch short-circuitiny piece 206' is switched to the dotted line position and battery 203 i5 charged by the half-wave rectified output of the output voltage from secondary winding 115, Consideriny that the number of secondary winding turns of secondary winding 115 is extremely small, say about 10 turns, the circuit i9 configured such that hattery 203 is charged with a half of the output voltage of inverter 100. Accordingly, the .,.

: . . . ... . .
. . ..

value of limiting resistor 204 in Figure 3 may be small. But the fluctuation of the battery chargi~g current by virtue of any fluctuations in the A/C voltage source is relatively large, as shown by the line X in Figure 14 . By contras t, according to : the embodiment shown, since battery 203 is adapted to be charged - with the ~ull output voltage of invertex 100 appearing across secondary winding 115, the value of limiting resistor 209 can be selected to be larger, say several times larger, as compared with tha~ of limi~ing resistor 204 in Figure 3. As a result, fluctuation of the battery ~harging current with respect to the fluctuation of the commercial power source voltag~ is reduced as shown by line Y in Figure 14.
Generally speaking, a diode ls structured such that a pellet is connected between two external lead wires. Therefore, if the diode is damaged, there is a good chance that the two external lead wires will be closed rather than opened. Accor~
dingly, if direct current motor 205 or battery 203 are short-circuited, a large short-circuit current will flow through secondary windlng 115, damaging the components of inverter 100~
20 To prevent this possibility, planar type transistors 207 and 208 ;are employed in place of conventional diodes, wherein base/
collector junctions having a rectifying capability are utilized.
Appropriate planar type transistors are structured such that a bonding~wire or internal lead wire of 30 microns thickness is :~.
: connected between the external lead wire and the pellet.
Accordingly, if a planar type transistor is employed, the amplitude of the collector current lS restricted by the thinness of the wire. It follows that should direct current motor 205 or . : ~
: battery 203 become short~circuited, the bonding wire will melt 30 and the circuit will be opened to prevent damage to the components of inverter 100.

Although the bas~/emitter junction o~ a transistor can be considered as being similar to an ordinary diode, ~he voltage VEBO, i.eO, the voltage between the base and emitter electrodes when the base electrode is opened, is lower and the transistor cannot be utilized as a diode when the induced voltage acxoss secondary winding 115 is large. In order to eliminate this inconvenience, the embodiment shown employs the base/collector junctions of transistors 207 and 208 for the purpose of rectification.
When load circuit 200 comprises a battery to be char~ed, it becomes important to avoid overcharging the battery.
Figures 15 through 17 show two embodiments for that purpose.
In Fig. 15, a switching device 306 is connected between the base and emitter electrodes of transistor 110 of inverter 100.
The battery 203 is shunted by a series connection of a resistor 302 and a constant voltage device 303 and a detector 301 for detecting whether the battery charge level is approaching a predetermined value. Detector 301 is structured to detect a point where the voltage of battery 203 associated with the battery-charge level reaches a reference voltage determined by constant voltage element 303O Detector 301 may comprise an operation amplifier, for example. Detector 301 is provided at the output thereof with an active element 305 for closing the above-described switching device 306. In the embodiment shown, a photocoupler 304 is employed wherein switching device 306 :: ~ may;be a ~phototransistor while active device 305 may be a light-emitting diode. i In operation, when direct current voltage source 2 .
and inverter 100 are operative as described above in relation ~30 to Fig. 3, battery ~03 is charged ~y the outpu~ from the inverter.
~'' ~ ' , ~ - 21 - ~

' ;
The output of the inverter is full-wave rectified by means of diodes 201 and 202. When b~ttery 203 is charged and the charged voltage increases to reach the reference voltage of constant voltage element 303, then active device 305 is rendered operative in response to the output of detector 301, thereby to emit light. Switching device 306 is shunted by the light to become conductive~ Accordingly, the base and emitter electrodes of transistor 110 are short-circuited and the voltage induced across feedback winding 112 is not applied between the base and emitter electrodes of transistor 110 any longer. Transistor 110 is thereby rendered non-conductive. Since transistor 110 is rendered non-conductive, the charging circuit of dividing capacitor 104 is interrupted, the dividing capacitor 104 is no longer charged and transistor 106 becomes non-conductive~
As a result, the oscillating operation of inverter 100 is stopped and hence the charging operation of battery 203 i9 also stopped.
Figure 16 is a schematic diagram of a further preferred embodiment of the present invention designed to prevent the overcharging of a battery. In comparison with the embodiment of Figure 15, th~ embodiment of Figure 16 comprises the following modifications7 As illustrated in Figure 15, switching device 306 was interposed between the base and emitter electrodes of transistor 110 so that transistor 110 of inverter 100, could be interrupted in response to the charge level of battery 203. By contrast, according to the Figure 16 embodiment, two active elements 305a and 30S~, ~uch as light-emitting diodes, are coupled to the output of detector 301 In addition, bidirectional switching device~ 3~6a and 306b, which may be phototransistors, are sn/o~f controlled in response to active devices 305a and 305b. Thus, each o~ switching devices 306a and ~ - 22 -, , 306b constitutes a series connection together with capacitors 307a and 307b respectively. The series connection of capacitor 307a and switching device 306a and the series connection of capacitor 307b and switching device 306b are each coupled between one end of feedback windings 108 and 112 and the junction a.
In operation, when switch 103 is turned on and inverter 100 is brought to an operating state, battery 203 is charged by the output from inverter 100 as rectified by diodes 201 and 202. When battery 203 is charged and the battery voltage increases to reach the reference vol~age of constant voltage device 3Q3, active devices 305a and 305b, comprising light-emitting diodes respond to the output of det ctor 301 by emitting light and bidirectional switching devices 306a and 306b (phototransistors) are shunted by the light from light emitting `
diodes 3~05a and 305b become conductive. Accordingly, because capacitors 307a and 307b are coupled in parallel with capacitors 116a and 116b, the composite capacitances of capacitors 116a and 307a and the capacitors 116b and 307b become -larger. Capacitors 116a, 307a, 116b, and 307b are charged due to the counter-electromotive force flowing throu~h feedback windings 108 and 112 and the base/collector junctions of transistors 106 and 110 respectively, More specifically, and with re~spect to capacitors 116b and 307b, in the case of a transition from a state where transistor 106 is turned on and tranRistor 110 is turned off to a situation where transistor 106 -~ ~ is turned off and transistor 110 is turned on, the capacitors 116b and 307b are charged due to the counter electromotive ~orce flowing throu~h feedback winding 112 and the base/collector junction of transistor 110. Since the composite capacitance of capacitors 116b and 307b in such a si tuation i5 larger than . ' .

the capacitance of capacitor 116b only, the reverse bias period of transistor 110, which is about to be turned on, increases.
As the peak value of the full-wave rectified output from the output terminals of direct current voltage source 2 increases, the base/emit~er junc~ion of transistor 110 is forward biased.
Therefore, the conduction period of transistor 110 commences when the peak value of the full-wave rectified output becomes high and the output from diodes 201 and ~02 is confined to the period T shown in Figure 17, resulting in a supplemental charging state. Accordingly, it is possible to detect whether the battery is being charged at a predetermined rate. Further, a reverse bias period of the transistor is provided by increasing the capacitor capacitance of the impedance circuit of inverter 100. Thus, the on period of transistors 106 and 110 is shortened and the charging state is switched from rapid charging to supplemental charging~ '~he usable range of a charging apparatus employing a half-bridge-type inverter can therefore be expanded.
It will be appreciated that the combination of active device 305 (305a and 305b~ and the switching device 306 (306a and 306b) may differ ~rom that described above to comprise, for example, a combination of a relay coil and the contacts thereof.
Generally, a power supply apparatus employing a push-pull type inverter yields a large oscillatory output from the inverter which is of a rectangular wave form. This type of output is likely to cause noi~e from the inverter. Assuming that audio equipment is placed near a small electrical device including a charging circuit, a ~pace capacitance or coupling may form between a battery in the load of the inverter and the .

;2 audio equipment. The noise transmitted through such a space capacitance is liable to adversely affect the audio equipment.
Such noise comprises a common mode noise which may be defined as noise occurring between ground and the electrical equiprnent.
Such noise is different from normal mode noise which occurs between two lines in the equipment. Common mode noise is difficult to remedy irrespective o~ an increased magnitude and is more difficult to overcome than normal mo~e noise.
The embodiment of Figure 18 is intended to overcome or mitigate the problem of common mode noise. To that end, a noise-absorbing capacitor 308 is provided. One end of noise-absorbing capacitor 30~ is connected to casing 203a of battery 203 or the casing ~05a of direct current motor 205. The other end of noise-absorbing capacitor 308 is connected to the rectifica~ion output terminal of the full-wave rectifyi~g circuit 22 or the output terminal of direct current voltage source 2. Capacitor 308 serves to isolate in a direct current manner the primary secondary side of the saturable transformer whil~ coupling them in an alternate current manner. Accordingly, noise appearing on primary winding 105 is transferred through secondary winding 115 to casing 203a of battery 203 or to casing 205a of direct current motor 205. The noise from casing 2~3a or casing 205a is absorbed on the primary side of the transfoxmer by means of -~
said capacitor 308. Experimentation shows that without capacitor 308, noise charac~eristics shown by the dotted line in Figure 19 are ob~ained, whereas with capacitor 308, noise characteristics as shown by the solid line are obtained. Use of~capacitor 30d results, therefore, in a decrease of noise levels by several deci~els.
Although the foregoing descriptions of various ... ..... ..

~/ .

9~

embodiments centre mainly on the characteristic features o~
each respective embodiment, it will be appreciated that a power supply apparatus suitable for any particular use can be provided by a proper combination of the above-described features.
It is suggested that the present invention is particularly suited as a power supply for a portable electric shaver. In general, an electric shaver comprises a casing having an opening, a shaver cutter assembly provided so as to be exposed through the opening, and a prime mover such as an electric motor for driving the shaver cutter assembly. Typically, the shaver cutter assembly comprises a stationary cutter mounted in the shaver casing so as to be exposed through the opening and a movable cutter provided to be movable with respect to the stationary cutter. The prime mover may comprise a direct current motor coupled to a direct current voltage supply. According ~o one prior art approach, the casing is formed of a space for housing a dry cell or a rechargeable battery for providing a direct current voltage output~ Alternatively, a prior art electric shaver is structured to be adaptably connected to a separate AC adapter, which is structured to convext an alternate current voltage from a commercial power supply into a direct current voltage suited for driving a direct current motor. Since prior art AC adapters are bulky, it was impossible to include both a rechargeable battery and an AC adapter within the shaver casing without increasing the bulkiness of the shaver casing.
~b~iQu~ly~ it would be de~irable to include both a reahargeable battery and an AC adapter within the sha~er casing. The - present power supply is suf~iciently compact and lightweight to afford this advantage. More speci~ically, referring to Figure 3, ` 30 for example, bat~ery 203 can be used as a rechargea~le battery for , 5~

energizing the razor's direct current motor. The direct current motor 205 can be used to drive ~he movable cutters of the shaver cutter assembly. The present power supply, including in~erter circuit 100, bridge circuit 22, rechargeable battery 203 and direct current mo~or 205 can all be housed within a shaver ca~ing of ordinary size.

~ ` "' ' .

~ : ~ ......

.
, ' - : . , : , , - . : ,.. . .. ..

Claims (29)

The embodiments of an invention in which an exclusive property or privilege is claimed are defined as follows:

1. A power supply apparatus comprising an inverter for converting a direct current output from a direct current voltage source into an alternating current voltage, the inverter comprising:
first and second dividing capacitors connected in series, said capacitors being arranged to be connected in parallel with said direct current voltage source;
first and second transistors having their main current-carrying paths connected in series, the series-connected paths also being arranged to be connected in parallel with said direct current voltage source;
a saturable transformer having a primary winding connected between the junction of said dividing capacitors and the junction of said main current-carrying paths of said transistors;
biasing means for each of said transistors, each biasing means comprising a respective feedback winding magnetically coupled to the primary winding and connected in such a manner that the voltage induced thereacross by current flowing through the primary winding when the respective trans-istor is conductive is operable to maintain the transistor in its conductive state until saturation occurs, the arrangement being such that in use the transistors are alternating conductive and thus cause an alternating current to flow through the primary winding; and the biasing means of the transistors being arranged to cause substantially symmetrical operation of the transistors.

2. A power supply apparatus as claimed in claim 1, wherein each said transistor has emitter, base and collector electrodes, and wherein the emitter electrodes of the transis-tors are arranged for connection to respective sides of the direct current voltage source.

3. A power supply apparatus as claimed in claim 2, wherein each of said biasing means is arranged for connection to a respective side of said direct current voltage source, the inverter further including impedance means connecting said biasing means.

4. A power supply apparatus as claimed in claim 3, wherein each of said biasing means comprises a biasing resistor connected in series with each of said feedback windings, said impedance means being connected between the junction of the feedback winding and the biasing resistor of one of the biasing means, and the junction between the feedback winding and the biasinq resistor of the other biasing means.

5. The power supply apparatus of claim 4, wherein said impedance means comprise a reactance component.

6. The power supply apparatus of claim 5, wherein said reactance component comprises a capacitive reactance component.

7. The power supply apparatus of claim 6, wherein said impedance means comprises a resistor and a capacitor connected in parallel.

8. The power supply apparatus of claim 4, wherein said impedance means comprise third and fourth capacitors con-nected in series and wherein the junction of said third and fourth capacitors is connected to the junction of said first and second dividing capacitors.

9. The power supply apparatus of claim 8, further comprising first and second diodes, each being connected in parallel with the main current-carrying path of a respective one of the transistors in the reverse-biased direction.

10. The power supply apparatus of claim 8, wherein each of said first and second transistors is selected such that the saturation voltage developed between the main current-carrying electrodes when each of said transistors is operating as a backward transistor is relatively small.

lI. rrhe powe.r supply apparatus of claim 10, wherein eacK of said biasing means includes a time constant circuit operable to compensate for differences in the current amplii-cation factors of the transistors which would otherwise cause differences in the conduction periods of the transistors.

12. The power supply apparatus of claim 11, including a pair of capacitors each connected in parallel with a respective one of said biasing resistors.

13. The power supply apparatus of claim 12, wherein said saturable transformer includes a saturable magnetic core having a secondary winding wound thereon, the apparatus further comprising means for rectifying the output voltage induced across said secondary winding for providing a direct current output.

14. A power supply apparatus in accordance with claim 13, wherein said rectifying means comprise a diode.

15. A power supply apparatus in accordance with claim 13, wherein said rectifying means comprise a collector/base junction of a planar transistor.

16. A power supply apparatus in accordance with claim 15 further comprising a storage battery chargeable by the output from said rectifying means.

17. A power supply apparatus in accordance With claim 16 further comprising:
means for detecting the charged state of said storage battery; and switching means responsive to the output from said detecting means for holding one of said first and second transistors in a non-conductive state.

18. A power supply apparatus in accordance with claim 16 further comprising:
means for detecting the charged state of said storage battery; and means responsive to the output from said detecting means for increasing the capacitance of said impedance means.

19. A power supply apparatus in accordance with claim 18, wherein said storage battery has a casing, the apparatus further comprising a noise absorbing capacitor interposed between said direct current voltage source and said casing of said storage battery.

20. A power supply apparatus in accordance with claim 19 including a direct current motor connected to the output from said rectifying means

21. A power supply apparatus in accordance with claim 20, wherein said direct current motor has a conductive casing, the apparatus further comprising a noise absorbing capacitor interposed between said direct current voltage source and said casing of said direct current motor.

22. An electric shaver including a power supply apparatus as claimed in claim 21.

23. An electric shaver, comprising.
a casing having an opening;
a shaver cutter assembly provided in said shaver cas-ing so as to be exposed through said opening, said shaver cutter assembly comprising a stationary cutter and a movable cutter;
a power supply apparatus provided within said shaver casing, said power supply apparatus comprising an inverter for converting a direct current output from a direct current voltage source into an alternating current voltage, the inverter comprising:
first and second dividing capacitors connected in series, said capacitors being arranged for connection in parallel with said direct current voltage source;
first and second transistors having their main current-carrying paths connected in series, the series-connected paths also being arranged for connection in parallel with said direct current voltage source;
a saturable transformer having a primary winding connected between the junction of said dividing capacitors and the junction of said main current-carrying paths of said transistors; and biasing means for each of said transistors, each biasing means comprising a respective feedback winding magnetically coupled to the primary winding and connected in such a manner that the voltage induced thereacross by current flowing through the primary winding when the respective trans-istor is conductive is operable to maintain the transistor in its conductive state until saturation occurs, the arrangement being such that in use the transistors are alternatively con-ductive and thus cause an alternating current to flow through the primary winding;
the biasing means of the transistors being arranged to cause substantially symmetrical operation of the transistors;

and said saturable transformer further including a secondary winding and the shaver further comprising motor means housed in said shaver casing and coupled to said secondary winding of said saturable transformer for driving said movable cutter of said shaver cutter assembly.

24. An electric shaver, comprising:
a shaver casing having an opening, a shaver cutter assembly provided in said shaver casing so as to be exposed through said opening, said shaver cutter assembly comprising a stationary cutter and a movable cutter;
a power supply apparatus provided in said shaver casing and comprising an inverter for converting a direct current output from a direct current voltage source into an alternating current voltage, said inverter comprising:
a pair of first and second dividing capacitors connected in series, said capacitors being arranged for connection in parallel with said direct current voltage source;
a pair of first and second transistors, each including emitter, base and collector electrodes, the main current-carrying paths of the transistors being connected in series and said series-connected paths being arranged for connection in parallel with said direct current voltage source;
a pair of first and second base biasing resistors;
a saturable transformer including a saturable magnetic core, a primary winding wound on said saturable magnetic core, a pair of first and second feedback windings wound on said saturable magnetic core and magnetically coupled to said primary winding, and a secondary winding wound on said saturable mag-netic core and magnetically coupled to said primary winding;
said primary winding being connected between the junction of said first and second dividing capacitors and the junction of said main current-carrying paths of said transistors;
each of said transistors having base biasing means comprising a respective one of said feedback windings connected via a respective one of said biasing resistors between the base and emitter electrodes of the transistor, the feedback winding being connected in such a manner that the voltage induced there-across by current flowing through the primary winding when the respective transistor is conductive is operable to maintain the transistor in its conductive state until saturation occurs, and the two biasing means of the transistors being arranged to cause substantially symmetrical operation of said transistors;
and impedance means connected between the junction of one of said feedback windings and one of said base biasing resistors and the junction between the other said feedback winding and the other said base biasing resistor; said electric shaver further comprising:
a rectifying circuit coupled to said secondary winding of said saturable transformer for rectifying the output from said secondary winding; and a direct current motor provided in said shaver casing and coupled to said rectifying circuit for driving said movable cutter of said shaver cutter assembly.

25. An electric shaver as claimed in claim 24, further including a rechargeable battery in said shaver casing, the battery being chargeable by said power supply apparatus.

26. An electric shaver as claimed in claim 25, further comprising, in said shaver casing, a direct current voltage source responsive to an alternating current input voltage for providing said direct current voltage.

27. A power supply apparatus as claimed in claim 1, wherein said first and second transistors are structured to be in a complementary manner.

28. An electric shaver as claimed in claim 3, wherein said first and second transistors are combined in complementary manner.

29. An electric shaver as claimed in claim 4, wherein said pair of first and second transistors are combined in complementary manner.