CA1102411A - Admittance measuring system for monitoring the condition of materials - Google Patents

Abstract

ADMITTANCE MEASURING SYSTEM FOR
MONITORING THE CONDITION OF MATERIALS

Abstract of the Disclosure A two-wire transmitter includes an admittance sensing probe adapted to sense the conditions and corresponding admit-tance of materials. The probe is coupled into an admittance responsive network which generates an admittance signal repre-senting the condition of materials. The output current from the transmitter is varied in response to the admittance signal.
In the embodiment of the invention, the admittance responsive network comprises a variable frequency oscillator whose frequency varies with the admittance of the materials. In another embodiment, the admittance responsive network comprises a ramp generator with a frequency which varies with the admittance of the materials. In another embodiment, the admittance responsive network comprises a bridge whose balance changes in response to the admittance of the materials.

Description

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Background of the Invention This invention relates to Rf admittance measuring systems for moni-toring the condition of materials, and more particularly~ to systems of this type which are adapted for use at remote locations.
~ lere~ofore, two-wire transmitters have been utilized to monitor various conditions at a remote location. Typically, a two-wire transmitter at a remote location is connected in series with a power supply and a load at an-other location through two transmission wires. As the condition being monitored at ~he transmitter varies, the effective series resistance across the transmitter varies so as to produce a change in the current drawn by the transmitter which represents ~e.g., is generally proportional to) the condition being monitored.
A two-wire transmitter of th$s type is designed for low power consumption since the amount of power available to the transmitter from the remotely located power supply may be limited. Furthermore, certain applications may require that the two-wire transmitter be "intrinsically safe" so as to permit its use in the monitoring of conditions in an explosive environment. Under these circumstances, low energy usually associated with low power consumption becomes important so as to preclude the possibility of ignition and explosion.
Although the state of the art in two-wire transmitters is ade-quate for monitoring various types of conditions, the M~56g-1 prior art technology with respect to the RF admittance measure-ment is deficient for two-wire transmitters for the following reasons.
When measuring the ~F admittance between a probe electrode ana a reference surface such as a grounde~ vessel, the resistance is parallel with the capacitance between the probe electrode and the grounded vessel becomes very important : from a power cor~sumption standpoint~ Heretofore, it has : generall~ been assumed that shunt resistance is sufficiently :~ lO small in a sufficiently large number o applications so as to .; ~ render the power provided by ths 4 milliamp c~rrent;:in a ~-20 milliamp two-wire transmitter system insufficient to power the two-wire transmitter. In other words, the shunt resistance alone might consume more power than is available at the 4 milli-. 15 amp condition leaving little or no power to operate the circuitry of the transmitter. Also, power limitations exist where the admittance measuring circuit is battery~powered.
. Moreover, in order for an admittance measurement - ?~ to be accurate, reliable detecti.on must be utilized. However, such reliability usually requires a sub-:.~ stantial source of power which is inconsistent with the low power requirements of a two-wire transmitter as discussed above and the available power because of the shunt resistance.
This combination of factors imposes severe restrictions on the power which is generally considered necessary to provide a reliable RF signal source~ Similar ~ restrictiGns are placed on the power generally considered : necessary to assure that the ~detect~r operates with a .high degree of reliability.

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Another problem which is somewhat unique to admit-tance measurements is the isolation of the admittance respon-si~e network in which the unknown admittance ~eing measured ~` is connected. Typically, the unknown admittance being measured is from a probe electrode to ground as disclosed in Maltby et al United Stated patent 3,781,672, issued Decsmber 27, 1977 and Maltby Uni~ed States patent 3,706,980, issued December 19, 1972 both of which are assigned to the assignee of this invention.
However, a power supply at a location remote frGm the admittance 10 responsive network as in the case of a two-wire transmitter, may - not be connected to ground in a manner compatible wi~h the network.
It is therefore necessary to isolate the admittance responsive network, or at least the admittance sensing probe, from the power supply so as to p-ermit the network to be connected ~o ground regardless of the power supply circuit. This is also true of the admittance responsive networks employing a variable frequency oscillator such as that disclosed in Spaw patent 3~807,231. More-over, if the voltage across the unknown admittance were reduced to minimize power comsumption, the signal representing the changes 20 in admittance of the admittance responsive network would require - amplification. Accordingly, the problem exists of providing an isolated source of power~`for such amplification.
Other problems exist in assuring linear and stable calibration of the admittance measuring system.~ It is also importan* ~ provide a system which will work with various types of probes and various lengths of cables associated with the probes without advers~ affecting the admi~tance measurement.

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Another problem which can be quite t~oublesome is the low level of analog signals which may be generated by an admittance measuring system.
Low level analog signals are particularly difficult to process if a high degree of accuracy is to be attained.
To a very large degree, the above-mentioned problems are encounter-ed when the system Eor monitoring the condition of materials comprises a battery-operatecl unit as well as a two-wire transmitter. Under these cir-cumstances, the available power is again limited.
Summary of the Invèntion ,;
; 10 It is an overall object of this invention to monitor the condition of materials a~ a remote location utilizing admittance measurements.
It is more specific object of this invention to minimize the power consumption necessary in making the admittance measurements.
It is also a more specific object of the invention to provide an intrinsically safe system for making the admittance measurements.

Arcording to a broad aspect of the present invention, there is provided in a two-wire transmi~*er system comprising a power supply and a load at one location and a two-wire transmitter at another location inter-connected by a pair of transmission lines carrying a variable signaling current, the impro~ement comprising: an aclmittance sensing probe including a probe electrode adapted to sense the condition and corresponding admittance of materials; an admittance responsi.ve network coupled to said probe repre-senting the condition of materials; and output means coupled to said admitt-ance responsive network for varying the signaling current in response to the condition of materials; wherein said admi~tance responsive network comprises:
first admittance means coupled to said sensing probe so as to include the admittance of said materials; second admittance means comprising a re~erence admittance; charge current means coupled to said ~irst admittance means and said second admittance means for charging thereof; discharge means coupled to said first admittance means and said second admittance means for discharg-ing thereof; and charge rate detection means for detecting the difference in charging rates between said first admittance means and said second admittance means.

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; The first admittance means and the second admittance means may comprise an admittance bridge where the first side of the bridge comprises the first admittance means and the second side of the bridge comprises the second admittance means and the charge rate detection means detects the difference in time to charge the first si.de as compared with the second side.
: The output means may comprise modulator means coupled to the charge rate detection means for generating a signal representing the differ-.. ..
ence in charge rates. A demodulator means is AC coupled to the modulator means and isolated therefrom for demodulating the modulated signa.l and apply-ing the demodulated signal to an output ampli~`i.er means which is coupled .

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to the pair of transmission lines so as to control the cùrrent drawn by the two-wire transmitter.
` In accordance with another important aspect of this embodiment, DC rather than RF operating controls are utilized to avoid the effects of stxay coupling. More specifically, the operating controls are located in the current source and adjust the DC flow of current to said admittance bridge.
Although specific objects~of the invention have been identified, other objects will be apparent from the drawings taken in conjunction with the specification.

Brief Description of the Drawings .
Fig. 1 is a block diagr~m of a two-wire transmitter embodying the invention;
Fig~ 2 is a schematic circuit diagram of an RF
signal generator embodying one important aspect of the invention;
Figs. 2(a-c) are waveform diagrams utilized in describing the operation of the circuit of Fig. 2;
Fig. 3 is a schematic circuit diagram of a chopper drive circuit embodying another important aspect of the invention;
Fig. 4 is a schematic circuit diagram of an output amplifier embbdying another impor~ant aspect of the invention;
Fig. S is a schematic representation of the bridge network including a mechanical representation of the probe;
- Fig. 6 is an equivalent circuit of the bridge net-work of Fig. 5;

Figs. 7(a-c~ are schematic representations of various probes immersed in various materials;

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' ; Figs. 8(a-c) are equivalent circuits of the admit-tance measured by the pxobes of Figs. 7~a-c~ respectively;
` . Fig. 9 is an equivalent circuit of khe admittance : of Figs. 8(a-c); . `~
Fig. 10 is a schematic diagram of a battery-powexed output amplifier;
Fig. 11 is a b~ock diagram of another two-wire transmitter representing an embodiment of the invention;
FigO 12 is a schematic diagram of circuitry shown in block form in Fig. 11;
- Figs. 12(a & b) are schematic diagrams o~ circuitry shown in block form in Fig. 11 where the diagram has been split along line x-xi . Fig4 1.3 is a block diagram of another two-wire ~ 15 transmitter representing another embodiment of the invention;
.'~ and Figs~ 14(a d) are schema~ic dia~rams of circuitry ; ~ shown in block form in Fig~ ~3 where the diagram has been sp it along 1 nes y-, and z-z.

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Detailed Description of a Particular ~mbodimcnk .
As s~lown in Fig. 1, a two-wire transmitter 10 i5 connected in series with a power supply 12 and a load repre-sented by a resistor 1~ through transmission wires 16 and 18connected to the terminals 20 and 22 of the two-wire trans-mi'~ter lO. In accordance with this invention, the trans~
mitter 10 is adapted to measure a~d draw a signal current representing an unknown measured admittance 24 which may represent the conaition of materials sensed by the probe.
; : The measured admittance 24 which represents the capacitance 24c and the resistance 24r ~rom a probe electrode to ground orms one arm of a briage ne~work 26 also comprising a capa-citor 28 and windings 30 and 32 of a secondary 3~ in a trans-former 36. The bridc3e network 26 is driven by an oscillator 38 havincg an output connected to the primary ~0 of the trans-former 36.
In accordance with this inYention~ the voltage across he admittance 24 is limited to a level so as to assure ade-quate power for the two-wire transmitter in view of the power consumption by the unknown resistance 24r. As will now be described in detail, the voltage is limited to less than ~
where V is the voltage across the two-wire transmitter and the current drawn by the two-wlre transmitter varies from 4-20 milliamps.
Hereto~ore, it has been assumed that the un~nown . resistance 24r o the unkno~n admittance 24 being measured may vary over. a wide r.ancJe. 0~ course, ~or a ~ixed voltage, i~ the resistance 2~r should become very small, a c300d deal , ~, .; . ~_ M- S G 9 ~ 4~
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of power would be consumed in that resistance. In a conven-tional two~wire txansmitter, the sole source of power is the current flow through the transmission wires 16 and 18 which is conventiona~l~ at levels of 4-20 milliampsO If it is assumed that the power supply produces an output voltage o 24 volts, the voltage across the terminals 20 and 22 o~ ~he two~w.ire transmitter may, for ex~nple, be 12 volts where the total voltage drop across the load 14 plus the drop across - each of the wires 16 and 18 is 12 volts. This means that when the two-wire transmitter is dxawing 4 milliamp~, the total power available to operate the two-wire transmitter is P = VI = 48 milliwatts. This would mean that extremely - small shunt resistances 24r would require extremely small ... . .
- ~oltages across the unknown admittance 24 to permit the two-` lS wire transmitter to operate from the available power at the ; 4 milliamp level.

I~ has however been discovered, as will be aescribed subsequently, that the resistance 24r~ in almost all appli-` ,f cations regardless of -~he t~pe of pxobe utilized, will not ~all below 500 ohms. Thus, by only moderately limiting the ~oltage across the unknown admittance 2~ and thus tlle voltage across the unknown xesistance 24r, suficient power is avail-able to the two--wire transmitter even at the 4 milli~np current level. Having once recognized that the magnitude of the resistance 24r will not, in almos~ all applications, all below 500 ohms, the magnitude of the voltage across the .resistance 2~r may be readily computed or a 4-20 milliamp ~7o wire transmitter rom the ollowiny equation:

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where ~ . ~ = -the voltage across the transmitter;
:~ v = the rms voltage across resistance 24r;
:~ 5I - ~he minimum current flow through the - m two-wire transmitter 10; and r24 ~ the.resistance in ohms o the resistance 24.
Fo~ Imequal 4 milliamps and r2~ equ~l 500 ohms, then v c ~ ~ (2) . ? 10 I~ V equals 12 volts, then v is less than ~ or less than approxi-.. ... .. _ . .. . . ............. . .
mately 5 volts rms. OP course, the two wire transmitter itself re~uires some power to operate. Therefore, in the preferred.em-bodiment where I~ - 4 milliamps an~. v = ~2 volts, v = approxi-~ . mately 2.2 volts.rms, or substantially less than : : 15 - }n ~urt~er accoxdance with this invention, th~
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oscillator 38 of the cIass C type, i.e., the co]lector ouxrent:
; of each of tne two transistors in the oscillator 38 which riv~ the tank circuit flows through an angle less than 180 of the 360 cycle of the RF sinusoidal signal applied to the -~. 20 bridge network 26. ~owever, class C operation ma~ produce distortion in the intended sinusoidal signal. Therefore, in further accordance with this invention, the oscillator 3~
: ~ comprises a resonant circuit in the form o~ a tank circuit in~lu~ing the trans~ormer 36 as well as the measured admittance 24 as will subse~uently bc descxibed in detail with reference to Fig. 2~ Since the admittance 24 is part of the resonant circuitr l.ittle additional curren~ is required to drive addi-: ~ional admitta)~ce between the probc and ground.

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As also sllown in Fig. 1, an AC error signal repr~-senting th~ unbalance of the bridge network 26 and thus the unknown measured admittance 24 is applied to an error ampli-fier 42~ ~he error amplifier 42 permits the use of relatively 5 low ~C voltages in the bridge network 26 in accordan~ ~7ith ~hi~ invention. The output from the error amplifier 42 is . then applied to a phase sensitive detector comprising a ~hopper.44 which is triggered by a chopper drive 46.
`. In~accordallce with another important aspect o the 10 invention, the brid~e netwoxk 26 and the error amplifier 42 are isola~ed from the power supply b~ the ~irst trans~oxmer ~ and the second transformer 48 which couples the output of the error amplifier 42 ko the input of the chopper 44. In ~er words, the power supply is allowed~to float with respect to $he probe. This permits the use of a probe for measurlng the admittance 24.between the probe electrode and ground with-.
out being concerned with the:manner in whlch the power supply12 is connected to ground. Note that this power supply 12 is at a remote location with respect to ~he tw~-wire transmitter ~ ~ 20 10 and the manner in which the power supply 12 is connected to ground may not be xeadily discernible at the two wixe - .transmitter ~0. The isolation provided by the txansformers 36 and 48 also allows either terminal 20 or 22 of the two-wire txansmitter 10 to be maintained at a vcxy substantial ~C or DC voltage with respect ko ground without any high voltage brea~down.

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, i M-SG~-l In ordcr to provide isolation for the bridge net-work 26 while still providing a DC power supply for the error amplifier ~2 which is directly coupled to the bridge network 24, diodes 50 and 52 are provided to rectify the RF
sinusoidal signal from the secondary 34 of the transfo.rmer

- 3~. Diodes 50 and 52 are then connecte~ to a terminal 54 of ~he amplifiex 42 so as to provide a DC power supply there-for which is isolated from the power supply 12.
In contrast, the DC power supply voltages for the RF osci~lator 38, the chopper drive 46, the chopper 44 and . an output amplifier 56 are provided by a voltage regulator J
58 with a positive power supply terminal -~V1O In addition, a negative power supp~ voltage is provided by a voltage : regulating circuit in the`RF oscillator 38 at a terminal -V2.
~ 15 The chopper arive 46, the chopper 44 and the output amplifier ;~ ~6 are also connected to the circuit common terminal C of the voltage regulator 58.
In order to permit the bridge to be zeroed~with a capa~itance 24c from probe to gr~und which is different from ~, ~ ,, .
- ' 20 the zerGing capacitance 28, the number of windings 30 differs ~rom the number of windings 32. For example, the number o ~ windings 30 may be three times as large as the number of windings 32 so as to allow the bridge to be zeroed when the measured capacitance 24c from probe to ground is three times - 25 as great as ~he zeroing capacitance 28. In addition, ~he . bridge networ]~ 26 includes a variable span capacitor 60. By : adjusting the span capacitor 60, the measured capacitance 2~c necessary to produce a predetermi.ncd current through the .

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transmission wires 16 and 18 may be varied. In addition, the output amplifier 56 may be provided with a gain adjust-ment which provides fine span con~rol~
. In order to provide spark protection for the trans-mittex lO, a pair of series connected, reversed poled Zener diodes 62 and 64 are connected between one terminal of the ~ .. .
: span capacitor 60 and ~round. A neon bulb 66 is connected be~ween the other terminal of th:~ span capacitor 60 and .. gxound~ The protection afforded by the diodes 62 and 64 and the bul~ 66 allow the transmitter 10 to wit~stand spikes of ~everal thousand volts across`the a~mittance 24 with no component failure or unbalancin~ of the ~ridge net~ork 26.
As also shown in Fig. 1, a ~ap on the primary 68 of the transformer 48 is connected to the input of -the error : . 15 amplifier 42. This connection provides feedback to the ampli-- fier 42 so as to control the gain thereof. O~ course, changing the location of the tap 68 will chailge the gain of i ~ the ampli~ier 42 and thus the magnitude o~ the output ~ .
applied to the chopper 44.
` ~ 20 As the output from the chopper 44 varies and is compared ~-ith the voltage across a resistor 57 connected to the wire 22, the signal current output from the amplifier 56 is txansmitted through the wires l~ and 1~. The current having a magnitude which represents the admittance 24 and the condition o~ the materials being measured is utili~ed to drive the load l~.
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In accordance with one aspect of the invention,the .input of t:h~ two wire transmitt~r 10 comprises a fullwave rectifying bridge network comprising diode pairs 70 and 72 which conduct the 4-20 milliamp current when the ter~inal 20 is positi~e with re~pect to the - terminal 22. Similarlyj the p~i~ o~ diodes 74 and 76 c~duct when the terminal 22 is positive with respect to the terminal 20 or 22 to be connected to either ~` transmission wire without damaging or af~ectiny the operation of the txansmittex.
The class C RF oscillat~r will ~ow ~e described in detail with reference to Fig. 2. The oscillator comprises a multivibrator such as a pulsed amplifier incIuding a pair or transistors 100 and 102 which are alternately conductive so as to drive a resonant tank circuit comprislng the transformer 36 and a capacitor 104 which is connected in parallel with the Frimary 40 or the transformer 36 as well as the measured admittance A in the bridge network 26. The base drive for the transistor 100 of the mul~ivibrator is provided by the capacitor 106 and resistors 108 and 110 where the resistor 110 is connected to a transistor 112 in a base current regulating circuit. Similarly, a capacitor 114 and resistors 116 and 118 provide a base drive for the transistor 102, The base current of the transistors 100 and 102 charge the capacitors 106 and 114 to a positive voltage higher than the supply vol~age thereby cutting off the transistors 100 and 102 during most of the cycle so as to achieve class C operation. Diodes 120 and 122 which are connected in the .

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M-5~9 ' base circuits of the transistors 100 and 102 respectively provide protection ~or the bases of the transistors by - ~lo~king current flow when the junction of the resistors ` ` 108 and 110 and the junction of the resistors 116 and 118 are driven positive~
~s mentioned previously, the transistor 112 is part of a regulating circuit. The regulation afforded by the transistor 112 maintains the amplitude of the RF sinu-; soidal signals substantially constant despite any change 1~ in the opera~ing characteristics o transistors within the oscillator and despite resistive loading due to the resist-- ` ance 24r. In this connection, the base of the transistor 11~ is connected to a tap in the voltage divider comprising resistors 124 and 126 with one terminal of the voltage i 15 divider connected to the ~Vl power supply terminal of the ; ~Dltage regulator and the other terminal of the voltage divider connected to a capacitor 128 which is connected to cixcuit sommon through a discharge resistor 130 which may be potted with the capacitor 128 to provide intrinsic safety.
- 20 The capacitor 128 is charged to a negative poten-~ial with respect to circuit cGmmon by full wave rectifying diodes 127 and 129 connected across the tank circuit such that the tap o the voltage divider connected to the base ~f the transistor 112 is maintained at an operating point of approximately ~ero volts which is just enough to render the collector-emitter circuit of ~he transistor 112 conduative.
The emitter o~ the transistor 112 is maintained slightly negative by a resistor 132 and a diod213~. Diode 13~
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compensates for the hase emitter voltage o~ the transistor 11~ and partlally compensates for changes in the base emitter volta~e of the transistor 112 with temperature so as to assure stable calibration. As clearly shown in Fig. 2, the negative voltage of the capacitor 12S is utilized to provide a negative power supply voltage -V2 for the chopper 44 and the output amplifier 5~ as shown in Fi~. 1.
The regulating circuit as previously descxibed including the transi~tor 112 operates in the ~ollowing manner to malntain the amplitude of the RF sinusoi~al signal at the transformer 36 substantially constant. The voltage across the transformer 36 whîch is the voltage across the tank circuit of the oscillator is, in efect, detected by the diodes 127 and 129 which charge the capàcitor 128. The resul~ing negative DC voltage on the capacitor is then compared to the voltage of the regulator 48 at the resistive voltaye divider comprising the resistors 124 and 126 so as -: to maintain the intermediate tap at approximately circuit common. As the characteristics o~ the transistors change wikh tem~erature and the probe is resistively loaded as respresented by the resistance 24r, the transistor 112 leaks bias of~ the capacitors 106 and 114 so as to maintain the - amplitude of the oscillator and the corxesponding voltage across the capacitor at tlle same potential.
2S In order to eliminate any distortion in the RF
sinusoidal signal, a relatively large choke inductor 136 provides a high impedallce load ~o tlle t.ank circuit thereb~
avoiding an~ sharp currenl: pul5e which migllt distort the RF
sinusoidal wavcform. ~n inductor 1l0 and a capacitor 1~2 providcs a power supply ilter netwol~.
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- The class C mode of operation for the oscillator 38 will now be des-cribed with reference to the waveforms of Figures 2 ~a-c~. As shGwn in Figure 2a, the output voltage from the collector to circuit common which is applied - across the primary 40 of the transformer 36 is substantially sinusoidal due to the resonant action of the primary 40 with the capacitor 104 and the image of the bridge capacitors 24C and 28 (shown on Figure 6) reflected through trans-` former 40. However, the diode 120 is biased off by the voltage on capacitor 106 for most of the cycle, producing a voltage pulse as shown in Figure 2C at the anode of diode 120. Thus, the collector current whn~ch flows through the - 10 transistor 100 is intermittent as shown in Figure 2b. In fact, only a brie~
surge of collector current flows as shown in Figure 2b during the 360 degree cycle depicted in Figure 2a. (In actuality, some current continues to flow during the remainder of the cycle but this current is small relative to the -~ surge of current flow and has not therefore been depicted in the drawing). As shown in Figure 2b, the su~stantial or surge of collector current flows for sub-stantially less than 90 degrees of the 360 degree cycle which is of course sub-stantially less th~n 180 degrees flow of current which still falls within the realm of class C operation. Note that the surge of current corresponds in time with the peak voltages for Figur~52a and ZC to assure that the maximum power is derived from the curren~ flow.
As shown in Figures 1 and 2, the tank circuit is connected to the chopper drive 46 through a switch 144 which is capable of connecting the chopperdrive to either terminal of the primary 40. By moving the switch from one position to the other, the phase of the chopper drive is reversed 180 degrees and the phase sensitive detection performed by the chopper 44 is changed by 180 degrees to permit the transmitter t~

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to operate in a high level or low level failsafe mode. As . will now be descxibed in detail with reference to Fig. 3, .~ the chopper drive 46 generates a square wave trigger signal for the chopper 44 while minimizing power consumption and optimizing stable, accurate calibration consistent with . . this invention~
To achieve these objectives, chopper drive 46 as shown in Fig. 3 comprises a first pair of ~ield effect transistors 200 and 202 having gate eIectrodes connected to the tank circui~ through a capacitor 204~ The first - channel (drain) electrodes of the transistors 200 and 202 .
- are interconnected and the second channel (source) electrodes are connected between circuit comm~n and the regulated supply voltage ~Vl. In accordance with the objectives of this invention, the second channel electrodes are connectea to the power supply voltage ~Vl and circuit common through resistors 206 and 208.
: ~he sinusoidal output from the oscillator 38 as shown in ~ig. 1 is applied to a capacitive divider net~ork 3 ~ including the capacitor 204 and capacitors 228 and 230 connected between the capacitor 204 and circuit common.
The capacitîvely divided sinusoidal signal across the capa-citors 228 and 230 is then applied to the gate electrodes :- of the transistors 200 and 20~ to alternately gate the tran-.
sistors bet~een the conductive s~ates~

I~ ~ill be understood that the resistors 206 and 208 play a particularly important role .in assuring low powcr . consumption and accuracy in the phase dctection at the : c)lopper 44. In this connection, it will be understood that ,::
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M-569-~_ the resistors 2~6 and 208 serve to limit the voltage across the channel electrodes o~ each of the transis~ors ~00 and ~ 202 which in ~urn sharpens the knee of the input volta~e-- output voltage transfer charactel-istics of the field efect transistors. As shown in curve a of Fig. 3a, large output voltages from channel-electrode-to-channel-electrode of a field effect transistor give a rounded knee to the output voltage-input voltage ~ransfer characteristic while limiting the output voltage as shown in cur~e b sharpens the knee of ~he output voltage-input voltage characteristic~ This tends to produce a more nearly sguare wave signal which is of the utmost importance in achieving reliability in the phase ~etection at the chopper 44.
Moreover, as shown in Fig. 3b, limiting the output voltage of channel electrode to channel electrode of the iield effect transistor tends to immunize the fiela effect transistor to changes in the output ~oltage-input voltage i txan~fer characteristic with temperature. As shown in wave-.`;~3 forms c and d of Fig~ 3b where curve c represents the output-input voltage characteristic at a temperature of -55C. and ~u~ve d represents the output-input voltage characteristic - at a temperature of ~25C. Thus, a large channel electrode~
to-channel-electrode voltage makes for a very substantial : difference in curves c and a which.affect the stability of ~5 the cal.ibrations for the system~ On the other hand, limiting e outp-it voltage as shown in curves e and f renders the -55~C. cur~re e s~bstantlally identlcal to the ~25C. curve ~' ' ..

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;- In addition, th~ channel resistors tend to limit ~urrent fl~ through the transi~tors 200 and 202 when tl~e transistors 200 and 202 ar~ simultaneously conductive be-tween the first and second channel electrodes. This assures that the power consumption by the transistors 200 and 202 will not be e~cessive as in the case where both of the ~- transistors 200 and 202 conduct simultaneously.
The output from the intexconnected firstchannel electrodes is a s~uare wave voltage riding above circuit ~ommon. In order to assure that the waveform is square, a fee~back resistor Z10 is provided between the first channel electrodes and the gate electrode so as to raise the gate electrode to the average DC voltage at the first channel electrodes. The resistor 210 as.sures a duty ~actor of 50~
lS ~hereby compensating for small differences in the threshold voltages of the field effect trans.istors. Capacitors 212 an~ 214 provide a low impedance to drive the gate capaci-tance of the succeeding stage with the square wave signal generated by the field effec-t transistors 200 and 202.
'.~ 20 ~hus, the irst state of the chopper drive generates . a voltage waveform which is square. However, the s~uare . . .
. voltage waveform is of insufficient peak-to-peak voltage ;~ . to drive the chopper because of the voltage drop across the channel resistors 206 and 208.
Therefore, the succeeding or second stage of the chopper drivel s~hich is ~C coupled to the preceding stage through capacitors 217 and ~19, comprises another or second pair of fiGld e~fect transi.stors 216 and 218 which are biased . ~

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near their respective threshold voltages by resistoxs 220, 222 and 224 which are connccted to the gate electrodes thereof. By biasing the transistors 216 and 218 near their thxeshold voltages the transistors turn on vexy near the zero 5 - ~rossing of the square wave signal gPnerated by the transis-t~rs 200 and 202. As a result, the duty factor of each of the transistors 216 and 21~ more closely approaches 50~
thereby eliminating any phase uncertainty so as to assure reliable phas~e detection at the chopper 44. Since the ~ransistors 216 and ~18 do not conduct simultaneously except for the instant of transition, there is little or no power wasted by the second stage.
Note that the transistors 216 and 218 are connectea directly across the power supply voltage ~Vl and circuit common so that the output to the chopper 44 is alternately switched between ~Vl and circuit commonn This produces a 1~J output impedance in the chopper drive to assure fast rise and fall times of the resulting sguare wave output signal without the necessity of dissipating large amounts of power ;~ 20 in the chopper drive. Accordingl~, the s~uare wave output signal generated by the field effect transistors 216 and 218 connected between the supply voltage Vl and circuit common ver~ closely approa~hes a perfect square wave so as to ~ssure phase stability in the phase sensitive detection without sacrificing efficiency of the chopper drive.
Where a probe is utilized to measure the level of liquids and the liquids tend to coat the probe, it is desirable to provide means by wllicll the phasing of the ~3 chopper drive square wave signal may be altered by a 45 lead.
In this connection, it will be understood that long coatings on a probe as described in the aforesaid patent 3,706,980 appear as an infinite transmission line and the conductive and - susceptive components of the coating are equal so to produc~ a 45 lag. By detecting at a 45 phase angle, the conductive component and the susceptive component will cancel leaving only the suscep-tance due to the change in capacitance of the liquid level being measured and nonsusceptance due to the coating intself.
In this connection, capacitor 226 and series resistor 234 or the capacitor 228 may be optionally connected in parallel with a capacitor 23~.
In accordance with another irnportant aspect of the invention, the output amplifier 56 comprises a voltage feed-back network connected to a resistor 57 as shown in Figure 1 through which the 4-20 milliamp DC current drawn by the two-wire transui~ter flows so as to stabilize the flow of the 4-20 milliamp DC current at all current levels. As shown in Fi$ure 4, the output amplifier 56 is divided into the following sections:
2Q a voltage feedback divider network 300, a first differential amplifier stage 302, a second differential stage 304, a volt-age to currenttgain stage 306 and an output amplifier stage 308 which is shown as including the resistor 57 connected between c~rcuit common and the terminal 22 in Figure 1.
The voltage feedback divider network 300 includes an independent point adjustment potentiometer 310 connected in series with resistors 312 and 314. A tap 316 on the ,. ,.2~/
.

.
- : :
' .

M-56~-~L

potentiometer 310 is set so that when the bridge network 26 shown in Fig. l is ~t balance, the current drawn by the two-wire transmitter is 4 milliamps when no current is flowing through the gain adjustment networ~ comprising a potentio-meter 318 in series wi~h a resis~or 320 and having a a~,ust-able tap 322 connected to the input o~ the irst differential stage 302 through a resistor 324. When there ls no current ~lowing thxough the ~ain adjustment ~etwork, the w l~a~e with respect to circui t common C at the tap 32~ remains at zero ;~ lO ~olts throughout the entire range of gain control.
:
The differential amplifier stage 302 compxises a irst transistor ~ having a base connected to the output rom the chopper 44 and the voltage feedback network 300.
~; The base of a second transîstor 330: is connected to circuit - 15 common C khrough a resistor 332. The differential amplifier stage 302 includes biasing resistors 334, 336 and 338 which are connected between the positive power s~pply terminal +V
and the nega~ive power supply terminal ~V
The second amplifier stage 304 comprises a first - Z ~Q transistor 34Q havin~ a base connected to khe collector of ~he transistor 32~ and a second transistor 342 ha~ing a base connected to the collector of the transistor 330. Biasing resistors 344, 346 and 3~8 are connected between the positive power suppl~ terminal -I-Vl and circuit common.
The collectoxs of the transistors 340 and 342 are connected to th~ bases of a pair of trans.istors 350 and 352 o ~he volta~e to currenk stage 306. The collector-emitter ci~cuits o the transistors 350 and 352 are connected in series with a resistor 354 ]~etweell the ~o~er suppl~ term:inal ~Vl an<~
the ne~ati.ve power supply terminal ~ 2 . .

~.r-~z~

The output sta~e comprises a pair of transistors 35Ç and ~58 where the base o~ the transistor 356 is connected to th~ junction of the resistor 35~ and the collector of the trsnsistor 352 in the ~oltage to curren~ gain st~ge 306. The output current from the output stage 308 is connected to the re~istor 57 through a resistor 360. Resistors 36~ and 364 ~r~ ~S JS~a~
co~nect the collec~or and cmitter o~ ~he~e-~-~t~s 356 and 358 x~spe~tively to the terminal 20 of the two-wire transmitter.
When an u~balance is created at the bxidge network 26, ~he ~olta~e output ~rom the chopp~r44 in~reas~s which tends to ma~e the ba~e of the transis~or 328 more positive. qlhis rende~s th~ kransistor 328 more conduc~ive and the transistor 330 less conductive which in turn causes ~he voltage at the c~llector o~ the transistor 328 to decrease.and the voltage o~ the collector of the transistor 330 ~o rise. The voltages at ~he collectors of the ~ransistors 328 and 330 are then applied as input ~o the b~ses of the transis~ors 340 and 342 causin~ the voltages ak the collecto_s-o the transistors 340 and 342 to increase and decrease respectively. This in turn.
aau~es ~he transistors 350 and 352 to becvme~more conduc~ive and incxease~the cuxrellt flow througll the resistor 354 thereby raisin~ th~ bas¢ o~ the transistor 356 to a more positive Yolta~ causing an increase in curren~ flow from the ou~put ~ran~i~tors 356 and 358.
S.ince all o~ the cur~^ent ~rom the output transistors 35G ~n~ 35~ ~lows thxou~ll the resi.s~or 57~ the voltage across ~hé rcs.is~ox ~5~ will incrcas~ with i.ncreasing current flow due ~o ~llc ullbalance oi.the bxidge ne~work thereby decreasinc3 , , 2~

the voltagc at the terminal 22 with respect to circuit common C.
This in turn incr~ases the negative voltage which is appli~d to the base o~ the transistor 328 through the voltage feedback divider ne~ork until that voltage is again zero volts thereby establishing a st~ble condition at the higher output current.
From the foregoing, ik should be understood that the output amplifier 56 may be analogized to an operational ampli-~i~r having one input at the base of transistor 328 acting as a summing junction for the voitage ~rom the output of the chopper 44 and the voltage of the voltage feedback divider network 300 and the other input at the base of the;transistor ` I connected to circuit common.
In accordance with another important aspec~ of t~e i~ention, the 1ength of the cables associated with the probe - 15 will not affect ~he admittance measurements.
As shown in Fig. 5, a probe 400 is connected into the brid~e net~qoxk 26. The probe 400 includes a guard elec-: , trode 410 juxtaposed to and surroun~:ng a probe electrode 412.
Insulation 414 surrounds the probe electrode 412 so as to insulate ~he guard electrode 410 rrom the probe electrode 412 ~ and the guard electrode 410 from a grounded conductive vessel ; 418, A coaY.ial cable is utilized to connect the probe 400 ~,,- , , 0~
into the bri~ge network~ here the shield of the cable ~20 is conn~ctea to the guard electrode 410 at one terminal of the span capacitor 60 al~d the axial conductor 422 connecks the prGl~c electrode 412 to the other terminal of the span capacitor 60.

.

.
~?
: ~ ~
..

24~

ReEerence to Figure 6~ wherein the equivalent circuit of Figure 5 is shown, reveals that a variation in the cable length will have no effect on the admittance measurement. As shown, the probe electrode to ground admit-tance 24 is represented by a capacitance 24c and a resistance 24r. Since the axial conductor 422 is surrounded by the coaxial shield 420 which is con-nected to the opposite terminal of the span capacitance 60, any admittance between the coaxial shield 420 and the axial conductor 422 will be connected across the span capacitance 60 and will not affect the balance or unbalance o the bridge network. ~imilarly) any admittance between the coaxial shield 420 and ground as represented by a capacitance 426c and a resistance 426r will have no effect on the balance of the bridge network 26 since this ad-mittance is in parallel with the secondary 34 of the transformer.
In accordance with another important aspect of the invention, ~- linear calibration of the admittance measuring system is achieved by making the span capacitance 60 large relative to the capacitance af the admittance ~ being measured as dlsclosed in United States patent 3,778,705 - ~altby, issued - December 11, 1973. Preferably, the capacitance of the span capacitor 408 or the span capacitor 26 is at least 10 times the capacitance of capacitance 424c or capacitance 24c. In a particularly preferred embodiment, the span capa-citance is 25 times the capacitance being measured.
`~ As shown in Figure 5, the probe 400 comprises a probe electrode 412 which is co~pletely surrounded with insulation 414. As also shown the insulation 414 is coated with materials 428 contained within the vessel 418. As will now be ., ~

: ' '- ' ~ ' , ' ' , ;
: ., ' . ' ' ' ' M-56~-t explained, the ~robe electrode-to-ground resistance ~4r will, in substantially all applications, be in excess of tha prev-iously mentioned 500 ohms even when the probe is covered with a coating 42S of conductive liquid 429 as shown in Fig~ 5.
Referring now to ~ig. 7a, a schematic representation of the coating 4~8 on the probe 400 o~ Fig. 5 illustrates the nature of the probe-to-ground xesistance. As shown there, the coating 428 may be represented.by a series of small resistors .
430 which are coextensive with the length of the coatin~. The junction o~ these resistors 430 are connected to the probe electrode 414 by shunt capacitors 43~ whlch repr6sent the capacitance o~ the insulation 414. An e~uivalent circuit corresponding to the probe and coating o Fig. 7a is illus-trated in Fig. 8a wherei~ the capacitor 432 is connected in lS shunt with the res.istor 430~ A capacitor 434 represents the capacitance through the insulation 414 from the conductive uid below the coating 428 to the probe electrode 412~ This e~uivalent circuit may in turn be represented as shown in Fig. 9 -.) by the shunt resistor 424r and the shunt capacitor 424c~ It ~ has been found that in substantially all applications where the resistance 424r as shown in Fig. 9 is contributed by the ..coating 428 as represented by the series of resistors 430 shown in Fig. 7a, the resistance 424r is more than 500 ohms.
Fig. 7b represents the insulated probe ~00 of Fig. 5 ~ 25 immersed in a s~mi-conductive liquid wherein the liquid itself :. is reprcsented by a number o~ shunt capacitors 436 and shunt xcsistors ~3~. The cquivalent circuit for the immersed probe o~ ~ig. 7b is shown in Fig. ~b wherein the shunt capacitors ~36 ;~
~ ? S' -and the shunt resistors 438 are connected in parallel and a capacitor 434 again represents the capacitance through the insulation from the materials to the probe electrode 412. The equivalent circuit of Pigure 8b may of course also be depicted as a shunt resistor-capacitor combination as shown in Figure 9.
Although the resis*or 438 is now contributed by the semi-conduc~ive material rather than the coating as in the immersed probe of ~igure 7a, it has never-theless been found that the equivalent resistance 424r as depicted in Figure 9 will, in substantially all cases, exceed 500 ohms for the immersed probe of Figure 7b.
Finally, Figure 7c depicts a bare probe 440 immersed in semi-conductive materials which may be represented b~ shunt capacitors 436 and shunt resistors 438 which a~e depicted in schematic circui~ form by a resistance 442 and a resistance 444 in Figure 8c. Once again, it has been found that the resistance444 which represents the resistance 424r of Figure 9 in thP bridge network will exceed 500 ohms for almost all applications.
As described in the foregoing, the invention may be utilized with insulated as well as bare immersions probes including guard electrodes of the type described in Maltby United States patent 3,879,644, issued April 22, 1975. It will of course be appreciated that the invention is equally applicable to two terminal probes without a guard electrode. It will also be understood that the invention is applicable to non-linear probes wherein the probe electrcde is characterized, i.e., ' -~2-., , ,~ ,. . . .
.
- . : ' ' ~:

24~L

the cross-sectional dimension of the probe elec~rode varies from one end of the probe electrode to the other. Probes of this type are disclosed in Schreiber United States patent 3J269,180, issued August 30, 1966 which dis-closes a non-linear probe wi~houti a guard electrode and a non-linear probe witha guard electrode as disclos0d in United States patent 4,06~,753 assigned to the assignee of this invention. Furthermore~ the invention is applicable to non-immersible probes which sense the condition of an admittance material when in close proximity therewith.
In the foregoing, the invention has been described in terms of a two-wire transmitter. It will of course be appreciated that many aspects of the inventlon may be embodied in other applications such as~ for example a battery powered system, wherein the power available is as limited if not more limited than the two-wire transmitter application.
In this connec~ion, another output amplifier 56 for use in a ~` battery powered system will now be described with reference to Figure 10.
As shown therei the output amplifier is in many respects similar to the output amplifier shown in Figure ~ and substantially identical circuit elements bear identical reference characters.
~` However, the output amplifier of Figure 9 differs in that the voltage feedback from the resistor 57 is not applied to a summing junction in the first differential amplifier stage but rather to the other input of the differential amplifier at the base of the transistor 330. Ihe output signal is represented by the current flow to and from oupput terminals 520 ` ' .
.

M-5~9-~

, and 522 a~ the terminals of a diode 524 in the collec~or~
-~ emitter circuit of the transistor 358.
In operation, a positive input at the base of the ~ransistor 328 and a first differenti~l amplifier stage tends S to increase the current flow through the resistor 57. l'his in turn raises the positive voltage applied to the base of the transistor 330 of the voltage divider network compri.sing the resistors 310, 312 and a resistor 526~ As a result, ~he current through the resistor 57 and the output current termi-nals s2a and 5~2 is stabilized a~ a higher current level.
; It should be understood ~hat the output amplifiex ~, i described is in effect an operational amplifier having one -~ input connected to the output of the chopper and the other input conneGted to a voltage feedback network as &ontrasted with the circuit of F.g. 4 wherein one input served as a summing junction ~onnected to the chopper output as well as the voltage feedback net~ork and the other input was connected to circuik common.
~lthough the chopper 44 has not been shown in detail, it will be understood that the chopper circuits and output amplifier circuits well known in the art are suitable for use in the two-wire transmitter system of this invention. For ~Aal~b~
- ~ e~ample, the chopper circuit disclosed in the aforesaid ~ e~e~
patent ~S, 3,778,705 may be utilized. The output ampliier may comprise any o a number of commerciall~ available differ-ential amplifiers It will also be understood tllat various resonant circuits May be utilized to replacc tlle tank circuit . .

M-5~
.

shown in Fig~ 1. Similarly, the voltage regulator circuit 58 may comprise a prior art voltage regulator well known in the : art.
Another embodiment of the invention will now be . described with reference to Fig. 11. As shown therein, terminals 20 and 22 of the two-wire transmitter are connected to the ~ull wave rectifying bridge comprising diodes 70, 72, 74 and 76 as described in conjunction with the embodiment of Fig. l. As in the previously described embodiment/ the diodes of the full wave rectifying bridge permit the polarity of the `~ terminals 20 and 22 to be reversed without risk of damaging the transmitter or affecting the operation thereof. A spark . protection Zener diode 502 is connected across the full wave : recti~ying bridge so as to limit the ~oltage.which can be - 15 applied to the signal processing circuitry~
The output from the fu11 wave recti~ing bridge i5 - connected to a voltage regulator 500 which supplies substan-tially constant voltages for various _omponents of the trans-- mitter thereby avoiding any inaccuracies in measurements due ~` ~ 20 to undesirable variations in the supply voltage of the . . , transmitter~
In accordance with this embodiment of the invention, the admittance responsive network comprises a probe oscillator : 504 having a frequency which is determined by the probe-to-~round admittance of the materials be.ing sensed as coupled into the probe oscillator 504 through a transformer 506. The frequency of the probe oscillator 504 is th~en compared at a frequency difference detector 507 with the ~requency generated ~ ..

.
. .

: ...

; ~-5~ 2~1 by a reference oscillator 508 having a frequency de~ermined by a reference admittance including step zero capacitors 510 (shown as a single variable capacitor) and a ~ine zero capacitor 512 which are coupled to the reference oscillator 508 through a transformer 514.
In accordance with one important feature of this embodiment, a compensate terminal is provided between the junction of the fine æero capacitor 512 and the step zero capacitor 510. This allows the use o~ matched cable sets which may be connected to the pxobe terminal and the compe~-sate te.rminal to eliminate the effect in variations in cable parameters on the measurements of the two-wire transmitter7 As shown in Fig. ll, the probe-to-ground admittance coupled into the probe oscillator 504 and the reference admittance coupled into the reference oscillator 508 form two sides or halves of an admittance bridge~ In effect, the bridge unbalance resulting from changes in the probe-to-ground admittance is measured by measuring the difference in frequency between the oscillators 504 and 508 at the frequency difference detector 507. The freguency difference detector 507 includes . , a multiplier 516 which is coupled to a low pass filter 518 so as to generate a signal representing the difference between the input frequencies of the probe oscillator 504 and the reference oscillator 508~ -25 . The output from the low pass filter 518 which repre-sents the frequency difference is applied to a squaring ampli-fier 52~ having a feedback path 522 so as to establish hysteresis which is substantially less than the amplituae of the frequency di~ference signal and subs~antially larger than ,~ ~

.~ .
: the amplitude of the carrier frequency components. The output of the ampli~ier 520 will be a square wave whose frequency is the difference between the frequencies of the probe oscillator 504 and the reference oscillator 508.
~- 5 . A di~ferentiating network 524 is coupled to the out-: put of the squaring amplifier 520 so as to generate pul6es having a frequency proporti~nal to the frequency difference ~etween the probe oscillator $04 and the reference oscillator 508 which are in turn coupled to a one-shot multivibrator 526.
; 10 The output from the one-shot multivibrator 526 is a train of pulses o~ constant width and having a pulse repetition rate ; equal to the difference in~frequencies between the probe : oscillator 504 and the reference oscillator 50~. Accordingly, the average DC value of the pulse train from the multivibrator 525 is proportional to its duty factor and this aver~ge value is determined by a low pass filter 528 which is coupled to an output amplifier 530 through a fine span potentiometer 531.
The output from the amplifier 53Q con rols the amount of current drawn through a transistor 532 and a resistor 534 0 connected in series with a resistor 536. As the current drawn.
through the resistor 536 changes, the feedback voltage applied to the output amplifier 530 through a resistor 538 varies so as to provide closed-loop control of the current flowing through the resistor 536 which in turn substantially represents the total current drawn by the instrument~
Re~erence will now be made to the detailed circuitry shown in Figs. 12a and 12b. As shown in Fig. 12a, the voltage regulatox 500 comprises transistors 540 and 542. The collector : : .

~i , i , . _~
:' ~.

M-56~

of the transistor 540 establishes a ~10 volt regulated supply -~ where the collector of the transistor 540 is connected to the emitter of the transistor 542 through a temperatur~ compensating diode 544 in series with a reverse poled diode 546. The emitter of the transistor 542 is connected to regulated circuit common through a resistor 548 and resistors 550 and 552 establish a bias ~or the base of the transistox S42. A capacitor 554 acts as a ~ilter for the voltage regulator. In addition, the voltage regulator 500 comprises a start-up resistor 900 between the B~
line and the ~10 volt line.
~he ~10 volt com~lon terminals of the voltage regulator are connected to the probe oscillator ~04 and the reference oscil-lator 508 shown in Fig~ 12b. Both the probe oscillator 504 and the reference oscillator 508 are of the Class C type for high.
lS efficiency and are respectively decoupled by choke coils 556 and 55~ ana capacitors 560 and 563. ~ pair of probe oscillator tran-sistors 562 and 564 have bases interconnected by a winding 566 which is transformer coupled to a winding 568 connected between proDe and ground, and the winding 568 is also coupled to a winding 570 which connects the collectors of the transistors 562 and ~64. A resistor 572 connects center taps of the windings 566 and 570.
Xf the oscillator 504 is not running, current flow through the resistor 572 will bias the transistors into the linear xegionO When the oscillator begins running, base rectification in the transistors 562 and 564 charges the -- capacitor 574 connected between the center tap of the winding 5~6 and the junction of the capacitor 560 and the coil 556 . resulting in a very eficient Class C mode of operation. A
resistor 576 connects the junction of the coil ~56 and the M-569-l ~24 ., ' , capacitor 560 to the emitters of the transistors 562 and 564 so as to reduce the amplitude of the resulting current pulses and spread their width thereby reducing the harmonic distcirtion present in the output waveform.
The reference oscillator 508 includes components which are comparable to those of the oscillator 504. In paxti.-cular~ the oscillator 508 includes transistors 578 and 580 having bases interconnected by a ~inding 582 coupled to a winding 584 connected between the compensate terminal and ground. The collectors of the transistors 578 and 580 are : connected by another winding 586 with the center ta~s of the : windings 582 and 586 being connected by a resistor 588. The reference oscillator 508 is also capable of Class C operation provided by the charging of the capacitor 590 which holds the transistors 578 and 580 off during most of the i~ycle. A
resistor 592 connected between the emitters of the transistors ~ 578 and 580 at the junction:of the coil 588 and the capacitor - 563 reduces the amplitude of the curr~nt pulses and spreads th~ir width as in the case of the resistor 576 in the probe oscillator 504.
In practice, the voltage appearing across the winding 570 and the winding 586 will be approximately 40 volts peak-to-peak with each end going plus and minus 10 volts. The base-to-base voltages of the transistors 562 and 564 and the tran-sistors 578 and 580 will be driven at 4 volts peak-to-peak,and since each base will be driven at 2 volts peak-to-peak, the center tap of the windings 566 and 582 will be approximate1y 1 volt positive with respect to the base of the conducting transistor or about 0.3 volts positive with xespect to the . emitter.

L- - ~7 ~: ' M-5~9~ 24~

The time constant of the resistor 572 and the capa-citor 574 and the time constant of the resistor 588 and the capacitor 590 are chosen so as to permit the capacitors to - discharge approximately 0.1 volts in each half cycle. This assures that there will be a pulse in the following half cycle if the Q of ~he tank circuit is at least S, It is necessary that every half cyc].e have a current pulse in order to prevent squegging or envelope modulation of the oscillators output waveform.
It will therefore be understood that the probe oscillator 5~4 and the reference oscillator 508 are substan-tially identical. However~ the probe oscillator includes the probe admittance in the tanh circuit whereas the reference oscillator includes the compensate admittance between the compensate terminal and ground, a fine zero capacitance 512 and a step zero capacitance 510. Moreover, the admittance of the probe oscillatox and the reference oscillator forms a bridge where the ratio of the inductance of the winding 568 to the inductance of the winding 584 is equal, at bridge balance, to the ratio of the combined fine zero capacitance 512, step zero capacitance 510 and the capacitive part of the compensating admittance to the capac.itive part of the probe admittance at bridge balance.
As the bridge moves off balance, the frequency of the probe oscillator 504 will change producing a difference in frequency between the reference oscillator and the probe oscillator. The voltages produced by these oscillators are applied to the multiplier 516 of the frequency dif~erence , .

4~

detector. Depending upon the balance of this multiplier circuit, components may appear in the output proportional to each of the input frequencies, the sum of the input frequencies, and the difference between the input frequencies. Of these, the difference between the input frequencies will be of a much lower frequency than any of the others. Thus, the difference frequency can be filtered off by a simple low pass filter network. As shown in Fig. 12b, ~he multiplier comprises a programmable ampli~ier 600 such as the RCA CA3080. The bias current ~or the amplifier is provided by resistors 602 and 604 and a capacitor 60~. These components are arranged so th~t the peak value of the component due to the probe oscillator 504, through the resistor 604 and the capa~itor 606, is approximately equal to the DC value from ~10 volts through the resistor 602.
A capacitor divider comprising capacitors 608 and 610 place a small raction of th voltage of the QUtpUt o the reference oscillator 508 on the positive input of the amplifier 600 whose negative input is maintainea at AC co~mon by a capacitor 628 The DC operating point of the amplifier 600 is determined by ~0 resistors 616, 618, 620 and 622. A capacitor 614 which is ; ~ connected across the transformer of the probe oscillator 504 is equal to the total capacitance represented by capacitors 608 and 610 connected across the transformer of the reference ; oscillator 508.
- 25 A low pass filter comprising resistors 624 and 626 and capacitors 628 and 630 is connected to circuit common so as to provide a very low frequency cuto~f filtex stabilize the DC operating point of the amplifier 600 at the voltage appearing at the junction of the resistors 616 and 618.

.:
- ~ 3~

Resistors 632 and 634 and capacitors 636, 638, and 640 form a ]ow pass filter 518 whose cutoff frequency lies between the highest output frequency desired, and the operating :Erequency o the probe oscillator 504 and the reference oscillator 508. The low pass filter 518 in conjunction with circuitry of the squaring amplifier 520 provides an AC si~nal at the difference frequency which is amplified by the squaring amplifier 520 without transmitting a significant amount of carrier frequency.
The squaring amplifier 520 will now be described in somewhat more detail with reference to Figure 12a. The output voltage of theclow pass filter lQ 518 is applied to the inverting or negative input terminal of an operational amplifier 642 of the squaring amplifier 520. A reference voltage is applied to the non-inverting or positive input of the operational amplifier 642 which is coupled to the junction of the resistors 616 and 618 in the frequency dif~
ference detector 506 through a resistor 644. The resistor 644 connected to ground through a capacitor 652 in conjunction with another resistor 646 forms a divider network which feeds back a small fraction of the output voltage of ~he operational amplifier 6~2 to provide hysteresis. The hysteresis of the squaring amplifier is substantiallyless than the amplitude o'f the frequency difference - si~p,al and substantial~larger than the amplitude o the carrier frequency com-ponents. Thus, the output of the ampli,fier 642 will be a square wave whose frequency is the difference between the frequencies of the reference oscillator 508 and the probe oscillator 504. The output from the operational amplifier 642 is coupled to the differentiating circuit 524 comprising a capacitor 648 anda resistor 650. Thc output from the differentiating circuit 524 is connected to the one-shot multivibrator 526 through a diode 65~.

, r ''''' .' ' ' ' M-569~ Z4 ~ ~
.

As shown in Fig. 12a, the one-shot multivibrator 5Z6 comprises field effect transistors 654, 656, and 658. A
positive pulse coupled through the diode 652 to the gate of the transistor 654 enhances it, driving the gates o~ the tran-sistors 655 and 658 negative. This causes the positive tran-~istor 656 to be enhanced, providing a positive out.put. At the same time, a step span capacitance 660 (shown as a variable capacitor or simplicity~ is selected by a switch and raises the voltage on the gate of the transistor 654 to a high level, e.g., approximately 10 voltsD As the step span capacitance charges through resistors 662 and 664, the voltage at the gate of the transistor 654 decays exponentially un~il it reaches the threshold voltage of the transistor 654. At that time, the transistor 654 turns off and the transistors 656 and 558 receive a positive gate voltage causing the output to return negativen The selected capacitor drives the gate of the transistor 654^hard negative, but the current supplied to the gate protection diodes is limited by ~he resistor 662 to prevent the capacitor discharge fro~ damaging the gate metali-.
zation. By making the resistor 662 considerably smaller than the resistor 664, the one-shot multivibrator is ready for another pulse in a small fraction of its operating time.
The time constant provided by the step span capacitor G60 and the resistors 662 and 664 is such as to permit the one shot multivibrator to have a duty factor in the range of 80-90 at the full-scale output. The output of the multivibrator 526 is a pulse train, whose pulses have a constant width determined by the step span capacitor 660 at a pulse repetition rate equal to the difference in frequencies between the probe oscillator . . .
' ,,~ '' S~/ ''' Z~

504 and the reference oscillator 508. The average DC value of this pulse train is directly proportional to its duty factorl which is in turn directly proportional to the pulse repetition rate since the pulses are of constant width~ and ~hus~ to the difference in frequency between the probe oscillator 504 and the reerence oscillator 508.
. The output from the multivibrator 526 is applied to the oukput amplifier 530 which will now be described in detail : with reference to Fig. 12a. The DC value from the multivibrator

5~6 is filtered off by resistors 668 and 670, and capacitors - - ` 672 and 674 `at the input o~ the output amplifier 530. A
resistor 676 connected in series with the resistors 668 and 670 raises the voltage at the junction of the resistor 670 and the resistor 676 to a value within the operating range of the - 15 operational amplifier 678. The junction of the resistor 670 and the resistor 676 is connected to the posit ve terminal of an operational amplifier 678 through a fine span potentiometer in series with a resistor 682~ A balance potentiometer 684 ~ - connected in series between resistors 686 and 688 is used to - 20 adjust the negative input of the operational amplifier 678 to be the same as th voltage at the junction of the resistor 682 and resistors 690 and 538 when there is no pulse train coming :~ from the multivibrator 526. As a re~ult, there is no voltage across the fine span potentiometer 680 at balance, and thus, the current drawn by the instrument is independent of fine pd-~Cn7!~ r 0~ e -~e r r~ ~pan setting. A ~e~R#s}.0me~e~ 694 is connected in series with the;resistor 690 for adjusting the current flow. This current `- ~low may be adjusted so as to establish 4 milliamperes in a ~, ...

M-569~ 24 ~

4-20 milliampere instrument when there is no pulse train from the multivibrator. In the alternative t another current may be established in a different current range.
As mentioned previously~ substantially all o the current drawn by the instrument flows through the resistor 536 so as to generate a voitage relative to common which is pro-portional to the total curr2nt drawn by the instrument. This ~oltage is fed back through the resistor 538 to the positive input of th~ operational amplifier 678. The operational ampli-fler 678 responds to a positi~e input by increasing the current drawn through the resistor 534 and the transistor 532 ther~by increasing the total current drawn by the instrument until the - volta~e drop across the resistor 536 brings the voltage at the positive input to the operational amplifier 678 down to the voltage at the negative input, thus giving closed-loop control o~ the total current drawn by the instrument.
- ~ two-wire transmitter embodiment is shown in Fig. 11 - ~ and Figs. 12a and 12b. However, the lnvention may be embodied ~ ;,f in a battery po~ered application where the squared pulses from the output of the squaring amplifier 520 are counted or other-wise integrated and displayed.
A further embodiment of the invention will now be described with reference to Fig. 13O As sho~n therein, terminals 20 and 22 o~ the two-wire transmitter are connected to the full ; 25 wave rectifying bridge comprising diodes 70, 72, 74 and 76 as described in conjunction with the embodiment o Fi~o 1 as well as the embodiment of Figs. 11, 12a and 12b. As in the previously described embodiments, the diodes o~ the full wave rectifying .
,'~,,~' ~
, ~5_ M-569~ 4~

bridge permit the polarity of the terminals 20 ana ~2 to be reversed without risk of damaging the transmitter or affecting the operation thereof~ The spark protection Zener diode 502 is connectea across the full wave rectifying bridge so as to : 5 limit the voltage which can be applied to the signal processing circuitry. The output from the full wave rectifying bridge is connected t~ the voltage regulator 500.
In accordance with this invention, the a~mittance responsive network comprise~ a ramp~type admittance bridge.
One side or half 790 of ~he bridge comprises a current source includlng a ~ixed zero current source 800 and a span current ~ source 802 where resistance is included in the zero and span : c~rrent sources. Both the zero current source and span current source are connected to the unknown admittance which is in ~eries with a capacitor &03. The æero current source estab-lishes a reerence value of the unknown admittance while_- the ~ span current source, ha~ing a magnitude controlled by an :- internally generated feedback voltage so as to rebalance the i' bridge, establishes the full scale range of the bridge. The re~erence side or half 792 of the bridge comprises a resistance . 804 in series with a capacitor 806.
The time required for the current sources 800 and , 802 to change the voltage across the unknown admittance between :- probe and ground in ramp-like fashion is compared to the same 2~ time required for the reference resistance 804 to change the voltage across the reerence capacitor by a fixed amount in the following manner~ A reset circuit 808 including a comparator 810 is connected across the capacitor 803 and the probe-to-ground ,.

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admittance. As shown in Fig. 13, the positive input to the comparator 810 is connecte~ to the junction of the capacitor 803 and the current sources 800 and 802. The negative input to the comparator 810 is connected to a reference voltage 812 .5 With switches 814 and 81Ç in the position shown, the unknown admittance from probe-to-ground and the capacitor 803 are free to charg2 in ramp-like ~ashion .in response to current flow from the sources ~00 and 802.
Simultaneously, a comparator 81B of the time . ~ 10 aifference detector ~ircuit 820 compares the voltage across - the xeference capacitor 806 with a reference voltage 822.
With a switch 824 of the reset circuit 808 in the position shown, the capacitor 806 is free to charge. By providing the reference resistor 804 and the reference comparator 806 with a shorter time constant than the time constant associated with the capacitor 803, the probe~to=ground admittance and : the resistance associated therewith, the comparator 818 will . ~ produce a change in state of this output before the comparato 810 produces a change in the state o~ its output. When the ~0 positive input to the comparator 810 rises to a sufficiently : high level, the state of the output from the comparator 810 will change which in turn changes the state of the switches 814, 816 and B24 to the opposite positions. When the switch 816 is in the opposite position, a reset voltage is applied to the negative input of the comparator 810. During the reset period, th* vol~age ~cross the probe-to-ground capacitance and the voltage across the re~erence capacitor 806 diminish until such time as the voltage applied ts the positive input of the comparator 81Q falls below the r~set voltage reference VRs :-r ` ~S~
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~t that time, the switches 814, 816 and 824 revert to the position shown and a new charging cycle is initiated. Upon reset, the output from the com-parator 818 changes states so as to produce a pulse output representing the magnitude of the unknown admittance from probe to ground. In other words9 the pulse width of the square wave represents the time difference in chargir,g of the reference capacitor 806 vis a vis the probe-to-ground-admittance.
Tile square waveoutput from the comparator 818 which is generated by switch means 826 is applied to a low pass filter 828 to obtain .m average DC voltage at the output of the Eilter which is proportional to the difference in charge ratesof the probe-torground admittance relative to the reference admittance 806. The output from the low pass filter 828 is applied to an amplifier 830 which produces a feedback voltage for controlling the span current source 802.
The output from the time difference detector 820 is then applied :- to a modulator 832. ln accordance with one important aspect of the invention, the modulator 832 which is directly connected to the probe circuitry is isolated from the remainder of the transmitter circuitry by an i501ating transformer 834 compr-i~sing a primary 836 and a secondary 838. Modulation i$ achieved by chopping the DC output from the amplifier 830 in response to the output from anoscillator 8~0. The chopping circuitry of the modulator 832 is depicted as an amplifier 842 in combination w`ith switch means 844.

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The oscillator 840 comprises a sguare wave oscillator section 846 ~hich is directly connected to the voltage regulator 500 and an isolated supply section 848 which is coupled to the square wave oscillator 846 by an isolating transformer 850.
The secondary of the isolating transfonmer 350 of the isolated . supply 848 provides the chopper drive for the modulator 832.
Tha isolated supply section 848 also pxovides a *11 and ~5 volt suppl~ to that portion of the ~wo-wire transmittex circuitry which is connected directly to the probe and ground. The . iO remainder of the transmitt~r circuitry including a demodulator 852 and an output circuit 854 are supplied by a +10 volt output ~rom the voltage regulator S00.
As shown in Fig. 13, the demodulator 852 co~prises a synchronous rectifier depicted by an amplifier 856 and switch mean5 858 which demodulate the square wave produced at the . secondary 838 of the transformer 834. The resulting ull wave . rectified voltage i5 app1ied to a low pass filter 860 to remove AC components prior to application ~o the output circuit 8540 -.Th~ output circuit 854 comprises the a~pli~ier 530 ' .
desc~ibed in connection with the embo~iment of Figs. 11, 12a and 12b as well as the transistor 532 and the resistors 534, 536 and 538. In addition, the output circuit 854 comprises a bias network 862 connected between common and the inverting . terminal of the amp~fier 530 and a resistor 8G4 connected between the low pass filter 86d and the non-inverting terminal . of the ampli~ier 530.

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In the embodiment of Fig. 13, a shield buffer 865 is provided for use in conjunction with a shield terminal which serves as a guard electrode to prevent long cables and coatings from - influencing the measurement of the admittance from probe to ground. The shield bufer 865 comprises an amplifiex 866 having a non-inverting terminal connected to the probe terminal and the output of the amp}ifier 866 connected to the shield terminal so as to drive the shield or guard electrode at substantially the same potential as the probe so as to eliminate `,~ 10 the effect of long cables and coatings on the measurement.
As also shown in Fig. 13, the unknown admittance side of the bridge 790 provides circuitry for protecting the probe and shield terminals. More particularly, a pair of parallel reverse poled diodes 86~ are connected between the ~; 15 probe and shield terminalsO In addition, a pair of reverse poled Zener diodes 870 are connected from the shield to ground.
In this configuration, the shield tends to break up any s~ray ~ ~ couplin~ path through the diodes 868 and 870.
- ` The embodiment o~ Fig~ 13 will now be described in further detail with reference to Figs. l~(a-d). As shown in Fig. 14a, the voltage regulator 5nO comprises substantially the same components as described with re~erence to the embodi-ments o~ Figs. 11, 12a and 12b. In additionl the voltage regulator 50~ comprises a start-up resistor 900 between the - 25 B+ line and the +10 volt line.
As also shown in Fig. 14a, the oscillator 840 compr.ises a multivibrator including transistors 902 and 904, capacitors 906 and 908, and resistors 910, 912, 914 and 315.

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The isolation transformer 850 which provides high voltage isolation between the portion 846 includes a transformer primary 918 which is directly connected to the two trans-mission lines and the portion of the oscillator circuit 848 -~ S comprising a secondary 920 which supplies the portion of the transmitter which is connected to the probe and ground. The output from the secondary 920 is rectified by diodes 922 and iltered by capacitors 924 so as to provide supply voltages ~or the ~11 and ~5 volt lines. A modulatiny signal is derived from a terminal 926 of the secondary 920 which is , . . .
grounded at the center tapO
- Reerring now to Fig. 14c, the side 790 of the bridge which incorporates the unknown admittance ~rom probe-to-ground will now be described in detail. As stated previously, the zero current source 800 and the span current source 802 are connected in series wilh the capacitor 803 and the unknown admittance from probe-to-ground. The zero current source 800 is controlled by a voltage picked off the ~5 volt supply line ~i~ by a fine zero potentiometer 928 which is connected in series with the resistors 930 and 932. The potentiometer 928 is connected to the non-inverting terminal ~f an operational amplifier 934 which has an output coupled to transistors 936 and 938 with the collector o~ the transistor 936 connec~ed to the capacitor 803 through a resistor 940. The emitter of the transistor 936 and the collector of the transistor 938 are connected to a step zero resistance 942 (whic~ has been shown as a potentiometer for simplicity). A feedback voltage is developed across the step resistance 942 which is applied to , . .

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the inverting terminal of the operational amplifier 934O The current flow from the operational amplifier 934 will increasè
or decrease in xesponse to changes in the variable resistance 94~ so as to achieve a balance between the input at the inverting terminal and the input at the non-inverting terminal - of the operational amplifier 934. ln this connection, it will be understood that as the voltage from the fine zero potentio-meter 928 goes more negative, a larger current will flow from ; the zero current source 800. The zero current source 800 further comprises a resistor 944 in series with a capacitor 946 which is conn?ected between the output of the operational amplifier 934 ana the ~5 volt supply line. A supply resistor 948 is connected between the ~5 volt supply line and the operational amplifier 934.
The span zero current source 802 comprises the sam~
- components as the zero current source 800. For the sake of brevity and simplicity, the same reerence characters on Fig.
14b have been utilized with the addit;on of the letter "~"
indicating a component of the span current source. The only difference between the span current source 802 and the zero current source 800 is the use of a feedback voltage at the non-inverting input of the operational amplifier 934s so as to maintain balance between the unknown admittance side 790 of the admittance bridge and the reference side 792 of the admittance bridge.
It will be noted that the operating controls for that portion of the transmitter which is connected to ground, i.e., fine zero, step zero, fine span and step span, are all direct current controls as contrasted with RF controls. More a~
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particularly, the operating controls comprise variable resistances in the zero curxent source and the span current source so a~ ~o adjust ~he charging rate of the unknown admittance probe ~o ground.
Referring to Fig~ 14d, the reset circuit 808 comprises a transistor 950 which sPrves as the switch 840 which is coupled to the positive terminal of the comparator 810~ A fie~d effect transistor 952 in conjllnction with a transistor 9S4 functions as the switch means 816 to control the negative input to the comparator 810. A transistor ~56 connected across the refexence capacitor 806 serves as the switch 824.
~he operation of the reset circuit 808 is as followsO
- The 2ero current source 800 and the span current source 802 charge the capacitor 803 and the unknown admittance as shown in Fig. 14c until the voltage thereacross is equal to the voltage at the negative input of the comparator amplifier 810 as determined by the +5 volt supply in conjunction with resistors 958, 960, 962, ; 964 and 966. At this time, th~ comparator amplifier ~10 turn~ on - i~ the field effect transistor 952 causing the reset function to be implemented and at the same time xeducing the voltage on the negative input of the comparator amplifier 810 via the resistor 962 to a small voltage. Simultaneously, the transistor 954 turns on the transistor 9S0 which discharges the admittance formed by the capacitor in series with the unknown admittance until the voltaye thereacross falls below the voltage present on the negative input of the comparator amplifier 810~ ~he reset function is then terminated and the charge cycle repeats. The reset circuit also comprises resistors 968, 970, 972 and 974 which .

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bias ~he transistors 950 and 956~ In additio~, a supply resi~tor 976 connects the comparator amplifier 810 to the ~5 volt supply . line.
. As shown in Fig 14d, the reference side of the ~ridge ~: 5 792 comprises the reference capacitor S06 and the reference ; resistor 804. By providing a ~ime constant for the reference side 792 of the bridge which is shorter than that of the current sources and the admittance formed by the capacitor 803 and the unknown admittance from probe to ground~ the comparator amplifier 818 will trip before the comparator amplifier 810. The ~oltage : across the capacitor 806 is compared with the voltage generated by the divider comprising resistors 978 and 980.
The switch 825 referred to in Fig. 13 compxises a field effect transistor g82 which is connected to the ~5 volt supply line through a resistor 984 and to the po~i~ive.:input o~ the amplifier 990 through a resistor 986-which is also connected to the ~5 volt supply line through a capacitor 988.
~-~ When the comparator amplifier 818 is tripped, the voltage at the ~i junction of the t:ransistor ~82 and resistors 986 and 984 will be 2Q pulled down toward ground. When the reset function is initiated the transistor 9S6 in the reset circuit 808 will dlscharge the capacitor 306 to reset the voltage at the junction of the tran-sistor 982 and resistors 986 and 984 will re~urn to ~5 volts.
~- The resistor 986 and the capacitor 988 fo~mthe low pass filter : 25 828 which filters the resulting square negative pulse in obta.ining DC voltage proportlonal to the charge time difference between the :. reference half of.the bridge 792 and the unknown reference side o~ the bridge 790.
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The voltage across the capacitor 988 is amplified by an amplifier 990 which is supplied by the ~5 volt supply line through a resistor 992. The gain of the amplifier 990 is pro-portional to the ratio of the sum of a feedback resistor 992 and : 5 a resistor 994 to the resistor 994 alone. The output from the amplifier 990 is ed back to the unknown side of the bridge 790, : and more particularly, to the span current source 802 so as to control the amplifier 934s.
The output from the amplifier 990 is also chopped in the modulator 832 by the switch means 844 comprising field effect transistors 996 and 998. The modulation is synchronous with the drive from the isolating transformer 920 shown in Fig. 14a at the terminal 926 which-is applied t,o the junction of the field effect transistors through a capacitor 1000. The bias at the juncti~n of the field effect transistors 996 and 998 is derived from sexies connected resistors 1002 and 1004~ The resulting . square wave produced by the field effect transistors 996 and 998 -- is coupled to the isolating transformPr 834 through a capacitc~
. '~ 1008.
The output rom the secondary 838 of the transformer 834 is coupled to the demodulator 852 which will now be described with xeference to Fig~ 14b.
- -- The square wa~e of varying amplitude which is coupled to the demodulator 852 i5 synchronously rectified by the switch means 858 comprising ield effect transistors 1010l 1012, 1014 and ~016.
The junction of the field efect txansistors is driven by a square wave generated at the oscillator 840 which is coupled through a capacitor 1018 to the ~ates of field effect transistors 1020 and 1022. Resistors 1024 and 1026 bias the ~ates of the `

transistors 1020 and 1022. The resulting full wave rectified volt-age is applied to the filter 860 comprising a resistor 1028 and a - capacitor 1030. The DC output voltage from the f~lter-1860 is fed to the output circuit 854 comprising ~he output amplifier 530, the transistor 532 and the resistors 864, 534, 536 and 538. The output circuit 854 also comprises a resistive bridge including resistors 1032, 1034 and 1036. The resistor 538 also forms part of` this resistive bridge which is ~mbalanced ln response to a positive voltage across the capacitor 1030. The resulting positive input to the amplifier 530 causes the output current to be increased and this output current is measured by the resistor 536 which develops a voltage proportional thereto. This voltage is placed in series with the resistor 538 thereby rebalancing the resistive bridge at the desired output current. In this way, the output current is held constant as a function of the voltage obtained from the de-modulator 852. The currenttdrawn by the output stage of the ampli-fier 530 is drawn through emitter follower transistors532 from the B~ line thereby avoiding any tendency of the output current to deregulate the 10 volt power supply. In this manner, any tendency of the output current to interfere h~ith the operation of the other circuits is eliminated. The output circuit 854 further comprises a supply resistor 1038 and a series RC combination including a resistor 1040 and a capacitor 1042. A capacitor 1044 is connected in parallel with the resistor 1034. Referring now to Figure 14c, the shield buffer 864 will be described. The base of a transistor 1046 forms a positive input to the shield buffer amplifier 866. The base receives the probe voltage through a capacitor 1048 where : `
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t~.e operating point of the transistor 1046 is established by the r~sistive div.ider comprising resi~tors 1050, 1052 and 1054 which is boot-strapped to the output of the amplifier by a capacitor 1056. The negative input of the amplifier 866 comprises the emitter of the transistor 1046~ The emitter is connected ~ directly to the output providing 100~ negative fee~back for the :` am~liier 8~6. Thus the shunting effect of the xesistor 1054 on the input of the ampli~ier 866 is reduced by the gain o~
the amplifier. The current drawn by the transistor 1~46 is proportional to the erxor voltage, i~e., the voltage at the ~ase minus the voltage at the emitter, times the fo~ward transfer admittance of the transistor 1046. This current ~enerates a voltage across the resistor 1060 and is ampli~ied by a transistor 1058~ The output voltage from the collector o~ the transistor 1058 is applied to the bases of transistors 1062 and 1064 which function as emitter followers so as to substantially reproduce the voltage at the output of ~he .~ transistor 1058 at a much lower imped.~lnce. The emitter follower transistors operate Class A/B, and the standby bias current is established by series connected diodes 1066 and 1068 and resistors 1070 and 1072. The diodes 1066 and 1068 compensate for the base emitter voltage of the transistors 1062 and 1064. The resistor 1070 establishes the voltage which the transistors will maintain across the resistor 1072. Since 2~ the diodes and th~ tran~Lstor base-emitter junctions have similar temperature coeffici~nts, the hias current will remain substan-tially unchanged as the temperature o~ the ampli~ier varies, ''`~ .
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~ . , A capacitor 1074 malntains the same drive voltage at the base of both transistors while a capacitox 1076 maintains a low output impedance for positive as well as negative output currents. A capacitor 1078 forms the dominant pole of th~
amplifier 866 allowing its gain to roll of~ to unity below the frequency at which 180 phase shift is obtained. In this manner, the amplifier 866 is prevented from parasitic oscillation.
The output of the amplifier 866 is coupled through a capacitor 1080 to the shield terminal so as to eliminate any DC from the shield and thereby prevent electrolytic corrosion of the shield electrode. A resistor 1082 is connected from the shield electrode to ground.
. ~lthough a pxeferred embodiment of the invention has been shown and described, it will be understood that various ~ 15 modifications may be made without departing from the true spirit : and scope of the invention as set forth in the appended claims.

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Claims (16)

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. In a two-wire transmitter system comprising a power supply and a load at one location and a two-wire transmitter at another location intercon-nected by a pair of transmission lines carrying a variable signaling current, the improvement comprising: an admittance sensing probe including a probe electrode adapted to sense the condition and corresponding admittance of materials; an admittance responsive network coupled to said probe representing the condition of materials; and output means coupled to said admittance respon-sive network for varying the signaling current in response to the condition of materials; wherein said admittance responsive network comprises: first ad-mittance means coupled to said sensing probe so as to include the admittance of said materials; second admittance means comprising a reference admittance;
charge current means coupled to said first admittance means and said second admittance means for charging thereof; discharge means coupled to said first admittance means and said second admittance means for discharging thereof;
and charge rate detection means for detecting the difference in charging rates between said first admittance means and said second admittance means.

2. The transmitter of claim 1 wherein said charge current means com-prises a first zero current source and a second span current source.

3. The transmitter of claim 1 wherein said first admittance means and said second admittance means comprises an admittance bridge.

4. The transmitter of claim 3 wherein said first admittance means forms a first side of said bridge and said second admittance means forms a second side of said bridge, said charge rate detection means detecting a difference in time to charge said first side as compared with said second side.

5. The transmitter of claim 4 wherein said bridge and said charge current means are DC isolated from said transmission lines.

6. The transmitter of claim 5 wherein charge current means comprises means for adjusting the DC charging current by adjusting DC current flow.

7. The transmitter of claim 4 further comprising feedback means coupling said charge rate detection means to said current source means for rebalancing said bridge.

8. The transmitter of claim 1 including guard means and guard amplifier means having an input coupled to said sensing probe and an output coupled to said guard means for driving said guard means at substantially the same poten-tial as said sensing probe.

9, The transmitter of claim 8 comprising parallel reverse poled diodes coupled between said sensing probe and said guard means and a pair of series reverse poled Zener diodes coupled between said shield means and ground.

10. The transmitter of claim 1 wherein said admittance responsive net-work is supplied by said power supply through said pair of transmission lines.

11. The transmitter of claim 10 wherein said output means comprises means for generating a feedback signal substantially proportionaly to the signal current.

12. The transmitter of claim 11 wherein said output means comprises modulator means coupled to said charge detection means for generating an AC sig-nal representing the feedback signal, demodulator means for demodulating the modulated AC signal, DC isolation means for coupling demodulator means to said modulator means, and output amplifier means coupled to said demodulator means, said output amplifier means being coupled to said pair of transmission lines so as to control the current drawn by said two-wire transmitter.

13. The transmitter of claim 1 including spark protection means coupled to said admittance sensing probe.

14, The transmitter of claim 13 including guard means associated with said probe, said spark protection means coupled between said probe and said guard means and said guard means and ground such that said guard means functions to break up the stray path comprised of the protecting devices.

15. The transmitter of claim 1 including spark protection means coupled to said transmission lines.

16. The transmitter of claim 1 further comprising: oscillator means coupled to said pair of transmission means; DC power supply means including rectifying means coupled to said admittance responsive network; and transformer means coupling said oscillator means to said DC power supply means.