WO2017145285A1 - Distortion compensation circuit - Google Patents

Distortion compensation circuit Download PDF

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Publication number
WO2017145285A1
WO2017145285A1 PCT/JP2016/055440 JP2016055440W WO2017145285A1 WO 2017145285 A1 WO2017145285 A1 WO 2017145285A1 JP 2016055440 W JP2016055440 W JP 2016055440W WO 2017145285 A1 WO2017145285 A1 WO 2017145285A1
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Prior art keywords
signal
filter
envelope
unit
calculation unit
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PCT/JP2016/055440
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French (fr)
Japanese (ja)
Inventor
安藤　暢彦
檜枝　護重
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三菱電機株式会社
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Priority to PCT/JP2016/055440 priority Critical patent/WO2017145285A1/en
Publication of WO2017145285A1 publication Critical patent/WO2017145285A1/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems

Definitions

  • the present invention relates to a distortion compensation circuit that compensates for nonlinear distortion generated in a power amplifier.
  • a power amplifier In a wireless communication device, a power amplifier is used to amplify a transmission signal.
  • a conventional distortion compensation circuit the distortion compensation circuit of Non-Patent Document 1 is known.
  • a conventional distortion compensation circuit stores a compensation table that compensates for nonlinear characteristics of a power amplifier in a memory, reads a compensation value corresponding to the instantaneous power of the transmission signal from the compensation table, and multiplies the transmission signal by the readout compensation value. Then, a transmission signal including a distortion compensation component is generated. The multiplied signal is input to the power amplifier, and the power is amplified.
  • the compensation table is created using an I / Q (In-phase / Quadrature) signal of the transmission signal and an I / Q signal of the output signal of the power amplifier.
  • An adaptive algorithm such as an LMS (Least Mean Square) algorithm is used to create the compensation table and updated as needed.
  • a conventional distortion compensation circuit estimates a nonlinear characteristic of a power amplifier by directly comparing a transmission signal and an output signal of the power amplifier, and performs distortion compensation. For this reason, in order to perform distortion compensation, it is necessary to observe at least the signal bandwidth of fifth-order distortion. This means that the operation speed of an A / D (Analog to Digital) converter (analog / digital converter) is required to be at least five times the bandwidth of the transmission signal.
  • a / D Analog to Digital
  • the compensation table is updated by digital signal processing, it is necessary to convert the output signal of the power amplifier into a digital signal by an A / D converter. Therefore, as the signal bandwidth of the transmission signal becomes wider, the operation speed of the A / D converter needs to be increased. However, when the operating speed of the A / D converter is increased, there is a problem that power consumption increases. In addition, increasing the operating speed of the A / D converter means using a high-speed A / D converter, leading to an increase in cost.
  • the present invention has been made in view of the above problems, and is a distortion compensation circuit capable of compensating for distortion of a power amplifier while suppressing the operation speed of an A / D converter even when the bandwidth of a transmission signal is wide.
  • the purpose is to obtain.
  • the distortion compensation circuit of the present invention includes an amplifier that amplifies a transmission signal, a detector that detects an envelope of the transmission signal amplified by the amplifier, and outputs the detected envelope as an envelope signal, and a signal band of the envelope signal
  • a first filter that has the following cutoff frequency and band-limits the envelope signal, an analog-digital converter that digitizes the envelope signal band-limited by the first filter, and an analog-digital converter digitized
  • a first calculation unit that calculates a first distribution with respect to the amplitude of the envelope signal, an absolute value calculation unit that calculates an absolute value of a transmission signal input to the amplifier, and outputs the calculated absolute value as a reference signal;
  • a second filter having a cutoff frequency equal to or lower than the signal band of the reference signal and band-limiting the reference signal; and a second filter for the amplitude of the reference signal band-limited by the second filter.
  • a second calculation unit that calculates the error, a comparison unit that calculates an error between the first distribution and the second distribution, and a transmission signal input to the amplifier so as to reduce the error calculated by the comparison unit. And a compensator that outputs the predistorted transmission signal to the amplifier.
  • FIG. 1 is a block diagram showing a configuration example of a distortion compensation circuit according to Embodiment 1 of the present invention.
  • the distortion compensation circuit includes a signal generation unit 100, a complex multiplication unit 101 (an example of a compensation unit), an absolute value calculation unit 102 (an example of an absolute value calculation unit), an operating point adjustment unit 103, and a LUT (Look Up Table) reading unit.
  • D / A (Digital to Analog) converter 105 D / A (Digital to Analog) converter 105, frequency conversion circuit 106, power amplifier 107 (an example of an amplifier), output terminal 108, envelope detector 109 (an example of a detector), LPF (Low Pass Filter) 110 (an example of a first filter), an A / D converter 111, a CCDF (Complementary Cumulative Distribution Function) calculation unit 112 (an example of a first calculation unit), a comparison unit 113 (an example of a comparison unit), a CCDF calculation unit 114 (an example of the second calculation unit) and LPF 115 (second It comprises one example of a filter).
  • CCDF Computer Cumulative Distribution Function
  • the signal generation unit 100 is a signal generation unit that generates a transmission signal.
  • the signal generator 100 is connected to the complex multiplier 101 and the absolute value calculator 102.
  • the signal generation unit 100 outputs the generated transmission signal to the complex multiplication unit 101 and the absolute value calculation unit 102.
  • the signal generation unit 100 includes a modulation processing circuit and a waveform shaping filter.
  • the complex multiplication unit 101 is a complex multiplication unit that multiplies the transmission signal output from the signal generation unit 100 and the output signal from the LUT reading unit 104.
  • the output signal of the LUT reading unit 104 is a compensation signal that compensates for the nonlinear characteristic of the power amplifier 107, and the complex multiplication unit 101 multiplies the transmission signal by the compensation signal to suppress distortion of the output signal of the amplifier 107. Thus, the transmission signal is predistorted.
  • the complex multiplication unit 101 is connected to the signal generation unit 100, the absolute value calculation unit 102, the LUT reading unit 104, and the D / A converter 105.
  • Complex multiplication section 101 outputs the multiplied transmission signal to D / A converter 105.
  • the complex multiplication unit 101 (an example of a compensation unit) may be configured as the complex multiplication unit 101 by combining the complex multiplication unit 101, the operating point adjustment unit 103, and the LUT reading unit 104.
  • the absolute value calculation unit 102 is an absolute value calculation unit that calculates the absolute value of the instantaneous time of the transmission signal.
  • the absolute value calculation unit 102 is connected to the signal generation unit 100, the complex multiplication unit 101, the operating point adjustment unit 103, and the LPF 115.
  • the absolute value calculation unit 102 outputs the calculated absolute value to the operating point adjustment unit 103 and the first LPF 115 as a reference signal.
  • calculating the absolute value of the transmission signal means detecting the envelope of the transmission signal.
  • the signal output from the absolute value calculation unit 102 is referred to as a reference signal.
  • the operating point adjusting unit 103 is an operating point adjusting unit that performs processing to increase or decrease the magnitude of the output signal (reference signal) of the absolute value calculating unit 102 in accordance with the adjustment value output from the comparing unit 113.
  • the operating point adjustment unit 103 is connected to the absolute value calculation unit 102, the LUT reading unit 104, the comparison unit 113, and the LPF 115.
  • the operating point adjustment unit 103 outputs a reference signal whose size is increased or decreased according to the adjustment value to the LUT reading unit 104.
  • the LUT reading unit 104 is a LUT reading unit that reads a compensation value corresponding to the magnitude of the reference signal from a table stored in a memory and outputs the read compensation value to the complex multiplication unit 101.
  • the LUT reading unit 104 includes a memory in which a compensation table that compensates for nonlinear characteristics of the power amplifier 107 is stored.
  • the LUT reading unit 104 is connected to the operating point adjustment unit 103 and the complex multiplication unit 101.
  • the D / A converter 105 is a converter that converts an input digital signal into an analog signal and outputs the analog signal to the frequency conversion circuit 106.
  • the D / A converter 105 is connected to the complex multiplication unit 101 and the frequency conversion circuit 106.
  • the D / A converter 105 may be integrated with an FPGA (Field (Programmable Gate Array).
  • the frequency conversion circuit 106 is a frequency conversion circuit that converts the frequency of the transmission signal and outputs the frequency-converted transmission signal to the power amplifier 107.
  • the frequency conversion circuit 106 is connected to the D / A converter 105 and the power amplifier 107.
  • a mixer is used for the frequency conversion circuit 106.
  • the power amplifier 107 is a power amplifier that amplifies the power of the transmission signal and outputs the amplified transmission signal to the output terminal 108 and the envelope detector 109.
  • the power amplifier 107 is connected to the frequency conversion circuit 106, the output terminal 108, and the envelope detector 109.
  • the power amplifier 107 may be a GaAs (Gallium Narside) amplifier, a GaN (Gallium Nitride) amplifier, a Si MOSEFET (Metal Oxide Semiconductor Field Effect Transistor) amplifier, or the like.
  • the output terminal 108 is a terminal that outputs an output signal of the power amplifier 107.
  • the output terminal 108 is connected to the power amplifier 107 and the envelope detector 109.
  • the envelope detector 109 is a detector that detects an envelope component from the output signal of the power amplifier 107 and outputs the detected envelope component to the LPF 110 as an envelope signal.
  • the envelope detector 109 is connected to the power amplifier 107, the output terminal 108, and the LPF 110.
  • a diode detector is used for the envelope detector 109.
  • the LPF 110 is a low-pass filter that performs band limitation processing on the output signal of the power amplifier 107 and outputs the band-limited signal to the A / D converter 111.
  • the cut-off frequency Fc of the LPF 110 is smaller than the bandwidth BW of the envelope signal, and Fc ⁇ BW.
  • the LPF 110 is connected to the envelope detector 109 and the A / D converter 111.
  • a CR filter, an LC filter, or the like is used for the LPF 110.
  • the A / D converter 111 is a converter that converts an analog envelope signal into a digital signal and outputs the digital signal to the CCDF calculation unit 112.
  • the A / D converter 111 is connected to the LPF 110 and the CCDF calculation unit 112.
  • the A / D converter 111 may be integrated with the FPGA.
  • the CCDF calculation unit 112 is a CCDF calculation unit that calculates the CCDF of the band-limited envelope signal and outputs the calculated CCDF to the comparison unit 113.
  • the CCDF calculation unit 112 is connected to the A / D converter 111 and the comparison unit 113.
  • the comparison unit 113 compares the output signal of the CCDF calculation unit 112 (an example of the first distribution) with the output signal of the CCDF calculation unit 114 (an example of the second distribution), and an error corresponding to the difference between the two CCDFs. It is a comparison unit that obtains a value, performs a process of comparing the obtained error value with a preset threshold value, and outputs an adjustment value according to the comparison result to the operating point adjustment unit 103.
  • the comparison unit 113 is connected to the CCDF calculation unit 112, the CCDF calculation unit 114, and the operating point adjustment unit 103.
  • the CCDF calculation unit 114 is a CCDF calculation unit that calculates the CCDF of the band-limited reference signal and outputs the calculated CCDF to the comparison unit 113.
  • the CCDF calculation unit 114 is connected to the LPF 115 and the comparison unit 113.
  • the LPF 115 is a low-pass filter that performs band limitation processing on the reference signal output from the absolute value calculation unit 102 and outputs the band-limited reference signal to the CCDF calculation unit 114.
  • the LPF 115 preferably has the same filter characteristics as the LPF 110.
  • the LPF 115 is connected to the absolute value calculation unit 102, the operating point adjustment unit 103, and the CCDF calculation unit 114.
  • the signal generation unit 100, the complex multiplication unit 101, the absolute value calculation unit 102, the operating point adjustment unit 103, the LUT reading unit 104, the CCDF calculation unit 112, the comparison unit 113, the CCDF calculation unit 114, and the LPF 115 are FPGA logics.
  • a circuit, an ASIC (Application Specific Integrated Circuit), a microcomputer (microcomputer), and the like are included.
  • the functions of the above constituent elements may be executed by hardware such as an FPGA, or may be executed by software so that the processor reads and executes a program indicating the function of each constituent element stored in the memory. May be.
  • the signal generation unit 100 outputs a transmission signal (I (t) + jQ (t)) expressed as a complex number at time t to the complex multiplication unit 101 and the absolute value calculation unit 102.
  • the absolute value calculation unit 102 calculates the absolute value of the transmission signal. If the output signal of the absolute value calculation unit 102 at time t is Mag (t), Mag (t) is expressed by the following equation.
  • the absolute value calculation unit 102 outputs Mag (t) in two directions, the LPF 115 and the operating point adjustment unit 103.
  • the LPF 115 performs band limitation processing on the reference signal and outputs the band-limited reference signal to the CCDF calculation unit 114.
  • the cutoff frequency Fc of the LPF 115 is lower than the bandwidth BW of the envelope component in the transmission signal, the bandwidth of the band-limited reference signal becomes narrower than the bandwidth BW of the envelope component in the transmission signal.
  • the cutoff frequency of the LPF 115 is the same as the cutoff frequency of the LPF 110 described later, and the filter characteristics of the LPF 115 and the filter characteristics of the LPF 110 are desirably the same.
  • the CCDF calculation unit 114 calculates the CCDF using the output signal of the LPF 115 and outputs the calculation result Ref_CCDF (n) to the comparison unit 113.
  • n represents the data number of the calculated CCDF.
  • the operating point adjustment unit 103 performs an operation represented by the following expression on the Mag (t) using the adjustment value G (i) output from the comparison unit 113, and outputs the operation result to the LUT reading unit 104.
  • G (i) corresponds to the gain.
  • i represents the number of comparisons in the comparison unit 113, and is a natural number of 0 or more.
  • the initial value of G (i) is G (0).
  • the operating point adjustment unit 103 uses G (0) as the initial value of the adjustment value.
  • LUT Index (t) is a value obtained by shifting the operating point of Mag (t) by G (i). That is, the operating point adjustment unit 103 adjusts the operating point of Mag (t) by changing the gain with respect to Mag (t).
  • the LUT reading unit 104 reads a compensation coefficient corresponding to the LUT Index (t) from the compensation table stored in the memory, and outputs the compensation coefficient to the complex multiplication unit 101.
  • the compensation coefficients are CompI (t) and CompQ (t). Note that when reading the compensation coefficient from the compensation table, the value may be obtained by interpolation or function fitting of the table value, and the value may be read out.
  • FIG. 2 is a diagram showing an example of a compensation table according to Embodiment 1 of the present invention.
  • the first column is LUT Index
  • the second column is CompI
  • the third column is CompQ.
  • the value of the LUT Index is discontinuous because the data is displayed by thinning to display the entire compensation table.
  • the LUT Index value is a continuous compensation table.
  • the compensation coefficients (CompI and CompQ) are values that compensate for the nonlinear characteristic of the power amplifier 107.
  • the LUT Index corresponds to the true value of the input power. Since it is difficult to understand in the form of a table, a graph illustrating a compensation table in the form of amplitude characteristics and phase characteristics is shown below. The amplitude is ⁇ (CompI 2 + CompQ 2 ), and the phase is tan ⁇ 1 (CompQ / CompI).
  • FIG. 3 is a diagram showing the compensation table according to the first embodiment of the present invention in the form of amplitude characteristics and phase characteristics.
  • the solid line is the amplitude characteristic
  • the broken line is the phase characteristic.
  • the amplitude characteristics and phase characteristics shown in FIG. 3 are inverse characteristics of the amplitude characteristics and phase characteristics of the power amplifier 107.
  • the complex multiplication unit 101 multiplies the output signal I (t) + jQ (t) of the signal generation unit 100 and the output signal CompI (t) + jCompQ (t) of the LUT reading unit 104 as represented by the following expression, and the multiplication result (I ′ (t)) + jQ ′ (t)) is output to the D / A converter 105.
  • the complex multiplication unit 101 multiplying the transmission signal by the compensation coefficient of the power amplifier 107 is called predistortion.
  • the D / A converter 105 converts the transmission signal obtained by complex multiplication of the compensation coefficient into an analog signal and outputs the analog signal to the frequency conversion circuit 106.
  • the frequency conversion circuit 106 converts the frequency of the input transmission signal and inputs the frequency-converted transmission signal to the power amplifier 107.
  • the power amplifier 107 amplifies the power of the transmission signal and outputs the amplified transmission signal to the output terminal 108 and also to the envelope detector 109. At this time, due to the nonlinear characteristic of the power amplifier 107, the output signal of the power amplifier 107 includes a distortion component.
  • the envelope detector 109 detects an envelope component (envelope signal) from the output signal of the power amplifier 107 and outputs the detected envelope signal to the LPF 110.
  • the LPF 110 performs band limiting processing on the envelope signal, and outputs the processed envelope signal to the A / D converter 111. At this time, since the cutoff frequency Fc of the LPF 110 is lower than the bandwidth BW of the envelope signal detected from the transmission signal, the bandwidth of the band-limited envelope signal is narrower than BW.
  • the A / D converter 111 digitizes (digitizes) the band-limited envelope signal.
  • FIG. 4 is a diagram showing the relationship between the frequency characteristic of the LPF 110 according to Embodiment 1 of the present invention and the bandwidth of the envelope signal.
  • the envelope signal includes a and distortion components up to a frequency F D, which is outside the band of the signal component and the signal component to frequency BW, if there is no LPF 110, all of the components operate at high speed A / Digitized by the D converter 111.
  • the operation speed Fs of the A / D converter needs to satisfy the following conditions in order to digitize the signal component of the envelope signal and the distortion component of the envelope signal.
  • the LPF 110 that cuts off Fc which is a frequency lower than the signal bandwidth BW, removes components in the frequency range higher than the frequency Fc, and digitizes components in the frequency range from 0 to Fc.
  • the operation speed Fs of the A / D converter should satisfy at least the following equation.
  • the operation speed of the A / D converter 111 can be reduced and the power consumption can be reduced.
  • the A / D converter 111 outputs the digitized signal to the CCDF calculation unit 112.
  • FIG. 5 is a flowchart showing an example of the operation of the digital portion in the distortion compensation circuit according to Embodiment 1 of the present invention.
  • step S 101 the CCDF calculation unit 112 calculates CCDF using the output signal of the A / D converter 111 and outputs the calculation result Test_CCDF (n) to the comparison unit 113.
  • FIG. 6 is a signal waveform diagram showing waveforms of the envelope signal and the reference signal when there is no band limitation according to Embodiment 1 of the present invention.
  • F S_OLD is the sampling frequency of the A / D converter 111
  • N OLD is the number of data points when determining the CCDF.
  • ⁇ P is a peak difference between the reference signal and the envelope signal.
  • the A / D converter 111 operates at high speed, so that the peak value of the envelope signal can be digitized.
  • the envelope signal includes a distortion component generated by the power amplifier 107, and the peak value close interference particularly includes a distortion component.
  • the reference signal does not pass through the power amplifier 107, a distortion component is not included. Therefore, when the CCDF value of the reference signal and the CCDF value of the envelope signal are compared, there is an error between the two.
  • FIG. 7 is a diagram showing the CCDF of the envelope signal and the reference signal when there is no band limitation according to Embodiment 1 of the present invention.
  • the vertical axis is the occurrence probability
  • the horizontal axis is the amplitude.
  • the difference in CCDF between the reference signal and the envelope signal in the region where the occurrence probability is low corresponds to the difference in peak value in FIG. 6 and is ⁇ P.
  • FIG. 8 is a signal waveform diagram showing waveforms of the envelope signal and the reference signal when there is a band limitation according to Embodiment 1 of the present invention.
  • F S_NEW is the sampling frequency of the A / D converter
  • N NEW is the number of data points when CCDF is obtained.
  • ⁇ P NEW is the peak difference between the reference signal and the envelope signal.
  • the band is limited by the LPF 110 and the LPF 115, so that the speed of the amplitude change becomes slow, but the operation speed of the A / D converter is also slow, so that the peak value of the envelope signal can be digitized. .
  • the distortion component is included in the BW of the envelope signal is that the envelope signal in the region where the amplitude is large even if the band is limited in FIG. 8 is a signal in the nonlinear region of the power amplifier 107. It is understood from the difference between the signal and the signal. The reason why the distortion component is included in BW will be explained a little more.
  • a is a coefficient indicating a linear characteristic.
  • b is a coefficient indicating nonlinear characteristics, and is related to distortion.
  • A-3b ⁇ cos ( ⁇ t) corresponds to the signal component in the signal band (the BW), 4b ⁇ cos (3 ⁇ t ) corresponds to the distortion component in the distortion-band (in F D). Even if the distortion band component is removed, since the signal component includes b, the signal in the signal band is affected by the nonlinear characteristic. As described above, even if the LPF 110 performs band limitation, the envelope signal includes a distortion component.
  • FIG. 9 is a diagram showing the CCDF of the envelope signal and the reference signal when there is a band limitation according to the first embodiment of the present invention.
  • the vertical axis is the occurrence probability
  • the horizontal axis is the amplitude.
  • the difference in CCDF between the reference signal and the envelope signal in the region where the occurrence probability is low corresponds to the difference in peak value in FIG. 8, and is ⁇ P NEW .
  • the number N of data points necessary for calculating the CCDF needs to satisfy at least the following expression using the sampling frequency Fs of the A / D converter 111 and the cut-off frequency Fc of the LPF 110.
  • the larger the number of data points the larger the required number of data points.
  • Fs corresponds to the operation speed of the A / D converter 111.
  • FIG. 10 is a diagram showing the relationship between the number of normalized data points and the amount of distortion compensation deterioration according to Embodiment 1 of the present invention.
  • the horizontal axis represents the number of normalized data points (N / (Fs / Fc)), and the vertical axis represents the deterioration amount of the distortion compensation amount.
  • the normalized data score is a data score obtained by normalizing the data score N with Fs / Fc.
  • the deterioration amount of the distortion compensation amount indicates an amount of deterioration from when the distortion compensation is ideally applied, that is, when the signal distortion is zero. When distortion compensation is ideally applied, the degradation amount of the distortion compensation amount becomes zero.
  • the deterioration amount of the distortion compensation amount deviates from the linear relationship and increases exponentially. This is because, when the number of normalized data points is small, the number of times of sampling the value near the peak of the envelope signal is small, and the accuracy of distortion compensation deteriorates in a region where the distortion is large.
  • the CCDF calculation unit 112 and the CCDF calculation unit 114 determine the normalized data score so that the normalized data score is 8000 or more, multiply the normalized data score by Fc / Fs, and determine the data score N.
  • the normalized data point 8000 corresponds to the distortion compensation amount deterioration amount 0.2 dB
  • the normalized data point number is determined so that the distortion compensation amount deterioration amount is within 0.2 dB, and the data point number N May be determined.
  • the number N of data points may be determined based on the CCDF error.
  • step S102 the comparison unit 113 compares the CCDF of the envelope signal calculated by the CCDF calculation unit 112 with the CCDF of the reference signal calculated by the CCDF calculation unit 114, and calculates a difference (error) between the CCDFs. .
  • the calculation represented by the following equation is performed to obtain the error Err.
  • M is the total number of CCDF values.
  • the comparison unit 113 compares the value of Err with a preset threshold value VTH .
  • the comparison unit 113 updates the adjustment value (gain) for the operating point adjustment unit 103 and decreases it by ⁇ G in step S103. That is, if the number of updates is i, the adjustment value before update is G (i), and the adjustment value after update is G (i + 1), the update is performed as shown in the following equation.
  • i represents the number of comparisons in the comparison unit 113, and is a natural number of 0 or more.
  • the comparison unit 113 updates the adjustment value for the operating point adjustment unit 103 and increases it by ⁇ G in step S104. That is, the comparison unit 113 updates the adjustment value as in the following equation.
  • the operating point adjusting unit 103 adjusts the operating point of the reference signal by increasing or decreasing the power of the reference signal by updating the gain with respect to the reference signal.
  • step S102 if the the Err and V TH satisfy the above equation, the comparing unit 113 does not change the adjustment value for the operating point adjustment unit 103, the process ends. That is, the adjustment value in this case is as follows.
  • the comparison unit 113 performs case analysis in relation to the Err and V TH, and outputs an adjustment value G (i + 1) to the operating point adjustment unit 103.
  • ⁇ G may not be a constant value, but may be variable depending on the number of comparisons, or may be a value proportional to Err.
  • step S ⁇ b> 105 the operating point adjustment unit 103 calculates LUT Index (t) from the adjustment value G (i + 1) and Expression (2), and outputs the LUT Index (t) to the LUT reading unit 104. .
  • step S106 the LUT reading unit 104 reads a compensation coefficient corresponding to the LUT Index (t) from the compensation table stored in the memory, and outputs the compensation coefficient to the complex multiplication unit 101.
  • step S107 the complex multiplication unit 101 multiplies the output signal I (t) + jQ (t) of the signal generation unit 100 by the compensation coefficient output from the LUT reading unit 104, and the multiplication result is D / A. Output to the converter 105.
  • the distortion compensation circuit repeats the series of steps (S101 to S107) described above until Expression (12) is satisfied.
  • Expression (12) is satisfied, the process of the flowchart in FIG. 5 ends.
  • the distortion compensation circuit can update the adjustment value of the operating point adjustment unit 103 so that the difference (Err) of the CCDF between the reference signal and the envelope signal is minimized. Therefore, this distortion compensation circuit can reduce the distortion generated in the power amplifier 107.
  • the distortion compensation circuit of the first embodiment even when the signal bandwidth of the transmission signal is wide, the band is limited by the LPF 110 and the CCDF is used to compare the reference signal and the envelope signal.
  • the distortion of the power amplifier 107 can be compensated while suppressing the operation speed of the / D converter 111.
  • the CCDF is used for the distribution for comparing the reference signal and the envelope signal.
  • PDF Probability Density Function
  • CDF Cumulative Distribution Function
  • FIG. 11 is a configuration diagram showing a configuration example of a distortion compensation circuit according to the second embodiment of the present invention. 11, the same reference numerals as those in FIG. 1 denote the same or corresponding parts. For this reason, the description of the same or corresponding parts is omitted. 1 is different from FIG. 1 in that an LPF 200 (an example of a third filter) is provided between the A / D converter 111 and the CCDF calculation unit.
  • LPF 200 an example of a third filter
  • LPF 200 is a low-pass filter that is arranged downstream of A / D converter 111 and further band-limits the envelope signal band-limited by LPF 110.
  • the LPF 200 band-limits the envelope signal output from the A / D converter 111 and outputs the band-limited envelope signal to the CCDF calculation unit 112.
  • the envelope signal output from the A / D converter 111 is a digital signal
  • the LPF 200 is a digital filter.
  • the cut-off frequency of the LPF 200 is smaller than that of the LPF 110 and is the same as the cut-off frequency of the LPF 115.
  • the LPF 200 has the same filter characteristics as the LPF 115.
  • the LPF 200 includes an FPGA logic circuit, an ASIC, a microcomputer, and the like.
  • the operations up to the A / D converter 111 are the same as those in the first embodiment.
  • the A / D converter 111 digitizes the envelope signal band-limited by the LPF 110, and outputs the digitized envelope signal to the LPF 200.
  • the LPF 200 Since the LPF 200 has a cutoff frequency lower than the cutoff frequency of the LPF 110, it further limits the band of the envelope signal band-limited by the LPF 110. That is, the cutoff frequency Fc3 of the LPF 115 and the cutoff frequency Fc2 of the LPF 110 are in the following relationship.
  • LPF 200 outputs a band-limited envelope signal to CCDF calculation unit 112.
  • the band of the envelope signal output from the LPF 200 is the same as the band of the reference signal output from the LPF 115. If the signals input to the LPF 200 and the LPF 115 are the same, the signals output from the LPF 200 and the LPF 115 are the same.
  • Embodiment 2 since the LPF 115 and the LPF 200 are digital filters, the same filter characteristics can be realized. Thereby, when calculating the CCDF of the CCDF calculation unit 112 and the CCDF calculation unit 114, an error does not occur in the calculated CCDF due to the difference in filter characteristics. Accordingly, when the comparison unit 113 compares both CCDFs, there is no CCDF calculation error, so that it is possible to prevent the optimum value of the operation point of the operation point adjustment unit 103 from being shifted due to a difference in filter characteristics.
  • the filter characteristics of the LPF 115 and the LPF 200 are different, even if the same signal is input to the LPF 115 and the LPF 200, there is a difference between the signal that has passed through the LPF 115 and the signal that has passed through the LPF 200. There is a difference between each CCDF that is calculated.
  • the CCDFs calculated from the same signal should be the same, but an error occurs between the CCDFs, and an error occurs in the operating point adjustment in the operating point adjustment unit 103. Therefore, the operating point is not optimized, and optimal distortion compensation cannot be performed for the power amplifier 107. Therefore, if the filter characteristics of the LPF 115 and the LPF 200 are different, the accuracy of distortion compensation deteriorates.
  • FIG. 12 is a diagram showing the relationship between the CCDF error and the distortion compensation deterioration amount according to the second embodiment of the present invention.
  • FIG. 12 shows that when the CCDF error increases, the distortion compensation amount deteriorates and the distortion compensation is not performed.
  • the deterioration amount of the distortion compensation amount indicates an amount of deterioration from when the distortion compensation is ideally applied, that is, when the signal distortion is zero.
  • the degradation amount of the distortion compensation amount becomes zero.
  • the LPF 115 and the LPF 200 are configured by digital filters, and the filter characteristics are the same. Therefore, no error occurs in the CCDF value, and the optimum operating point adjustment is performed. It can be carried out.

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Abstract

Conventional distortion compensation circuits pose a problem in which power consumption by the A/D converter increases when the transmission signal bandwidth increases. The distortion compensation circuit according to the present invention is provided with a detector which detects an envelope signal from a transmission signal that has been amplified by an amplifier, a first filter which limits the bandwidth of the envelope signal to no more than the signal band of the envelope signal, an A/D converter which digitizes the envelope signal that has been subjected to bandwidth limitation by the first filter, a first calculation unit which calculates a first distribution relative to the amplitude of the digitized envelope signal, an absolute value calculation unit which calculates the absolute value of the transmission signal input into the amplifier and outputs the calculated absolute value as a reference signal, a second filter which limits the bandwidth of the reference signal to no more than the signal band of the reference signal, a second calculation unit which calculates a second distribution relative to the amplitude of the reference signal that has been subjected to bandwidth limitation by the second filter, a comparison unit which calculates the difference between the first distribution and the second distribution, and a compensation unit which applies pre-distortion to the transmission signal so as to minimize the difference.

Description

歪み補償回路Distortion compensation circuit
　本発明は、電力増幅器で発生する非線形歪みを補償する歪み補償回路に関する。 The present invention relates to a distortion compensation circuit that compensates for nonlinear distortion generated in a power amplifier.
　無線通信装置では、送信信号を増幅するために電力増幅器が用いられるが、増幅すると電力増幅器の非線形歪みが生じ、送信信号の信号品質が劣化するため、非線形歪みを補償する必要がある。従来の歪み補償回路として、非特許文献１の歪み補償回路が知られている。従来の歪み補償回路は、電力増幅器の非線形特性を補償する補償テーブルをメモリに保存しており、送信信号の瞬時電力に応じた補償値を補償テーブルから読出し、読み出した補償値を送信信号に乗算し、歪み補償成分が含まれた送信信号を生成する。乗算後の信号は電力増幅器へ入力され、その電力が増幅される。補償テーブルは、送信信号のＩ／Ｑ（Ｉｎ－ｐｈａｓｅ／Ｑｕａｄａｒａｔｕｒｅ）信号と電力増幅器の出力信号のＩ／Ｑ信号とを用いて作成される。補償テーブルの作成にはＬＭＳ（Ｌｅａｓｔ　Ｍｅａｎ　Ｓｑｕａｒｅ）アルゴリズムなどの適応アルゴリズムが用いられ随時更新される。 In a wireless communication device, a power amplifier is used to amplify a transmission signal. However, when amplified, nonlinear distortion of the power amplifier occurs and signal quality of the transmission signal deteriorates, so it is necessary to compensate for the nonlinear distortion. As a conventional distortion compensation circuit, the distortion compensation circuit of Non-Patent Document 1 is known. A conventional distortion compensation circuit stores a compensation table that compensates for nonlinear characteristics of a power amplifier in a memory, reads a compensation value corresponding to the instantaneous power of the transmission signal from the compensation table, and multiplies the transmission signal by the readout compensation value. Then, a transmission signal including a distortion compensation component is generated. The multiplied signal is input to the power amplifier, and the power is amplified. The compensation table is created using an I / Q (In-phase / Quadrature) signal of the transmission signal and an I / Q signal of the output signal of the power amplifier. An adaptive algorithm such as an LMS (Least Mean Square) algorithm is used to create the compensation table and updated as needed.
　従来の歪み補償回路は、送信信号と電力増幅器の出力信号とを直接比較することにより電力増幅器の非線形特性を推定し、歪み補償を行う。このため、歪み補償を行うには少なくとも５次歪みの信号帯域幅まで観測する必要がある。これは、Ａ／Ｄ（Ａｎａｌｏｇ ｔｏ　Ｄｉｇｉｔａｌ）変換器（アナログディジタル変換器）の動作速度が、少なくとも送信信号の帯域幅に対して５倍以上必要であることを意味する。 A conventional distortion compensation circuit estimates a nonlinear characteristic of a power amplifier by directly comparing a transmission signal and an output signal of the power amplifier, and performs distortion compensation. For this reason, in order to perform distortion compensation, it is necessary to observe at least the signal bandwidth of fifth-order distortion. This means that the operation speed of an A / D (Analog to Digital) converter (analog / digital converter) is required to be at least five times the bandwidth of the transmission signal.
　Ｙａｎｇ Ｊｕｎ, “Ｄｉｇｉｔａｌ Ｐｒｅｄｉｓｔｏｒｔｉｏｎ ｗｉｔｈ ｉｍｐｒｏｖｅｄ ＬＵＴ ａｎｄ ＬＭＳ ｍｅｔｈｏｄ,” ＣＳＱＲＷＣ,　２０１１． Yang Jun, “Digital Predistortion with improved LUT and LMS method,” CSRQRW, 2011.
　補償テーブルの更新は、ディジタル信号処理によって行われるため、Ａ／Ｄ変換器によって電力増幅器の出力信号をディジタル信号に変換する必要がある。そのため、送信信号の信号帯域幅が広くなると、Ａ／Ｄ変換器の動作速度は高速化する必要がある。しかしながら、Ａ／Ｄ変換器の動作速度を高速化すると、消費電力が増加する課題が生じる。その上、Ａ／Ｄ変換器の動作速度を高速化することは、高速なＡ／Ｄ変換器を用いることを意味し、コストアップにつながる。 Since the compensation table is updated by digital signal processing, it is necessary to convert the output signal of the power amplifier into a digital signal by an A / D converter. Therefore, as the signal bandwidth of the transmission signal becomes wider, the operation speed of the A / D converter needs to be increased. However, when the operating speed of the A / D converter is increased, there is a problem that power consumption increases. In addition, increasing the operating speed of the A / D converter means using a high-speed A / D converter, leading to an increase in cost.
　本発明は、上記の課題を鑑みてなされたものであり、送信信号の帯域幅が広い場合においても、Ａ／Ｄ変換器の動作速度を抑制しつつ、電力増幅器の歪みを補償できる歪み補償回路を得ることを目的とする。 The present invention has been made in view of the above problems, and is a distortion compensation circuit capable of compensating for distortion of a power amplifier while suppressing the operation speed of an A / D converter even when the bandwidth of a transmission signal is wide. The purpose is to obtain.
本発明の歪み補償回路は、送信信号を増幅する増幅器と、増幅器が増幅した送信信号の包絡線を検出し、検出した包絡線を包絡線信号として出力する検出器と、包絡線信号の信号帯域以下のカットオフ周波数をもち、包絡線信号を帯域制限する第１のフィルタと、第１のフィルタが帯域制限した包絡線信号をディジタル化するアナログディジタル変換器と、アナログディジタル変換器がディジタル化した包絡線信号の振幅に対する第１の分布を計算する第１の計算部と、増幅器に入力される送信信号の絶対値を算出し、算出した絶対値を参照信号として出力する絶対値計算部と、参照信号の信号帯域以下のカットオフ周波数をもち、参照信号を帯域制限する第２のフィルタと、第２のフィルタが帯域制限した参照信号の振幅に対する第２の分布を計算する第２の計算部と、第１の分布と第２の分布との誤差を算出する比較部と、比較部が算出した誤差が小さくなるように、増幅器に入力される送信信号をプレディストーションし、プレディストーションした送信信号を増幅器に出力する補償部とを備える。 The distortion compensation circuit of the present invention includes an amplifier that amplifies a transmission signal, a detector that detects an envelope of the transmission signal amplified by the amplifier, and outputs the detected envelope as an envelope signal, and a signal band of the envelope signal A first filter that has the following cutoff frequency and band-limits the envelope signal, an analog-digital converter that digitizes the envelope signal band-limited by the first filter, and an analog-digital converter digitized A first calculation unit that calculates a first distribution with respect to the amplitude of the envelope signal, an absolute value calculation unit that calculates an absolute value of a transmission signal input to the amplifier, and outputs the calculated absolute value as a reference signal; A second filter having a cutoff frequency equal to or lower than the signal band of the reference signal and band-limiting the reference signal; and a second filter for the amplitude of the reference signal band-limited by the second filter. A second calculation unit that calculates the error, a comparison unit that calculates an error between the first distribution and the second distribution, and a transmission signal input to the amplifier so as to reduce the error calculated by the comparison unit. And a compensator that outputs the predistorted transmission signal to the amplifier.
　本発明によれば、Ａ／Ｄ変換器の動作速度を抑制しつつ、電力増幅器の歪みを補償できるという効果がある。 According to the present invention, there is an effect that the distortion of the power amplifier can be compensated while suppressing the operation speed of the A / D converter.
この発明の実施の形態１に係る歪み補償回路の一構成例を示す構成図である。It is a block diagram which shows one structural example of the distortion compensation circuit which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る補償テーブルの一例を示す図である。It is a figure which shows an example of the compensation table which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る補償テーブルを振幅特性及び位相特性の形式で表示した図である。It is the figure which displayed the compensation table which concerns on Embodiment 1 of this invention in the format of an amplitude characteristic and a phase characteristic. この発明の実施の形態１に係るＬＰＦ１１０の周波数特性と包絡線信号の帯域幅との関係を示す図である。It is a figure which shows the relationship between the frequency characteristic of LPF110 which concerns on Embodiment 1 of this invention, and the bandwidth of an envelope signal. この発明の実施の形態１に係る歪み補償回路におけるディジタル部分の動作の一例を示すフローチャートである。It is a flowchart which shows an example of operation | movement of the digital part in the distortion compensation circuit which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る帯域制限がない場合における包絡線信号及び参照信号の波形を示す信号波形図である。It is a signal waveform diagram which shows the waveform of an envelope signal and a reference signal when there is no band limitation which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る帯域制限がない場合における包絡線信号及び参照信号のＣＣＤＦを示す図である。It is a figure which shows CCDF of an envelope signal and a reference signal in case there is no band limitation which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る帯域制限がある場合における包絡線信号及び参照信号の波形を示す信号波形図である。It is a signal waveform diagram which shows the waveform of an envelope signal and a reference signal in case there exists a zone | band limitation which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る帯域制限がある場合における包絡線信号及び参照信号のＣＣＤＦを示す図である。It is a figure which shows CCDF of an envelope signal and a reference signal in case there exists a zone | band limitation which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る規格化データ点数と歪み補償量の劣化量との関係を示す図である。It is a figure which shows the relationship between the number of normalized data points and the amount of degradation of distortion compensation amount concerning Embodiment 1 of this invention. この発明の実施の形態２に係る歪み補償回路の一構成例を示す構成図である。It is a block diagram which shows the example of 1 structure of the distortion compensation circuit which concerns on Embodiment 2 of this invention. この発明の実施の形態２に係るＣＣＤＦの誤差と歪み補償量の劣化量との関係を示す図である。It is a figure which shows the relationship between the error of CCDF which concerns on Embodiment 2 of this invention, and the deterioration amount of distortion compensation amount.
実施の形態１.
　図１は、この発明の実施の形態１に係る歪み補償回路の一構成例を示す構成図である。
　本歪み補償回路は、信号生成部１００、複素乗算部１０１（補償部の一例）、絶対値計算部１０２（絶対値計算部の一例）、動作点調整部１０３、ＬＵＴ（Ｌｏｏｋ　Ｕｐ　Ｔａｂｌｅ）読み出し部１０４、Ｄ／Ａ（Ｄｉｇｉｔａｌ　ｔｏ　Ａｎａｌｏｇ）変換器１０５、周波数変換回路１０６、電力増幅器１０７（増幅器の一例）、出力端子１０８、包絡線検出器１０９（検出器の一例）、ＬＰＦ（Ｌｏｗ　Ｐａｓｓ　Ｆｉｌｔｅｒ）１１０（第１のフィルタの一例）、Ａ／Ｄ変換器１１１、ＣＣＤＦ（Ｃｏｍｐｌｅｍｅｎｔａｒｙ　Ｃｕｍｕｌａｔｉｖｅ　Ｄｉｓｔｒｉｂｕｔｉｏｎ　Ｆｕｎｃｔｉｏｎ）計算部１１２（第１の計算部の一例）、比較部１１３（比較部の一例）、ＣＣＤＦ計算部１１４（第２の計算部の一例）、及びＬＰＦ１１５（第２のフィルタの一例）を備える。 Embodiment 1.
FIG. 1 is a block diagram showing a configuration example of a distortion compensation circuit according to Embodiment 1 of the present invention.
The distortion compensation circuit includes a signal generation unit 100, a complex multiplication unit 101 (an example of a compensation unit), an absolute value calculation unit 102 (an example of an absolute value calculation unit), an operating point adjustment unit 103, and a LUT (Look Up Table) reading unit. 104, D / A (Digital to Analog) converter 105, frequency conversion circuit 106, power amplifier 107 (an example of an amplifier), output terminal 108, envelope detector 109 (an example of a detector), LPF (Low Pass Filter) 110 (an example of a first filter), an A / D converter 111, a CCDF (Complementary Cumulative Distribution Function) calculation unit 112 (an example of a first calculation unit), a comparison unit 113 (an example of a comparison unit), a CCDF calculation unit 114 (an example of the second calculation unit) and LPF 115 (second It comprises one example of a filter).
　信号生成部１００は、送信信号を生成する信号生成部である。信号生成部１００は、複素乗算部１０１及び絶対値計算部１０２に接続される。信号生成部１００は、生成した送信信号を複素乗算部１０１及び絶対値計算部１０２に出力する。例えば、信号生成部１００は、変調処理回路及び波形整形フィルタで構成される。 The signal generation unit 100 is a signal generation unit that generates a transmission signal. The signal generator 100 is connected to the complex multiplier 101 and the absolute value calculator 102. The signal generation unit 100 outputs the generated transmission signal to the complex multiplication unit 101 and the absolute value calculation unit 102. For example, the signal generation unit 100 includes a modulation processing circuit and a waveform shaping filter.
　複素乗算部１０１は、信号生成部１００の出力した送信信号とＬＵＴ読み出し部１０４の出力信号とを乗算する複素乗算部である。ＬＵＴ読み出し部１０４の出力信号は、電力増幅器１０７の非線形特性を補償する補償信号であり、複素乗算部１０１は、送信信号に補償信号を乗算することにより、増幅器１０７の出力信号の歪みを抑制するように、送信信号をプレディストーションする。複素乗算部１０１は、信号生成部１００、絶対値計算部１０２、ＬＵＴ読み出し部１０４、及びＤ／Ａ変換器１０５に接続される。複素乗算部１０１は、乗算した送信信号をＤ／Ａ変換器１０５に出力する。複素乗算部１０１（補償部の一例）は、複素乗算部１０１、動作点調整部１０３、及びＬＵＴ読み出し部１０４を合わせて複素乗算部１０１としても良い。 The complex multiplication unit 101 is a complex multiplication unit that multiplies the transmission signal output from the signal generation unit 100 and the output signal from the LUT reading unit 104. The output signal of the LUT reading unit 104 is a compensation signal that compensates for the nonlinear characteristic of the power amplifier 107, and the complex multiplication unit 101 multiplies the transmission signal by the compensation signal to suppress distortion of the output signal of the amplifier 107. Thus, the transmission signal is predistorted. The complex multiplication unit 101 is connected to the signal generation unit 100, the absolute value calculation unit 102, the LUT reading unit 104, and the D / A converter 105. Complex multiplication section 101 outputs the multiplied transmission signal to D / A converter 105. The complex multiplication unit 101 (an example of a compensation unit) may be configured as the complex multiplication unit 101 by combining the complex multiplication unit 101, the operating point adjustment unit 103, and the LUT reading unit 104.
　絶対値計算部１０２は、送信信号の瞬時時間の絶対値を計算する絶対値計算部である。絶対値計算部１０２は、信号生成部１００、複素乗算部１０１、動作点調整部１０３、及びＬＰＦ１１５に接続される。絶対値計算部１０２は、計算した絶対値を参照信号として動作点調整部１０３及び第１のＬＰＦ１１５に出力する。なお、送信信号の絶対値を計算することは、送信信号の包絡線を検出することを意味する。ここでは、絶対値計算部１０２が出力する信号を参照信号と呼ぶ。 The absolute value calculation unit 102 is an absolute value calculation unit that calculates the absolute value of the instantaneous time of the transmission signal. The absolute value calculation unit 102 is connected to the signal generation unit 100, the complex multiplication unit 101, the operating point adjustment unit 103, and the LPF 115. The absolute value calculation unit 102 outputs the calculated absolute value to the operating point adjustment unit 103 and the first LPF 115 as a reference signal. Note that calculating the absolute value of the transmission signal means detecting the envelope of the transmission signal. Here, the signal output from the absolute value calculation unit 102 is referred to as a reference signal.
　動作点調整部１０３は、比較部１１３が出力した調整値に応じて、絶対値計算部１０２の出力信号（参照信号）の大きさを増減させる処理を行う動作点調整部である。動作点調整部１０３は、絶対値計算部１０２、ＬＵＴ読み出し部１０４、比較部１１３、及びＬＰＦ１１５に接続される。動作点調整部１０３は、調整値に応じて大きさを増減した参照信号をＬＵＴ読み出し部１０４へ出力する。 The operating point adjusting unit 103 is an operating point adjusting unit that performs processing to increase or decrease the magnitude of the output signal (reference signal) of the absolute value calculating unit 102 in accordance with the adjustment value output from the comparing unit 113. The operating point adjustment unit 103 is connected to the absolute value calculation unit 102, the LUT reading unit 104, the comparison unit 113, and the LPF 115. The operating point adjustment unit 103 outputs a reference signal whose size is increased or decreased according to the adjustment value to the LUT reading unit 104.
　ＬＵＴ読み出し部１０４は、メモリに保存されたテーブルから参照信号の大きさに応じた補償値を読み出し、読み出した補償値を複素乗算部１０１へ出力するＬＵＴ読み出し部である。ＬＵＴ読み出し部１０４は、電力増幅器１０７の非線形特性を補償する補償テーブルが保存されるメモリを有する。ＬＵＴ読み出し部１０４は、動作点調整部１０３及び複素乗算部１０１に接続される。 The LUT reading unit 104 is a LUT reading unit that reads a compensation value corresponding to the magnitude of the reference signal from a table stored in a memory and outputs the read compensation value to the complex multiplication unit 101. The LUT reading unit 104 includes a memory in which a compensation table that compensates for nonlinear characteristics of the power amplifier 107 is stored. The LUT reading unit 104 is connected to the operating point adjustment unit 103 and the complex multiplication unit 101.
　Ｄ／Ａ変換器１０５は、入力されたディジタル信号をアナログ信号へ変換し、周波数変換回路１０６に出力する変換器である。Ｄ／Ａ変換器１０５は、複素乗算部１０１及び周波数変換回路１０６に接続される。なお、Ｄ／Ａ変換器１０５は、ＦＰＧＡ（Ｆｉｅｌｄ Ｐｒｏｇｒａｍｍａｂｌｅ Ｇａｔｅ Ａｒｒａｙ）と一体構成でも良い。 The D / A converter 105 is a converter that converts an input digital signal into an analog signal and outputs the analog signal to the frequency conversion circuit 106. The D / A converter 105 is connected to the complex multiplication unit 101 and the frequency conversion circuit 106. The D / A converter 105 may be integrated with an FPGA (Field （Programmable Gate Array).
　周波数変換回路１０６は、送信信号の周波数を変換し、周波数変換した送信信号を電力増幅器１０７に出力する周波数変換回路である。周波数変換回路１０６は、Ｄ／Ａ変換器１０５及び電力増幅器１０７に接続される。例えば、周波数変換回路１０６には、ミクサが用いられる。 The frequency conversion circuit 106 is a frequency conversion circuit that converts the frequency of the transmission signal and outputs the frequency-converted transmission signal to the power amplifier 107. The frequency conversion circuit 106 is connected to the D / A converter 105 and the power amplifier 107. For example, a mixer is used for the frequency conversion circuit 106.
　電力増幅器１０７は、送信信号の電力を増幅し、増幅した送信信号を出力端子１０８及び包絡線検出器１０９に出力する電力増幅器である。電力増幅器１０７は、周波数変換回路１０６、出力端子１０８、及び包絡線検出器１０９に接続される。例えば、電力増幅器１０７は、ＧａＡｓ（Ｇａｌｌｉｕｍ Ａｒｓｅｎｉｄｅ）増幅器、ＧａＮ（Ｇａｌｌｉｕｍ Ｎｉｔｒｉｄｅ）増幅器、Ｓｉ　ＭＯＳＥＦＥＴ（Ｍｅｔａｌ　Ｏｘｉｄｅ　Ｓｅｍｉｃｏｎｄｕｃｔｏｒ Ｆｉｅｌｄ　Ｅｆｆｅｃｔ Ｔｒａｎｓｉｓｔｏｒ）増幅器などが用いられる。 The power amplifier 107 is a power amplifier that amplifies the power of the transmission signal and outputs the amplified transmission signal to the output terminal 108 and the envelope detector 109. The power amplifier 107 is connected to the frequency conversion circuit 106, the output terminal 108, and the envelope detector 109. For example, the power amplifier 107 may be a GaAs (Gallium Narside) amplifier, a GaN (Gallium Nitride) amplifier, a Si MOSEFET (Metal Oxide Semiconductor Field Effect Transistor) amplifier, or the like.
　出力端子１０８は、電力増幅器１０７の出力信号を出力する端子である。出力端子１０８は、電力増幅器１０７、及び包絡線検出器１０９に接続される。 The output terminal 108 is a terminal that outputs an output signal of the power amplifier 107. The output terminal 108 is connected to the power amplifier 107 and the envelope detector 109.
　包絡線検出器１０９は、電力増幅器１０７の出力信号から包絡線成分を検出し、検出した包絡線成分を包絡線信号としてＬＰＦ１１０に出力する検出器である。包絡線検出器１０９は、電力増幅器１０７、出力端子１０８、及びＬＰＦ１１０に接続される。例えば、包絡線検出器１０９には、ダイオード検波器が用いられる。 The envelope detector 109 is a detector that detects an envelope component from the output signal of the power amplifier 107 and outputs the detected envelope component to the LPF 110 as an envelope signal. The envelope detector 109 is connected to the power amplifier 107, the output terminal 108, and the LPF 110. For example, a diode detector is used for the envelope detector 109.
　ＬＰＦ１１０は、電力増幅器１０７の出力信号に対し帯域制限処理を行い、帯域制限した信号をＡ／Ｄ変換器１１１へ出力するローパスフィルタである。ＬＰＦ１１０のカットオフ周波数Ｆｃは、包絡線信号の帯域幅ＢＷより小さく、Ｆｃ＜ＢＷである。ＬＰＦ１１０は、包絡線検出器１０９及びＡ／Ｄ変換器１１１に接続される。例えば、ＬＰＦ１１０には、ＣＲフィルタ、ＬＣフィルタなどが用いられる。 The LPF 110 is a low-pass filter that performs band limitation processing on the output signal of the power amplifier 107 and outputs the band-limited signal to the A / D converter 111. The cut-off frequency Fc of the LPF 110 is smaller than the bandwidth BW of the envelope signal, and Fc <BW. The LPF 110 is connected to the envelope detector 109 and the A / D converter 111. For example, a CR filter, an LC filter, or the like is used for the LPF 110.
　Ａ／Ｄ変換器１１１は、アナログの包絡線信号をディジタル信号へ変換し、ＣＣＤＦ計算部１１２に出力する変換器である。Ａ／Ｄ変換器１１１は、ＬＰＦ１１０及びＣＣＤＦ計算部１１２に接続される。なお、Ａ／Ｄ変換器１１１は、ＦＰＧＡと一体構成でも良い。 The A / D converter 111 is a converter that converts an analog envelope signal into a digital signal and outputs the digital signal to the CCDF calculation unit 112. The A / D converter 111 is connected to the LPF 110 and the CCDF calculation unit 112. The A / D converter 111 may be integrated with the FPGA.
　ＣＣＤＦ計算部１１２は、帯域制限された包絡線信号のＣＣＤＦを計算し、計算したＣＣＤＦを比較部１１３に出力するＣＣＤＦ計算部である。ＣＣＤＦ計算部１１２は、Ａ／Ｄ変換器１１１及び比較部１１３に接続される。 The CCDF calculation unit 112 is a CCDF calculation unit that calculates the CCDF of the band-limited envelope signal and outputs the calculated CCDF to the comparison unit 113. The CCDF calculation unit 112 is connected to the A / D converter 111 and the comparison unit 113.
　比較部１１３は、ＣＣＤＦ計算部１１２の出力信号（第１の分布の一例）とＣＣＤＦ計算部１１４の出力信号（第２の分布の一例）とを比較し、両者のＣＣＤＦの差に対応する誤差値を求め、求めた誤差値と事前に設定された閾値と比較する処理を行い、比較結果に応じた調整値を動作点調整部１０３に出力する比較部である。比較部１１３は、ＣＣＤＦ計算部１１２、ＣＣＤＦ計算部１１４、及び動作点調整部１０３に接続される。 The comparison unit 113 compares the output signal of the CCDF calculation unit 112 (an example of the first distribution) with the output signal of the CCDF calculation unit 114 (an example of the second distribution), and an error corresponding to the difference between the two CCDFs. It is a comparison unit that obtains a value, performs a process of comparing the obtained error value with a preset threshold value, and outputs an adjustment value according to the comparison result to the operating point adjustment unit 103. The comparison unit 113 is connected to the CCDF calculation unit 112, the CCDF calculation unit 114, and the operating point adjustment unit 103.
　ＣＣＤＦ計算部１１４は、帯域制限された参照信号のＣＣＤＦを計算し、計算したＣＣＤＦを比較部１１３に出力するＣＣＤＦ計算部である。ＣＣＤＦ計算部１１４は、ＬＰＦ１１５及び比較部１１３に接続される。 The CCDF calculation unit 114 is a CCDF calculation unit that calculates the CCDF of the band-limited reference signal and outputs the calculated CCDF to the comparison unit 113. The CCDF calculation unit 114 is connected to the LPF 115 and the comparison unit 113.
　ＬＰＦ１１５は、絶対値計算部１０２が出力する参照信号に対し帯域制限処理を行い、帯域制限した参照信号をＣＣＤＦ計算部１１４へ出力するローパスフィルタである。ＬＰＦ１１５は、ＬＰＦ１１０と同じフィルタ特性を持つことが望ましい。ＬＰＦ１１５は、絶対値計算部１０２、動作点調整部１０３、ＣＣＤＦ計算部１１４に接続される。 The LPF 115 is a low-pass filter that performs band limitation processing on the reference signal output from the absolute value calculation unit 102 and outputs the band-limited reference signal to the CCDF calculation unit 114. The LPF 115 preferably has the same filter characteristics as the LPF 110. The LPF 115 is connected to the absolute value calculation unit 102, the operating point adjustment unit 103, and the CCDF calculation unit 114.
　例えば、信号生成部１００、複素乗算部１０１、絶対値計算部１０２、動作点調整部１０３、ＬＵＴ読み出し部１０４、ＣＣＤＦ計算部１１２、比較部１１３、ＣＣＤＦ計算部１１４、及びＬＰＦ１１５は、ＦＰＧＡの論理回路、ＡＳＩＣ（Ａｐｐｌｉｃａｔｉｏｎ　Ｓｐｅｃｉｆｉｃ　Ｉｎｔｅｇｒａｔｅｄ　Ｃｉｒｃｕｉｔ）、マイコン（マイクロコンピュータ）などで構成される。また、上記の構成要素の機能は、ＦＰＧＡなどのハードウェアで実行されても良いし、メモリに記憶された各構成要素の機能を示すプログラムをプロセッサが読みだして実行するようにソフトウェアで実行されても良い。 For example, the signal generation unit 100, the complex multiplication unit 101, the absolute value calculation unit 102, the operating point adjustment unit 103, the LUT reading unit 104, the CCDF calculation unit 112, the comparison unit 113, the CCDF calculation unit 114, and the LPF 115 are FPGA logics. A circuit, an ASIC (Application Specific Integrated Circuit), a microcomputer (microcomputer), and the like are included. Further, the functions of the above constituent elements may be executed by hardware such as an FPGA, or may be executed by software so that the processor reads and executes a program indicating the function of each constituent element stored in the memory. May be.
　次に、この発明の実施の形態１に係る歪み補償回路の動作を説明する。 Next, the operation of the distortion compensation circuit according to the first embodiment of the present invention will be described.
　信号生成部１００は、時刻ｔにおいて複素数で表現される送信信号（Ｉ(t)＋ｊＱ(t)）を複素乗算部１０１及び絶対値計算部１０２に出力する。 The signal generation unit 100 outputs a transmission signal (I (t) + jQ (t)) expressed as a complex number at time t to the complex multiplication unit 101 and the absolute value calculation unit 102.
　絶対値計算部１０２は、送信信号の絶対値を計算する。時刻tにおける絶対値計算部１０２の出力信号をＭａｇ(t)とすると、Ｍａｇ(t)は以下の式で表される。 The absolute value calculation unit 102 calculates the absolute value of the transmission signal. If the output signal of the absolute value calculation unit 102 at time t is Mag (t), Mag (t) is expressed by the following equation.
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
　絶対値計算部１０２は、Ｍａｇ(t)をＬＰＦ１１５及び動作点調整部１０３の２方向に出力する。 The absolute value calculation unit 102 outputs Mag (t) in two directions, the LPF 115 and the operating point adjustment unit 103.
　まず、ＬＰＦ１１５に出力された信号の流れを説明する。ＬＰＦ１１５は、参照信号に対し帯域制限処理を行い、帯域制限した参照信号をＣＣＤＦ計算部１１４へ出力する。このとき、ＬＰＦ１１５のカットオフ周波数Ｆｃは、送信信号における包絡線成分の帯域幅ＢＷより低いため、帯域制限された参照信号の帯域幅は、送信信号における包絡線成分の帯域幅ＢＷより狭くなる。また、ＬＰＦ１１５のカットオフ周波数は、後述するＬＰＦ１１０のカットオフ周波数と同じであり、ＬＰＦ１１５のフィルタ特性とＬＰＦ１１０のフィルタ特性とは、同じであることが望ましい。 First, the flow of the signal output to the LPF 115 will be described. The LPF 115 performs band limitation processing on the reference signal and outputs the band-limited reference signal to the CCDF calculation unit 114. At this time, since the cutoff frequency Fc of the LPF 115 is lower than the bandwidth BW of the envelope component in the transmission signal, the bandwidth of the band-limited reference signal becomes narrower than the bandwidth BW of the envelope component in the transmission signal. Further, the cutoff frequency of the LPF 115 is the same as the cutoff frequency of the LPF 110 described later, and the filter characteristics of the LPF 115 and the filter characteristics of the LPF 110 are desirably the same.
　ＣＣＤＦ計算部１１４は、ＬＰＦ１１５の出力信号を用いてＣＣＤＦを計算し、計算結果 Ｒｅｆ_ＣＣＤＦ（ｎ）を比較部１１３へ出力する。ここで、ｎは、計算したＣＣＤＦのデータ番号を示す。 The CCDF calculation unit 114 calculates the CCDF using the output signal of the LPF 115 and outputs the calculation result Ref_CCDF (n) to the comparison unit 113. Here, n represents the data number of the calculated CCDF.
　次に、動作点調整部１０３に出力された信号の流れを説明する。動作点調整部１０３は、Ｍａｇ(t)に対して、比較部１１３が出力した調整値Ｇ（ｉ）を用いて次式で表す演算を行い、演算結果をＬＵＴ読み出し部１０４へ出力する。ここで、Ｇ（ｉ）は利得に対応する。ｉは比較部１１３での比較回数を表し、０以上の自然数である。なお、Ｇ（ｉ）の初期値は、Ｇ（０）である。動作点調整部１０３は、調整値の初期値としてＧ（０）を用いる。 Next, the flow of signals output to the operating point adjustment unit 103 will be described. The operating point adjustment unit 103 performs an operation represented by the following expression on the Mag (t) using the adjustment value G (i) output from the comparison unit 113, and outputs the operation result to the LUT reading unit 104. Here, G (i) corresponds to the gain. i represents the number of comparisons in the comparison unit 113, and is a natural number of 0 or more. Note that the initial value of G (i) is G (0). The operating point adjustment unit 103 uses G (0) as the initial value of the adjustment value.
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000002
　なお、上式は、真値による記載であり、対数表示ではＧ（ｉ）とＭａｇ（ｔ）との足し算の形で表される。したがって、対数表示で考えると、ＬＵＴＩｎｄｅｘ（ｔ）は、Ｍａg（ｔ）の動作点が、Ｇ（ｉ）分、シフトした値になる。つまり、動作点調整部１０３は、Ｍａｇ（ｔ）に対する利得を変化させることで、Ｍａg（ｔ）の動作点を調整する。 Note that the above expression is a description by a true value, and is expressed in the form of addition of G (i) and Mag (t) in logarithmic display. Therefore, when considering logarithmic display, LUT Index (t) is a value obtained by shifting the operating point of Mag (t) by G (i). That is, the operating point adjustment unit 103 adjusts the operating point of Mag (t) by changing the gain with respect to Mag (t).
　ＬＵＴ読み出し部１０４は、メモリに保存された補償テーブルからＬＵＴＩｎｄｅｘ (t)に応じた補償係数を読み出し、複素乗算部１０１へ出力する。ここで、補償係数は、ＣｏｍｐＩ（ｔ）、ＣｏｍｐＱ（ｔ）である。なお、補償テーブルから補償係数を読み出す際に、テーブルの値を内挿、または関数フイッテイングして値を求め、その値を読み出すようにしても良い。 The LUT reading unit 104 reads a compensation coefficient corresponding to the LUT Index (t) from the compensation table stored in the memory, and outputs the compensation coefficient to the complex multiplication unit 101. Here, the compensation coefficients are CompI (t) and CompQ (t). Note that when reading the compensation coefficient from the compensation table, the value may be obtained by interpolation or function fitting of the table value, and the value may be read out.
　図２は、この発明の実施の形態１に係る補償テーブルの一例を示す図である。
　１列目がＬＵＴＩｎｄｅｘ であり、２列目がＣｏｍｐＩであり、３列目がＣｏｍｐＱである。図２において、ＬＵＴＩｎｄｅｘの値は不連続であるが、これは、補償テーブルの全体を表示するためにデータを間引いで表示しているためである。実際は、ＬＵＴＩｎｄｅｘの値が連続の補償テーブルである。ここで、補償係数（ＣｏｍｐＩ及びＣｏｍｐＱ）は、電力増幅器１０７の非線形性特性を補償する値である。ＬＵＴＩｎｄｅｘ は、入力電力の真値に対応している。表の形では分かりにくいので、振幅特性及び位相特性の形で補償テーブルを図示したグラフを次に示す。振幅は、√（ＣｏｍｐＩ２＋ＣｏｍｐＱ２）であり、位相は、ｔａｎ－１（ＣｏｍｐＱ／ＣｏｍｐＩ）である。 FIG. 2 is a diagram showing an example of a compensation table according to Embodiment 1 of the present invention.
The first column is LUT Index , the second column is CompI, and the third column is CompQ. In FIG. 2, the value of the LUT Index is discontinuous because the data is displayed by thinning to display the entire compensation table. Actually, the LUT Index value is a continuous compensation table. Here, the compensation coefficients (CompI and CompQ) are values that compensate for the nonlinear characteristic of the power amplifier 107. The LUT Index corresponds to the true value of the input power. Since it is difficult to understand in the form of a table, a graph illustrating a compensation table in the form of amplitude characteristics and phase characteristics is shown below. The amplitude is √ (CompI 2 + CompQ 2 ), and the phase is tan −1 (CompQ / CompI).
　図３は、この発明の実施の形態１に係る補償テーブルを振幅特性及び位相特性の形式で表示した図である。
　実線が振幅特性であり、破線が位相特性である。図３に示す振幅特性及び位相特性は、電力増幅器１０７の振幅特性及び位相特性の逆特性になっている。 FIG. 3 is a diagram showing the compensation table according to the first embodiment of the present invention in the form of amplitude characteristics and phase characteristics.
The solid line is the amplitude characteristic, and the broken line is the phase characteristic. The amplitude characteristics and phase characteristics shown in FIG. 3 are inverse characteristics of the amplitude characteristics and phase characteristics of the power amplifier 107.
　複素乗算部１０１は、信号生成部１００の出力信号Ｉ(t)＋ｊＱ(t)と、ＬＵＴ読み出し部１０４の出力信号ＣｏｍｐＩ（ｔ）＋ｊＣｏｍｐＱ（ｔ）を次式で表すように乗算し、乗算結果（Ｉ’（ｔ））＋ｊＱ’（ｔ））をＤ／Ａ変換器１０５へ出力する。なお、複素乗算部１０１が送信信号に電力増幅器１０７の補償係数を乗算することをプレディストーションという。 The complex multiplication unit 101 multiplies the output signal I (t) + jQ (t) of the signal generation unit 100 and the output signal CompI (t) + jCompQ (t) of the LUT reading unit 104 as represented by the following expression, and the multiplication result (I ′ (t)) + jQ ′ (t)) is output to the D / A converter 105. The complex multiplication unit 101 multiplying the transmission signal by the compensation coefficient of the power amplifier 107 is called predistortion.
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000003
　Ｄ／Ａ変換器１０５は、補償係数が複素乗算された送信信号をアナログ信号に変換し、周波数変換回路１０６へ出力する。 The D / A converter 105 converts the transmission signal obtained by complex multiplication of the compensation coefficient into an analog signal and outputs the analog signal to the frequency conversion circuit 106.
　周波数変換回路１０６は、入力された送信信号の周波数を変換し、周波数変換した送信信号を電力増幅器１０７へ入力する。 The frequency conversion circuit 106 converts the frequency of the input transmission signal and inputs the frequency-converted transmission signal to the power amplifier 107.
　電力増幅器１０７は、送信信号の電力を増幅し、増幅した送信信号を出力端子１０８へ出力するとともに、包絡線検出器１０９へ出力する。このとき、電力増幅器１０７の非線形特性により電力増幅器１０７の出力信号には歪み成分が含まれる。 The power amplifier 107 amplifies the power of the transmission signal and outputs the amplified transmission signal to the output terminal 108 and also to the envelope detector 109. At this time, due to the nonlinear characteristic of the power amplifier 107, the output signal of the power amplifier 107 includes a distortion component.
　包絡線検出器１０９は、電力増幅器１０７の出力信号から包絡線成分（包絡線信号）を検出し、検出した包絡線信号をＬＰＦ１１０に出力する。 The envelope detector 109 detects an envelope component (envelope signal) from the output signal of the power amplifier 107 and outputs the detected envelope signal to the LPF 110.
　ＬＰＦ１１０は、包絡線信号に対し帯域制限処理を行い、処理後の包絡線信号をＡ／Ｄ変換器１１１へ出力する。このとき、ＬＰＦ１１０のカットオフ周波数Ｆｃは、送信信号から検出した包絡線信号の帯域幅ＢＷより低いため、帯域制限された包絡線信号の帯域幅は、ＢＷより狭くなる。 The LPF 110 performs band limiting processing on the envelope signal, and outputs the processed envelope signal to the A / D converter 111. At this time, since the cutoff frequency Fc of the LPF 110 is lower than the bandwidth BW of the envelope signal detected from the transmission signal, the bandwidth of the band-limited envelope signal is narrower than BW.
　Ａ／Ｄ変換器１１１は、帯域制限された包絡線信号をディジタイズ（デジタル化）する。 The A / D converter 111 digitizes (digitizes) the band-limited envelope signal.
　図４は、この発明の実施の形態１に係るＬＰＦ１１０の周波数特性と包絡線信号の帯域幅との関係を示す図である。
　包絡線信号には、周波数ＢＷまでの信号成分と信号成分の帯域外にある周波数ＦＤまでの歪み成分とが含まれており、ＬＰＦ１１０がない場合、これらすべての成分が高速に動作するＡ／Ｄ変換器１１１によりディジタイズされる。 FIG. 4 is a diagram showing the relationship between the frequency characteristic of the LPF 110 according to Embodiment 1 of the present invention and the bandwidth of the envelope signal.
The envelope signal includes a and distortion components up to a frequency F D, which is outside the band of the signal component and the signal component to frequency BW, if there is no LPF 110, all of the components operate at high speed A / Digitized by the D converter 111.
　したがって、ＬＰＦ１１０が無い場合は、包絡線信号の信号成分と包絡線信号の歪み成分をディジタイズするために、Ａ／Ｄ変換器の動作速度Ｆｓは、以下の条件を満たす必要がある。 Therefore, when the LPF 110 is not provided, the operation speed Fs of the A / D converter needs to satisfy the following conditions in order to digitize the signal component of the envelope signal and the distortion component of the envelope signal.
Figure JPOXMLDOC01-appb-M000004
Figure JPOXMLDOC01-appb-M000004
　一方、本構成では、信号帯域幅ＢＷより低い周波数であるＦｃをカットオフとするＬＰＦ１１０により、周波数Ｆｃより大きな周波数範囲にある成分は除去され、０からＦｃまでの周波数範囲にある成分がディジタイズされる。 On the other hand, in this configuration, the LPF 110 that cuts off Fc, which is a frequency lower than the signal bandwidth BW, removes components in the frequency range higher than the frequency Fc, and digitizes components in the frequency range from 0 to Fc. The
　したがって、本構成では、ＬＰＦ１１０を挿入することにより、ディジタイズする信号の周波数範囲を狭くできるため、Ａ／Ｄ変換器の動作速度Ｆｓは、少なくとも次式を満たせば良い。 Therefore, in this configuration, since the frequency range of the signal to be digitized can be narrowed by inserting the LPF 110, the operation speed Fs of the A / D converter should satisfy at least the following equation.
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000005
　このように、ＬＰＦ１１０を設けることにより、Ａ／Ｄ変換器１１１の動作速度を低速化させ、消費電力を低減できる。 Thus, by providing the LPF 110, the operation speed of the A / D converter 111 can be reduced and the power consumption can be reduced.
　Ａ／Ｄ変換器１１１は、ディジタイズした信号をＣＣＤＦ計算部１１２へ出力する。 The A / D converter 111 outputs the digitized signal to the CCDF calculation unit 112.
　以下、フローチャートを参照しながらディジタル部分の動作を説明する。
　図５は、この発明の実施の形態１に係る歪み補償回路におけるディジタル部分の動作の一例を示すフローチャートである。 The operation of the digital part will be described below with reference to the flowchart.
FIG. 5 is a flowchart showing an example of the operation of the digital portion in the distortion compensation circuit according to Embodiment 1 of the present invention.
　まず、ステップＳ１０１において、ＣＣＤＦ計算部１１２は、Ａ／Ｄ変換器１１１の出力信号を用いてＣＣＤＦを計算し、計算結果Ｔｅｓｔ_ＣＣＤＦ（ｎ）を比較部１１３へ出力する。 First, in step S 101, the CCDF calculation unit 112 calculates CCDF using the output signal of the A / D converter 111 and outputs the calculation result Test_CCDF (n) to the comparison unit 113.
　ここで、帯域制限した場合と帯域制限しない場合とにおける包絡線信号及び参照信号の違いを説明する。
　図６は、この発明の実施の形態１に係る帯域制限がない場合における包絡線信号及び参照信号の波形を示す信号波形図である。
　ここで、ＦＳ＿ＯＬＤは、Ａ／Ｄ変換器１１１のサンプリング周波数であり、ＮＯＬＤは、ＣＣＤＦを求めるときのデータ点数である。ΔＰは、参照信号と包絡線信号とのピーク差である。
　帯域制限が施されない場合、Ａ／Ｄ変換器１１１は高速に動作するため、包絡線信号のピーク値をディジタイズすることができる。包絡線信号には、電力増幅器１０７で発生した歪み成分が含まれており、ピーク値近妨には特に歪み成分が含まれる。一方、参照信号は、電力増幅器１０７を通過しないため、歪み成分は含まれていない。そのため、参照信号のＣＣＤＦ値と包絡線信号のＣＣＤＦ値とを比較すると、両者の間に誤差がある。 Here, the difference between the envelope signal and the reference signal when the band is limited and when the band is not limited will be described.
FIG. 6 is a signal waveform diagram showing waveforms of the envelope signal and the reference signal when there is no band limitation according to Embodiment 1 of the present invention.
Here, F S_OLD is the sampling frequency of the A / D converter 111, N OLD is the number of data points when determining the CCDF. ΔP is a peak difference between the reference signal and the envelope signal.
When the band limitation is not performed, the A / D converter 111 operates at high speed, so that the peak value of the envelope signal can be digitized. The envelope signal includes a distortion component generated by the power amplifier 107, and the peak value close interference particularly includes a distortion component. On the other hand, since the reference signal does not pass through the power amplifier 107, a distortion component is not included. Therefore, when the CCDF value of the reference signal and the CCDF value of the envelope signal are compared, there is an error between the two.
　図７は、この発明の実施の形態１に係る帯域制限がない場合における包絡線信号及び参照信号のＣＣＤＦを示す図である。
　縦軸が発生確率であり、横軸が振幅である。発生確率が低い領域で参照信号と包絡線信号とのＣＣＤＦの差は、図６のピーク値の差に対応しており、ΔＰである。 FIG. 7 is a diagram showing the CCDF of the envelope signal and the reference signal when there is no band limitation according to Embodiment 1 of the present invention.
The vertical axis is the occurrence probability, and the horizontal axis is the amplitude. The difference in CCDF between the reference signal and the envelope signal in the region where the occurrence probability is low corresponds to the difference in peak value in FIG. 6 and is ΔP.
　図８は、この発明の実施の形態１に係る帯域制限がある場合における包絡線信号及び参照信号の波形を示す信号波形図である。
　ここで、ＦＳ＿ＮＥＷは、Ａ／Ｄ変換器のサンプリング周波数であり、ＮＮＥＷは、ＣＣＤＦを求めるときのデータ点数である。ΔＰＮＥＷは、参照信号と包絡線信号とのピーク差である。
　ＬＰＦ１１０及びＬＰＦ１１５による帯域制限がされることにより、振幅の変化の速度が遅くなるが、Ａ／Ｄ変換器の動作速度も低速になっているため、包絡線信号のピーク値をディジタイズすることができる。ＬＰＦ１１０により包絡線信号に帯域制限が行われても、０からＦｃまでの周波数範囲にも歪み成分が存在するため、ＣＣＤＦを計算することにより、参照信号と包絡線信号との間の誤差を検出することができる。包絡線信号のＢＷ内に歪み成分が含まれることは、図８において、帯域制限されても振幅が大きい領域の包絡線信号は電力増幅器１０７の非線形領域の信号であるため、参照信号と包絡線信号との間に差が生じることから理解される。ＢＷ内に歪み成分が含まれる理由をもう少し説明する。電力増幅器１０７の非線形特性をｙ＝ａ・ｘ＋ｂ・ｘ３と仮定し、ｘ＝ｃｏｓ（ωｔ）を代入すると、ｙ＝（ａ－３ｂ）・ｃｏｓ（ωｔ）＋４ｂ・ｃｏｓ（３ωｔ）となる。ここで、ａは、線形特性を示す係数である。ｂは、非線形特性を示す係数であり、歪みに関係する。（ａ－３ｂ）・ｃｏｓ（ωｔ）が信号帯域内（ＢＷ内）の信号成分に対応し、４ｂ・ｃｏｓ（３ωｔ）が歪み帯域内（ＦＤ内）の歪み成分に対応する。歪み帯域の成分を除去しても、信号成分にｂが含まれるため、信号帯域内の信号は、非線形特性の影響を受ける。このように、ＬＰＦ１１０が帯域制限を行っても、包絡線信号には歪み成分が含まれる。 FIG. 8 is a signal waveform diagram showing waveforms of the envelope signal and the reference signal when there is a band limitation according to Embodiment 1 of the present invention.
Here, F S_NEW is the sampling frequency of the A / D converter, and N NEW is the number of data points when CCDF is obtained. ΔP NEW is the peak difference between the reference signal and the envelope signal.
The band is limited by the LPF 110 and the LPF 115, so that the speed of the amplitude change becomes slow, but the operation speed of the A / D converter is also slow, so that the peak value of the envelope signal can be digitized. . Even if the band limit is applied to the envelope signal by the LPF 110, there is also a distortion component in the frequency range from 0 to Fc, so the error between the reference signal and the envelope signal is detected by calculating the CCDF. can do. The fact that the distortion component is included in the BW of the envelope signal is that the envelope signal in the region where the amplitude is large even if the band is limited in FIG. 8 is a signal in the nonlinear region of the power amplifier 107. It is understood from the difference between the signal and the signal. The reason why the distortion component is included in BW will be explained a little more. Assuming that the nonlinear characteristic of the power amplifier 107 is y = a · x + b · x 3 and substituting x = cos (ωt), y = (a−3b) · cos (ωt) + 4b · cos (3ωt). Here, a is a coefficient indicating a linear characteristic. b is a coefficient indicating nonlinear characteristics, and is related to distortion. (A-3b) · cos ( ωt) corresponds to the signal component in the signal band (the BW), 4b · cos (3ωt ) corresponds to the distortion component in the distortion-band (in F D). Even if the distortion band component is removed, since the signal component includes b, the signal in the signal band is affected by the nonlinear characteristic. As described above, even if the LPF 110 performs band limitation, the envelope signal includes a distortion component.
　図９は、この発明の実施の形態１に係る帯域制限がある場合における包絡線信号及び参照信号のＣＣＤＦを示す図である。
　縦軸が発生確率であり、横軸が振幅である。発生確率が低い領域で参照信号と包絡線信号とのＣＣＤＦの差は、図８のピーク値の差に対応しており、ΔＰＮＥＷである。 FIG. 9 is a diagram showing the CCDF of the envelope signal and the reference signal when there is a band limitation according to the first embodiment of the present invention.
The vertical axis is the occurrence probability, and the horizontal axis is the amplitude. The difference in CCDF between the reference signal and the envelope signal in the region where the occurrence probability is low corresponds to the difference in peak value in FIG. 8, and is ΔP NEW .
　ここで、ＣＣＤＦを計算するために必要なデータ点数Ｎは、Ａ／Ｄ変換器１１１のサンプリング周波数ＦｓとＬＰＦ１１０のカットオフ周波数Ｆｃとを用いて、少なくとも以下の式を満たす必要がある。式（６）においてＦｓが大きいほど、必要なデータ点数が多くなるが、これは、Ｆｓが大きい場合、サンプリング間隔が狭くなるため、データ点数が多くしないと、Ｆｃで帯域制限された包絡線信号のピーク付近の値をサンプリングできないからである。なお、Ｆｓは、Ａ／Ｄ変換器１１１の動作速度に対応する。 Here, the number N of data points necessary for calculating the CCDF needs to satisfy at least the following expression using the sampling frequency Fs of the A / D converter 111 and the cut-off frequency Fc of the LPF 110. In Formula (6), the larger the number of data points, the larger the required number of data points. However, when Fs is large, the sampling interval is narrowed. Therefore, if the number of data points is not increased, the envelope signal band-limited by Fc. This is because the value near the peak cannot be sampled. Note that Fs corresponds to the operation speed of the A / D converter 111.
Figure JPOXMLDOC01-appb-M000006
Figure JPOXMLDOC01-appb-M000006
　例えば、ＣＣＤＦ計算部１１２及びＣＣＤＦ計算部１１４は、式（６）を満たしたうえで、以下のようにデータ点数Ｎを決定する。
　図１０は、この発明の実施の形態１に係る規格化データ点数と歪み補償量の劣化量との関係を示す図である。
　横軸は、規格化データ点数（Ｎ／（Ｆｓ／Ｆｃ））であり、縦軸は、歪み補償量の劣化量である。規格化データ点数とは、データ点数ＮをＦｓ／Ｆｃで規格化したデータ点数である。歪み補償量の劣化量とは、理想的に歪み補償がかかった場合、つまり信号の歪みがゼロの場合からの劣化量を示す。理想的に歪み補償がかかった場合、歪み補償量の劣化量はゼロになる。 For example, the CCDF calculation unit 112 and the CCDF calculation unit 114 satisfy the equation (6) and determine the number N of data points as follows.
FIG. 10 is a diagram showing the relationship between the number of normalized data points and the amount of distortion compensation deterioration according to Embodiment 1 of the present invention.
The horizontal axis represents the number of normalized data points (N / (Fs / Fc)), and the vertical axis represents the deterioration amount of the distortion compensation amount. The normalized data score is a data score obtained by normalizing the data score N with Fs / Fc. The deterioration amount of the distortion compensation amount indicates an amount of deterioration from when the distortion compensation is ideally applied, that is, when the signal distortion is zero. When distortion compensation is ideally applied, the degradation amount of the distortion compensation amount becomes zero.
　図１０において、規格化データ点数が８０００を下回ると、歪み補償量の劣化量は、線形関係からはずれ、指数関数的に大きくなっている。これは、規格化データ点数が小さいと、包絡線信号のピーク付近の値をサンプリングする回数が少なくなるため、歪みが大きい領域で歪み補償の精度が劣化するためである。
　ＣＣＤＦ計算部１１２及びＣＣＤＦ計算部１１４は、規格化データ点数が８０００以上となるように規格化データ点数を決定し、規格化データ点数にＦｃ／Ｆｓを乗算し、データ点数Ｎを決定する。なお、規格化データ点数８０００は、歪み補償量の劣化量０．２ｄＢに対応するため、歪み補償量の劣化量が０．２ｄＢ以内になるように、規格化データ点数を決定し、データ点数Ｎを決定しても良い。また、ＣＣＤＦの誤差を基準にデータ点数Ｎを決定しても良い。 In FIG. 10, when the number of normalized data points is less than 8000, the deterioration amount of the distortion compensation amount deviates from the linear relationship and increases exponentially. This is because, when the number of normalized data points is small, the number of times of sampling the value near the peak of the envelope signal is small, and the accuracy of distortion compensation deteriorates in a region where the distortion is large.
The CCDF calculation unit 112 and the CCDF calculation unit 114 determine the normalized data score so that the normalized data score is 8000 or more, multiply the normalized data score by Fc / Fs, and determine the data score N. Since the normalized data point 8000 corresponds to the distortion compensation amount deterioration amount 0.2 dB, the normalized data point number is determined so that the distortion compensation amount deterioration amount is within 0.2 dB, and the data point number N May be determined. The number N of data points may be determined based on the CCDF error.
　以下、フローチャートの説明に戻る。
　ステップＳ１０２において、比較部１１３は、ＣＣＤＦ計算部１１２が計算した包絡線信号のＣＣＤＦと、ＣＣＤＦ計算部１１４が計算した参照信号のＣＣＤＦとを比較し、両者のＣＣＤＦの差（誤差）を算出する。例えば、次式で表す演算を行い、誤差Ｅｒｒを求める。 Hereinafter, the description returns to the flowchart.
In step S102, the comparison unit 113 compares the CCDF of the envelope signal calculated by the CCDF calculation unit 112 with the CCDF of the reference signal calculated by the CCDF calculation unit 114, and calculates a difference (error) between the CCDFs. . For example, the calculation represented by the following equation is performed to obtain the error Err.
Figure JPOXMLDOC01-appb-M000007
Figure JPOXMLDOC01-appb-M000007
　ここで、ＭはＣＣＤＦ値の全データ数である。 Where M is the total number of CCDF values.
　比較部１１３は、Ｅｒｒの値を予め設定された閾値ＶＴＨと比較する。 The comparison unit 113 compares the value of Err with a preset threshold value VTH .
Figure JPOXMLDOC01-appb-M000008
Figure JPOXMLDOC01-appb-M000008
　上式を満たす場合、比較部１１３は、ステップＳ１０３において、動作点調整部１０３に対する調整値（利得）を更新し、ΔＧだけ低下させる。即ち、更新回数をｉ、更新前の調整値をＧ（ｉ）、更新後の調整値をＧ（ｉ＋１）とすると、以下の式のように更新する。ここで、ｉは比較部１１３での比較回数を表し、０以上の自然数である。 If the above equation is satisfied, the comparison unit 113 updates the adjustment value (gain) for the operating point adjustment unit 103 and decreases it by ΔG in step S103. That is, if the number of updates is i, the adjustment value before update is G (i), and the adjustment value after update is G (i + 1), the update is performed as shown in the following equation. Here, i represents the number of comparisons in the comparison unit 113, and is a natural number of 0 or more.
Figure JPOXMLDOC01-appb-M000009
Figure JPOXMLDOC01-appb-M000009
Figure JPOXMLDOC01-appb-M000010
Figure JPOXMLDOC01-appb-M000010
　一方、ＥｒｒとＶＴＨとが上式を満たす場合、比較部１１３は、ステップＳ１０４において、動作点調整部１０３に対する調整値を更新し、ΔGだけ増加させる。即ち、比較部１１３は、以下の式のように調整値を更新する。 On the other hand, when Err and V TH satisfy the above equation, the comparison unit 113 updates the adjustment value for the operating point adjustment unit 103 and increases it by ΔG in step S104. That is, the comparison unit 113 updates the adjustment value as in the following equation.
Figure JPOXMLDOC01-appb-M000011
Figure JPOXMLDOC01-appb-M000011
　このように、動作点調整部１０３は、参照信号に対する利得を更新することで、参照信号の電力を増減し、参照信号の動作点を調整する。 As described above, the operating point adjusting unit 103 adjusts the operating point of the reference signal by increasing or decreasing the power of the reference signal by updating the gain with respect to the reference signal.
Figure JPOXMLDOC01-appb-M000012
Figure JPOXMLDOC01-appb-M000012
　ステップＳ１０２において、ＥｒｒとＶＴＨとが上式を満たす場合、比較部１１３は、動作点調整部１０３に対する調整値を変化させず、処理は終了する。即ち、この場合の調整値は、以下の式になる。 In step S102, if the the Err and V TH satisfy the above equation, the comparing unit 113 does not change the adjustment value for the operating point adjustment unit 103, the process ends. That is, the adjustment value in this case is as follows.
Figure JPOXMLDOC01-appb-M000013
Figure JPOXMLDOC01-appb-M000013
　上記で述べたように、比較部１１３は、ＥｒｒとＶＴＨとの関係で場合分けを行い、調整値Ｇ（ｉ＋１）を動作点調整部１０３へ出力する。なお、ΔＧは一定値でなく、比較回数により可変にしても良いし、Ｅｒｒに比例する値にしても良い。 As described above, the comparison unit 113 performs case analysis in relation to the Err and V TH, and outputs an adjustment value G (i + 1) to the operating point adjustment unit 103. Note that ΔG may not be a constant value, but may be variable depending on the number of comparisons, or may be a value proportional to Err.
　次に、ステップＳ１０５において、動作点調整部１０３は、調整値Ｇ（ｉ＋１）と式（２）とから、ＬＵＴＩｎｄｅｘ（ｔ）を計算し、ＬＵＴＩｎｄｅｘ（ｔ）をＬＵＴ読み出し部１０４に出力する。 Next, in step S <b> 105, the operating point adjustment unit 103 calculates LUT Index (t) from the adjustment value G (i + 1) and Expression (2), and outputs the LUT Index (t) to the LUT reading unit 104. .
　次に、ステップＳ１０６において、ＬＵＴ読み出し部１０４は、メモリに保存された補償テーブルからＬＵＴＩｎｄｅｘ (t)に応じた補償係数を読み出し、複素乗算部１０１へ出力する。 Next, in step S106, the LUT reading unit 104 reads a compensation coefficient corresponding to the LUT Index (t) from the compensation table stored in the memory, and outputs the compensation coefficient to the complex multiplication unit 101.
　次に、ステップＳ１０７において、複素乗算部１０１は、信号生成部１００の出力信号Ｉ(t)＋ｊＱ(t)と、ＬＵＴ読み出し部１０４の出力する補償係数とを乗算し、乗算結果をＤ／Ａ変換器１０５へ出力する。 Next, in step S107, the complex multiplication unit 101 multiplies the output signal I (t) + jQ (t) of the signal generation unit 100 by the compensation coefficient output from the LUT reading unit 104, and the multiplication result is D / A. Output to the converter 105.
　本歪み補償回路は、以下、式（１２）が満たされるまで、上で説明した一連のステップ（Ｓ１０１～Ｓ１０７）を繰り返す。式（１２）を満たした場合、図５のフローチャートの処理は終了する。これにより、本歪み補償回路は、参照信号と包絡線信号とのＣＣＤＦの差（Ｅｒｒ）が最小になるように動作点調整部の１０３の調整値を更新することができる。よって、本歪み補償回路は、電力増幅器１０７で生じる歪みを小さくすることができる。 The distortion compensation circuit repeats the series of steps (S101 to S107) described above until Expression (12) is satisfied. When Expression (12) is satisfied, the process of the flowchart in FIG. 5 ends. As a result, the distortion compensation circuit can update the adjustment value of the operating point adjustment unit 103 so that the difference (Err) of the CCDF between the reference signal and the envelope signal is minimized. Therefore, this distortion compensation circuit can reduce the distortion generated in the power amplifier 107.
　以上のように、実施の形態１の歪み補償回路によれば、送信信号の信号帯域幅が広い場合でもＬＰＦ１１０により帯域制限を行い、参照信号と包絡線信号との比較にＣＣＤＦを用いるので、Ａ／Ｄ変換器１１１の動作速度を抑制しつつ、電力増幅器１０７の歪み補償を行うことができる。 As described above, according to the distortion compensation circuit of the first embodiment, even when the signal bandwidth of the transmission signal is wide, the band is limited by the LPF 110 and the CCDF is used to compare the reference signal and the envelope signal. The distortion of the power amplifier 107 can be compensated while suppressing the operation speed of the / D converter 111.
　なお、本実施の形態１では、参照信号と包絡線信号とを比較する分布にＣＣＤＦを用いたが、本発明はこれに限られるものではなく、ＰＤＦ（Ｐｒｏｂａｂｉｌｉｔｙ Ｄｅｎｓｉｔｙ Ｆｕｎｃｔｉｏｎ）やＣＤＦ（Ｃｕｍｕｌａｔｉｖｅ Ｄｉｓｔｒｉｂｕｔｉｏｎ Ｆｕｎｃｔｉｏｎ）などのように、信号の瞬時電力の確率分布または度数分布を示す分布であれば、どのような分布を用いても良い。 In the first embodiment, the CCDF is used for the distribution for comparing the reference signal and the envelope signal. However, the present invention is not limited to this, and PDF (Probability Density Function) or CDF (Cumulative Distribution Function) Any distribution may be used as long as it is a distribution indicating the probability distribution or frequency distribution of the instantaneous power of the signal.
実施の形態２.
　図１１は、この発明の実施の形態２に係る歪み補償回路の一構成例を示す構成図である。図１１において、図１と同一符号は、同一または相当部分を示している。このため、同一または相当部分は、説明を省略する。図１とは、Ａ／Ｄ変換器１１１とＣＣＤＦ計算部との間にＬＰＦ２００（第３のフィルタの一例）が設けられている点が異なる。 Embodiment 2.
FIG. 11 is a configuration diagram showing a configuration example of a distortion compensation circuit according to the second embodiment of the present invention. 11, the same reference numerals as those in FIG. 1 denote the same or corresponding parts. For this reason, the description of the same or corresponding parts is omitted. 1 is different from FIG. 1 in that an LPF 200 (an example of a third filter) is provided between the A / D converter 111 and the CCDF calculation unit.
　ＬＰＦ２００は、Ａ／Ｄ変換器１１１の後段に配置され、ＬＰＦ１１０で帯域制限された包絡線信号をさらに帯域制限するローパスフィルタである。ＬＰＦ２００は、Ａ／Ｄ変換器１１１が出力する包絡線信号を帯域制限し、帯域制限した包絡線信号をＣＣＤＦ計算部１１２に出力する。ここで、Ａ／Ｄ変換器１１１が出力する包絡線信号は、ディジタル信号であり、ＬＰＦ２００は、ディジタルフィルタである。ＬＰＦ２００のカットオフ周波数は、ＬＰＦ１１０より小さく、かつＬＰＦ１１５のカットオフ周波数と同じである。また、ＬＰＦ２００は、ＬＰＦ１１５と同じフィルタ特性を持つ。例えば、ＬＰＦ２００は、ＦＰＧＡの論理回路、ＡＳＩＣ、マイコンなどで構成される。 LPF 200 is a low-pass filter that is arranged downstream of A / D converter 111 and further band-limits the envelope signal band-limited by LPF 110. The LPF 200 band-limits the envelope signal output from the A / D converter 111 and outputs the band-limited envelope signal to the CCDF calculation unit 112. Here, the envelope signal output from the A / D converter 111 is a digital signal, and the LPF 200 is a digital filter. The cut-off frequency of the LPF 200 is smaller than that of the LPF 110 and is the same as the cut-off frequency of the LPF 115. The LPF 200 has the same filter characteristics as the LPF 115. For example, the LPF 200 includes an FPGA logic circuit, an ASIC, a microcomputer, and the like.
　次に、この発明の実施の形態２に係る歪み補償回路の動作について説明する。実施の形態１と同様の動作は説明を省略し、実施の形態１と異なる動作を説明する。 Next, the operation of the distortion compensation circuit according to the second embodiment of the present invention will be described. Description of operations similar to those of the first embodiment will be omitted, and operations different from those of the first embodiment will be described.
　実施の形態２においてＡ/Ｄ変換器１１１までの動作は、実施の形態１と同じである。Ａ/Ｄ変換器１１１は、ＬＰＦ１１０で帯域制限された包絡線信号をディジタイズし、ディジタイズした包絡線信号をＬＰＦ２００に出力する。 In the second embodiment, the operations up to the A / D converter 111 are the same as those in the first embodiment. The A / D converter 111 digitizes the envelope signal band-limited by the LPF 110, and outputs the digitized envelope signal to the LPF 200.
　ＬＰＦ２００は、ＬＰＦ１１０のカットオフ周波数より低いカットオフ周波数をもつので、ＬＰＦ１１０が帯域制限した包絡線信号をさらに帯域制限する。つまり、ＬＰＦ１１５のカットオフ周波数Ｆｃ３と、ＬＰＦ１１０のカットオフ周波数Ｆｃ２とは以下の関係にある Since the LPF 200 has a cutoff frequency lower than the cutoff frequency of the LPF 110, it further limits the band of the envelope signal band-limited by the LPF 110. That is, the cutoff frequency Fc3 of the LPF 115 and the cutoff frequency Fc2 of the LPF 110 are in the following relationship.
Figure JPOXMLDOC01-appb-M000014
Figure JPOXMLDOC01-appb-M000014
　ＬＰＦ２００は、帯域制限した包絡線信号をＣＣＤＦ計算部１１２に出力する。このとき、ＬＰＦ２００は、ＬＰＦ１１５と同じカットオフ周波数を持ち、フィルタ特性も同じであるため、ＬＰＦ２００が出力する包絡線信号の帯域は、ＬＰＦ１１５が出力する参照信号の帯域と同じになる。仮に、ＬＰＦ２００とＬＰＦ１１５に入力される信号が同じであれば、ＬＰＦ２００とＬＰＦ１１５とから出力される信号は同じである。 LPF 200 outputs a band-limited envelope signal to CCDF calculation unit 112. At this time, since the LPF 200 has the same cutoff frequency as the LPF 115 and the filter characteristics are the same, the band of the envelope signal output from the LPF 200 is the same as the band of the reference signal output from the LPF 115. If the signals input to the LPF 200 and the LPF 115 are the same, the signals output from the LPF 200 and the LPF 115 are the same.
　ＣＣＤＦ計算部１１２以降の動作は、実施の形態１と同じであるため、説明を省略する。 Since the operation after the CCDF calculation unit 112 is the same as that of the first embodiment, the description thereof is omitted.
　実施の形態２では、ＬＰＦ１１５及びＬＰＦ２００はディジタルフィルタであるため、同一のフィルタ特性が実現可能である。これにより、ＣＣＤＦ計算部１１２及びＣＣＤＦ計算部１１４のＣＣＤＦを計算する際に、フィルタ特性の差により計算するＣＣＤＦに誤差が生じることがない。したがって、比較部１１３において、両者のＣＣＤＦを比較するときに、ＣＣＤＦの計算誤差が生じないので、フィルタ特性の差により動作点調整部１０３の動作点の最適値がずれることを防ぐことができる。 In Embodiment 2, since the LPF 115 and the LPF 200 are digital filters, the same filter characteristics can be realized. Thereby, when calculating the CCDF of the CCDF calculation unit 112 and the CCDF calculation unit 114, an error does not occur in the calculated CCDF due to the difference in filter characteristics. Accordingly, when the comparison unit 113 compares both CCDFs, there is no CCDF calculation error, so that it is possible to prevent the optimum value of the operation point of the operation point adjustment unit 103 from being shifted due to a difference in filter characteristics.
　仮に、ＬＰＦ１１５及びＬＰＦ２００のフィルタ特性が異なっていると、ＬＰＦ１１５及びＬＰＦ２００に同じ信号が入力されても、ＬＰＦ１１５を通過した信号とＬＰＦ２００を通過した信号とには差が生じるため、それぞれの信号を元に計算するそれぞれのＣＣＤＦの間には差が生じる。このように、本来、同じ信号から計算するＣＣＤＦ同士は同じになるはずであるが、ＣＣＤＦの間に誤差が生じることになるので、動作点調整部１０３における動作点調整に誤差が生じる。よって、動作点が最適にならず、電力増幅器１０７に対して最適な歪み補償を行うことができなくなる。ゆえに、ＬＰＦ１１５及びＬＰＦ２００のフィルタ特性が異なっていると歪み補償の精度が劣化する。 If the filter characteristics of the LPF 115 and the LPF 200 are different, even if the same signal is input to the LPF 115 and the LPF 200, there is a difference between the signal that has passed through the LPF 115 and the signal that has passed through the LPF 200. There is a difference between each CCDF that is calculated. Thus, the CCDFs calculated from the same signal should be the same, but an error occurs between the CCDFs, and an error occurs in the operating point adjustment in the operating point adjustment unit 103. Therefore, the operating point is not optimized, and optimal distortion compensation cannot be performed for the power amplifier 107. Therefore, if the filter characteristics of the LPF 115 and the LPF 200 are different, the accuracy of distortion compensation deteriorates.
　図１２は、この発明の実施の形態２に係るＣＣＤＦの誤差と歪み補償量の劣化量との関係を示す図である。
　図１２は、上記で説明したように、ＣＣＤＦの誤差が大きくなると、歪み補償量の劣化量は大きくなり、歪み補償がかからなくなることを示している。ここで、歪み補償量の劣化量とは、理想的に歪み補償がかかった場合、つまり信号の歪みがゼロの場合からの劣化量を示す。理想的に歪み補償がかかった場合、歪み補償量の劣化量はゼロになる。 FIG. 12 is a diagram showing the relationship between the CCDF error and the distortion compensation deterioration amount according to the second embodiment of the present invention.
As described above, FIG. 12 shows that when the CCDF error increases, the distortion compensation amount deteriorates and the distortion compensation is not performed. Here, the deterioration amount of the distortion compensation amount indicates an amount of deterioration from when the distortion compensation is ideally applied, that is, when the signal distortion is zero. When distortion compensation is ideally applied, the degradation amount of the distortion compensation amount becomes zero.
　以上のように、実施の形態２の歪み補償回路によれば、ＬＰＦ１１５とＬＰＦ２００とをディジタルフィルタで構成し、フィルタ特性を同じにするので、ＣＣＤＦ値に誤差が生じず、最適な動作点調整を行うことができる。 As described above, according to the distortion compensation circuit of the second embodiment, the LPF 115 and the LPF 200 are configured by digital filters, and the filter characteristics are the same. Therefore, no error occurs in the CCDF value, and the optimum operating point adjustment is performed. It can be carried out.
１００  信号生成部、１０１  複素乗算部、１０２　絶対値計算部、１０３　動作点調整部、１０４　ＬＵＴ読み出し部、１０５　Ｄ／Ａ変換器、１０６　周波数変換回路、１０７　電力増幅器、１０８　出力端子、１０９　包絡線検出器、１１０　ＬＰＦ、１１１　Ａ／Ｄ変換器、１１２　ＣＣＤＦ計算部、１１３　比較部、１１４　ＣＣＤＦ計算部、１１５　ＬＰＦ、２００　ＬＰＦ。 100 signal generation unit, 101 complex multiplication unit, 102 absolute value calculation unit, 103 operating point adjustment unit, 104 LUT reading unit, 105 D / A converter, 106 frequency conversion circuit, 107 power amplifier, 108 output terminal, 109 envelope Detector, 110 LPF, 111 A / D converter, 112 CCDF calculation unit, 113 comparison unit, 114 CCDF calculation unit, 115 LPF, 200 LPF.

Claims (7)

  1. 送信信号を増幅する増幅器と、
    前記増幅器が増幅した前記送信信号の包絡線を検出し、検出した前記包絡線を包絡線信号として出力する検出器と、
    前記包絡線信号の信号帯域以下のカットオフ周波数をもち、前記包絡線信号を帯域制限する第１のフィルタと、
    　前記第１のフィルタが帯域制限した前記包絡線信号をディジタル化するアナログディジタル変換器と、
    前記アナログディジタル変換器がディジタル化した前記包絡線信号の振幅に対する第１の分布を計算する第１の計算部と、
    前記増幅器に入力される前記送信信号の絶対値を算出し、算出した前記絶対値を参照信号として出力する絶対値計算部と、
    前記参照信号の信号帯域以下のカットオフ周波数をもち、参照信号を帯域制限する第２のフィルタと、
    第２のフィルタが帯域制限した前記参照信号の振幅に対する第２の分布を計算する第２の計算部と、
    前記第１の分布と前記第２の分布との誤差を算出する比較部と、
    前記比較部が算出した前記誤差が小さくなるように、前記増幅器に入力される前記送信信号をプレディストーションし、プレディストーションした前記送信信号を前記増幅器に出力する補償部と、
    を備えたことを特徴とする歪み補償回路。     An amplifier for amplifying the transmission signal;
    A detector that detects an envelope of the transmission signal amplified by the amplifier and outputs the detected envelope as an envelope signal;
    A first filter having a cut-off frequency equal to or lower than a signal band of the envelope signal and band-limiting the envelope signal;
    An analog-to-digital converter for digitizing the envelope signal band-limited by the first filter;
    A first calculation unit for calculating a first distribution with respect to the amplitude of the envelope signal digitized by the analog-digital converter;
    An absolute value calculation unit that calculates an absolute value of the transmission signal input to the amplifier and outputs the calculated absolute value as a reference signal;
    A second filter having a cut-off frequency equal to or lower than the signal band of the reference signal and band-limiting the reference signal;
    A second calculation unit for calculating a second distribution for the amplitude of the reference signal band-limited by a second filter;
    A comparator that calculates an error between the first distribution and the second distribution;
    Predistorting the transmission signal input to the amplifier so that the error calculated by the comparison unit is reduced, and a compensation unit outputting the predistorted transmission signal to the amplifier;
    A distortion compensation circuit comprising:
  2. 　前記第１のフィルタのカットオフ周波数と前記第２のフィルタのカットオフ周波数とが等しいことを特徴とする請求項１に記載の歪み補償回路。 The distortion compensation circuit according to claim 1, wherein a cutoff frequency of the first filter is equal to a cutoff frequency of the second filter.
  3. 　前記第１の分布及び前記第２の分布は確率分布であって、前記第１の計算部と前記第２の計算部とがそれぞれ前記第１の分布と前記第２の分布とを計算するときのデータ点数Ｎが、前記アナログディジタル変換器のサンプリング周波数Ｆｓと第１のフィルタのカットオフ周波数Ｆｃとを用いて、Ｎ＞Ｆｓ／Ｆｃを満たすことを特徴とする請求項１に記載の歪み補償回路。 When the first distribution and the second distribution are probability distributions, and the first calculation unit and the second calculation unit calculate the first distribution and the second distribution, respectively. 2. The distortion compensation according to claim 1, wherein the number N of data points satisfies N> Fs / Fc using the sampling frequency Fs of the analog-digital converter and the cutoff frequency Fc of the first filter. circuit.
  4. 　前記データ点数Ｎが、Ｆｓ／Ｆｃの整数倍であることを特徴とする請求項３に記載の歪み補償回路。 The distortion compensation circuit according to claim 3, wherein the number N of data points is an integer multiple of Fs / Fc.
  5. 　前記補償部は、前記増幅器の非線形特性を補償する補償係数が保存されており、前記比較部が算出した前記誤差の絶対値が閾値以上の場合、前記絶対値計算部が検出した前記参照信号に乗算する利得を増加または減少させ、前記利得を乗算した前記参照信号の電力に対応する前記補償係数を用いて、前記増幅器に入力される前記送信信号をプレディストーションすることを特徴とする請求項１に記載の歪み補償回路。 The compensation unit stores a compensation coefficient that compensates for the nonlinear characteristic of the amplifier. When the absolute value of the error calculated by the comparison unit is equal to or greater than a threshold value, the compensation unit detects the reference signal detected by the absolute value calculation unit. The transmission signal input to the amplifier is predistorted using the compensation coefficient corresponding to the power of the reference signal multiplied or increased by increasing or decreasing the gain to be multiplied. The distortion compensation circuit according to 1.
  6. 　前記第１のフィルタのカットオフ周波数より低いカットオフ周波数を有し、前記アナログディジタル変換器より後段に設けられ、前記第１のフィルタが帯域制限した前記包絡線信号を帯域制限する第３のフィルタを備え、
    　前記第２のフィルタ及び前記第３のフィルタがディジタルフィルタであって、前記第２のフィルタ及び前記第３のフィルタのカットオフ周波数が等しいことを特徴とする請求項３に記載の歪み補償回路。     A third filter having a cutoff frequency lower than the cutoff frequency of the first filter, provided downstream from the analog-digital converter, and band-limiting the envelope signal band-limited by the first filter. With
    4. The distortion compensation circuit according to claim 3, wherein the second filter and the third filter are digital filters, and the cutoff frequencies of the second filter and the third filter are equal.
  7. 　前記第２のフィルタのフィルタ特性と前記第３のフィルタのフィルタ特性とが等しいことを特徴とする請求項６に記載の歪み補償回路。 The distortion compensation circuit according to claim 6, wherein a filter characteristic of the second filter and a filter characteristic of the third filter are equal.
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Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2008146355A1 (en) * 2007-05-28 2008-12-04 Panasonic Corporation Distortion compensator
JP2014146963A (en) * 2013-01-29 2014-08-14 Mitsubishi Electric Corp Distortion compensation circuit

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2008146355A1 (en) * 2007-05-28 2008-12-04 Panasonic Corporation Distortion compensator
JP2014146963A (en) * 2013-01-29 2014-08-14 Mitsubishi Electric Corp Distortion compensation circuit

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