WO2014016955A1 - Optical modulation and demodulation method and optical transceiver - Google Patents

Optical modulation and demodulation method and optical transceiver Download PDF

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Publication number
WO2014016955A1
WO2014016955A1 PCT/JP2012/069143 JP2012069143W WO2014016955A1 WO 2014016955 A1 WO2014016955 A1 WO 2014016955A1 JP 2012069143 W JP2012069143 W JP 2012069143W WO 2014016955 A1 WO2014016955 A1 WO 2014016955A1
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phase
polarization
optical
signal
slip state
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PCT/JP2012/069143
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French (fr)
Japanese (ja)
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吉田　剛
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三菱電機株式会社
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Priority to PCT/JP2012/069143 priority Critical patent/WO2014016955A1/en
Priority to JP2014501343A priority patent/JP5653561B2/en
Publication of WO2014016955A1 publication Critical patent/WO2014016955A1/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/40—Transceivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38—Demodulator circuits; Receiver circuits

Definitions

  • the present invention relates to an optical modulation / demodulation method and an optical transceiver, and particularly to an optical modulation / demodulation method and an optical transceiver using a digital coherent method.
  • BPSK binary phase-shift keying
  • OOK on-off keying
  • QPSK Phase-Shift Keying
  • a method for high-density wavelength multiplexing, a method is used in which the number of transmission bits per symbol is doubled by polarization multiplexing in which independent signals are assigned to two orthogonal polarization components.
  • a method of increasing the signal multiplicity and increasing the number of transmission bits per symbol such as QPSK or 16-value quadrature amplitude modulation (16 quadrature Amplitude Modulation: 16QAM).
  • QPSK and 16QAM are transmitted by assigning signals to the same phase axis (In-Phase axis: I axis) and quadrature phase axis (Quadrature-Phase axis: Q axis) in the optical transmitter.
  • Non-Patent Document 1 chromatic dispersion generated in a transmission line and polarization mode dispersion (Polarization-polarization-) are achieved by linear photoelectric conversion by synchronous detection and fixed, semi-fixed, and adaptive linear equalization by digital signal processing. It is possible to achieve excellent equalization characteristics and excellent noise tolerance against linear waveform distortion caused by Mode Dispersion (PMD).
  • PMD Mode Dispersion
  • an M-th power method (see, for example, Non-Patent Document 2) or a provisional determination type algorithm (for example, see Patent Document 1) has been used.
  • a slip of an angle (2 ⁇ / M) ⁇ N may occur when the carrier phase is restored under conditions where noise and waveform distortion are large, and a large-scale continuous error may occur (M: PSK phase).
  • M PSK phase
  • differential encoding / decoding has been generally used as a method for preventing phase slip.
  • the period count (cycle count) value is increased or decreased on the condition that the transition of the estimated carrier phase between symbols exceeds a positive or negative threshold value. It is disclosed that carrier phase estimation is accurately performed based on the period count value.
  • Patent Document 3 a reference signal (pilot signal) is inserted between data signals on the transmission side in coherent optical communication.
  • the carrier phase is estimated from the reference signal.
  • the carrier phase estimated based on the reference signal is linearly interpolated to guarantee the estimation accuracy of the carrier phase.
  • Patent Document 4 discloses that in coherent optical communication, a reference signal (SYNC burst) is inserted between data signals on the transmission side, and a continuous error that may occur due to phase slip is relieved by error correction.
  • phase slip can be generated by checking whether the decoding results of the same symbol decoded in each direction are inconsistent after overlapping and collating the two-way data decoding in the time series order and the time series reverse order. Is also detected and corrected by error correction.
  • Patent Document 2 has a problem that the detection accuracy of the phase slip is insufficient and a large-scale continuous error due to the missed phase slip is unavoidable.
  • the present invention has been made in order to solve the above-described problem, an optical modulation / demodulation method capable of preventing transmission quality deterioration due to phase slip, improving transmission quality, and improving noise immunity, and The purpose is to obtain an optical transceiver.
  • an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them.
  • the receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency
  • a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase
  • an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them.
  • the receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency
  • a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase
  • Embodiments of an optical transceiver and an optical modulation / demodulation method according to the present invention will be described below in detail with reference to the drawings.
  • the embodiment described below is an embodiment for embodying the present invention, and is not intended to limit the present invention to its category.
  • FIG. 1 is a diagram illustrating a configuration example of an optical transceiver according to Embodiment 1 of the present invention.
  • the optical transceiver according to the first embodiment includes an optical transmission unit 100, a transmission channel 200, and an optical reception unit 300.
  • the optical transmission unit 100 includes a light source 101, a symbol mapper 102, and an optical modulation unit 103.
  • the optical receiver 300 includes a light source 301, an optical demodulator 302, a waveform equalizer 303, a frequency / phase estimator 304 (X / Y combination), a decoder 305 (X / Y simultaneously), a phase slip
  • the state estimation unit 306 and the selection unit 307 are configured.
  • the optical transmitter 100 transmits an optical signal to a counterpart optical transceiver (not shown) via the transmission channel 200.
  • the optical receiving unit 300 receives an optical signal transmitted from a partner optical transceiver (not shown) via the transmission channel 200.
  • the counterpart optical transmitter / receiver is not shown in FIG. 1, but has the same configuration as the optical transmitter / receiver shown in FIG.
  • the light source 101 generates a carrier light signal that oscillates at a predetermined wavelength set in advance, and outputs the carrier wave signal to the light modulator 103.
  • the symbol mapper 102 performs signal point arrangement based on transmission data (two series of codes A and B, see FIG. 4) input from the outside (not shown), and outputs a 4-lane binary electrical signal.
  • FIG. 2 is a signal point arrangement in the polarization phase space of the polarization-multiplexed BPSK (DP-BPSK: Dual-Polarization BPSK) method.
  • the horizontal axis represents the optical phase of X polarization, which is one polarization component when orthogonal polarization multiplexing is performed, and the vertical axis is the optical phase of the Y polarization, which is the other polarization component when orthogonal polarization multiplexing is performed.
  • Ordinary BPSK is represented by binary values of phase 0 and phase ⁇ . In FIG. 2, these are indicated by phase ⁇ / 4 and phase -3 ⁇ / 4 obtained by rotating them by ⁇ / 4.
  • each of the X polarization and the Y polarization takes a binary phase, four signal points (symbols) 0, 1, 2, and 3 can be taken in FIG.
  • the arrangement of these signal points is symmetrical (point symmetry, line symmetry) between the X polarization and the Y polarization.
  • FIG. 3 is a signal point arrangement in the polarization phase space according to Embodiment 1 of the present invention.
  • the signal point arrangement is an asymmetric arrangement between the X polarization and the Y polarization, and the signal point arrangement differs between the X polarization and the Y polarization.
  • the phase of X polarization is ⁇ 3 ⁇ / 4 (signal points 1 and 3), from the signal point arrangement of FIG. Shift the phase by + ⁇ / 2.
  • the phase of the Y polarization is the same as the signal point arrangement in FIG.
  • the arrangement of the signal points 0, 1, 2, and 3 is neither point-symmetric nor line-symmetric.
  • the signal point arrangement of both X / Y polarization is binary phase shift keying, and one side phase of one binary phase shift keying of X / Y polarization is performed. Only in this case, the other signal point of the X / Y polarization is shifted by ⁇ / 2.
  • the arrangement of the signal points is asymmetric between the X polarization and the Y polarization.
  • the symbol mapper 102 receives two series of codes A and B as transmission data from the outside (not shown), and is a 4-lane binary (+/ ⁇ ) electrical signal XI, XQ. , YI, YQ are generated and output to the light modulation unit 103.
  • the Y polarization is apparently QPSK, but the QPSK side is not gray-coded, as can be seen from FIG. 3, between 0-1 and 0-3 rather than between symbols 0-2 and 1-3. Since the polarization phase space distance is longer between 2-1 and 2-1 and 2-3, this is due to the intention of increasing the hamming distance on the far side.
  • This processing can be realized by table processing in which A and B are input and XI, XQ, YI, and YQ are output. Alternatively, it can be obtained by the following logical operation.
  • the symbol mapper 102 generates a four-lane binary signal with XI “+”, XQ “+”, YI “+”, and YQ “+”.
  • the symbols 1, 2, and 3 and the codes A and B shown in FIG. 4 are inputted, respectively, and based on them, a binary signal of 4 lanes shown in FIG. 4 is generated.
  • the optical modulation unit 103 modulates the carrier optical signal input from the light source 101 with the 4-lane binary electrical signals (XI, XQ, YI, YQ) input from the symbol mapper 102, multiplexes them, and FIG.
  • An optical signal having the signal point arrangement shown in FIG. A signal (binary electrical signal) input from the symbol mapper 102 is generally amplified by a driver (not shown) to drive the light modulator 103, but the driver is omitted in the configuration of FIG. Yes.
  • the optical modulation unit 103 divides the carrier wave input from the light source 101 into two systems, respectively performs quadrature phase (I / Q) modulation with the binary electric signal from the symbol mapper 102, and performs orthogonal polarization multiplexing.
  • the transmission unit 100 may be provided with a digital / analog conversion unit (not shown) to perform digital signal processing such as predistortion of waveform distortion at the transmission end.
  • the transmission channel 200 transmits an optical signal input from the optical modulation unit 103 in the optical transmission unit 100 and outputs the optical signal to the optical reception unit 300 of the counterpart optical transceiver.
  • the transmission channel 200 includes devices and components generally used for transmitting optical signals, such as a wavelength multiplexing / demultiplexing device, an optical amplifying device, and a transmission line optical fiber.
  • the light source 301 generates a local oscillation optical signal that oscillates at a predetermined wavelength set in advance, and outputs the local oscillation optical signal to the optical demodulation unit 302.
  • the local oscillation optical signal is oscillated at substantially the same wavelength as the carrier optical signal generated by the light source 101.
  • the optical demodulator 302 decomposes the optical signal transmitted via the transmission channel 200 and the local oscillation optical signal input from the light source 301 into orthogonal polarization components and orthogonal phase components. Next, the optical demodulator 302 performs coherent detection that mixes the decomposed optical signal and the local oscillation optical signal, performs photoelectric conversion, and converts the signal into a four-lane electrical signal. Further, these electric signals are converted into digital signals by analog / digital conversion, and quantized and sampled. The 4-lane digital signal thus obtained is output to the waveform equalization unit 303. In the analog-digital conversion, quantization is usually performed with a resolution of 6 bits or more, and sampling is performed at a sampling rate that is twice or more the baud rate.
  • the waveform equalization unit 303 receives a 4-lane digital signal from the optical demodulation unit 302 and generates waveform distortion caused by chromatic dispersion, polarization rotation, polarization mode dispersion, fiber nonlinear optical effect, or the like generated in the transmission channel 200. And a 4-lane digital signal separated by orthogonal polarization is output to the frequency / phase estimator 304.
  • the frequency / phase estimation unit 304 receives 4-lane digital signals from the waveform equalization unit 303 and combines them between orthogonal polarizations (between X / Y polarizations) to estimate and compensate for the carrier frequency / phase. . That is, the frequency / phase estimation unit 304 compensates for the center frequency difference between the carrier optical signal and the local oscillation optical signal, which is present in the 4-lane digital signal input from the waveform equalization unit 303, and the M multiplication method. Also, carrier frequency estimation and carrier phase estimation are performed based on the provisional determination method and the like, and the 4-lane digital signal after frequency / phase compensation is output to the decoding unit 305 and the phase slip state estimation unit 306. Here, the carrier frequency estimation and the carrier phase estimation are not performed independently between the orthogonal polarizations, but are performed simultaneously by combining between the orthogonal polarizations (between the X / Y polarizations).
  • the phase slip state includes (1) a state in which the phase is not slipped in both X / Y polarization, and (2) X / Y polarization.
  • There can be a total of four states: a state where both slip +90 degrees, a state where (3) both X / Y polarizations slip 180 degrees, and a state where both (4) X / Y polarizations slip -90 degrees. If the carrier frequency and phase are estimated independently for X / Y polarization, slips will occur independently for each polarization. Therefore, the slip does not fit in these four ways, but the square of 4 (4 2 16), and phase slip detection described later becomes difficult.
  • FIG. 5 shows signal point arrangements after frequency / phase compensation for four types of phase slip states obtained by combining with X / Y polarization and simultaneously performing carrier frequency / phase estimation.
  • the phase slip state State-0 is when there is no phase slip
  • the phase slip state State-1 is when the phase slip +90 degrees
  • the phase slip state State-2 is when the phase slip +180 degrees
  • the phase slip state State- 3 is a case where the phase slip is -90 degrees.
  • the signal points do not overlap between the phase slip states, and the signal point arrangement is unique for each phase slip state.
  • these signal point arrangements are signal point arrangements that cannot be taken when there is no phase slip, it is possible to detect and compensate for a phase slip state.
  • the decoding unit 305 performs decoding for all possible phase slip states for each symbol. That is, the decoding unit 305 decodes the 4-lane digital signal input from the frequency / phase estimation unit 304 based on the threshold value indicated by the dotted line in the signal point arrangement of each State in FIG. Is output to the selection unit 307.
  • FIG. 6 illustrates a configuration example of the decoding unit 305.
  • the decoding unit 305 includes four two-dimensional phase identifiers (2-D2-Phase Slicers) 500, 501, 502, and 503.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally input to the two-dimensional phase identifiers 500, 501, 502, and 503.
  • the two-dimensional phase discriminator 500 performs decoding for each 4-lane digital signal corresponding to State-0, and outputs the decoding result to the selection unit 307 (see FIG. 1).
  • the two-dimensional phase identifier 501 performs decoding for each signal of the 4-lane digital signal corresponding to State-1, and outputs the decoding result to the selection unit 307.
  • the two-dimensional phase discriminator 502 performs decoding for each signal of the 4-lane digital signal corresponding to State-2, and outputs the decoding result to the selection unit 307.
  • the two-dimensional phase discriminator 503 performs decoding for each digital lane signal corresponding to State-3 and outputs the decoding result to the selection unit 307.
  • Symbols Z ′, Z ′′, and Z ′′ ′′ are decoded as symbols Z, respectively.
  • a soft value may be passed to the selection unit 307 as a soft decision to give reliability information according to the distance from the signal point center in the polarization phase space.
  • the phase slip state estimation unit 306 calculates likelihoods for all possible phase slip states for each signal (symbol), compares them, and estimates which phase slip state has the maximum likelihood. That is, the phase slip state estimation unit 306 estimates the maximum likelihood phase slip state based on the 4-lane digital signal input from the frequency / phase estimation unit 304, and outputs it to the selection unit 307.
  • FIG. 7 illustrates a configuration example of the phase slip state estimation unit 306.
  • the phase slip state estimation unit 306 includes four likelihood generation units (MetricMeGenerators) 600, 601, 602, and 603 and one maximum likelihood state estimation unit (Maximum Likelihood State Estimator) 610.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally output to the likelihood generation units 600, 601, 602, and 603.
  • the likelihood generation unit 600 calculates the likelihood that the phase slip state is State-0, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 601 calculates the likelihood that the phase slip state is State-1, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 602 calculates the likelihood that the phase slip state is State-2, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 603 calculates the likelihood that the phase slip state is State-3, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the maximum likelihood state estimation unit 610 is based on the likelihood information of each phase slip state State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603.
  • the most likely phase slip state is obtained from the slip states State-0, State-1, State-2, and State-3, and is output to the selection unit 307 (not shown in FIG. 7).
  • FIG. 8 illustrates a configuration example of the likelihood generation units 600, 601, 602, and 603 in the phase slip state estimation unit 306. Since the likelihood generation units 600, 601, 602, and 603 each have the same functional block, the case of the likelihood generation unit 600 that calculates the likelihood of State-0 will be described here.
  • the likelihood generation unit 600 includes four polarization phase distance calculation units 700, 701, 702, and 703, one selection unit 710, n delay units 720A, 720B,. , 721X and one adder (sum) 722.
  • the delay units 720A, 720B,..., 720X are connected in series as shown in FIG.
  • the multipliers 721A, 721B,..., 721X are connected to delay units 720A, 720B,.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally output to the polarization phase distance calculation units 700, 701, 702, and 703.
  • the polarization phase space distance calculation unit 700 applies the polarization phase from the central signal point 0 to the input 4-lane digital signal based on the State-0 signal point arrangement rule in which there is no phase slip. Find the spatial distance. For example, the sum of squares of the phase distance D_ph_x (0) in the X polarization direction and the phase distance D_ph_y (0) in the Y polarization direction ((D_ph_x (0)) 2 + (D_ph_y (0)) 2 ) is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 701 obtains the polarization phase space distance from the signal point 1 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (1)) 2 + (D_ph_y (1)) 2 ) of the phase distance D_ph_x (1) in the X polarization direction and the phase distance D_ph_y (1) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 702 obtains the polarization phase space distance from the signal point 2 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (2)) 2 + (D_ph_y (2)) 2 ) of the phase distance D_ph_x (2) in the X polarization direction and the phase distance D_ph_y (2) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 703 obtains the polarization phase space distance from the signal point 3 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (3)) 2 + (D_ph_y (3)) 2 ) of the phase distance D_ph_x (3) in the X polarization direction and the phase distance D_ph_y (3) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the selection unit 710 selects the minimum polarization phase space distance from the four polarization phase space distances input from the polarization phase space distance calculation units 700, 701, 702, and 703, and outputs the selected polarization phase space distance to the delay unit 720A. To do.
  • the delay unit 720A holds the digital signal input from the selection unit 710 for one symbol time, and outputs the digital signal to the delay unit 720B and the multiplication unit 721A.
  • Multiplication unit 721A outputs the result of multiplying the digital signal input from delay unit 720A by coefficient C [0] to addition unit 722.
  • the delay unit 720B holds the digital signal input from the delay unit 720A for one symbol time, and outputs the digital signal to the delay unit 720C (not shown) and the multiplication unit 721B connected in series to the delay unit 720B.
  • the multiplier 721B outputs the result of multiplying the digital signal input from the delay unit 720B by the coefficient C [1] to the adder 722.
  • n-th delay unit 720X holds the digital signal input from the (n ⁇ 1) -th delay unit 720W (not shown) for one symbol time, and outputs the digital signal to the multiplication unit 721X. .
  • the multiplication unit 721X outputs the result of multiplying the digital signal input from the delay unit 720X by the coefficient C [n ⁇ 1] to the addition unit 722.
  • the adding unit 722 calculates the sum of the n multiplication results input from the multiplying units 721A, 721B,..., 721X, and outputs the value of the sum to the maximum likelihood state estimating unit 610 (see FIG. 7).
  • the smaller the sum value the higher the likelihood.
  • the delay units 720A, 720B,..., 720X, the multipliers 721A, 721B,..., 721X, and the adder 722 constitute a finite impulse response filter.
  • the instantaneous likelihood at the time is averaged. Therefore, delay units 720A, 720B,..., 720X, multiplication units 721A, 721B,..., 721X, and addition unit 722 constitute an averaging unit that averages the signal output from selection unit 710. is doing.
  • the filter constituting the averaging unit is not limited to a finite impulse response filter, and a forgetting infinite impulse response filter may be used. In that case, maximum likelihood sequence estimation and posterior probability maximization can be performed more complicatedly.
  • the same processing as the processing in the likelihood generation unit 600 is performed. Here, the description is omitted.
  • the maximum likelihood state estimation unit 610 includes a total sum of phase slip states State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603,
  • the phase slip state having the smallest sum value is obtained as the “maximum likelihood phase slip state” and is output to the selection unit 307 (see FIG. 1).
  • the selection unit 307 compares the four decoding results input from the decoding unit 305 based on the phase slip state estimated by the phase slip state estimation unit 306, and corresponds to the phase slip state from those decoding results.
  • the decoding result to be selected is selected as the maximum likelihood decoding result, and is output to an external device (not shown) connected to the optical transceiver.
  • the simulation of the bit error rate characteristic when this embodiment is used was performed. Simulation conditions are shown in FIG.
  • the bit rate was 128 Gb / s, and the 11-stage pseudo-random binary sequence was used as the code sequence.
  • the number of symbols was 16384, and the calculation was repeated with the noise pattern changed 100 times.
  • the transmission line was an additive white Gaussian Noise (AWGN) channel, and the noise bandwidth when defining the optical signal power to noise power ratio was 0.1 nm.
  • AWGN additive white Gaussian Noise
  • the phase noise between the carrier optical signal and the local oscillation optical signal was 0, and the frequency difference between the carrier optical signal and the local oscillation optical signal was 0. Polarization demultiplexing is ideally performed.
  • the fourth power method was used, the phase errors detected in each polarization were averaged, and equal phase compensation was performed for each polarization.
  • the averaging of the carrier phase estimation was a 17-symbol moving average, and an unwrapping process was performed so that the phase change from the carrier phase one symbol before was within ⁇ ⁇ / 4.
  • the averaging of the phase slip state estimation was performed by a 17 symbol moving average. After obtaining the code error rate, it was converted into a Q value through a complementary error function and evaluated.
  • Figure 10 shows the simulation results.
  • a DP-DE (Differential Encoded) QPSK signal obtained by applying differential encoding / decoding to the DP-QPSK method and a case where there is no phase slip (No Slip) in the DP-QPSK signal were also calculated.
  • the horizontal axis is OSNR
  • the vertical axis is Q value (Q-factor).
  • the Q value of the present embodiment in the region where the Q value is 9 dB or more, a Q value equivalent to that without slip is obtained, and good characteristics are exhibited.
  • the Q value of the present embodiment is lower than the Q value without the phase slip, but the DP-DEQPSK signal However, the Q value still shows an excellent Q value of 0.5 dB or more.
  • signal point arrangement that is asymmetrical between orthogonally polarized waves as shown in FIG. 3 is performed in optical transmitter 100, and carrier frequency / phase estimation is performed between orthogonally polarized waves in receiver 300.
  • the phase slip can be detected by carrying out simultaneously. Thereby, the phase slip state can be estimated, and the most likely slip state can be estimated.
  • phase slip compensation can be performed by selecting a decoding result based on a decoding rule corresponding to the estimated slip state.
  • phase slip state can be detected and the phase slip compensation can be performed. Therefore, the phase slip state can be estimated on a signal basis without using differential encoding / decoding and without increasing the redundancy. Also, decoding according to the estimated slip state can be performed. As a result, transmission quality deterioration due to phase slip can be avoided, noise tolerance can be improved, and transmission quality can be improved.
  • the optical modulation / demodulation method according to the present invention is useful for an optical transmission system using a digital coherent method, and is particularly suitable for an optical transmission system with a low code error rate.

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Abstract

Provided is an optical transceiver provided with an optical sending unit (100) and an optical receiving unit (300), in which signal points of a sent optical signal are distributed asymmetrically through X polarization and Y polarization by the optical sending unit (100) and the frequency and phase of a carrier wave of a received optical signal are simultaneously estimated between the X polarization and the Y polarization by the optical receiving unit (300), thereby detecting phase-slipping. Because the distribution of signal points of the received optical signal is specific to the phase-slipping state, it is possible to estimate the phase-slipping state and carry out decoding in accordance with the estimated phase-slipping state, for each signal point.

Description

光変復調方法および光送受信器Optical modulation / demodulation method and optical transceiver
　本発明は光変復調方法および光送受信器に関し、特に、デジタルコヒーレント方式を用いた光変復調方法および光送受信器に関する。 The present invention relates to an optical modulation / demodulation method and an optical transceiver, and particularly to an optical modulation / demodulation method and an optical transceiver using a digital coherent method.
　大容量光伝送のためには、光信号（optical signal）対雑音電力（noise power）の限界の克服と、高密度波長多重化とが、課題である。 For high-capacity optical transmission, overcoming the limitations of optical signal versus noise power and high-density wavelength multiplexing are challenges.
　光信号対雑音電力の限界を克服する技術として、従来のオンオフキーイング（Ｏｎ－Ｏｆｆ　Ｋｅｙｉｎｇ：ＯＯＫ）の代わりに、２値位相偏移変調（Ｂｉｎａｒｙ　Ｐｈａｓｅ－Ｓｈｉｆｔ　Ｋｅｙｉｎｇ：ＢＰＳＫ）や４値ＰＳＫ（Ｑｕａｔｅｒｎａｒｙ　Ｐｈａｓｅ－Ｓｈｉｆｔ　Ｋｅｙｉｎｇ：ＱＰＳＫ）が用いられる。 As a technique for overcoming the limitations of optical signal-to-noise power, binary phase-shift keying (BPSK) and quaternary PSK (Quaternary) are used instead of conventional on-off keying (OOK). Phase-Shift Keying (QPSK) is used.
　高密度波長多重化のために、直交する２つの偏波成分に独立の信号を割り当てる偏波多重によって、１シンボル当たりの伝送ビット数を２倍に増やす方式が用いられている。あるいは、別の方式として、ＱＰＳＫや１６値直交振幅変調（１６　Ｑｕａｄｒａｔｕｒｅ　Ａｍｐｌｉｔｕｄｅ　Ｍｏｄｕｌａｔｉｏｎ：１６ＱＡＭ）のように、信号多重度を上げて、１シンボル当たりの伝送ビット数を増やす方式が知られている。ＱＰＳＫや１６ＱＡＭは、光送信器において、同位相軸（Ｉｎ－Ｐｈａｓｅ軸：Ｉ軸）と、直交位相軸（Ｑｕａｄｒａｔｕｒｅ－Ｐｈａｓｅ軸：Ｑ軸）とに信号を割り当てて伝送する。 For high-density wavelength multiplexing, a method is used in which the number of transmission bits per symbol is doubled by polarization multiplexing in which independent signals are assigned to two orthogonal polarization components. Alternatively, as another method, there is known a method of increasing the signal multiplicity and increasing the number of transmission bits per symbol, such as QPSK or 16-value quadrature amplitude modulation (16 quadrature Amplitude Modulation: 16QAM). QPSK and 16QAM are transmitted by assigning signals to the same phase axis (In-Phase axis: I axis) and quadrature phase axis (Quadrature-Phase axis: Q axis) in the optical transmitter.
　また、同期検波方式にデジタル信号処理を組み合わせて、これらの光変調信号を受信するデジタルコヒーレント方式が注目されている（例えば、非特許文献１参照）。この方式では、同期検波による線形な光電気変換と、デジタル信号処理による固定的または半固定的、および、適応的な線形等化により、伝送路で生じる波長分散や、偏波モード分散（Ｐｏｌａｒｉｚａｔｉｏｎ－Ｍｏｄｅ　Ｄｉｓｐｅｒｓｉｏｎ：ＰＭＤ）等に起因する線形な波形歪みに対する優れた等化特性や優れた雑音耐力を実現できる。 Also, a digital coherent method for receiving these optical modulation signals by combining digital signal processing with the synchronous detection method has attracted attention (for example, see Non-Patent Document 1). In this method, chromatic dispersion generated in a transmission line and polarization mode dispersion (Polarization-polarization-) are achieved by linear photoelectric conversion by synchronous detection and fixed, semi-fixed, and adaptive linear equalization by digital signal processing. It is possible to achieve excellent equalization characteristics and excellent noise tolerance against linear waveform distortion caused by Mode Dispersion (PMD).
　デジタルコヒーレント方式において受信部で搬送波位相オフセット補償を行う方法として、Ｍ乗法（例えば、非特許文献２参照）や、仮判定型アルゴリズム（例えば、特許文献１参照）が用いられてきた。これらの方法では、雑音や波形歪みが大きい条件において、搬送波の位相の復元時に、角度(２π／Ｍ)×Ｎのスリップが生じ、大規模連続誤りが生じる可能性がある（Ｍ：ＰＳＫの位相数を示す自然数、Ｎ：自然数）。従来、位相スリップを防ぐ方法として、差動符号化・復号化が一般的に利用されてきた。 As a method for performing carrier phase offset compensation at the receiver in the digital coherent method, an M-th power method (see, for example, Non-Patent Document 2) or a provisional determination type algorithm (for example, see Patent Document 1) has been used. In these methods, a slip of an angle (2π / M) × N may occur when the carrier phase is restored under conditions where noise and waveform distortion are large, and a large-scale continuous error may occur (M: PSK phase). A natural number indicating a number, N: a natural number). Conventionally, differential encoding / decoding has been generally used as a method for preventing phase slip.
　また、差動符号化・復号化を用いずに、位相スリップを回避する方法も検討されている。 Also, a method for avoiding phase slip without using differential encoding / decoding has been studied.
　特許文献２では、コヒーレント光受信器の搬送波の位相の推定において、シンボル間の推定搬送波位相の遷移が正もしくは負のしきい値を超えることを条件に、周期カウント（サイクルカウント）値を増減させ、周期カウント値をもとに正確に搬送波位相推定を行うことが開示されている。 In Patent Document 2, in estimating the phase of a carrier wave of a coherent optical receiver, the period count (cycle count) value is increased or decreased on the condition that the transition of the estimated carrier phase between symbols exceeds a positive or negative threshold value. It is disclosed that carrier phase estimation is accurately performed based on the period count value.
　特許文献３では、コヒーレント光通信において、送信側で、データ信号の合間に参照信号（パイロット信号）を挿入する。受信側で、参照信号から、搬送波位相を推定する。データ信号のうち位相変動が大きいシンボルについては、参照信号に基づいて推定した搬送波位相を線形補間することにより、搬送波位相の推定確度を保証することが開示されている。 In Patent Document 3, a reference signal (pilot signal) is inserted between data signals on the transmission side in coherent optical communication. On the receiving side, the carrier phase is estimated from the reference signal. For symbols having a large phase variation in the data signal, it is disclosed that the carrier phase estimated based on the reference signal is linearly interpolated to guarantee the estimation accuracy of the carrier phase.
　特許文献４では、コヒーレント光通信において、送信側でデータ信号の合間に参照信号（ＳＹＮＣバースト）を挿入し、位相スリップにより発生しうる連続誤りは誤り訂正により救済することが開示されている。また、時系列順及び時系列逆順の両方向データ復号を重ね合わせて照合し、その後、各方向で復号される同一シンボルの復号結果に不整合があるかどうかを判定することで、位相スリップの発生を検知し、誤り訂正で認識して訂正することも開示されている。 Patent Document 4 discloses that in coherent optical communication, a reference signal (SYNC burst) is inserted between data signals on the transmission side, and a continuous error that may occur due to phase slip is relieved by error correction. In addition, phase slip can be generated by checking whether the decoding results of the same symbol decoded in each direction are inconsistent after overlapping and collating the two-way data decoding in the time series order and the time series reverse order. Is also detected and corrected by error correction.
ＵＳ　２０１１／０２１７０４３　Ａ１US 2011/0217043 A1 ＵＳ　２００６／０２４５７６６　Ａ１US 2006/0245766 A1 ＵＳ　２０１１／０２７４４４２　Ａ１US 2011/0274442 A1 ＵＳ　７５２２８４１　Ｂ２US 7522841 B2
　しかしながら、差動符号化・復号化を用いる従来技術によれば、伝送路での１シンボル誤りが伝搬しておよそ２シンボル誤りになり、符号誤り率を劣化させるため、誤り伝搬により雑音耐力が損われるという課題があった。 However, according to the prior art using differential encoding / decoding, one symbol error in the transmission channel propagates to about two symbol errors, which degrades the code error rate. There was a problem of being.
　特許文献２に記載の方法では、位相スリップの検出確度が不十分であり、見逃した位相スリップによる大規模連続誤りが不可避であるという課題があった。 The method described in Patent Document 2 has a problem that the detection accuracy of the phase slip is insufficient and a large-scale continuous error due to the missed phase slip is unavoidable.
　特許文献３及び特許文献４に記載の方法では、参照信号の挿入が必要であり、冗長度を増大させ、伝送速度の高速化を招くという課題があった。 In the methods described in Patent Document 3 and Patent Document 4, it is necessary to insert a reference signal, and there is a problem in that redundancy is increased and transmission speed is increased.
　特許文献４に記載の方法では、誤り訂正の連続誤り耐性を高めるために、誤り訂正符号の符号則に制約が加わり、ランダム誤りに対する耐性を低下させてしまうという課題があった。 In the method described in Patent Document 4, there is a problem that in order to increase the resistance to consecutive errors in error correction, a restriction is added to the coding rule of the error correction code and the resistance to random errors is reduced.
　本発明は、上記の課題を解決するためになされたものであって、位相スリップによる伝送品質劣化を防止し、伝送品質の向上および雑音耐力の改善を行うことを可能とする、光変復調方法および光送受信器を得ることを目的とする。 The present invention has been made in order to solve the above-described problem, an optical modulation / demodulation method capable of preventing transmission quality deterioration due to phase slip, improving transmission quality, and improving noise immunity, and The purpose is to obtain an optical transceiver.
　この発明は、光送信部により、Ｘ偏波およびＹ偏波の各直交偏波で非対称となる信号点配置を有する光信号を生成して、それらを多重化するステップと、前記光送信部により、多重化された前記光信号を送信するステップと、光受信部により、多重化された前記光信号を受信するステップと、前記光受信部により、所定の波長で発振する局部発振光を生成するステップと、前記光受信部により、受信した前記光信号と前記局部発振光とを混合して光／電気変換を行うとともに、アナログ／デジタル変換を行って、デジタル信号を出力するステップと、前記光受信部により、前記デジタル信号に基づいて、受信した前記光信号の搬送波周波数及び搬送波位相を前記Ｘ偏波および前記Ｙ偏波で同時に推定するステップと、推定された搬送波周波数及び搬送波位相に基づき、シンボル毎にとり得る位相スリップ状態ごとの周波数・位相補償後のデジタル信号を出力するステップと、前記周波数・位相補償後のデジタル信号に基づいて、前記位相スリップ状態ごとの尤度を演算するステップと、前記周波数・位相補償後のデジタル信号を、前記位相スリップ状態ごとに予め設定された閾値に基づいて復号し、前記位相スリップ状態ごとの復号結果を出力するステップと、前記位相スリップ状態ごとの復号結果の中から、前記尤度の中で最大の尤度の位相スリップ状態に対応する復号結果を選択して出力するステップとを備えたことを特徴とする光変復調方法である。 According to the present invention, an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them. A step of transmitting the multiplexed optical signal; a step of receiving the multiplexed optical signal by an optical receiver; and a local oscillation light that oscillates at a predetermined wavelength by the optical receiver. A step of mixing the received optical signal and the local oscillation light by the optical receiving unit to perform optical / electrical conversion, performing analog / digital conversion, and outputting a digital signal; and The receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency And a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase And a step of selecting and outputting a decoding result corresponding to a phase slip state having the maximum likelihood among the decoding results for each slip state. .
　この発明は、光送信部により、Ｘ偏波およびＹ偏波の各直交偏波で非対称となる信号点配置を有する光信号を生成して、それらを多重化するステップと、前記光送信部により、多重化された前記光信号を送信するステップと、光受信部により、多重化された前記光信号を受信するステップと、前記光受信部により、所定の波長で発振する局部発振光を生成するステップと、前記光受信部により、受信した前記光信号と前記局部発振光とを混合して光／電気変換を行うとともに、アナログ／デジタル変換を行って、デジタル信号を出力するステップと、前記光受信部により、前記デジタル信号に基づいて、受信した前記光信号の搬送波周波数及び搬送波位相を前記Ｘ偏波および前記Ｙ偏波で同時に推定するステップと、推定された搬送波周波数及び搬送波位相に基づき、シンボル毎にとり得る位相スリップ状態ごとの周波数・位相補償後のデジタル信号を出力するステップと、前記周波数・位相補償後のデジタル信号に基づいて、前記位相スリップ状態ごとの尤度を演算するステップと、前記周波数・位相補償後のデジタル信号を、前記位相スリップ状態ごとに予め設定された閾値に基づいて復号し、前記位相スリップ状態ごとの復号結果を出力するステップと、前記位相スリップ状態ごとの復号結果の中から、前記尤度の中で最大の尤度の位相スリップ状態に対応する復号結果を選択して出力するステップとを備えたことを特徴とする光変復調方法であるので、位相スリップによる伝送品質劣化を防止し、伝送品質の向上および雑音耐力の改善を行うことができる。 According to the present invention, an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them. A step of transmitting the multiplexed optical signal; a step of receiving the multiplexed optical signal by an optical receiver; and a local oscillation light that oscillates at a predetermined wavelength by the optical receiver. A step of mixing the received optical signal and the local oscillation light by the optical receiving unit to perform optical / electrical conversion, performing analog / digital conversion, and outputting a digital signal; and The receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency And a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase And a step of selecting and outputting a decoding result corresponding to a phase slip state having the maximum likelihood among the decoding results for each slip state. Therefore, transmission quality deterioration due to phase slip can be prevented, and transmission quality can be improved and noise tolerance can be improved.
本発明の実施の形態１にかかる光送受信構成を示す図である。It is a figure which shows the optical transmission / reception structure concerning Embodiment 1 of this invention. 偏波多重ＢＰＳＫ方式の偏波位相空間信号点配置を示す図である。It is a figure which shows polarization | polarized-light phase space signal point arrangement | positioning of a polarization multiplexing BPSK system. 本発明の実施の形態１にかかる偏波位相空間信号点配置を示す図である。It is a figure which shows polarization | polarized-light phase space signal point arrangement | positioning concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかる信号点配置を実現するシンボルマッパの入出力テーブルを示す図である。It is a figure which shows the input / output table of the symbol mapper which implement | achieves the signal point arrangement | positioning concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかる位相スリップ状態ごとの信号点配置を示す図である。It is a figure which shows signal point arrangement | positioning for every phase slip state concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかる復号部の詳細機能ブロック構成例を示す図である。It is a figure which shows the detailed functional block structural example of the decoding part concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかる位相スリップ状態推定部の詳細機能ブロック構成例を示す図である。It is a figure which shows the detailed functional block structural example of the phase slip state estimation part concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかる位相スリップ状態推定部内の尤度生成部の構成例を示す図である。It is a figure which shows the structural example of the likelihood production | generation part in the phase slip state estimation part concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかるシミュレーション条件を示す図である。It is a figure which shows the simulation conditions concerning Embodiment 1 of this invention. 本発明の実施の形態１にかかるシミュレーション結果を示す図である。It is a figure which shows the simulation result concerning Embodiment 1 of this invention.
　以下に、本発明に係る光送受信器及び光変復調方法の実施の形態を図面に基づいて詳細に説明する。なお、以下に説明する実施の形態は、本発明を具体化する際の一形態であって、本発明をその範疇に限定するためのものではない。 Embodiments of an optical transceiver and an optical modulation / demodulation method according to the present invention will be described below in detail with reference to the drawings. The embodiment described below is an embodiment for embodying the present invention, and is not intended to limit the present invention to its category.
　実施の形態１．
　図１は、本発明の実施の形態１に係る光送受信器の構成例を示す図である。図１に示すように、本実施の形態１の光送受信器は、光送信部１００と、伝送チャネル２００と、光受信部３００とで構成される。光送信部１００は、光源１０１と、シンボルマッパ１０２と、光変調部１０３とで構成される。光受信部３００は、光源３０１と、光復調部３０２と、波形等化部３０３と、周波数・位相推定部３０４（Ｘ／Ｙ結合）と、復号部３０５（Ｘ／Ｙ同時）と、位相スリップ状態推定部３０６と、選択部３０７とで構成される。光送信部１００は伝送チャネル２００を介して相手側の光送受信器（図示せず）に光信号を送信する。光受信部３００は伝送チャネル２００を介して相手側の光送受信器（図示せず）から送信されてきた光信号を受信する。なお、相手側の光送受信器は、図１では図示していないが、図１に示す光送受信器と同じ構成を有しているものとする。 Embodiment 1 FIG.
FIG. 1 is a diagram illustrating a configuration example of an optical transceiver according to Embodiment 1 of the present invention. As illustrated in FIG. 1, the optical transceiver according to the first embodiment includes an optical transmission unit 100, a transmission channel 200, and an optical reception unit 300. The optical transmission unit 100 includes a light source 101, a symbol mapper 102, and an optical modulation unit 103. The optical receiver 300 includes a light source 301, an optical demodulator 302, a waveform equalizer 303, a frequency / phase estimator 304 (X / Y combination), a decoder 305 (X / Y simultaneously), a phase slip The state estimation unit 306 and the selection unit 307 are configured. The optical transmitter 100 transmits an optical signal to a counterpart optical transceiver (not shown) via the transmission channel 200. The optical receiving unit 300 receives an optical signal transmitted from a partner optical transceiver (not shown) via the transmission channel 200. The counterpart optical transmitter / receiver is not shown in FIG. 1, but has the same configuration as the optical transmitter / receiver shown in FIG.
　以下、本実施の形態１に係る光変復調方法および光送受信器について説明する。 Hereinafter, the optical modulation / demodulation method and the optical transceiver according to the first embodiment will be described.
　まず、光送信部１００について説明する。光源１０１は、予め設定された所定の波長で発振する搬送波光信号を生成し、光変調部１０３に出力する。 First, the optical transmitter 100 will be described. The light source 101 generates a carrier light signal that oscillates at a predetermined wavelength set in advance, and outputs the carrier wave signal to the light modulator 103.
　シンボルマッパ１０２は、外部（図示せず）から入力される送信データ（２系列の符号Ａ，Ｂ、図４参照）に基づき、信号点配置を行い、４レーンの２値電気信号を出力する。 The symbol mapper 102 performs signal point arrangement based on transmission data (two series of codes A and B, see FIG. 4) input from the outside (not shown), and outputs a 4-lane binary electrical signal.
　図２は、偏波多重ＢＰＳＫ（ＤＰ－ＢＰＳＫ：Ｄｕａｌ－Ｐｏｌａｒｉｚａｔｉｏｎ　ＢＰＳＫ）方式の偏波位相空間における信号点配置である。横軸は直交偏波多重する際の一方の偏波成分であるＸ偏波の光位相を示し、縦軸は同じく直交偏波多重する際の他方の偏波成分であるＹ偏波の光位相を示す。通常のＢＰＳＫは、位相０および位相πの２値で表示されるが、図２では、それらをπ／４回転させた位相π／４および位相－３π／４で示す。Ｘ偏波、Ｙ偏波それぞれ２値の位相をとるため、図２上で、０、１、２、３の４つの信号点（シンボル）をとり得る。これらの信号点の配置は、Ｘ偏波とＹ偏波とで対称（点対称、線対称）となっている。 FIG. 2 is a signal point arrangement in the polarization phase space of the polarization-multiplexed BPSK (DP-BPSK: Dual-Polarization BPSK) method. The horizontal axis represents the optical phase of X polarization, which is one polarization component when orthogonal polarization multiplexing is performed, and the vertical axis is the optical phase of the Y polarization, which is the other polarization component when orthogonal polarization multiplexing is performed. Indicates. Ordinary BPSK is represented by binary values of phase 0 and phase π. In FIG. 2, these are indicated by phase π / 4 and phase -3π / 4 obtained by rotating them by π / 4. Since each of the X polarization and the Y polarization takes a binary phase, four signal points (symbols) 0, 1, 2, and 3 can be taken in FIG. The arrangement of these signal points is symmetrical (point symmetry, line symmetry) between the X polarization and the Y polarization.
　図３は、本発明の実施の形態１に係る、偏波位相空間における信号点配置である。図３に示すように、信号点配置が、Ｘ偏波とＹ偏波とで非対称配置となっており、Ｘ偏波とＹ偏波とで互いに異なる信号点配置となっている。具体的には、図３に示すように、ＤＰ－ＢＰＳＫにおいて、Ｘ偏波の位相が－３π／４の場合（信号点１、３）のみ、図２の信号点配置から、Ｙ偏波の位相を＋π／２シフトさせる。なお、Ｘ偏波の位相がπ／４の場合（信号点０、２）は、図２の信号点配置と同じＹ偏波の位相である。これにより、各信号点０，１，２，３の配置は、点対称でもなく、線対称でもなくなる。このように、本実施の形態１においては、Ｘ／Ｙ偏波の両方の信号点配置が２値位相偏移変調であり、Ｘ／Ｙ偏波の一方の２値位相偏移変調の片側位相の場合のみ、Ｘ／Ｙ偏波の他方の信号点をπ／２ずらす。こうして、本実施の形態１では、各信号点の配置が、Ｘ偏波とＹ偏波とで、非対称になっている。 FIG. 3 is a signal point arrangement in the polarization phase space according to Embodiment 1 of the present invention. As shown in FIG. 3, the signal point arrangement is an asymmetric arrangement between the X polarization and the Y polarization, and the signal point arrangement differs between the X polarization and the Y polarization. Specifically, as shown in FIG. 3, in DP-BPSK, only when the phase of X polarization is −3π / 4 (signal points 1 and 3), from the signal point arrangement of FIG. Shift the phase by + π / 2. When the phase of the X polarization is π / 4 (signal points 0 and 2), the phase of the Y polarization is the same as the signal point arrangement in FIG. As a result, the arrangement of the signal points 0, 1, 2, and 3 is neither point-symmetric nor line-symmetric. As described above, in the first embodiment, the signal point arrangement of both X / Y polarization is binary phase shift keying, and one side phase of one binary phase shift keying of X / Y polarization is performed. Only in this case, the other signal point of the X / Y polarization is shifted by π / 2. Thus, in the first embodiment, the arrangement of the signal points is asymmetric between the X polarization and the Y polarization.
　本発明の実施の形態１に係る信号点配置の入出力関係を図４に示す。光送信部１００において、シンボルマッパ１０２は、外部（図示せず）から、２系列の符号Ａおよび符号Ｂが送信データとして入力され、４レーンの２値（＋／－）の電気信号ＸＩ、ＸＱ、ＹＩ、ＹＱを生成し、光変調部１０３に対して出力する。Ｙ偏波は見かけ上ＱＰＳＫとなるが、ＱＰＳＫ側をグレイ符号していないのは、図３から分かるように、シンボル０―２間や１―３間よりも、０－１間、０－３間、２－１間、２－３間の方が偏波位相空間距離が遠いので、遠い側のハミング距離を大きくする意図による。ただし、本発明の範疇を、この符号化則に限定するものではない。この処理は、Ａ、Ｂを入力としてＸＩ、ＸＱ、ＹＩ、ＹＱを出力とするテーブル処理により実現可能である。あるいは、以下の論理演算により求めることも可能である。
ＸＩ＝ＮＯＴ｛Ｂ｝
ＸＱ＝ＮＯＴ｛Ｂ｝
ＹＩ＝ＥＸＯＲ｛Ｂ｝
ＹＱ＝ＮＯＴ｛Ａ｝ The input / output relationship of the signal point arrangement according to Embodiment 1 of the present invention is shown in FIG. In the optical transmitter 100, the symbol mapper 102 receives two series of codes A and B as transmission data from the outside (not shown), and is a 4-lane binary (+/−) electrical signal XI, XQ. , YI, YQ are generated and output to the light modulation unit 103. The Y polarization is apparently QPSK, but the QPSK side is not gray-coded, as can be seen from FIG. 3, between 0-1 and 0-3 rather than between symbols 0-2 and 1-3. Since the polarization phase space distance is longer between 2-1 and 2-1 and 2-3, this is due to the intention of increasing the hamming distance on the far side. However, the scope of the present invention is not limited to this encoding rule. This processing can be realized by table processing in which A and B are input and XI, XQ, YI, and YQ are output. Alternatively, it can be obtained by the following logical operation.
XI = NOT {B}
XQ = NOT {B}
YI = EXOR {B}
YQ = NOT {A}
　図４に示すように、シンボル０では、符号Ａが“０”、符号Ｂが“０”で、Ｘ位相がπ／４、Ｙ位相がπ／４となっている。このとき、シンボルマッパ１０２は、ＸＩが“＋”、ＸＱが“＋”、ＹＩが“＋”、ＹＱが“＋”の４レーンの２値信号を生成する。シンボル１，２，３についても、同様であり、それぞれ、図４に示す符号Ａ，Ｂが入力され、それらの基づき、図４に示す４レーンの２値信号を生成する。 As shown in FIG. 4, in the symbol 0, the code A is “0”, the code B is “0”, the X phase is π / 4, and the Y phase is π / 4. At this time, the symbol mapper 102 generates a four-lane binary signal with XI “+”, XQ “+”, YI “+”, and YQ “+”. The same applies to the symbols 1, 2, and 3, and the codes A and B shown in FIG. 4 are inputted, respectively, and based on them, a binary signal of 4 lanes shown in FIG. 4 is generated.
　光変調部１０３は、光源１０１から入力する搬送波光信号を、シンボルマッパ１０２から入力する４レーンの２値電気信号（ＸＩ、ＸＱ、ＹＩ、ＹＱ）で変調し、それらを多重化して、図３に示す信号点配置の光信号を生成する。シンボルマッパ１０２から入力される信号（２値電気信号）は、一般に、ドライバ（図示せず）で増幅されて、光変調部１０３を駆動されるが、図１の構成では、ドライバは省略されている。光変調部１０３は、光源１０１から入力される搬送波を２系統に分け、それぞれ、シンボルマッパ１０２からの２値電気信号で直交位相（Ｉ／Ｑ）変調し、直交偏波多重化する。また、送信部１００に、デジタル・アナログ変換部（図示せず）を備えて、送信端で波形歪みの予補償等のデジタル信号処理を行うようにしてもよい。 The optical modulation unit 103 modulates the carrier optical signal input from the light source 101 with the 4-lane binary electrical signals (XI, XQ, YI, YQ) input from the symbol mapper 102, multiplexes them, and FIG. An optical signal having the signal point arrangement shown in FIG. A signal (binary electrical signal) input from the symbol mapper 102 is generally amplified by a driver (not shown) to drive the light modulator 103, but the driver is omitted in the configuration of FIG. Yes. The optical modulation unit 103 divides the carrier wave input from the light source 101 into two systems, respectively performs quadrature phase (I / Q) modulation with the binary electric signal from the symbol mapper 102, and performs orthogonal polarization multiplexing. Further, the transmission unit 100 may be provided with a digital / analog conversion unit (not shown) to perform digital signal processing such as predistortion of waveform distortion at the transmission end.
　伝送チャネル２００は、光送信部１００内の光変調部１０３から入力される光信号を伝送し、相手側の光送受信器の光受信部３００に出力する。伝送チャネル２００は、波長合分波装置、光増幅装置、伝送路光ファイバ等、一般に光信号の伝送に用いられる装置および部品を備える。 The transmission channel 200 transmits an optical signal input from the optical modulation unit 103 in the optical transmission unit 100 and outputs the optical signal to the optical reception unit 300 of the counterpart optical transceiver. The transmission channel 200 includes devices and components generally used for transmitting optical signals, such as a wavelength multiplexing / demultiplexing device, an optical amplifying device, and a transmission line optical fiber.
　次に、光受信部３００について説明する。光受信部３００において、光源３０１は、予め設定された所定の波長で発振する局部発振光信号を生成し、光復調部３０２に出力する。一般に、局部発振光信号は、光源１０１で生成される搬送波光信号と概略同一の波長で発振させる。 Next, the optical receiver 300 will be described. In the optical receiving unit 300, the light source 301 generates a local oscillation optical signal that oscillates at a predetermined wavelength set in advance, and outputs the local oscillation optical signal to the optical demodulation unit 302. In general, the local oscillation optical signal is oscillated at substantially the same wavelength as the carrier optical signal generated by the light source 101.
　光復調部３０２は、伝送チャネル２００を介して伝送されてきた光信号と、光源３０１から入力される局部発振光信号とを、直交偏波成分及び直交位相成分に分解する。次に、光復調部３０２は、分解した光信号と局部発振光信号とを混合して光電気変換を行って、４レーンの電気信号に変換するコヒーレント検波を行う。さらに、それらの電気信号を、アナログ・デジタル変換によりデジタル信号に変換して、量子化・標本化する。そうして得られた４レーンのデジタル信号を波形等化部３０３に出力する。前記アナログ・デジタル変換は通常６ビット以上の分解能で量子化を行い、ボーレートの２倍以上のサンプリングレートで標本化を行う。 The optical demodulator 302 decomposes the optical signal transmitted via the transmission channel 200 and the local oscillation optical signal input from the light source 301 into orthogonal polarization components and orthogonal phase components. Next, the optical demodulator 302 performs coherent detection that mixes the decomposed optical signal and the local oscillation optical signal, performs photoelectric conversion, and converts the signal into a four-lane electrical signal. Further, these electric signals are converted into digital signals by analog / digital conversion, and quantized and sampled. The 4-lane digital signal thus obtained is output to the waveform equalization unit 303. In the analog-digital conversion, quantization is usually performed with a resolution of 6 bits or more, and sampling is performed at a sampling rate that is twice or more the baud rate.
　波形等化部３０３は、光復調部３０２から４レーンのデジタル信号が入力され、伝送チャネル２００で生じる波長分散、偏波回転、偏波モード分散、または、ファイバ非線形光学効果等に起因する波形歪みを補償し、直交偏波分離した４レーンのデジタル信号を、周波数・位相推定部３０４に出力する。 The waveform equalization unit 303 receives a 4-lane digital signal from the optical demodulation unit 302 and generates waveform distortion caused by chromatic dispersion, polarization rotation, polarization mode dispersion, fiber nonlinear optical effect, or the like generated in the transmission channel 200. And a 4-lane digital signal separated by orthogonal polarization is output to the frequency / phase estimator 304.
　周波数・位相推定部３０４は、波形等化部３０３から４レーンのデジタル信号が入力され、それらを直交偏波間（Ｘ／Ｙ偏波間）で結合して、搬送波周波数・位相を推定して補償する。すなわち、周波数・位相推定部３０４は、波形等化部３０３から入力される４レーンのデジタル信号に存在する、前記搬送波光信号と前記局部発振光信号との中心周波数差を補償し、前記Ｍ乗法や前記仮判定法等に基づき、搬送波周波数推定および搬送波位相推定を行い、周波数・位相補償後の４レーンのデジタル信号を復号部３０５及び位相スリップ状態推定部３０６に出力する。ここで、これらの搬送波周波数推定や搬送波位相推定は、直交偏波間で独立に行うのではなく、直交偏波間（Ｘ／Ｙ偏波間）で結合して同時に行うことに特徴がある。 The frequency / phase estimation unit 304 receives 4-lane digital signals from the waveform equalization unit 303 and combines them between orthogonal polarizations (between X / Y polarizations) to estimate and compensate for the carrier frequency / phase. . That is, the frequency / phase estimation unit 304 compensates for the center frequency difference between the carrier optical signal and the local oscillation optical signal, which is present in the 4-lane digital signal input from the waveform equalization unit 303, and the M multiplication method. Also, carrier frequency estimation and carrier phase estimation are performed based on the provisional determination method and the like, and the 4-lane digital signal after frequency / phase compensation is output to the decoding unit 305 and the phase slip state estimation unit 306. Here, the carrier frequency estimation and the carrier phase estimation are not performed independently between the orthogonal polarizations, but are performed simultaneously by combining between the orthogonal polarizations (between the X / Y polarizations).
　搬送波周波数推定および搬送波位相推定をＸ／Ｙ偏波で同時に行った場合、位相スリップ状態には、（１）Ｘ／Ｙ偏波ともに位相がスリップしていない状態、（２）Ｘ／Ｙ偏波ともに＋９０度スリップした状態、（３）Ｘ／Ｙ偏波ともに１８０度スリップした状態、（４）Ｘ／Ｙ偏波ともに－９０度スリップした状態の、計４通りの状態があり得る。もし、Ｘ／Ｙ偏波で独立に搬送波周波数・位相推定してしまうと、それぞれの偏波でスリップが独立に生じてしまうため、この４通りに収まらず、４の２乗通り（４2＝１６通り）となり、後述する位相スリップ検出が困難となる。 When carrier frequency estimation and carrier phase estimation are performed simultaneously with X / Y polarization, the phase slip state includes (1) a state in which the phase is not slipped in both X / Y polarization, and (2) X / Y polarization. There can be a total of four states: a state where both slip +90 degrees, a state where (3) both X / Y polarizations slip 180 degrees, and a state where both (4) X / Y polarizations slip -90 degrees. If the carrier frequency and phase are estimated independently for X / Y polarization, slips will occur independently for each polarization. Therefore, the slip does not fit in these four ways, but the square of 4 (4 2 = 16), and phase slip detection described later becomes difficult.
　Ｘ／Ｙ偏波で結合して搬送波周波数・位相推定を同時に行うことにより得られる４通りの位相スリップ状態に対し、周波数・位相補償後の信号点配置は、それぞれ、図５に示す通りとなる。図５において、位相スリップ状態Ｓｔａｔｅ－０は位相スリップがない場合、位相スリップ状態Ｓｔａｔｅ－１は位相スリップ＋９０度の場合、位相スリップ状態Ｓｔａｔｅ－２は位相スリップ＋１８０度の場合、位相スリップ状態Ｓｔａｔｅ－３は位相スリップ－９０度の場合である。各位相スリップ状態間で信号点に重なりがなく、各位相スリップ状態ごとの固有の信号点配置となることがわかる。また、これらの信号点配置は、位相スリップ無しの場合には、とり得ない信号点配置であるため、位相スリップ状態の検出・補償が可能になる。Ｓｔａｔｅ－１におけるシンボルＺ’（Ｚ’＝０’、１’、２’、３’）は、Ｓｔａｔｅ－０におけるシンボルＺ（Ｚ＝０、１、２、３）が＋９０度スリップしたものである。Ｓｔａｔｅ－２におけるシンボルＺ’’（Ｚ’’＝０’’、１’’、２’’、３’’）は、シンボルＺが１８０度スリップしたものである。Ｓｔａｔｅ－３におけるシンボルＺ’’’（Ｚ’’’＝０’’’、１’’’、２’’’、３’’’）は、シンボルＺが－９０度スリップしたものである。 FIG. 5 shows signal point arrangements after frequency / phase compensation for four types of phase slip states obtained by combining with X / Y polarization and simultaneously performing carrier frequency / phase estimation. . In FIG. 5, the phase slip state State-0 is when there is no phase slip, the phase slip state State-1 is when the phase slip +90 degrees, the phase slip state State-2 is when the phase slip +180 degrees, and the phase slip state State- 3 is a case where the phase slip is -90 degrees. It can be seen that the signal points do not overlap between the phase slip states, and the signal point arrangement is unique for each phase slip state. In addition, since these signal point arrangements are signal point arrangements that cannot be taken when there is no phase slip, it is possible to detect and compensate for a phase slip state. Symbol Z ′ (Z ′ = 0 ′, 1 ′, 2 ′, 3 ′) in State-1 is obtained by slipping +90 degrees on symbol Z (Z = 0, 1, 2, 3) in State-0. . A symbol Z ″ (Z ″ = 0 ″, 1 ″, 2 ″, 3 ″) in State-2 is a symbol Z slipped 180 degrees. The symbol Z ″ ″ (Z ″ ″ = 0 ″ ″, 1 ″ ″, 2 ″ ″, 3 ″ ″) in State-3 is obtained by slipping the symbol Z by −90 degrees.
　復号部３０５では、シンボル毎にとり得るすべての位相スリップ状態について復号を行う。すなわち、復号部３０５では、図５の各Ｓｔａｔｅの信号点配置に点線で示すしきい値に基づき、周波数・位相推定部３０４から入力される４レーンのデジタル信号を復号し、４通りの復号結果を選択部３０７に出力する。 The decoding unit 305 performs decoding for all possible phase slip states for each symbol. That is, the decoding unit 305 decodes the 4-lane digital signal input from the frequency / phase estimation unit 304 based on the threshold value indicated by the dotted line in the signal point arrangement of each State in FIG. Is output to the selection unit 307.
　図６は、復号部３０５の構成例を図示したものである。復号部３０５は、４つの二次元位相識別器（2-D Phase Slicer）５００、５０１、５０２、５０３で構成される。周波数・位相推定部３０４から入力される４レーンのデジタル信号は、等しく二次元位相識別器５００、５０１、５０２、５０３に入力される。二次元位相識別器５００は、Ｓｔａｔｅ－０に対応して４レーンのデジタル信号の信号毎に復号を行い、復号結果を選択部３０７（図１参照）に出力する。二次元位相識別器５０１は、Ｓｔａｔｅ－１に対応して４レーンのデジタル信号の信号毎に復号を行い、復号結果を選択部３０７に出力する。二次元位相識別器５０２は、Ｓｔａｔｅ－２に対応して４レーンのデジタル信号の信号毎に復号を行い、復号結果を選択部３０７に出力する。二次元位相識別器５０３は、Ｓｔａｔｅ－３に対応して４レーンのデジタル信号の信号毎に復号を行い、復号結果を選択部３０７に出力する。シンボルＺ’，Ｚ’’，Ｚ’’’はそれぞれシンボルＺとして復号する。前記復号は、硬判定のみならず、偏波位相空間における信号点中心からの距離に応じて、信頼度情報を付与する軟判定として、軟値を選択部３０７に渡してもよい。 FIG. 6 illustrates a configuration example of the decoding unit 305. The decoding unit 305 includes four two-dimensional phase identifiers (2-D2-Phase Slicers) 500, 501, 502, and 503. The 4-lane digital signals input from the frequency / phase estimation unit 304 are equally input to the two- dimensional phase identifiers 500, 501, 502, and 503. The two-dimensional phase discriminator 500 performs decoding for each 4-lane digital signal corresponding to State-0, and outputs the decoding result to the selection unit 307 (see FIG. 1). The two-dimensional phase identifier 501 performs decoding for each signal of the 4-lane digital signal corresponding to State-1, and outputs the decoding result to the selection unit 307. The two-dimensional phase discriminator 502 performs decoding for each signal of the 4-lane digital signal corresponding to State-2, and outputs the decoding result to the selection unit 307. The two-dimensional phase discriminator 503 performs decoding for each digital lane signal corresponding to State-3 and outputs the decoding result to the selection unit 307. Symbols Z ′, Z ″, and Z ″ ″ are decoded as symbols Z, respectively. In the decoding, not only a hard decision but also a soft value may be passed to the selection unit 307 as a soft decision to give reliability information according to the distance from the signal point center in the polarization phase space.
　位相スリップ状態推定部３０６は、信号（シンボル）毎にとり得るすべての位相スリップ状態について尤度を演算し、それらを比較し、最大尤度を有する位相スリップ状態がいずれであるかを推定する。すなわち、位相スリップ状態推定部３０６は、周波数・位相推定部３０４から入力される４レーンのデジタル信号に基づき、最大尤度の位相スリップ状態を推定し、選択部３０７に出力する。 The phase slip state estimation unit 306 calculates likelihoods for all possible phase slip states for each signal (symbol), compares them, and estimates which phase slip state has the maximum likelihood. That is, the phase slip state estimation unit 306 estimates the maximum likelihood phase slip state based on the 4-lane digital signal input from the frequency / phase estimation unit 304, and outputs it to the selection unit 307.
　図７は、位相スリップ状態推定部３０６の構成例を図示したものである。位相スリップ状態推定部３０６は、４つの尤度生成部（Metric Generator）６００、６０１、６０２、６０３と、１つの最尤状態推定部（Maximum Likelihood State Estimator）６１０とで構成される。周波数・位相推定部３０４（図１参照）から入力される４レーンのデジタル信号は、等しく尤度生成部６００、６０１、６０２、６０３に出力される。尤度生成部６００では、位相スリップ状態がＳｔａｔｅ－０であることの尤度を計算し、尤度情報を最尤状態推定部６１０に出力する。尤度生成部６０１では、位相スリップ状態がＳｔａｔｅ－１であることの尤度を計算し、尤度情報を最尤状態推定部６１０に出力する。尤度生成部６０２では、位相スリップ状態がＳｔａｔｅ－２であることの尤度を計算し、尤度情報を最尤状態推定部６１０に出力する。尤度生成部６０３では、位相スリップ状態がＳｔａｔｅ－３であることの尤度を計算し、尤度情報を最尤状態推定部６１０に出力する。最尤状態推定部６１０は、尤度生成部６００、６０１、６０２、６０３から入力される各位相スリップ状態Ｓｔａｔｅ－０，Ｓｔａｔｅ－１，Ｓｔａｔｅ－２，Ｓｔａｔｅ－３の尤度情報に基づき、位相スリップ状態Ｓｔａｔｅ－０，Ｓｔａｔｅ－１，Ｓｔａｔｅ－２，Ｓｔａｔｅ－３の中から、最も尤度の高い位相スリップ状態を求めて、選択部３０７（図７では図示省略）に出力する。 FIG. 7 illustrates a configuration example of the phase slip state estimation unit 306. The phase slip state estimation unit 306 includes four likelihood generation units (MetricMeGenerators) 600, 601, 602, and 603 and one maximum likelihood state estimation unit (Maximum Likelihood State Estimator) 610. The 4-lane digital signals input from the frequency / phase estimation unit 304 (see FIG. 1) are equally output to the likelihood generation units 600, 601, 602, and 603. The likelihood generation unit 600 calculates the likelihood that the phase slip state is State-0, and outputs the likelihood information to the maximum likelihood state estimation unit 610. The likelihood generation unit 601 calculates the likelihood that the phase slip state is State-1, and outputs the likelihood information to the maximum likelihood state estimation unit 610. The likelihood generation unit 602 calculates the likelihood that the phase slip state is State-2, and outputs the likelihood information to the maximum likelihood state estimation unit 610. The likelihood generation unit 603 calculates the likelihood that the phase slip state is State-3, and outputs the likelihood information to the maximum likelihood state estimation unit 610. The maximum likelihood state estimation unit 610 is based on the likelihood information of each phase slip state State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603. The most likely phase slip state is obtained from the slip states State-0, State-1, State-2, and State-3, and is output to the selection unit 307 (not shown in FIG. 7).
　図８は、位相スリップ状態推定部３０６内の尤度生成部６００、６０１、６０２、６０３の構成例を図示したものである。尤度生成部６００、６０１、６０２、６０３は、それぞれ同様の機能ブロックを備えるため、ここではＳｔａｔｅ－０の尤度を計算する尤度生成部６００の場合について説明する。 FIG. 8 illustrates a configuration example of the likelihood generation units 600, 601, 602, and 603 in the phase slip state estimation unit 306. Since the likelihood generation units 600, 601, 602, and 603 each have the same functional block, the case of the likelihood generation unit 600 that calculates the likelihood of State-0 will be described here.
　尤度生成部６００は、４つの偏波位相距離計算部７００、７０１、７０２、７０３と、１つの選択部７１０と、ｎ個の遅延部７２０Ａ、７２０Ｂ、・・・、７２０Ｘと、ｎ個の乗算部７２１Ａ、７２１Ｂ、・・・、７２１Ｘと、１つの加算部（sum）７２２とで構成される。遅延部７２０Ａ、７２０Ｂ、・・・、７２０Ｘは、図８に示すように、順に、直列に接続されている。乗算部７２１Ａ、７２１Ｂ、・・・、７２１Ｘは、図８に示すように、それぞれ、遅延部７２０Ａ、７２０Ｂ、・・・、７２０Ｘに接続されている。周波数・位相推定部３０４から入力される４レーンのデジタル信号は、等しく偏波位相距離計算部７００、７０１、７０２、７０３に出力される。 The likelihood generation unit 600 includes four polarization phase distance calculation units 700, 701, 702, and 703, one selection unit 710, n delay units 720A, 720B,. , 721X and one adder (sum) 722. The multipliers 721A, 721B,. The delay units 720A, 720B,..., 720X are connected in series as shown in FIG. The multipliers 721A, 721B,..., 721X are connected to delay units 720A, 720B,. The 4-lane digital signals input from the frequency / phase estimation unit 304 are equally output to the polarization phase distance calculation units 700, 701, 702, and 703.
　偏波位相空間距離計算部７００は、入力された４レーンのデジタル信号に対し、位相スリップのない状態であるＳｔａｔｅ－０の信号点配置規則に基づき、中心となる信号点０からの偏波位相空間距離を求める。例えば、Ｘ偏波方向の位相距離Ｄ＿ｐｈ＿ｘ（０）とＹ偏波方向の位相距離Ｄ＿ｐｈ＿ｙ（０）との二乗和（（Ｄ＿ｐｈ＿ｘ（０））2＋（Ｄ＿ｐｈ＿ｙ（０））2）をとり、演算結果を選択部７１０に出力する。偏波位相空間距離計算部７０１は、４レーンのデジタル信号に対し、位相スリップのない状態であるＳｔａｔｅ－０の信号点配置規則に基づき、信号点１からの偏波位相空間距離を求める。例えば、Ｘ偏波方向の位相距離Ｄ＿ｐｈ＿ｘ（１）とＹ偏波方向の位相距離Ｄ＿ｐｈ＿ｙ（１）との二乗和（（Ｄ＿ｐｈ＿ｘ（１））2＋（Ｄ＿ｐｈ＿ｙ（１））2）をとり、演算結果を選択部７１０に出力する。偏波位相空間距離計算部７０２は、４レーンのデジタル信号に対し、位相スリップのない状態であるＳｔａｔｅ－０の信号点配置規則に基づき、信号点２からの偏波位相空間距離を求める。例えば、Ｘ偏波方向の位相距離Ｄ＿ｐｈ＿ｘ（２）とＹ偏波方向の位相距離Ｄ＿ｐｈ＿ｙ（２）との二乗和（（Ｄ＿ｐｈ＿ｘ（２））2＋（Ｄ＿ｐｈ＿ｙ（２））2）をとり、演算結果を選択部７１０に出力する。偏波位相空間距離計算部７０３は、４レーンのデジタル信号に対し、位相スリップのない状態であるＳｔａｔｅ－０の信号点配置規則に基づき、信号点３からの偏波位相空間距離を求める。例えば、Ｘ偏波方向の位相距離Ｄ＿ｐｈ＿ｘ（３）とＹ偏波方向の位相距離Ｄ＿ｐｈ＿ｙ（３）との二乗和（（Ｄ＿ｐｈ＿ｘ（３））2＋（Ｄ＿ｐｈ＿ｙ（３））2）をとり、演算結果を選択部７１０に出力する。 The polarization phase space distance calculation unit 700 applies the polarization phase from the central signal point 0 to the input 4-lane digital signal based on the State-0 signal point arrangement rule in which there is no phase slip. Find the spatial distance. For example, the sum of squares of the phase distance D_ph_x (0) in the X polarization direction and the phase distance D_ph_y (0) in the Y polarization direction ((D_ph_x (0)) 2 + (D_ph_y (0)) 2 ) is calculated. The result is output to the selection unit 710. The polarization phase space distance calculation unit 701 obtains the polarization phase space distance from the signal point 1 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (1)) 2 + (D_ph_y (1)) 2 ) of the phase distance D_ph_x (1) in the X polarization direction and the phase distance D_ph_y (1) in the Y polarization direction is calculated. The result is output to the selection unit 710. The polarization phase space distance calculation unit 702 obtains the polarization phase space distance from the signal point 2 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (2)) 2 + (D_ph_y (2)) 2 ) of the phase distance D_ph_x (2) in the X polarization direction and the phase distance D_ph_y (2) in the Y polarization direction is calculated. The result is output to the selection unit 710. The polarization phase space distance calculation unit 703 obtains the polarization phase space distance from the signal point 3 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (3)) 2 + (D_ph_y (3)) 2 ) of the phase distance D_ph_x (3) in the X polarization direction and the phase distance D_ph_y (3) in the Y polarization direction is calculated. The result is output to the selection unit 710.
　選択部７１０は、偏波位相空間距離計算部７００、７０１、７０２、７０３から入力される４つの偏波位相空間距離の中から、最小の偏波位相空間距離を選択し、遅延部７２０Ａに出力する。 The selection unit 710 selects the minimum polarization phase space distance from the four polarization phase space distances input from the polarization phase space distance calculation units 700, 701, 702, and 703, and outputs the selected polarization phase space distance to the delay unit 720A. To do.
　遅延部７２０Ａは、選択部７１０から入力されるデジタル信号を１シンボル時間保持し、遅延部７２０Ｂ及び乗算部７２１Ａに出力する。 The delay unit 720A holds the digital signal input from the selection unit 710 for one symbol time, and outputs the digital signal to the delay unit 720B and the multiplication unit 721A.
　乗算部７２１Ａは、遅延部７２０Ａから入力されるデジタル信号に、係数Ｃ［０］を掛け合わせた結果を加算部７２２に出力する。 Multiplication unit 721A outputs the result of multiplying the digital signal input from delay unit 720A by coefficient C [0] to addition unit 722.
　遅延部７２０Ｂは、遅延部７２０Ａから入力されるデジタル信号を１シンボル時間保持し、遅延部７２０Ｂに直列接続された遅延部７２０Ｃ（図示せず）及び乗算部７２１Ｂに出力する。 The delay unit 720B holds the digital signal input from the delay unit 720A for one symbol time, and outputs the digital signal to the delay unit 720C (not shown) and the multiplication unit 721B connected in series to the delay unit 720B.
　乗算部７２１Ｂは、遅延部７２０Ｂから入力されるデジタル信号に、係数Ｃ［１］を掛け合わせた結果を加算部７２２に出力する。 The multiplier 721B outputs the result of multiplying the digital signal input from the delay unit 720B by the coefficient C [1] to the adder 722.
　以降、同様の処理を繰り返し、ｎ番目の遅延部７２０Ｘは、（ｎ－１）番目の遅延部７２０Ｗ（図示せず）から入力されるデジタル信号を１シンボル時間保持し、乗算部７２１Ｘに出力する。 Thereafter, the same processing is repeated, and the n-th delay unit 720X holds the digital signal input from the (n−1) -th delay unit 720W (not shown) for one symbol time, and outputs the digital signal to the multiplication unit 721X. .
　乗算部７２１Ｘは、遅延部７２０Ｘから入力されるデジタル信号に、係数Ｃ［ｎ－１］を掛け合わせた結果を加算部７２２に出力する。 The multiplication unit 721X outputs the result of multiplying the digital signal input from the delay unit 720X by the coefficient C [n−1] to the addition unit 722.
　加算部７２２は、乗算部７２１Ａ、７２１Ｂ、・・・、７２１Ｘから入力されるｎ個の乗算結果の総和をとり、当該総和の値を最尤状態推定部６１０（図７参照）に出力する。ここで、総和の値が小さいほど尤度が高いことに注意を要する。 The adding unit 722 calculates the sum of the n multiplication results input from the multiplying units 721A, 721B,..., 721X, and outputs the value of the sum to the maximum likelihood state estimating unit 610 (see FIG. 7). Here, it should be noted that the smaller the sum value, the higher the likelihood.
　遅延部７２０Ａ、７２０Ｂ、・・・、７２０Ｘ、乗算部７２１Ａ、７２１Ｂ、・・・、７２１Ｘ、および、加算部７２２は、有限インパルス応答フィルタを構成しており、当該有限インパルス応答フィルタにより、各シンボル時点の瞬時尤度を平均化している。従って、遅延部７２０Ａ、７２０Ｂ、・・・、７２０Ｘ、乗算部７２１Ａ、７２１Ｂ、・・・、７２１Ｘ、および、加算部７２２は、選択部７１０から出力される信号を平均化する平均化部を構成している。なお、平均化部を構成するフィルタは、有限インパルス応答フィルタに限らず、忘却型の無限インパルス応答フィルタを用いることも可能である。その場合には、より複雑に、最尤系列推定や事後確率最大化を行うことも可能である。 The delay units 720A, 720B,..., 720X, the multipliers 721A, 721B,..., 721X, and the adder 722 constitute a finite impulse response filter. The instantaneous likelihood at the time is averaged. Therefore, delay units 720A, 720B,..., 720X, multiplication units 721A, 721B,..., 721X, and addition unit 722 constitute an averaging unit that averages the signal output from selection unit 710. is doing. The filter constituting the averaging unit is not limited to a finite impulse response filter, and a forgetting infinite impulse response filter may be used. In that case, maximum likelihood sequence estimation and posterior probability maximization can be performed more complicatedly.
　尤度生成部６０１、６０２、６０３においても、尤度生成部６００における処理と同様の処理を行う。ここでは、その説明は省略する。 In the likelihood generation units 601, 602, and 603, the same processing as the processing in the likelihood generation unit 600 is performed. Here, the description is omitted.
　こうして、尤度生成部６００、６０１、６０２、６０３から４つの総和の値が最尤状態推定部６１０に入力される。最尤状態推定部６１０は、尤度生成部６００、６０１、６０２、６０３から入力される各位相スリップ状態Ｓｔａｔｅ－０，Ｓｔａｔｅ－１，Ｓｔａｔｅ－２，Ｓｔａｔｅ－３の総和の値の中から、最も総和の値が小さい位相スリップ状態を、「最尤の位相スリップ状態」として求めて、選択部３０７（図１参照）に出力する。 Thus, four total sum values are input to the maximum likelihood state estimation unit 610 from the likelihood generation units 600, 601, 602, and 603. The maximum likelihood state estimation unit 610 includes a total sum of phase slip states State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603, The phase slip state having the smallest sum value is obtained as the “maximum likelihood phase slip state” and is output to the selection unit 307 (see FIG. 1).
　選択部３０７では、位相スリップ状態推定部３０６が推定する位相スリップ状態に基づき、復号部３０５から入力される４つの復号結果を比較して、それらの復号結果の中から、当該位相スリップ状態に対応する復号結果を、最大尤度の復号結果として選択し、光送受信器に接続された外部装置（図示せず）に出力する。 The selection unit 307 compares the four decoding results input from the decoding unit 305 based on the phase slip state estimated by the phase slip state estimation unit 306, and corresponds to the phase slip state from those decoding results. The decoding result to be selected is selected as the maximum likelihood decoding result, and is output to an external device (not shown) connected to the optical transceiver.
　本実施の形態を用いた場合の符号誤り率特性のシミュレーションを行った。シミュレーション条件を図９に示す。ビットレートを１２８Ｇｂ／ｓとし、符号系列は１１段疑似ランダム２進系列を用いた。シンボル数は１６３８４とし、１００回雑音パタンを変えて繰り返し計算した。また、伝送路は、加法性白色ガウス雑音（ＡＷＧＮ：Ａｄｄｉｔｉｖｅ　Ｗｈｉｔｅ　Ｇａｕｓｓｉａｎ　Ｎｏｉｓｅ）チャネルとし、光信号電力対雑音電力比を定義する際の雑音帯域幅は０．１ｎｍとした。搬送波光信号及び局部発振光信号の位相雑音は０とし、搬送波光信号及び局部発振光信号の周波数差は０とした。偏波多重分離は理想的に行えるものとした。搬送波位相推定には、４乗法を用い、各偏波で検出された位相誤差を平均化して、それぞれの偏波で等量の位相補償を行った。搬送波位相推定の平均化は、１７シンボル移動平均とし、１シンボル前の搬送波位相からの位相変化が±π／４以内となるようＵｎｗｒａｐｐｉｎｇ処理を行った。位相スリップ状態推定の平均化は１７シンボル移動平均により行った。符号誤り率を求めた後、補誤差関数を介してＱ値に変換して評価した。 The simulation of the bit error rate characteristic when this embodiment is used was performed. Simulation conditions are shown in FIG. The bit rate was 128 Gb / s, and the 11-stage pseudo-random binary sequence was used as the code sequence. The number of symbols was 16384, and the calculation was repeated with the noise pattern changed 100 times. The transmission line was an additive white Gaussian Noise (AWGN) channel, and the noise bandwidth when defining the optical signal power to noise power ratio was 0.1 nm. The phase noise between the carrier optical signal and the local oscillation optical signal was 0, and the frequency difference between the carrier optical signal and the local oscillation optical signal was 0. Polarization demultiplexing is ideally performed. For the carrier phase estimation, the fourth power method was used, the phase errors detected in each polarization were averaged, and equal phase compensation was performed for each polarization. The averaging of the carrier phase estimation was a 17-symbol moving average, and an unwrapping process was performed so that the phase change from the carrier phase one symbol before was within ± π / 4. The averaging of the phase slip state estimation was performed by a 17 symbol moving average. After obtaining the code error rate, it was converted into a Q value through a complementary error function and evaluated.
　図１０にシミュレーション結果を示す。比較のため、ＤＰ－ＱＰＳＫ方式に差動符号化・復号化を適用したＤＰ－ＤＥ（Ｄｉｆｆｅｒｅｎｔｉａｌ　Ｅｎｃｏｄｅｄ）ＱＰＳＫ信号と、ＤＰ－ＱＰＳＫ信号で位相スリップのない場合（Ｎｏ　Ｓｌｉｐ）についても計算した。図１０において、横軸をＯＳＮＲとし、縦軸をＱ値（Q-factor）とする。本実施の形態は、図１０に示されるように、Ｑ値が９ｄＢ以上の領域では、スリップなしの場合と同等のＱ値が得られ、良好な特性を示す。Ｑ値が７ｄＢ以下の領域では、本実施の形態による位相スリップ推定に誤りが生じるため、本実施の形態のＱ値は、位相スリップなしの場合のＱ値に比べて低いが、ＤＰ－ＤＥＱＰＳＫ信号に対しては、依然として、０．５ｄＢ以上優れたＱ値を示す。 Figure 10 shows the simulation results. For comparison, a DP-DE (Differential Encoded) QPSK signal obtained by applying differential encoding / decoding to the DP-QPSK method and a case where there is no phase slip (No Slip) in the DP-QPSK signal were also calculated. In FIG. 10, the horizontal axis is OSNR, and the vertical axis is Q value (Q-factor). In the present embodiment, as shown in FIG. 10, in the region where the Q value is 9 dB or more, a Q value equivalent to that without slip is obtained, and good characteristics are exhibited. In the region where the Q value is 7 dB or less, an error occurs in the phase slip estimation according to the present embodiment. Therefore, the Q value of the present embodiment is lower than the Q value without the phase slip, but the DP-DEQPSK signal However, the Q value still shows an excellent Q value of 0.5 dB or more.
　以上のように、本実施の形態では、光送信部１００において、図３に示すような直交偏波間で非対称となる信号点配置を行い、受信部３００において、直交偏波間で搬送波周波数・位相推定を同時に行うことで、位相スリップを検出できるようにした。これにより、位相スリップ状態を推定することが可能であり、最も尤もらしいスリップ状態を推定できる。また、推定したスリップ状態に応じた復号規則による復号結果を選択することで、位相スリップ補償を行うことが可能になる。このように、本実施の形態によれば、位相スリップが起きたとき、位相スリップ状態に応じて受信信号点が固有の配置をとり、位相スリップ無しの場合には、とり得ない信号点の配置となる。これにより、位相スリップ状態の検出が可能となり、位相スリップ補償を行うことが可能となる。従って、差動符号化・復号化を用いることなく、また、冗長度を増大させることもなく、信号単位で位相スリップ状態を推定することができる。また、推定したスリップ状態に応じた復号を行うことができる。その結果、位相スリップによる伝送品質劣化を回避し、雑音耐力の改善を行い、かつ、伝送品質を向上させることができる。 As described above, in the present embodiment, signal point arrangement that is asymmetrical between orthogonally polarized waves as shown in FIG. 3 is performed in optical transmitter 100, and carrier frequency / phase estimation is performed between orthogonally polarized waves in receiver 300. The phase slip can be detected by carrying out simultaneously. Thereby, the phase slip state can be estimated, and the most likely slip state can be estimated. In addition, phase slip compensation can be performed by selecting a decoding result based on a decoding rule corresponding to the estimated slip state. Thus, according to the present embodiment, when a phase slip occurs, the received signal point has a unique arrangement according to the phase slip state, and in the case of no phase slip, the arrangement of signal points that cannot be taken It becomes. Thereby, the phase slip state can be detected and the phase slip compensation can be performed. Therefore, the phase slip state can be estimated on a signal basis without using differential encoding / decoding and without increasing the redundancy. Also, decoding according to the estimated slip state can be performed. As a result, transmission quality deterioration due to phase slip can be avoided, noise tolerance can be improved, and transmission quality can be improved.
産業上の利用の可能性Industrial applicability
　以上のように、本発明にかかる光変復調方式は、デジタルコヒーレント方式を用いた光伝送システムに有用であり、特に、符号誤り率の悪い光伝送システムに適している。 As described above, the optical modulation / demodulation method according to the present invention is useful for an optical transmission system using a digital coherent method, and is particularly suitable for an optical transmission system with a low code error rate.
　１００　光送信部、１０１　光源、１０２　シンボルマッパ、１０３　光変調部、２００　伝送チャネル、３００　光受信部、３０１　光源、３０２　光復調部、３０３　波形等化部、３０４　周波数・位相推定部、３０５　復号部、３０６　位相スリップ状態推定部、３０７　選択部、５００，５０１，５０２，５０３　二次元位相識別器、６００，６０１，６０２，６０３　尤度生成部、６１０　最尤状態推定部、７００，７０１，７０２，７０３　偏波位相空間距離計算部、７１０　選択部、７２０Ａ，７２０Ｂ，７２０Ｃ，７２０Ｘ　遅延部、７２１Ａ，７２１Ｂ，７２１Ｘ　乗算部、７２２　加算部。 100 optical transmission unit, 101 light source, 102 symbol mapper, 103 optical modulation unit, 200 transmission channel, 300 optical reception unit, 301 light source, 302 optical demodulation unit, 303 waveform equalization unit, 304 frequency / phase estimation unit, 305 decoding unit 306, phase slip state estimation unit, 307 selection unit, 500, 501, 502, 503 two-dimensional phase discriminator, 600, 601, 602, 603 likelihood generation unit, 610 maximum likelihood state estimation unit, 700, 701, 702 703 Polarization phase space distance calculation unit, 710 selection unit, 720A, 720B, 720C, 720X delay unit, 721A, 721B, 721X multiplication unit, 722 addition unit.

Claims (8)

  1. 　光送信部により、Ｘ偏波およびＹ偏波の各直交偏波で非対称となる信号点配置を有する光信号を生成して、それらを多重化するステップと、
    　前記光送信部により、多重化された前記光信号を送信するステップと、
    　光受信部により、多重化された前記光信号を受信するステップと、
    　前記光受信部により、所定の波長で発振する局部発振光を生成するステップと、
    　前記光受信部により、受信した前記光信号と前記局部発振光とを混合して光／電気変換を行うとともに、アナログ／デジタル変換を行って、デジタル信号を出力するステップと、
    　前記光受信部により、前記デジタル信号に基づいて、受信した前記光信号の搬送波周波数及び搬送波位相を前記Ｘ偏波および前記Ｙ偏波で同時に推定するステップと、
    　推定された搬送波周波数及び搬送波位相に基づき、シンボル毎にとり得る位相スリップ状態ごとの周波数・位相補償後のデジタル信号を出力するステップと、
    　前記周波数・位相補償後のデジタル信号に基づいて、前記位相スリップ状態ごとの尤度を演算するステップと、
    　前記周波数・位相補償後のデジタル信号を、前記位相スリップ状態ごとに予め設定された閾値に基づいて復号し、前記位相スリップ状態ごとの復号結果を出力するステップと、
    　前記位相スリップ状態ごとの復号結果の中から、前記尤度の中で最大の尤度の位相スリップ状態に対応する復号結果を選択して出力するステップと
    　を備えたことを特徴とする光変復調方法。     A step of generating an optical signal having a signal point arrangement that is asymmetrical in each of the orthogonal polarizations of the X polarization and the Y polarization by the optical transmission unit, and multiplexing them;
    Transmitting the multiplexed optical signal by the optical transmitter; and
    Receiving the multiplexed optical signal by an optical receiver;
    Generating local oscillation light that oscillates at a predetermined wavelength by the optical receiver;
    The optical receiving unit mixes the received optical signal and the local oscillation light to perform optical / electrical conversion, performs analog / digital conversion, and outputs a digital signal;
    Estimating the carrier frequency and the carrier phase of the received optical signal simultaneously with the X-polarized wave and the Y-polarized wave by the optical receiver based on the digital signal;
    Outputting a digital signal after frequency / phase compensation for each phase slip state that can be taken for each symbol based on the estimated carrier frequency and carrier phase;
    Based on the frequency / phase compensated digital signal, calculating a likelihood for each phase slip state;
    Decoding the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state;
    An optical modulation / demodulation method comprising: selecting a decoding result corresponding to a phase slip state having a maximum likelihood from the decoding results for each phase slip state; .
  2. 　前記Ｘ偏波および前記Ｙ偏波の両方の信号点配置が２値位相偏移変調であり、
    　前記Ｘ偏波および前記Ｙ偏波の一方の前記２値位相偏移変調の片側位相の場合のみ、前記Ｘ偏波および前記Ｙ偏波の他方の信号点をπ／２ずらす
    　ことを特徴とする請求項１に記載の光変復調方法。     The signal point arrangement of both the X polarization and the Y polarization is binary phase shift keying,
    The other signal point of the X polarization and the Y polarization is shifted by π / 2 only in the case of one side phase of the binary phase shift keying of one of the X polarization and the Y polarization. The optical modulation / demodulation method according to claim 1.
  3. 　光送信部と光受信部とを備えた光送受信器であって、
    　前記光送信部は、
    　光信号を出力する光源と、
    　外部から入力される符号データに基づき、直交偏波多重化のための電気信号を出力するシンボルマッパと、
    　前記光源から出力された光信号を前記シンボルマッパから出力された電気信号で変調して直交偏波多重化し、Ｘ偏波およびＹ偏波の各直交偏波で非対称となる信号点配置を有する光信号を生成する光変調部と
    　を備え、
    　前記光受信部は、
    　局部発振光信号を出力する局部発振光源と、
    　光信号を受信し、受信した前記光信号と前記局部発振光信号とを混合して光／電気変換を行うとともに、アナログ／デジタル変換を行って、デジタル信号を出力する光復調部と、
    　前記デジタル信号の波形歪みを補償する波形等化部と、
    　前記光受信部により、前記デジタル信号に基づいて、受信した前記光信号の搬送波周波数及び搬送波位相を前記Ｘ偏波および前記Ｙ偏波で同時に推定して補償する搬送波周波数・位相推定部と、
    　前記周波数・位相補償後のデジタル信号を、前記位相スリップ状態ごとに予め設定された閾値に基づいて復号し、前記位相スリップ状態ごとの復号結果を出力する復号部と、
    　前記周波数・位相補償後のデジタル信号に基づいて、前記位相スリップ状態ごとの尤度を演算し、前記尤度を比較することにより、最大尤度の位相スリップ状態を推定する位相スリップ状態推定部と、
    　前記位相スリップ状態推定部が推定する位相スリップ状態に基づき、前記位相スリップ状態ごとの復号結果の中から、前記最大尤度の位相スリップ状態に対応する復号結果を選択して出力する選択部と
    　を備えている
    　ことを特徴とする光送受信器。     An optical transceiver including an optical transmitter and an optical receiver,
    The optical transmitter is
    A light source that outputs an optical signal;
    A symbol mapper that outputs an electrical signal for orthogonal polarization multiplexing based on code data input from the outside;
    Light having a signal point arrangement in which the optical signal output from the light source is modulated by the orthogonal polarization multiplexed by the electrical signal output from the symbol mapper and is asymmetrical in each of the X polarization and the Y polarization. An optical modulation unit for generating a signal,
    The optical receiver is
    A local oscillation light source that outputs a local oscillation light signal;
    An optical demodulator that receives an optical signal, mixes the received optical signal and the local oscillation optical signal to perform optical / electrical conversion, performs analog / digital conversion, and outputs a digital signal;
    A waveform equalizer for compensating for waveform distortion of the digital signal;
    A carrier frequency / phase estimator for simultaneously estimating and compensating the carrier frequency and the carrier phase of the received optical signal with the X polarization and the Y polarization based on the digital signal by the optical receiver;
    Decoding the frequency / phase compensated digital signal based on a preset threshold for each phase slip state, and outputting a decoding result for each phase slip state;
    A phase slip state estimator that calculates a likelihood for each phase slip state based on the frequency / phase compensated digital signal and compares the likelihoods to estimate a maximum likelihood phase slip state; ,
    A selection unit that selects and outputs a decoding result corresponding to the phase slip state of the maximum likelihood from decoding results for each phase slip state based on the phase slip state estimated by the phase slip state estimation unit; An optical transceiver characterized by comprising.
  4. 　前記シンボルマッパは、
    　２系列の２値の電気信号系列が入力され、
    　出力する４系列の２値の電気信号が、前記Ｘ／Ｙ偏波の両方の信号点配置が２値位相偏移変調であり、前記Ｘ／Ｙ偏波の一方の前記２値位相偏移変調の片側位相の場合のみ、前記Ｘ／Ｙ偏波の他方の信号点をπ／２ずらす
    　ことを特徴とする請求項３に記載の光送受信器。     The symbol mapper is
    Two series of binary electrical signal series are input,
    The four-sequence binary electric signal to be output is such that the signal point arrangement of both of the X / Y polarizations is binary phase shift keying, and one of the X / Y polarizations is the binary phase shift keying. 4. The optical transceiver according to claim 3, wherein the other signal point of the X / Y polarized wave is shifted by π / 2 only in the case of one side phase.
  5. 　前記復号部は、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共にない場合、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に＋９０度の場合、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に＋１８０度の場合、及び、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に－９０度の場合
    のそれぞれの位相スリップ状態について復号を行うことで、各位相スリップ状態に対して１つずつの合計４つの復号結果を得る
    　ことを特徴とする請求項３または４に記載の光送受信器。     The decoding unit
    When there is no phase slip in the X polarization and the Y polarization,
    When the phase slip is both +90 degrees in the X polarization and the Y polarization,
    When the phase slip is both +180 degrees in the X polarization and the Y polarization, and
    Decoding is performed for each phase slip state when the phase slip is -90 degrees in both the X polarization and the Y polarization, so that a total of four decoding results are obtained, one for each phase slip state. The optical transceiver according to claim 3 or 4,
  6. 　前記位相スリップ状態推定部は、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共にない場合、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に＋９０度の場合、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に＋１８０度の場合、及び、
    　前記Ｘ偏波および前記Ｙ偏波において位相スリップが共に－９０度の場合
    のそれぞれの位相スリップ状態であることの尤度を計算する尤度生成部を備え、
    　前記尤度に基づいて、最大尤度の位相スリップ状態がいずれであるかを推定する
    　ことを特徴とする請求項３ないし５のいずれか１項に記載の光送受信器。     The phase slip state estimator is
    When there is no phase slip in the X polarization and the Y polarization,
    When the phase slip is both +90 degrees in the X polarization and the Y polarization,
    When the phase slip is both +180 degrees in the X polarization and the Y polarization, and
    A likelihood generation unit that calculates the likelihood that each phase slip state is -90 degrees in both the X polarization and the Y polarization;
    The optical transceiver according to any one of claims 3 to 5, wherein the phase slip state of the maximum likelihood is estimated based on the likelihood.
  7. 　前記尤度生成部は、
    　偏波位相空間におけるシンボル中心からの偏波位相空間距離を計算する偏波位相空間距離計算部と、
    　前記偏波位相空間距離の最小値を選択する選択部と、
    　前記選択部から出力される信号を平均化する平均化部と
    　を備えることを特徴とする請求項６に記載の光送受信器。     The likelihood generator is
    A polarization phase space distance calculation unit for calculating a polarization phase space distance from the symbol center in the polarization phase space;
    A selector for selecting a minimum value of the polarization phase space distance;
    The optical transceiver according to claim 6, further comprising: an averaging unit that averages a signal output from the selection unit.
  8. 　前記平均化部は、有限インパルス応答フィルタまたは忘却型の無限インパルス応答フィルタで構成される
    　ことを特徴とする請求項７に記載の光送受信器。     The optical transceiver according to claim 7, wherein the averaging unit includes a finite impulse response filter or a forgetting type infinite impulse response filter.
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