KR20010080248A - Method and apparatus for pseudonoise spreading in a cdma communication system - Google Patents

Abstract

The present invention relates to a method of PN spreading a CDMA signal using a truncated PN sequence. The present invention also relates to a method of generating the truncated PN sequence, a method of generating a second truncated PN sequence by masking a first truncated sequence, and a method of generating a phase shifted version of the truncated PN sequence.

Description

METHOD AND APPARATUS FOR PSEUDONOISE SPREADING IN A CDMA COMMUNICATION SYSTEM}

As wireless communication technology develops, the demand for high-speed data services in a wireless environment becomes greater. The use of code division multiple access (CDMA) modulation schemes is one of several technologies that provide digital wireless transmission suitable for the transmission of digital data. Other methods of digital wireless transmission include time division multiple access (TDMA) and frequency division multiple access (FDMA).

However, CDMA spread spectrum modulation techniques have significant advantages over other digital modulation techniques. The use of CDMA technology in a multiple access communication system is disclosed in US Pat. No. 4,901,307, entitled "SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS", assigned to the assignee of the present invention and cross-referenced herein. The use of CDMA technology in a multiple access communication system is disclosed in US Patent No. 5,103,459, entitled "SYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM", assigned to the assignee of the present invention and cross-referenced herein. . A method for providing digital wireless communications using CDMA modulation can be found in the Telecommunications Industry Association (TIA) regarding TIA / EIA / IS-95-A Mobile Station-Base Station Compatibility Standard (hereafter IS-95) for dual-mode wideband spread spectrum cellular systems. Standardized by

At the receiver, the multipath characteristic of the land channel produces a signal with several individual propagation paths in motion. The characteristic of a multipath channel is that time spread is introduced into the signal transmitted through the channel. Spread-spectrum pseudonoise (PN) modulation used in CDMA systems allows different propagation paths of the same signal to be distinguished and combined if the difference in path delay exceeds PN chip time. If a PN chip rate of approximately 1 MHz is used in the CDMA system, the full spread spectrum processing gain is equal to the spread bandwidth versus the system data rate, and can be used in paths with delays of more than 1 microsecond. One microsecond path delay difference corresponds to a path distance of approximately 300 meters. Urban environments typically provide path delay differences in excess of 1 microsecond. Another characteristic of a multipath channel is that each path through the channel can result in different attenuation elements. For example, if an ideal impulse is transmitted in a multipath channel, each pulse of the received pulse stream generally has a different signal strength than other received pulses.

Another characteristic of multipath channels is that each path through the channel causes a different phase in the signal. For example, if an ideal impulse is transmitted in a multipath channel, the pulse of each stream of received pulses has a different phase than the other received pulses. This results in signal fading.

Fading occurs when the multipath vector is added destructively, yielding a received signal that is smaller than the individual element of the two. For example, if a sine wave has a time delay of d with attenuation component of X dB and a phase shift of Q radians and a second path with phase damping of Q + _ radians and attenuation of X dB If transmitted over a multipath channel with two paths with a time delay of d, no signal is received at the output of the channel.

As mentioned above, in a typical CDMA demodulator architecture, the PN chip spacing represents the minimum separation that two paths must be combined in order. Before the individual paths are demodulated, the relative arrival time (or offset) of the paths of the received signals must first be determined. The demodulator accomplishes this function by "navigating" the sequence of offsets and measures the energy received at each offset. If the energy associated with the potential offset exceeds a certain threshold, a demodulation element or "finger" may be assigned to that offset. The signal at the path offset can be added with the contribution of the other finger at each offset.

Methods and apparatus for finger assignment based on searcher and finger energy levels are disclosed in US Pat. No. 5,490,165, entitled “FINGER ASSIGNMENT IN A SYSTEM CAPABLE OF RECEIVING MULTIPLE SIGNALS”, filed Oct. 28, 1993, Assigned to the assignee and cross-referenced herein. In a typical embodiment, the CDMA signal is transmitted in accordance with the Telecommunications Industry Association TIA / EIA / IS-95-A, which is "MOBILE STATION-BASE STATION COMPATIBILITY STANDARD FOR DUAL-MODE WIDEBAND SPECTRUM CELLULAR SYSTEM". The signal transmitted from the base station to the mobile station is referred to as a forward link signal, and the signal transmitted from the mobile station to the base station is referred to as a reverse link signal.

A typical embodiment of a circuit capable of demodulating an IS-95 forward link signal is disclosed in detail in US Pat. No. 5,764,592, entitled "MOBILE DEMODULATOR ARCHITECTURE FOR A SPREAD SPECTRUM MULTIPLE ACCESS SYSTEM," and is assigned to the assignee of the present invention. Cross-referenced herein. A typical embodiment of a circuit for demodulating an IS-95 reverse link signal is disclosed in detail in US Pat. No. 5,654,979, entitled "CELL SITE DEMODULATOR ARCHITECTURE FOR A SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM," and is assigned to the assignee of the present invention. , Cross-referenced herein.

1 shows a typical set of base station signals arriving at a mobile station. It is apparent to those skilled in the art that Figure 1 can be used equally to the signal of a mobile station that has reached a base station. The vertical axis represents power received in decibel (dB) scale. The horizontal axis represents the delay in the arrival time of the signal due to the multipath delay. The axis leading to the page (not shown) represents a segment of time. Signals in the common plane travel along different paths to the receiver at the same time, but are transmitted at different times.

In common terms, the peak for the right side is transmitted by the base station earlier than the peak for the left side. For example, the left-most peak spike 2 corresponds to the most recently transmitted signal. Each signal spike 2-7 travels a different path and thus exhibits different time delays and different amplitude responses.

Six different signal spikes, represented by 2-7, represent a representative severe multipath environment. Typical urban environments create fewer available routes. The noise floor of the system is represented by peaks and dips with low energy levels.

The job of the seeker is to identify the delay measured by the horizontal axis of the signal spikes 2-7 for potential finger assignment. The task of the finger is to demodulate one of the set of multipath peaks for combining with a single output. The task of the finger is also to track peaks that can move properly once they are assigned to multipath peaks.

The horizontal axis may also have units of PN offset. At any given time, the mobile station receives several signals from the base station, each of which travels in a different path and may have different delays. The signal of the base station is modulated by the PN sequence. A local copy of the PN sequence is generated at the mobile station. Each multipath signal in the mobile station is also individually demodulated with a PN sequence code assigned to its reception time offset. Horizontal axis coordinates correspond to PN sequence code offsets that can be used to demodulate a signal at the coordinates.

Each multipath peak varies in size as a function of time and is represented as an irregular floor of each multipath peak. There is no significant change in multipath peaks at the limited time. In the extended time range, multipath peaks are not visible and new paths are generated over time. This peak can also slide with an early or slow offset as the path distance changes as the mobile station moves relative to the base station. Each finger tracks the change in said small signal assigned to it.

In narrowband systems, the presence of multiple paths in the wireless channel can result in severe fading through the narrow frequency bands used. The system is a capacity constrained by the extra transmit power needed to overcome deep fade. As mentioned above, the CDMA signal path can be identified and diversity is combined in the demodulation process.

There are three main types of diversity: time diversity, frequency diversity, and space / path diversity. Time diversity can be best obtained using detection coding that generates repetition, time interleaving, and error correction and redundancy. The system may utilize each of the above techniques in the form of time diversity.

CDMA provides a form of frequency diversity by spreading signal energy over a wide bandwidth because of its inherent wideband nature. Frequency selective fading, which can cause deep fading through the frequency bandwidth of a narrowband system, generally affects only a portion of the frequency band used by the CDMA spread spectrum signal.

Rake receivers provide path diversity by the ability to combine multipath delayed signals. That is, all paths with assigned fingers must fade together before the combined signal degrades. Additional path diversity is obtained through a process known as "soft hand-off" in which several simultaneous redundant links from two or more base stations can be established with the mobile station. It supports robust linking in environments that subscribe in cell boundary areas. Examples of path diversity include U.S. Patent No. 5,101,501, entitled "SOFT HAND-OFF IN A CDMA CELLULAR TELEPHONE SYSTEM," filed March 21, 1992, and "DIVERSITY RECEIVER IN A CDMA, filed April 28, 1992." CELLULAR TELEPHONE SYSTEM ", US Patent No. 5,109,390, all of which are assigned to the assignee of the present invention and are cross-referenced herein.

The cross-correlation between different PN sequences and autocorrelation of PN sequences all have nearly zero mean values for all time shifts that are different from zero values. It is possible for different user signals to be identified upon reception. Autocorrelation and cross-correlation require logic "1" at similar mappings or values of "-1" and logic "0" at values of "1" such that a zero mean value is obtained.

However, the PN signal is not orthogonal. Cross-correlation is basically averaged to zero over the entire sequence length for short time intervals, such as information bit times, but cross-correlation is randomly variable as a binomial distribution. As such, these signals interfere with each other as if they were wide bandwidth Gaussians at the same power spectral density.

A set of n orthogonal binary sequences, each of length n, can be constructed as having a power of 2 for n (page 45- of Digital Communications with Space Applications, published by Prentice-Hall, 1964, SW Golomb, etc.). 64). In fact, orthogonal binary sequence sets are well known for most lengths of less than two hundred and a multiple of four. One class of the sequence that is advantageous to generate is called a Walsh function. The nth Walsh function can be defined inductively as

(One)

Where W￢ represents the logical complement of W, where W (1) = │0│.

The Walsh sequence or code is one of the columns of Walsh function metrics. The nth Walsh function matrix includes n sequences for each of the n Walsh chips. The nth Walsh function matrix (also another orthogonal function of n) has the property that the cross-correlation between all the different sequences in the set is zero over an interval of n bits. Each sequence in the set differs from another in exactly half the bit. There is always one sequence containing all zero values, all other sequences are half one and half zero.

In the system disclosed in the '459 patent, the call signal starts out as a 9600-bit information source per second that is converted into a 19,200 symbol output stream per second by a half rate forward error correction encoder. Each call signal broadcast from a cell is covered with one of 64 orthogonal Walsh sequences, each with 64 Walsh chips or one symbol. Although the symbol is covered, the orthogonality of all Walsh sequences ensures all interference from other user signals that the cell is canceled during symbol integration. Non-orthogonal disorders from multiple paths in addition to other cells limit the capacity in the forward link.

In IS-95, all user signals transmitted by the base station are quadrature phase shift key (QPSK) spread using the same in-phase (I) channel PN sequence and quadrature (Q) channel PN sequence. Each base station in a CDMA system transmits on the same frequency band using the same PN sequence, but at a single offset associated with the unshifted PN sequence assigned to the universal time reference. The PN diffusion rate is equal to the Walsh cover rate, i.e. 1,2288 MHz or 64 PN chips per symbol. In a preferred embodiment, each base station transmits a pilot reference, and in the description of the present invention, different information is transmitted in the I and Q channels which substantially increases the capacity of the system.

The pilot channel is a “beacon” that transmits a constant zero symbol and spreads to the same I and Q PN sequences used by the traffic bearing signal. In a preferred embodiment, the pilot channel is covered with Walsh sequence 0 which is all zeros. During initial system recognition, the mobile station searches for all possible shifts in the PN sequence and can synchronize itself with the system time if the pilot of the base station is found. As will be described in detail below, the pilot plays a fundamental role in the mobile modulator rake receiver architecture than is used for initial synchronization.

2 shows a typical Rake receiver modulator 10 of a radio for receiving and modulating forward link signal 20 arriving at antenna 18. Analog transmitter and receiver 16 include a QPSK downconverter chain that outputs digital I and Q channel samples 32 at baseband. The sampling clock, CHIPX8 40, is used to digitize the receive waveform and is derived from a voltage controlled temperature compensated local oscillator (TCXO).

The demodulator 10 is managed by the microprocessor 30 via the data bus 34. Within the demodulator, I and Q samples 32 are provided to multiple fingers 12a-c and searcher 14. The searcher 14 searches for a window of offsets to include multipath signal peaks suitable for the assignment of fingers 12a-c. For each offset of the search window, searcher 14 reports that pilot energy was found at the offset to the microprocessor. Fingers 12a-c are examined and unallocated or traced paths are assigned an offset that includes the strong path identified by searcher 14 by microprocessor 30.

When fingers 12a-c are locked with a multipath signal at the assigned offset, the path is tracked by magnetic force until the path fades or until it is reallocated using an internal time tracking loop. The finger time tracking loop above measures the energy on one side of the peak at the offset at which the finger is currently demodulated. The difference between the energies forms a metric that is then filtered and integrated.

The output of the integrator controls the decimator to select one of the input samples for the chip spacing for use in the demodulator. If the peak moves, the finger adjusts the position of the dataator to move. The decimated sample stream is spread with a PN sequence whose fingers match the assigned offset. Despread I and Q samples are added to the symbols to produce a pilot vector (P _I , P _Q ). The same despread I and Q samples are uncovered Walsh using Walsh code assignments unique to the mobile user, and the uncovered despread I and Q samples are used to generate symbol data vectors D _I , D _Q. It is added to the symbol. The dope product operator is defined as

P (n) D (n) = P _I (n) D _I (n) + P _Q (n) D _Q (n) (2)

Where P _I (n) and P _Q (n) are the I and Q components of the pilot vector P for symbol n, respectively, and D _I (n) and D _Q (n) are the data vectors D for symbol n, respectively. I and Q components.

Since the pilot signal vector is much stronger than the data signal vector, it can be used as an accurate phase reference for coherent demodulation. That is, the dot product calculates the magnitude of the data vector component in phase with the pilot vector. As disclosed in co-pending US Patent No. 5,506,865, "PILOT CARRIER DOT PRODUCT CIRCUIT", the dot product effectively combines each finger symbol output 42a-c by the relative strength of the pilot received by the finger. Weights the contribution of the finger to scale. The dot product therefore performs the dual role of finger symbol weighting and phase projection required for a coherent Rake receiver demodulator.

Each finger has a lock detector circuit that masks the symbol output to combiner 42 if the long average energy does not exceed the minimum threshold. This ensures that only the fingers tracking the reliable path will contribute to the combined output, thus enhancing the demodulator performance.

Due to the relative difference in arrival times of the paths to which each finger 12a-c is assigned, each finger 12a-c allows the symbol combiner 22 to combine with each other to produce a "soft decision" demodulated symbol. It has a deskew buffer for allocating finger symbol streams 42a-c. The symbol is weighted by the reliability of correctly identifying the original transmitted symbol. This symbol is a deinterleaver / decoder circuit whose forward error correction decodes the symbol stream using a maximum likelihood Viterbi algorithm after the first frame is deinterleaved. Is sent to. The decoded data is used in microprocessor 30 or in other components such as speech vocoder for subsequent processing.

To correctly demodulate, the mechanism required to assign the local oscillator frequency to the clock used in the cell to modulate the data is required. Each finger estimates the frequency error by measuring the rotation rate of the pilot vector in the QPSK I, Q space using a vector product operator (Equation 3).

P (n) XP (n-1) = P _I (n) P _Q (n-1)-P _I (n-1) P _Q (n) (3)

Frequency error estimates from each finger 44a-c are combined and integrated at frequency error combiner 26. The integrator output, LO_ADJ 36, is supplied to the voltage control of the TCXO of the analog transmitter and receiver 16 to adjust the clock frequency of the CHIPX8 clock 40, thus providing a closed loop mechanism to compensate for the frequency error of the local oscillator. to provide.

If the transmission power of the signal transmitted by the base station to the mobile station is too high, problems such as interference with other mobile stations may occur. Optionally, the mobile station may receive multiple error frames if the transmit power of the signal transmitted by the base station is too low. Land channel fading and other known factors can affect the received power of a signal transmitted by a base station. As a result, each base station can adjust the transmit power of the signal transmitted to the mobile station quickly and accurately.

In a useful method of controlling the transmit power of a signal transmitted by a base station, the mobile station sends a signal or message to the base station when the power of the received frame of data deviates from the threshold or is received as an error. In response to this message, the base station increases the transmit power of the signal transmitted by the base station. A method and apparatus for controlling transmit power is disclosed in U.S. Patent No. 5,056,109, entitled "METHOD AND APPARATUS FOR CONTROLLING TRANSMISSION POWER IN A CDMA CELLULAR TELEPHONE SYSTEM," assigned to the assignee of the present invention and cross-referenced herein. .

Recently, there has been a growing interest in providing high speed digital data in wireless communication links. In providing such a high speed data link, the system has been developed as using a larger bandwidth. One method of providing high speed data involves the use of CDMA technology based on advances made in the deployment of IS-95 systems. The system proposed by the Electronics and Telecommunications Industry Association for providing high speed data in wireless communication systems is named "CDMA 2000".

In the transmission of the CDMA signal, it is desirable to reduce the peak to the average ratio of transmit power passing through the power amplifier of the wireless communication device. The method of reducing the peak to average ratio is described by the hybrid PN diffusion as disclosed in "REDUCED PEAK TO AVERAGE TRANSMIT POWER HIGH DATA RATE IN A CDMA WIRELESS COMMUNICATION SYSTEM," filed April 9, 1996. It is about use. In the hybrid PN, the information signals I 'and Q' are PN spread according to the following equation.

I = I'PN _I -Q'PN _{Q '} (4)

Q = I'PN _Q + Q'PN _{I '} (5)

Where PN _I and PN _Q are each PN spreading codes. This hybrid PN spreading is very useful when the amount of information is above all the information signal channel I 'or Q'. Hybrid PN diffusion serves to balance the load, thus reducing the peak to average ratio. U.S. Patent Application No. 08 / 856,428, above, also discloses a method for despreading hybrid PN spreading data.

CDMA 2000 systems use orthogonal code channels that are substantially spread by similar noise sequences. The system is designed to provide diffusion for a given set of chip rates, including 1.2288 Mcps, 3.6864 Mcps, and 11.0592 Mcps. It may also preferably be equipped to function at each chip rate. In addition, a search method for pilot signals that provides fast and effective acquisition is desirable.

TECHNICAL FIELD The present invention relates to spread spectrum communications, and in particular, to novel and improved apparatus and methods for spreading similar noise in a direct access code division multiple access (CDMA) communication system.

1 illustrates a multipath signal in a CDMA environment.

2 shows a rake receiver for receiving a CDMA signal.

3 shows a block diagram of the initial processing of a CDMA signal in a third generation CDMA communication system.

4 shows a block diagram of the final processing of a CDMA signal in a third generation CDMA communication system.

5 is a block diagram illustrating the generation of a masked PN sequence used in an IS-95 system.

6 is a diagram of a typical PN generator.

7 is a diagram of the state of an offset truncated PN sequence and the state of the truncated PN sequence.

8 is a block diagram of an exemplary embodiment of a circuit used to generate a truncated PN sequence.

9 is an exemplary embodiment of a register used in the LFSR of FIG.

10 is a block diagram of an exemplary embodiment of a second truncated PN sequence using a single PN generator and circuitry used to generate the truncated PN sequence.

11A-11B illustrate problems in phase shifting truncated PN sequences.

12 is a block diagram of a circuit for providing a phase shifted version of a truncated PN sequence.

Figure 13 illustrates an exemplary embodiment of the present invention where the initial phase of the PN sequence is the same despite the final chip rate.

The present invention relates to a novel and improved method of PN spreading CDMA signals. The present invention shows a method of generating a truncated PN sequence. The invention also shows a method of masking a first truncated sequence to produce a second truncated PN sequence and a method of generating a phase shifted version of the truncated PN sequence.

The invention is described in detail below with reference to the drawings.

Spectrally effective multiple access systems (see cdmaOne systems), as specified in IS-95, can handle current capacity requirements, while producers and operators can increase the demands on systems such as interest in wireless data communications. Predict the rise of popularity as wireless telephones increase. In anticipating increasing demands, the International Telecommunication Union proposed a program to standardize third generation wireless communication systems.

The Radiocommunication Technology (RTT) candidate of the Telecommunications Industry Association submitted to the ITU was named "The cdma2000 ITU-R Candidate Submission (0.18)" (hereinafter referred to as the cdma2000 submission). The cdma2000 submission is provided for operation at a set of different chip rates so that the system can increase reasonably with capacity requirements. In particular, the cdma2000 submission is offered to operate at 1.2288 Mcps (chip rate of current cdmaOne system), 3.6864 Mcps, 7.3728 Mcps, 11.0592 Mcps and 14.7475 Mcps.

In the cdmaOne system, as planned in the cdma2000 system, each base station spreads forward link transmissions using a common PN spreading code offset by a predetermined amount from other nearby base stations. The amount of offset required between base stations is a function of the maximum propagation path predicted by the system designer.

As disclosed in US Pat. No. 5,764,592, the CDMA receiver generates a local version of the PN sequence to despread the received signal. The PN sequence generated at the receiver will be the offset of the PN sequence used to perform spreading at the base station due to the time the signal propagates from the base station to the receiver of the mobile station. If the PN offsets of the base stations in a given area are close to each other, the base station PN spreading will not appear unique to the mobile station and will prevent the mobile station from identifying signals received from different base stations. Therefore, the amount of PN offset between base stations is a function of the maximum predicted propagation time of the signal to reach the mobile station.

Since the required PN offset is a function of the propagation time, the minimum amount of PN chip offset between base stations is the product of the maximum propagation time and the PN spreading ratio. In addition, the number of states required to be generated by PN spreading is equal to the number of base stations that can interfere with each other or communicate with the mobile station and the minimum amount of PN chip offset between the base stations. Therefore, a shorter PN code would be required for a system running at 1.2288Mcps than a system running at 3.6864Mcps.

3 shows a downlink (forward link) transmission proposal proposed in the cmda2000 submission. The frame of data is provided to the tail bit generator 102 and the CRC to generate a set of parity bits for the frame referred to as cyclic redundancy bits. Methods for generating cyclic redundancy check bits are known in the art, and methods for generating CRC bits are disclosed in US Pat. No. 5,504,773, entitled "METHOD AND APPARATUS FOR THE FORMATTING OF DATA FOR TRANSMISSION," which is an assignee of the present invention. And cross-referenced herein. The CRC and tail bit generator 102 adds a set of tail bits to the frame used to clear the memory of the decoder and receiver.

The data packet is then provided to the decoder 104. Encoder 104 may be a convolutional encoder or a turbo encoder. Convolutional encoders are known in the art. Convolutional encoders enable the use of trellis decoders in receivers that greatly reduce the amount of energy needed to accurately transmit data to remote stations. Optionally, the encoder 4 may be a turbo encoder, the design of which is US Patent No. 5,466,747 named "ERROR CORRECTION CODING METHOD WITH AT LEAST TWO SYSTEMATIC CONVOLUTIONAL CODINGS IN PARALLEL, CORRESPONDING ITERATIVE DECODING METHOD, DECODING MODULE AND DECODER." Known examples and those skilled in the art.

The encoded symbol is provided to the interleaver 106 by the encoder 104. Interleaver 106 reorders the symbols to provide protected time diversity against burst type errors common in a wireless environment. In a typical embodiment, interleaver 106 is a block interleaver in which data is read into a memory element in a column and read in a row.

The rearranged symbol is provided to the scrambling element 112. The scrambling element 112 scrambles the interleaved data along the decimated long PN sequence. This scrambling is accomplished by performing modulo-2 addition of interleaver output symbols with binary values of long PN chips. Long code is generated by a linear feedback shift register (LFSR) that is passed through a masking function. The masking function is typically a user's identification function based on the user's electronic serial number (ESN). The long code generated by the LFSR is provided to the bit selector 110 which masks and decimates the sequence at the proper rate. The decimated sequence is provided to the scrambling element 112 which provides the user with an additional level of call security.

The scrambled symbols are provided to the multiplexer and signal point mapping element 114. In a typical embodiment, each two bit set is supplied to a signal point mapping element 114 and a multiplexer that maps a binary bit set, the arrangement of which is point (1, 1), (1, -1), (-1). , 1) and (-1, -1). One of the points of the batch mapping is provided to the data channel gain element 116 at the first output, and a second point of this batch is provided to the data channel gain element 118 at the second output.

Gain-adjusted packets are provided to punching elements 122 and 28 that adjust the transmission gain of the packets. The gain adjusted packet is then provided to punching elements 122 and 124. Power control bits (not shown) for adjusting the transmit power of the remote station are provided to the power control channel gain element 120.

The power control gain element 120 adjusts the gain of the power control bits and provides the gain adjusted power control bits to the punching elements 122 and 124. The punching elements 122 and 124 punch the power control bits into packets at predetermined locations. This packet is provided to the covering elements 128 and 130.

The packet contains a symbol of ± 1 value. This symbol is provided to the covering elements 128 and 130, which multiply the symbol by an orthogonal sequence containing ± 1 values. This orthogonal sequence is for traffic transmission to the user of a particular remote station. An orthogonal sequence of a typical embodiment is a Walsh sequence, the generation of which is known in the art and disclosed in US Pat. No. 5,103,459, entitled “SYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM”, which is Is assigned to the assignee of and is cross-referenced herein.

The Walsh cover sequence is provided to a forward link channel adder that sums the data of the covering elements 128 and 130 with other similarly modulated data packets for transmission to other users in addition to common channel data.

Referring back to FIG. 2, the final added data is provided to the PN diffusion elements 134, 136, 138 and 140. Each base station is identified by a unique PN offset to a set of nearby base stations. In the present invention, two PN sequences are used to spread the data PN _I , PN _Q. Each PN sequence contains a sequence of ± 1 values. The PN diffusion operation shown in FIG. 2 is performed to provide the following results.

I = I'PN _I -Q'PN _{Q '} (6)

Q = I'PN _Q + Q'PN _{I '} (7)

The data sequence I 'is provided to the multipliers 134 and 138. The multiplier 134 multiplies the data I 'by the similar noise sequence PN _I and provides the result to the summing input of the subtractor 142. The multiplier 138 multiplies the data I 'by the similar noise sequence PN _Q and provides the result to the first adder input of the adder 144.

The data sequence Q 'is provided to the first input of the multipliers 136 and 140. The multiplier 140 multiplies the data Q 'by the similar noise sequence PN _I and provides the result to the subtraction input of the subtractor 142. The multiplier 136 multiplies the data Q 'by the similar noise sequence PN _Q and provides the result to the second adder input of the adder 144.

Subtractor 142 subtracts the output of multiplier 140 from the output of multiplier 134 and supplies the result to baseband filter BBF 146. Adder 144 adds the output of multiplier 138 to the output of multiplier 136 and provides the result to baseband filter BBF 148. Baseband filters 146 and 148 filter the PN spreading sequences and provide the filtered sequences to upconverters 150 and 152, respectively.

Upconverters 150 and 152 upconvert the input data according to quadrature phase shift keyed (QPSK) modulation as is known in the art. Upconverter 150 upconverts signal I for transmission along carrier modulation cos (2_fc). Upconverter 152 upconverts signal Q according to carrier modulation sin (2_fc). The two orthogonal signals are then added in addition element 154 which is responsible for amplification and transmission.

In the present invention, a method of spreading both I and Q with a single PN sequence spread using a single PN generator is proposed. To provide a single spread of the I and Q channels using a single PN generator, the PN sequence used to spread the Q channel data is offset from the PN sequence used to spread the I channel data.

Linear Feedback Shift Registers (LFSRs) are typically used to generate PN sequences for CDMA spreading. The output of the LFSR is cycled through a single state with a predetermined number before its initial state is returned and then resumes the cycle of output. In the full length LFSR sequence, the shift register passes through all possible states of that memory before its initial state is returned. This is valid when the polynomial is chosen so that the feedback is at its maximum length. The output of the LFSR is a function of the state of the LFSR.

The PN sequences PN _I and PN _Q may be generated by different polynomials (ie, PN generators) or by different phases of the same PN generator. In a first embodiment of the PN generator of the I and Q channels, the I and Q channels are generated using the same PN generator having a sequence offset with respect to each other.

As is known, the phase of the PN generator output can be changed using a masking operation performed in the output state of the LFSR. 5 shows a long code PN generator used in an IS-95 system. The LFSR 175 provides the masking element 177 with the maximum length of sequence. Masking element 177 is a series of AND gates that change the phase of the output. The mask in this case is a 42-bit input sequence where turn on and tone off tap the output of the LFSR. Mask operation changes the phase of the output sequence. The masked sequence is then provided to a modulo-2 adder 179 which outputs a long code sequence.

6 shows a generalized version of LFSR with a feedback path determined according to the following polynomial.

G = g ₀ + g ₁ x + g ₂ x ² + ... + g _r x ^r-1 (8)

Where g ₀ is the maximum rightmost feedback and is always equal to "1", where gn is the binary value that enables or disables the corresponding feedback path of the LFSR, and x represents the position of the feedback. The feedback polynomial (G) can be harwired where no AND gate is required. The LFSR 206 includes a set of adders 202, registers 200 and an AND gate 204. The state of the LFSR is determined by the value to the adder 202. The LFSR starts in the initial state set by loading register 200.

As the clock is processed in the circuit, the value stored in register 200 is the output for the first input of adder 202. The value output by the register 200a is the output of the LFSR 206 and is also fed back in the feedback line 208 as one input to the AND gate 204. The other input to AND gate 204 is the binary values g ₀ , g ₁ ... g _r .

In the first embodiment, the sequence generated for the spreading of the I and Q channels is a phase shifted version of the large PN sequence. In order for a single PN generator to produce a non-overlapped PN sequence, the PN generator must generate a sequence that is at least twice the sequence used to spread the I and Q channels, and the sequence must be truncated.

7 illustrates the phase or environment of an LFSR in an environment of circles. The circle achieves an effective visualization of the output of the LFSR because, like the circle, the state of the LFSR repeats starting from the initial state circle and returning to the initial state through all output states.

In the first embodiment of the present invention, the sequence used to spread the I and Q channels is generated using truncated PN sequences. In Figure 7, the LFSR is shown to have an N unique state. The sequence used to spread the I channel PN _I is truncated to k state (k <N <2). Therefore, a PN generator designed to cycle through all N-capable states is truncated once it reaches the k + 1 state after the initial state of time.

In a preferred version of the first embodiment, the sequence used to spread the Q channel is an offset from the sequence used to spread the I channel by an amount such that the two sequences do not overlap. As shown in FIG. 7, the sequence used to spread the Q channel PN _Q includes states between m and m + k (m> k and m + k <N). In a typical embodiment, PN _Q is generated by masking the PN _I sequence.

In alternative embodiments, PN _I and PN _Q may be generated as PN sequences truncated by different polynomials. The feedback polynomial G can be Hawaiianed where no AND gate is needed. The initial state S _I of the PN generator 306 is provided at the bus line 312 to the register 300. In a typical embodiment, register 300 is known as a D-flip-flop, but other memory structures may equally be used within the scope of the present invention. The output of PN generator 300a is fed back in line 308 to the first output of AND gate 304. Polynomial G is provided as the second output of the AND gate at input line 314.

Polynomial G may enable or disable the feedback provided at line 308. If the input to AND gate 304 of input line 314 is high, feedback to the modulo-2 adder 302 is enabled, and the output bits are fed back to this adder. If the input to AND gate 304 of input line 314 is low, feedback to the modulo-2 adder 302 is disabled, and output bits are blocked from reaching the adder. The output of each register 300b to 300r is provided to the first input of the corresponding modulo-2 adder 302a to 302 (r-1). The second input to the modulo-2 adders 302a through 302 (r-1) is the feedback output bit from the register 300a at a position where the feedback path can be enabled by polynomial G.

In order to truncate the PN sequence, the state identification element 310 is used to compare the state of the LFSR with the final state of the truncated PN sequence. Although several methods may be used to perform the state detection element 310, the earliest scheme includes a number of XOR gates, each of which receives one input and one second final bits of the current state. Has input The output of each XOR gate will be provided to the r-input AND gate and the result will be the output of line 318.

Returning to FIG. 7, which simply generates an I-sequence, PN generator 306 is first set to its initial state (designated a 0). Upon reaching the final state of the cutting sequence (state k + 1), state identification element 310 and transmits a signal to the register 300 to load the initial state (S _I) into a register. Upon detection by the state detection element 310 of the initial state S _f, the state detection element 310 outputs a binary signal indicating the identification of the final state. Signal in line 318 causes the switch to output the initial state S _i from having each register 300 outputs a maximum synthesis of a preceding adder element.

For example, in most cases register 300c takes an output from adder 302c and provides this output to modulo-2 adder 302b. However, when the signal on line 318 goes high, it is detected that the final state is indicated, and register 300c outputs Si ₂ to modulo-2 adder 302b. At the next clock interval, the state detection element 310 will detect the difference between the state of the PN generator and the final state, set the signal on line 318 low, which will cause the operation to resume, and register ( 300c will provide the output of adder 302c to the first input of adder 302b.

9 illustrates a typical implementation of register 300 using a loadable D-type flip floc 320. The A input receives the output of that adder. The B input receives the bits of the initial state S _i . The value of line 318 is provided to the LD, which determines whether the output Q will provide input A or input B. The clock runs at the PN chip rate.

10 shows an apparatus for generating two pseudo noise sequences PN _I and PN _Q that are phase offset with respect to each other. The pseudo noise sequence PNI is generated by the PN generator 426 in the same manner as described for the PN generator 426 of FIG. The mask (M) and feedback (G) polynomials are Hawaiianed where no AND gate is needed. The state of the PN generator 426 is provided to the masking element 428 which phase shifts the output PNQ with respect to the sequence PN _I.

The PN generator 426 of the present invention produces a truncated PN sequence. The initial state SI of the PN generator 426 is provided at the bus line 412 to the register 400. In a typical embodiment, register 400 is disclosed as a D-flip-flop, but other memory structures may equally be used within the scope of the present invention. The output of the PN generator 426 is the output of the register 400a. In addition, the output of register 400a is feedback at line 408 to AND gate 404. Polynomial G is provided as a second input of AND gate 404 at input line 414.

Polynomial G may enable or disable the feedback provided at line 408. If the input to the AND gate 404 of the input line 414 is high, feedback to the modulo-2 adder 402 is enabled and the output bits are fed back to this adder. If the input to the AND gate 404 of the input line 414 is low, feedback to the modulo-2 adder 402 is disabled and output bits are blocked from reaching the adder. The output of each register 400b-400r is provided to the first input of the corresponding modulo-2 adders 402a-402 (r-1). The second input to the modulo-2 adders 402a through 402 (r-1) is the feedback output bit from the register 400a at a position where the feedback path can be enabled by polynomial G.

To truncate the PN sequence, the status identification element 410 is used to compare the status of the truncated PN sequence with the status of the LFSR. Although several methods may be used to perform the state detection element 410, the earliest scheme includes a number of XOR gates, each of which receives one input and one second final bits of the current state. Has input The output of each XOR gate will be provided to the r-input AND gate, and the result will be the output of line 418.

Returning to FIG. 7, which simply generates an I-sequence, PN generator 406 is first set to its initial state (designated a 0). Upon reaching the final state of the cutting sequence (state k + 1), state identification element 410 transmits a signal to the register 400 to load the initial state (S _I) into a register. Upon detection by the state detection element 410 of the initial state S _f, the state detection element 410 outputs a binary signal indicating the identification of the final state. Signal in line 418 causes the switch to output the initial state S _i from having each register 400 outputs a maximum synthesis of a preceding adder element.

For example, in most cases register 400c takes an output from adder 402c and provides this output to modulo-2 adder 402b. However, when the signal on line 418 goes high, an indication of the final state is detected, and register 400c outputs Si ₂ to modulo-2 adder 402b. At the next clock interval, the state detection element 410 will detect the difference between the state of the PN generator and the final state, set the signal on line 418 to low, which will cause the operation to resume, and register ( 400c will provide the output of adder 402c to the first input of adder 402b.

The state of PN generator 426 (S = [S ₀ ... S _r-1 ]) is provided in line 430 for masking element 428. Masking element 428 includes a set of AND gates 422 and a modulo-2 adder 424. Each bit of register 400 is provided to a first input of a corresponding AND gate 422. The second input of the AND gate is provided by masking polynomial M provided at bus line 420. The result of the AND operation between masking polynomial M and state S _I is provided to modulo-2 adder 424. Modulo-2 adder 424 performs modulo-2 summation at the input of AND gate 422 and outputs the final sequence to modulate Q channel PN _Q.

Using the present invention, a single generator polynomial can be used to generate a PN spreading sequence despite the desired final chip rate until the polynomial can produce a sufficient number of states for the highest chip rate. This is achieved by simply adjusting the truncation of the sequence according to the desired number of states for a given chip rate. For example, when using a 3.6864 Mcps diffusion rate, the sequence will be truncated after a certain number of states.

In a preferred embodiment of the present invention, the same initial phase offset is used at all chip rates. As shown in FIG. 13, the initial phase of the PN sequence is the same despite the final chip rate. As shown in Fig. 13, a PN sequence large enough for any given chip rate is selected. In this case, a maximum length sequence of 2 ²⁰ -1 is used. In this example, the state of the PN sequence used to spread the I channel is not overlapped with the state of the PN sequence used to spread the Q channel. However, all sequences contain a common initial state. Having a common initial phase, an initial state (or final state) that automatically leads to fixing the offset between the I and Q channels, enabling the use of a common mask for all chip rates for identification between the I and Q spreading sequences. You can save hardware by allocating.

Each base station will spread the forward link signal according to the offset version of the same truncation sequence. To achieve a chip rate of 3.6864 Mcps, the sequence will not be truncated until approximately three times later than many states using the same hardware and the same generator polynomial.

As mentioned above, in cdmaOne and the proposed cdma2000 system, each base station transmits a phase offset version of a common PN sequence. In the case of truncated PN sequences, this leads to a unique problem with the masking operation used to generate the offset truncated sequence. 11A and 11B illustrate problems in generating offset truncated PN sequences.

11A illustrates the generation of an uncut offset PN sequence. The sequence shown as circle 500 begins in state 501 and repeats returning to state 501. The offset sequence shown by circle 502 starts at PN offset position 503 and returns to initial state 503 and repeats this. When the sequence is not truncated, this can be achieved by using a simple masking operation as described above, which masks the sequence towards creation or slows down.

11B, however, when the sequence is truncated, the traditional masking operation is inappropriate. Arc 504 shows an unshifted PN sequence. The generator begins at initial state 505 and moves to final state 507 after returning to state 505 as disclosed. Arc 506 represents the truncated PN sequence 504 shifted using traditional masking techniques. When using the traditional masking method, the shift sequence begins at state 508 and continues to 509 before returning to state 508. This is not an exactly phase shifted version of the first PN sequence. The correctly phase shifted version, the truncated PN sequence, begins in the starting state 508 and proceeds to the final state 507, returns to the starting state 505, and passes through the state 508 to the final state 507. Repeat.

12 illustrates a method for providing a correctly phase shifted version of an offset PN sequence. Truncated sequence generator 600 generates a truncated PN sequence as disclosed for PN generator 306. The truncated PN sequence is provided to the mask operator 606 at line 608. Mask operator 608 performs a masking operation along a mask provided from multiplexer 604 as disclosed for masking element 428. Mask operator 608 shifts the truncated PN sequence as described above.

First, the mask operator 606 uses the first mask, denoted M1, to provide a shift in the sequence corresponding to the shift indicated by the PN offset in FIG. 11B. When the final state of the truncated sequence is reached (505 of FIG. 11B), the second mask is used by the mask operator 606 to phase shift the truncated sequence S to provide the first part of the truncated sequence.

Phase detector 602 is used to search for time for a switch between masks. Referring to Fig. 11B, phase detector 602 detects the points of mask transitions 507 and 508. Mask 1 is the state (508) from state (508). 507. Upon detection of state 507, mask 2 is used to generate a delayed sequence between states 505 and 508. On detection of state 508, a mask is used. 1 is used again.

In a typical embodiment, the state of truncated sequence generator 600 is provided with phase detector 602 at bus line 609. The phase detector 602 compares this state with the appropriate state to make the next mask change as described above. In a typical embodiment, the state at which the mask change is to be made is provided by the microprocessor 610 at the bus line to the multiplexer 612 which is stored in the memory element 608 and provides phase information to the phase detector 602. .

When detecting a request for a mask change, phase detector 602 sends a signal to multiplexer 604. In response to the signal from the phase detector 602, the multiplexer 604 switches the mask provided to the mast operator 606. In a typical embodiment, the masking value is stored in memory 608 and provided through microprocessor 610 at a common microprocessor bus line to register 612. Register 612 provides mask 1 to the first input of multiplexer 604 and mask 2 to the second input of multiplexer 604.

Although described in the context of LFSR PN sequences, the present invention may use other classes of PN sequences as other configurations, such as Gold Code and Fibonacci.

The previous descriptions of the preferred embodiments are provided to enable any person skilled in the art to make use of the invention. Many modifications to the above embodiments will be readily apparent to those skilled in the art, and the foregoing principles can be applied to other embodiments without the use of inventive equipment. Therefore, the invention is not limited to the embodiments described herein but is to the fullest extent encompassing the described principles and novel features.

Claims (1)

A device for providing a truncated PN sequence: A PN sequence generator for generating a PN sequence comprising a plurality of states greater than the truncated PN sequence; End state detection means for detecting a final state in said state of said PN sequence generator and supplying a signal indicative of detection of said final state; And Initial state loading means for loading an initial state into said PN sequence generator in response to said signal indicative of detection of said final state.