JP2002528951A - Pseudo noise diffusion method and apparatus in CDMA communication system - Google Patents

Abstract

(57) [Summary] A short PN sequence is generated. A method of PN spreading that spreads an SCMA signal using a shortened PN sequence. A method for generating the shortened PN sequence. A method of generating a second shortened PN sequence by masking the first shortened sequence. A method for generating a phase shifted version of a shortened PN sequence.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

TECHNICAL FIELD OF THE INVENTION

The present invention relates to spread spectrum communication. More particularly, the present invention relates to a new and improved pseudo-spreading method and apparatus in a direct sequence code division multiple access (CDMA) communication system.

[0002]

[Prior art]

As wireless communication technology advances, the demand for high-speed data services in a wireless environment has increased dramatically. Code division multiple access (CDMA) modulation is one of several techniques that provide digital wireless transmission well suited for transmitting digital data. Other methods of digital wireless transmission include time division multiple access (TDMA) and frequency division multiple access (FDMA).

[0003] However, the spread spectrum modulation technique of CDMA has significant advantages over other digital modulation techniques. The use of CDMA technology in multiple access communication systems is disclosed in U.S. Pat. No. 4,901,307, entitled "Spread Spectrum Multiple Access Communication Systems Using Satellites or Terrestrial Repeaters," Assigned to the assignee and incorporated herein by reference. The use of CDMA technology in multiple access communication systems is further described in U.S. Pat.
, 103,459, assigned to the assignee of the present invention and incorporated herein by reference. A method for providing digital wireless communication using CDMA modulation is described in TIA / E of Dual Mode Wideband Spread Spectrum Cellular System.
An IA / IS-95-A (hereinafter IS-95) mobile station-base station compatible standard, standardized by the Telecommunications Industry Association (TIA).

[0004] The multipath characteristics of a terrestrial channel produce a signal at the receiver that has several distinct transit propagation paths. One of the characteristics of multipath is the time spread introduced in the signal transmitted over the channel. Spread spectrum pseudo noise (PN) modulation used in CDMA systems allows different propagation paths of the same signal to be identified and combined if the path delay difference exceeds the PN chip modulation time period. If a PN chip rate of approximately 1 MHz is used in a CDMA system, a total spread spectrum processing gain equal to the ratio of spreading bandwidth to system data rate is used for paths with delays that differ by more than 1 ms. A path delay difference of 1 ms corresponds to a path distance difference of approximately 300 m. Urban environments typically provide path delay differences in excess of 1 ms.

[0005] Another property of a multipath channel is that each path through that channel can create a different attenuation factor. For example, if the ideal impulse is transmitted on a multipath channel, each pulse of the received stream of pulses typically has a different signal strength than the other received pulses.

Another characteristic of multipath is that each path through the channel can produce a different signal phase. For example, if an ideal impulse is transmitted over a multipath channel, each pulse of the stream of received pulses will typically have a different phase than other received pulses. This is due to fading of the signal.

[0007] Attenuation occurs when multipath vectors are added destructively, resulting in a smaller received signal than either individual vector. For example, if the sine wave is transmitted over a multipath channel having two paths, the first path may be XdB
If the second path has an attenuation factor of d with a phase shift of Q radians and a time delay of d with an attenuation factor of X dB and a phase shift of Q + _ radians, then the signal at the output of the channel is Not received.

As described above, in the conventional CDMA demodulation structure, the PN chip time defines that the two smallest paths to be separated must be normally combined. Before this separate path can be demodulated, the relative arrival time (or offset) of that path of the received signal must first be determined. The demodulator performs this function by "searching" for a sequence of offsets and measuring the energy received at each offset. When the energy associated with a potential offset exceeds a certain threshold, a demodulation factor, or "finger", is assigned to that offset. The signal present at the path offset is summed by the contribution of the other fingers of their respective offset.

A searcher and a method and apparatus for finger assignment based on finger energy levels are disclosed in US Pat. No. 5,528,993, filed Oct. 28, 1993, entitled "Finger Assignment in Systems Allowing the Receiving of Multiple Signals." No. 490,165, assigned to the assignee of the present invention and incorporated herein by reference. In an exemplary embodiment, the CDMA signal is a telecommunications industry association T.T.R. entitled "Mobile-Base Station Compatible Standard for Dual Mode Wideband Spread Spectrum Cellular Systems".
Sent according to IA / EIA / IS-95-A. The signal transmitted from the base station to the mobile station is referred to herein as a forward link signal, and the signal transmitted from the mobile station to the base station is referred to herein as a reverse link signal.

An exemplary embodiment of a circuit that enables demodulation of an IS-95 forward link signal is described in US Pat. No. 5,764,592, entitled “Mobile Demodulator Structure for Spread Spectrum Multiple Access Systems”. Disclosed in detail, assigned to the assignee of the present invention and incorporated herein by reference. An exemplary embodiment of a circuit that enables demodulation of an IS-95 reverse link signal is described in detail in U.S. Pat. No. 5,654,979 entitled "Cell Site Demodulator Structure for Spread Spectrum Multiple Access Communication Systems." And assigned to the assignee of the present invention and incorporated herein by reference.

FIG. 1 shows a representative set of signals arriving at a mobile station from a base station. It will be appreciated by those skilled in the art that FIG. 1 is equally applicable to signals arriving at a base station from a mobile station. The vertical axis shows the received power on a scale of decibels (dB). The horizontal axis indicates the delay of the arrival time of the signal due to the multipath delay. An axis (not shown) in the direction of entering the paper indicates a fraction of time. The signals in the common plane follow different paths that are simultaneously received by the receiver, but are transmitted at different times.

In the common plane, peaks from the right are transmitted earlier by the base station than peaks from the left. For example, the leftmost peak spike 2 corresponds to the most recently transmitted signal. Each signal spike 2-7 follows a different path and therefore exhibits a different time delay and a different amplitude response.

The six different signal spikes, indicated by spikes 2-7, are representative of a harsh multipath environment. In a typical urban environment, fewer available paths are created. The noise floor of the system is indicated by peaks and slopes having lower energy levels.

The searcher's task is to identify the delay measured by the horizontal axis of the signal spikes 2-7 of the potential finger assignment. The finger's task is to combine one of the sets of multipath peaks and demodulate it into a single output. Tracking peaks as they move over time is also a task of a finger once assigned to a multipath peak.

The horizontal axis can be considered as having units of PN offset. At any given time, the mobile station receives a variety of signals from base stations having different sequences, each following a different path than the other. The signal of the base station is demodulated by the PN sequence. A local copy of the PN sequence is also generated at the mobile station. In the mobile station, each multipath signal is individually demodulated with a PN sequence code aligned with its reception time offset. The horizontal axis coordinates may be considered to correspond to the PN sequence code offset used for demodulating the signal at that coordinate.

As indicated by the non-uniform ridges of each multipath peak, the amplitude of each multipath peak varies as a function of time. There is no major variation of the multipath peak for that time limit shown. Over time, over an extended time range, the multipath peaks disappear and new paths are created. Also, as the mobile station moves relative to the base station, the peak may slide to an earlier or later offset as the path distance varies. Each finger tracks these small variations in the signal assigned to it.

In a narrowband system, the presence of multiple paths in a wireless channel can cause severe fading beyond the narrow frequency band utilized. In such a system,
Performance is constrained by the extra transmit power needed to overcome severe attenuation. As mentioned above, the CDMA signal paths are identified and diversity combined in the demodulation process.

There are three main types of diversity: time diversity, frequency diversity, and space / path diversity. Time diversity is
It is best obtained with the use of repetition, time interleaving, and error correction and detection coding that results in redundancy. The system can use each of these technologies as a form of time diversity.

Due to its inherent wideband properties, CDMA provides a form of frequency diversity by spreading signal energy over a wide bandwidth. Frequency selective attenuation, which can cause severe attenuation over the frequency bandwidth of a narrowband system, only affects the few frequency bands normally used by CDMA spread spectrum signals.

The rake receiver provides path diversity to combine multipath delay signals through its capabilities. All paths with their respective assigned fingers attenuate each other before the combined signal drops. Additional path diversity is obtained through a process known as "soft handoff" where multiple simultaneous, redundant links from two or more base stations can be established at the mobile station. This supports robust links in challenging environments in the cell boundary area. An example of path diversity is U.S. Pat. No. 5, issued March 21, 1992, entitled "Soft Handoff in CDMA Cellular Telephone Systems."
U.S. Pat. No. 5,109,501, issued Apr. 28, 1992 and entitled "Diversity Receivers in CDMA Cellular Telephone Systems".
No. 390, both of which are assigned to the assignee of the present invention.

The cross-correlation between the different PN sequences and the auto-correlation of the PN sequences for all non-zero time shifts has a nearly zero mean value. This allows different user signals to be identified on reception. Autocorrelation and cross-correlation require a similar mapping such that a logical "0" takes on a value of "1", a logical "1" takes on a value of "-1", or a zero mean value is obtained. I do.

However, such PN signals are not orthogonal. Although the cross-correlation is essentially averaged to zero over the entire short sequence length, such as the information bit time, the cross-correlation is randomly variable with a binomial distribution. Thus, like Gaussian noise of a wide bandwidth at the same power spectral density, the signals interfere with each other in substantially the same manner.

A set of n orthogonal binary sequences n, each of length n, which may be any power of 2
Can be built (Digital Communications for Spatial Applications, SW Golomb)
Prentice-Hall inc. Pp. 1964 45-64
reference). In fact, sets of orthogonal binary sequences are also known for most multiplexed lengths less than 4 and 200. One layer of such a sequence that is easily generated is called a Walsh function. The Walsh function of order n is functionally defined as

(Equation 1)

Here, W ′ is a logical complement of W, and W (1) = | 0 |.

A Walsh sequence or code is one of the columns of a Walsh function matrix. A Walsh function matrix of order n contains n sequences, each length n of Walsh chips. A Walsh function matrix of order n (as well as other orthogonal functions of length n) has the property that over a time interval of n bits, the cross-correlation between all the different sequences of the set is zero. Every sequence in the set is different from every other sequence in just half that bit. There is always one sequence containing all zeros, and all other sequences are half one
It is noteworthy that half contains zero.

In the system described in the '459 patent, the call signal begins as a 9600 bit / sec source and is then converted to a 19,200 symbol / sec output stream by レ ー ト rate forward error correction encoding. . Each call signal broadcast from the cell is covered by 64 orthogonal Walsh sequences, one of each 64 Walsh chips, or one symbol during the time period. Despite the symbols being covered, the orthogonality of all sequences means that any interference from other user signals in that cell is canceled during symbol aggregation. Non-orthogonal interference from other cells, as well as multipath, limits the capabilities of the forward link.

In IS-95, all user signals transmitted by a base station have the same in-phase (
I) Quadrature phase modulation (QPSK) spreading using a channel PN sequence and a quadrature (Q) channel PN sequence. Each base station in a CDMA system transmits on the same frequency band using the same PN sequence, but with a unique offset associated with the unshifted PN sequence aligned with the universal time base. The PN spreading rate is the same as the Walsh cover rate of 1.2288 MHz, or 64 PN chips / symbol. In a preferred embodiment, each base station transmits a pilot reference. In the description of the present invention, different information is transmitted on I and Q channels which inherently increases the capacity of the system.

The pilot channel is a “beacon” that transmits a constant zero symbol and is spread with the same I and Q PN sequences utilized by the transistor carrier signal. In the preferred embodiment, all pilot channels are covered with zero Walsh sequence 0. During initial system acquisition, the mobile station may search for all possible PN sequence shifts and, once it finds it a base station pilot, may itself synchronize to system time. As will be described in detail below, the pilot plays the fundamental role of a mobile demodulator rake receiver well beyond its specification in initial synchronization.

FIG. 2 shows a wireless general rake receiver demodulator 10 that receives and demodulates a forward link signal 20 arriving at an antenna 18. Analog transmitter and receiver 16
Includes a series of QPSK downconverters that output I and Q channel samples 32 digitized at baseband. The sampling clock CHIPX840 for digitizing the received waveform is obtained from a voltage-controlled temperature-compensated local oscillator (TCXO).

The demodulator 10 is monitored by the microprocessor 30 via the data bus 34. Within the demodulator, I and Q samples 32 are provided to a plurality of fingers 12a-c and searcher 14. Searcher 14 searches for an offset window likely to contain a multipath signal peak suitable for finger 12a-c assignment. For each offset in the searched window, searcher 14 reports to the microprocessor the pilot energy found at that offset. Finger 12
a-c are then examined and those unassigned or tracking weaker paths are assigned by microprocessor 30 to the stronger paths identified by searcher 14.

Once the fingers 12a-c are locked to the multipath signal at their assigned offsets, the paths themselves are relocated until there are no more paths or until they are reassigned using their internal time tracking loop. Chase. This finger time tracking loop measures the energy on each side of the peak at the offset where the finger is currently being demodulated. The difference between these energies forms a distance,
It is then filtered and integrated.

The output of the integrator controls a decimator that selects one of the input samples across the chip internet for use in demodulation. As the peak moves, the finger adjusts its decimator position to move it. The decimated sample stream is then despread with a PN sequence that matches the offset to which the finger was assigned. The despread I and Q samples are aggregated into symbols for generating a pilot vector (P _I , P _Q ). These same despread I and Q samples are Walsh decovered, decovered, and despread I and Q with Walsh code assignments specific to the mobile user.
The samples are aggregated into symbols for generating a symbol data vector (D _I , D _Q ). The dot product operator is defined by

P (n) · D (n) = P _I (n) D _I (n) + P _Q (n) D _Q (n) (2) where P _I (n) and P _Q (n) Are the I and Q components of pilot vector P for symbol n, respectively, and D _I (n) and D _Q (n) are the I and Q components of data vector D for symbol n, respectively.

Since the pilot signal vector is stronger than the data signal vector, it can be used as an accurate phase reference for coherent demodulation. The inner product calculates the magnitude of the data vector in phase with the pilot vector. Assigned to the assignee of the present invention,
As described in pending US Pat. No. 5,506,865, entitled "Pilot Carrier Dot Product Circuit," the dot product weights the finger contributions for efficient coupling and, in effect, Each finger symbol output 42a-c is scaled by the relative strength of the received pilot. Thus, the dot product performs the dual role of both phase projection and finger symbol weighting required for a coherent rake receiver demodulator.

Each finger has a lock detector circuit that masks the symbol output to combiner 42 if its long-term average energy does not exceed a minimum threshold. This ensures that the fingers that follow the stable path only contribute to the combined output and thus enhance the performance of the demodulator.

Due to the relative differences in the arrival times of the paths to which each finger 12a-c is assigned, each finger 12a-c is placed in a finger symbol stream for symbol combiner 22 to generate a "soft-decision" demodulated symbol. It has a deskew buffer that establishes finger symbol streams 42a-c so that the 42a-c aggregates each other. This symbol is weighted by the assurance that it correctly identifies the originally transmitted symbol. The symbols are sent to a deinterleaver / decoder circuit 28 which first frame deinterleaves the symbol stream using a maximum likelihood Viterbi algorithm and then forward error correction decodes. The decoded data is then processed by the microprocessor 3 for further processing.
It is made available to other components such as 0 and conversation vocoder.

For accurate demodulation, a mechanism for aligning the local oscillator frequency with a clock used in a cell for demodulating data is required. Each finger evaluates a frequency error by measuring a pilot vector circulation rate in QPSK I and Q using a cross product vector operator.

P (n) × P (n−1) = P _I (n) P _Q (n−1) −P _I (n−1) P _Q (n) (3) Fingers 44 joined and integrated by the device 26
It is evaluated from ac. The integrator output LO_ADJ36 is a CHIPX8 clock 4
To adjust the clock frequency to zero, the analog transmitter TCXO voltage control and flow to the receiver 16 and provide a closed loop mechanism for compensating for local oscillator frequency errors.

If the transmission power of the signal transmitted by the base station to the mobile station is very high, problems such as interference with other mobile stations may occur. Alternatively, if the transmission power of the signal transmitted by the base station is very low, the mobile station may receive multiple erroneous frames.
Terrestrial channel fading and other known factors can affect the received power of signals transmitted by base stations. As a result, each base station must quickly and accurately adjust the transmission power of the signal transmitted to the mobile station.

A useful method of controlling the transmission power of a signal transmitted by a base station is to allow the mobile station to signal to the base station when the power of the received frame of data falls below a threshold or is received with an error. Or send a message. In response to this message, the base station increases its transmission power of the signal transmitted by the base station.
A transmission power control method and apparatus is disclosed in US Pat. No. 5,056,109 entitled "Transmission Power Control Method and Apparatus in a CDMA Cellular Telephone System," which is assigned to the assignee of the present invention, Incorporated here for

Recently, there has been a great deal of interest in providing high speed digital data over wireless communication links. With the provision of these high-speed data links, systems have been developed that utilize wider bandwidths. One method of providing high-speed data involves the use of CDMA techniques, which are based on approaches similar to the developments made in IS-95 system deployments. A system proposed by the Telecommunications Industry Association that provides high-speed data in wireless communication systems is entitled "cdma2000".

In transmitting a CDMA signal, it is desirable to reduce the peak to the average ratio of transmission power via the power amplifier of the wireless communication device. A method for reducing peaks to an average ratio is described in pending co-pending April 9, 1996 entitled "Peak Decreased in Average Transmit Power High Data Rates in CDMA Wireless Communication Systems" and assigned to the assignee of the present invention. U.S. Patent Application No. 08, assigned to and incorporated herein by reference.
/ 856,428, through the use of complex PN spreading. For a complex PN, the information signals I 'and Q' are PN spread according to the following equation:

[0043]

(Equation 2)

[0044] Here, PN _I and PN _Q are distinct PN spreading codes. This complex PN spreading is very useful when the information content of the information signal channels (I 'and Q') is substantially higher than others. Complex PN spreading adjusts the load and thus reduces the peaks to the average ratio. The aforementioned US patent application Ser. No. 08 / 856,428 also describes a method for despreading complex PN spread data.

The cdma2000 system uses an orthogonal code channel that is substantially spread by a pseudo-noise sequence. This system has 1.2288 Mcps, 3.6
It is designed to provide spreading for a given set of chip rates, including 864 Mcps, 7.3728 Mcps and 11.0592 Mcps. Providing a different set of PN generators for each chip rate requires additional hardware and increases maintenance costs. It is also desirable that maintenance can function at each of the chip rates. Further, a method of searching for a pilot signal is as follows.
It is desirable to provide fast and efficient acquisition.

[0046]

[Means for Solving the Problems]

The above and further features, objects, and advantages of the present invention will become more apparent from the following detailed description of embodiments of the invention, when considered in conjunction with the drawings. In the drawings, like reference numbers are identified as similar.

SUMMARY OF THE INVENTION The present invention is a new and improved PN spreading method for CDMA signals. The present invention describes a method for generating a shortened PN sequence. The invention further describes a method of masking the first shortened sequence to generate a second shortened PN sequence. A method for generating a phase-shifted version of a shortened PN sequence is also described.

[0048]

BEST MODE FOR CARRYING OUT THE INVENTION

The features, objects and advantages of the present invention will become more apparent in the detailed description set forth below in connection with the drawings. In the drawings, the same reference numerals refer to the same throughout the drawings.

While it is possible to handle today's demands with a spectrally efficient multiple access system as defined in IS-95 (referred to as the cdmaOne system), manufacturers and operators are not Are interested in the growing demand for these systems and the increasing popularity of wireless telephones. In anticipation of this growing demand, the International Telecommunication Union (ITU) has launched a program for standardizing third generation wireless communication systems.

The Telecommunications Industry Association has proposed “cdma2000 ITU-R Standard Proposal (0.18
) "(Hereinafter cdma2000 Proposal) submitted to the ITU a proposed Radio Transmission Technology (RTT) standard. The cdma2000 proposal operates at a set of different chip rates that can scale the system to meet the required capacity. CD
The chip rate of the MA system determines the amount of data that can be transmitted by the system through the required spreading gain factor. Especially cdma2000 proposal is 1.2288 Mc
ps (chip rate of current cdmaOne system), 3.6864 Mcps
, 7.3728 Mcps, 11.0592 Mcps, and 14.7456 Mc
Provides operation in ps.

In the cdmaOne system, each base station transmits its forward link transmissions using a common PN spreading code offset by a predetermined amount from other nearby base stations, similar to those considered in the cdma2000 system. Spread. The amount of offset required between base stations is a function of the maximum propagation path predicted by the system designer.

As described in the aforementioned US Pat. No. 5,764,592, a CDMA receiver generates a local version of a PN sequence to despread the received signal. The PN sequence generated at the receiver is based on the PN sequence used to perform spreading at the base station due to the time required to propagate the signal from the base station to the mobile station receiver.
Offset from N sequences. If the PN offsets of the base stations within a given area are very close to each other, the base station PN spreading is not unique to the mobile station, so that the mobile station can It cannot be distinguished from the received signal. Therefore, the amount of PN offset between base stations is a function of the maximum predicted propagation time of a signal reaching the mobile station.

Since the required PN offset is a function of the propagation time, the minimum amount of PN chip offset between base stations is the product of the maximum propagation time and the PN spreading rate. Furthermore, PN
The number of required states created by spreading is equal to the minimum amount of PN chip offset between the base station and the number of base stations that can interfere with or communicate with the mobile station. Thus, a shorter PN code requires system operation at 1.2288 Mcps, rather than system operation at 3.6864 Mcps.

FIG. 3 shows an outline of a downlink (forward link) transmission proposed in the cdma2000 proposal. A frame of data is a CRC and a tail bit generator 1 that generates a set of parity bits for the frame, called a cyclic redundancy bit.
02. Methods for generating cyclic redundancy check bits are known in the art, and methods for generating CRC bits are described in detail in US Pat. No. 5,504,773, entitled "Method and Apparatus for Formatting Transmission Data." And this patent is assigned to the assignee of the present invention and incorporated herein by reference. At this time, the CRC and tail bits generator 102 adds a set of tail bits to the frame used to clear the decoder and receiver memory.

Next, a packet of data is provided to encoder 104. Encoder 104 may be either a convolutional encoder or a turbo encoder. Convolutional encoders are known in the art. A convolutional encoder allows the use of a trellis decoder at the receiver, greatly reducing the amount of energy required to accurately transmit data to a remote station. Alternatively, the encoder 4 may be a turbo encoder, the design of which is known in the art, an example being "interactive decoding method, at least two systematic convolutional codings corresponding to the decoding module and the decoder. US Patent No. 5,466,747, entitled "Error Correction Coding Method With", which is incorporated herein by reference.

The encoded symbols are provided by encoder 104 to interleaver 106. Interleaver 106 provides time diversity that organizes symbols to prevent burst-type errors common in wireless environments. In an exemplary embodiment, interleaver 106 is a block interleaver, in which data is read into memory elements in rows and read in columns.

The organized symbols are provided to a scramble element 112. The scramble element 112 scrambles the interleaved data according to the decimated length PN sequence. The scrambling is completed by adding modulo-2 to the interleaver output code having the binary value of the long PN chip. The long code is generated by a linear feedback shift register (LFSR) that passes through a mask function.
The mask function is a user identification function, and is generally a user's electronic serial number (
ESN). The long code generated at the LFSR is masked and provided to bit selector 110, where it decimates the sequence for the proper rate. The decimating sequence is provided to a scrambling element 112, which provides an additional level of call security for the user.

The scrambled symbols are passed to the multiplexer and signal point mapping element 114
Provided to In an exemplary embodiment, each set of 2 bits points to a multiplexer and a set of binary bits (1,1), (1, -1), (-1,1).
) Signal point mapping element 1 that maps to an array consisting of (-1, -1)
14 is provided. One of the points of this constellation mapping is provided on a first output to a data channel gain element 116, and a second point of the constellation is provided on a second output to a data channel gain element 118.

The gain adjusted packets are provided to puncture elements 122 and 28 that adjust the transmission gain of the packets. Next, the gain-adjusted packet is puncture element 1
22 and 124. Power control bits that control the transmission power of the remote station (not shown) are provided to a power control channel gain element 120.

Power control gain element 120 adjusts the gain of the power control bits and provides gain adjusted power control bits to puncture elements 122 and 124. Puncture elements 122 and 124 have their power control bits punctured in packets in place. At this time, the packet is provided to cover elements 128 and 130.

The packet is composed of ± 1 value symbols. Symbols are cover elements 128 and 1
30, where the symbols are multiplied by an orthogonal sequence comprised of ± 1 values. This orthogonal sequence is assigned to traffic transmission to a particular remote user. The orthogonal sequence of the exemplary embodiment is a Walsh sequence, the generation of which is known in the art, and is described in U.S. Pat. No. 5, entitled "Systems and Methods for Generating Signal Waveforms in CDMA Cellular Telephone Systems." 103, 4
No. 59, assigned to the assignee of the present invention and incorporated herein by reference.

Next, a Walsh cover sequence is provided to the forward link channel adder, where the data from cover elements 128 and 130 and other similar modulated data packets are added to provide other user and common channel data. Is transmitted.

Returning to FIG. 2, next, the obtained addition data is stored in the PN spreading elements 134, 136 and 13.
8 and 140. Each base station is identified by a PN offset that is unique to each base station in the set of closely located base stations. In the present invention, two P
Spread the data using N sequences (PN ₁ and PN _Q ). Each PN sequence is composed of a sequence of ± 1 values. The following result is obtained by performing the PN spreading operation shown in FIG.

[0064]

(Equation 3)

The data sequence I ′ is provided to multipliers 134 and 138. The multiplier 134 multiplies the data (I ′) by the pseudo noise sequence PN ₁ and outputs the result to the subtractor 14.
Provide 2 for the add input. Multiplier 138 multiplies data (I ′) by pseudo-noise sequence PN _Q and provides the result to a first summing input of adder 144.

The data sequence Q ′ is provided to first inputs of multipliers 136 and 140. The multiplier 140 multiplies the data (Q ′) by the pseudo noise sequence PN ₁ and provides the result to a subtraction input of a subtractor 142. Multiplier 136 multiplies data (Q ′) by pseudo noise sequence PN _q and provides the result to a second summing input of adder 144.

The subtractor 142 subtracts the output of the multiplier 140 from the output of the multiplier 134, and provides the result to the baseband filter (BBF) 146. Adder 14
4 adds the output of multiplier 138 to the output of multiplier 136 and provides the result to baseband filter (BBF) 148. Baseband filter 146,
148 filters the PN spreading sequence and provides the filtered sequence to upconverters 150, 152, respectively.

Upconverters 150, 152 upconvert input data by quadrature phase shift (QPSK) modulation as is known in the art. Upconverter 150 according carrier modulation cos (2_f _c), the signal for transmission (
Upconvert I). The up-converter 152 has a carrier modulation sin
In accordance with (2_f _c), up-converts the signal (Q). The two orthogonal signals are then summed in a summing element 154 which is amplified and transmitted.

In the present invention, a method is proposed for spreading data with a unique PN sequence that spreads for both the I and Q channels using a single PN generator. The PN sequence used to spread the Q channel data is used to spread the I channel to provide the inherent spreading of the I and Q channels using a single PN generator. Offset from the PN sequence.

A linear feedback shift register (LFSR) is commonly used to generate a PN sequence for CDMA spreading. The output of the LFSR circulates through a predetermined number of unique states before returning to its initial state, and then the output cycle begins again. In a maximum length LFSR sequence, the shift register goes through all possible states of its memory before returning to the initial state. This is the case when the polynomial is chosen such that the feedback is of maximum length. The output of the LFSR is a function of the state of the LFSR.

The PN sequences (PN _I , PN _Q ) may be generated either by different polynomials (ie, PN generators) or by different phases of the same PN generator. In a first exemplary embodiment of the I and Q channel PN generators, the I and Q channels are generated using the same PN generator as the sequences offset from each other.

As is known in the art, the phase of the output of the PN generator can be changed using a masking operation performed on the output state of the LFSR. FIG. 5 shows a long code PN generator used in an IS-95 system. LFSR
175 provides the maximum length sequence to the masking element 177. Masking element 177 is a series of AND gates that change the phase of the output. The mask in this case is a 42-bit input sequence that turns the LFSR tap output on and off. The masking operation changes the phase of the output sequence. The masked sequence is then provided to a modulo-2 adder 179 that outputs a long code sequence.

FIG. 6 shows a generalized version of the LFSR with a feedback path determined by a polynomial. G = g ₀ + g ₁ X + g ₂ X ² +... G _r X ^r−1 (8) where g ₀ is the rightmost feedback and is always equal to “1”, and g _n is the LFSR Is a binary value that either enables or disables the corresponding feedback path, and X indicates the position of the feedback. It will be appreciated by those skilled in the art that the feedback polynomial (G) can be hardwired where an AND gate is not required. The LFSR 206 includes a set of adders 202, a register 200,
And an AND gate 204. The state of the LFSR is determined by this value to adder 202. The LFSR starts with an initial state set by loading register 200.

When the clock is managed by the circuit, the value stored in register 200 is output to a first input of adder 202. The value output by the register 200a is
The output of LFSR 206 and is further fed back on feedback line 208 as one input to AND gate 204. Another input to AND gate 204 is a binary value (g ₀ , g ₁ ... G _r ).

In the first embodiment, the sequence generated for spreading the I and Q channels is a phase shifted version of the large PN sequence. In order to allow a single PN generator to generate non-overlapping PN sequences, the PN generator generates a sequence that is at least twice as long as the sequence used to spread the I and Q channels. Must be generated,
And the sequence needs to be shortened.

FIG. 7 shows the state or phase of the LFSR around the circle. The circular shape provides an effective visualization of the output of the LFSR, as the state of the LFSR starts at the initial state, circulates through all output states in a circle, returns to the initial state, and repeats.

In a first embodiment of the present invention, the sequence used to spread the I and Q channels is generated using a shortened PN sequence. In FIG. 7, the LFSR is shown to have N unique states. The sequence used to spread the I channel (PN _I ) is reduced to k states (k <N / 2). Thus, a PN generator that is configured to cycle through all N possible states is shortened when it reaches the k + 1 state since it was pushed to the initial state.

In a first preferred version, the sequence used to spread the Q channel is offset from the sequence used to spread the I channel by an amount such that the two sequences do not overlap. As shown in FIG. 7, the sequence used to spread the Q channel (PN _Q ) consists of states between m and m + k, where m> k and m + k <N. In an exemplary embodiment, PN _Q is generated by masking a PN _I sequence.

In another embodiment, PN _I and PN _Q are shortened PNs with different polynomials
It can be generated as a sequence. The description using a single polynomial is sufficient to enable one embodiment where two different polynomials are used. The generation of different PN sequences using multiple polynomials is known in the art.

FIG. 8 shows a PN generator 306 of the present invention for generating a shortened PN sequence. It will be appreciated by those skilled in the art that the feedback polynomial (G) can be hardwired where an AND gate is not required. P
The initial state (S _I ) of N generator 306 is provided to register 300 on bus line 312. In an exemplary embodiment, the register 300 has the following structure:
Applicable to the same extent and within the scope of the present invention, they are disclosed as D-flip-flops. The output of the PN generator 308 is output from the register 300a. In addition, the output from register 300a is fed back on line 308 to a first input of AND gate 304. The polynomial G is represented by the input line 31
4 is provided as the second output of AND gate 304.

The polynomial G enables or disables the feedback provided at line 308. If the input from the input line 314 to the AND gate 304 is high, then feedback to the corresponding modulo-2 adder 302 is enabled,
Then, the output bit is fed back to the adder. Input line 314
If the input to AND gate 304 is low, then the corresponding modulo-
Feedback to the adder 302 is disabled and output bits are blocked from reaching the adder. The output of each of the registers 330b through 300r is provided to a first input of a corresponding modulo-2 adder 302a through 302 (rl). The second input to modulo-2 adders 302a through 302 (r-1) is the feedback output bit from register 300a where the feedback path is enabled by polynomial G.

To shorten the PN sequence, a state identification element 310 is used to compare the state of the LFSR with the final state of the shortened PN sequence. Various methods can be used to implement the state sensing element 310, with the easiest concepts being one input each receiving one bit from the current state and one input from the final state. And a second input receiving one bit. The output of each of the XOR gates is provided to an r-input AND gate, and the result of this operation is output on line 318.

Returning temporarily to the generation of the I-sequence of FIG. 7, the PN generator 306 is initially set to an initial state (indicated by 0). The final state of the shortened sequence (state k +
When 1) is reached, the state identification element 310 signals the register 300 which causes the register to load the initial state (S _I ). Once detected by the state detecting element 310 of the final state S _f, the state detection element 310 outputs a binary signal indicative of the identity of the final state. Signal on line 318, the respective registers 300, to switch to output the initial state S _I from outputting the sum resulting from the sum elements previously described.

For example, register 300c often takes an output from adder 302c and provides the output to modulo-2 adder 302b. However, when a signal for displaying the final state detected by the line 318 is high, then the register 300c outputs the Si ₂ to modulo-2 adder 302b. At the next clock interval, state sensing element 310 detects the difference between the state of the PN generator and the final state, and sets the signal low on line 318 to resume operation, and register 300c adds The output of summer 302c is provided to a first input of summer 302b.

FIG. 9 shows an exemplary implementation of a register 300 that uses a loadable D-type flip-flop 320. The A input receives the output of the corresponding adder. B input receives the bits of the initial state _{S i.} The value on line 318 is the output (Q
) Is provided to the LD which determines whether to provide input A or input B. The clock runs at the PN chip rate.

FIG. 10 shows two pseudo-noise sequences PN _I , phase offset from each other.
4 shows an apparatus for generating a PN _Q. Pseudo-noise sequence PN _I is, P in FIG. 9
Generated by PN generator 426 in the same manner as described for N generator 306. The mask (M) polynomial and the feedback (G) polynomial are represented by AN
It will be appreciated by those skilled in the art that the D-gate can be hardwired where it is not needed. The state of the PN generator 426 is the sequence PN
Are provided to a masking element 428 that phase shifts the output PNQ to

The PN generator 426 of the present invention generates a shortened PN sequence. PN generator 4
The initial state of 26 (S _I ) is provided to the register 400 on the bus line 412.
In an exemplary embodiment, register 400 is disclosed as a D-flip-flop, although other memory structures are equally applicable and within the scope of the present invention. The output of PN generator 426 is output from register 400a. In addition, the output from register 400a is fed back on line 408 to a first input of AND gate 404. Polynomial G is provided on input line 414 as a second input of AND gate 404.

The polynomial G either enables or disables the feedback provided at line 408. If the input from bus line 414 to AND gate 404 is high, then feedback to the corresponding adder 402 is enabled and the output bits are fed back to that adder. Input line 41
4 to the AND gate 404 is low, then the corresponding adder 40
Feedback to 2 is disabled, and the output bits are blocked from reaching the adder. The output of each of the registers 400b through 400r is provided to a first input of a corresponding adder 402a through 402 (r-1) (not shown). The second input from adder 402a to 402 (r-1) (not shown) is a register 400a whose feedback path is at a location enabled by polynomial G.
This is the feedback output bit from.

To shorten the PN sequence, the state identification element 410 is used to compare the state of the LFSR with the final state of the shortened PN sequence. Various methods can be used to implement the condition sensing element 410, the easiest concept being:
Each consists of a plurality of XOR gates having one output receiving bits from the current state and a second input receiving bits from the final state.
The output of each of the XOR gates is provided to an r-input AND gate, and
The result of this operation is output on line 418.

Returning temporarily to the generation of the I-sequence of FIG. 7, the PN generator 406 is initially set to the zero state. Upon reaching the final state of the shortened sequence (state k), the state identification element 410 causes the register 40 to load the register with the initial state (S _I ).
Send a signal to 0. When detected by the state detecting element 410 of the final state S _f, the state detection element 410 outputs a binary signal indicative of the identity of the final state. Signal on line 418, the respective registers 400 from outputting the sum resulting from the sum elements described above, to switch to output the initial state S _i.

For example, in many cases, register 400c takes an output from adder 402c and provides the output to adder 402b. However, when the signal indicating the detected final state goes high at line 418, then register 400c
It outputs Si ₂ to the adder 402b. At the next clock interval, the state detection element 310
Detects the difference between the state of the PN generator and the final state. And line 41
The signal is set low at 8 and operation resumes, and register 400c provides the output of adder 402c to the final input of adder 402b.

The state of the PN generator 426 (S = [S ₀ ... S _r−1 ]) is provided on line 430 to the masking element 428. Masking element 428 comprises a set of AND gates 422 and a modulo-2 adder 424. Each of the bits from register 400 is provided to a first input of a corresponding AND gate 422. The second input of the AND gate is provided by a masking polynomial M provided on bus line 420. Result of AND operation between the masking polynomial M and state _{S I} is provided to the modulo-2 adder 424. Modulo-2 adder 424 performs a modulo-2 addition on the input from AND gate 422 and outputs the resulting sequence for modulating the Q channel (PN _Q ).

Using the present invention, a single generator polynomial is independent of the desired resulting chip rate as long as the polynomial can generate a sufficient amount of state for the highest chip rate. Instead, it can be used to generate a PN spreading sequence. This is easily achieved by adjusting the sequence shortening according to the desired amount of conditions for a given chip rate. For example, 3.68
When using a 64 Mcps spreading rate, the sequence is shortened after a predetermined amount of state.

In the preferred embodiment of the present invention, the same initial phase offset is used for all chip rates. As shown in FIG. 13, the initial phase of the PN sequence is the same regardless of the resulting chip rate. As shown in FIG. 13, a PN sequence that is large enough for all of the predetermined chip rates is selected. In this case, a maximum length sequence of 2 ²⁰ −1 is used. In the example, the state of the PN sequence used to spread the I channel does not overlap with the state of the PN sequence used to spread the Q channel. However, all sequences contain a common initial state. By having a common initial phase, hardware preservation automatically
Can be achieved by aligning the initial (or final phase) that results in fixing the offset between the channel and the Q channel;
Thereby, it is possible to use a common mask for all chip rates to distinguish between I and Q spreading sequences.

[0095] Each base station spreads the forward link signal with an offset version of this same shortened sequence. Thus, to achieve a chip rate of 3.6864 Mcps, the sequence is not shortened until it is almost three times as large using the same hardware and the same generator polynomial.

As described above, in the cdmaOne system and the proposed cdma2000 system, each base station transmits a phase offset version of a common PN sequence. For a shortened PN sequence, this is an inherent problem for the masking operation used to generate the shortened offset sequence. FIG.
) And FIG. 11 (b) illustrate the problem of generating a shortened offset PN sequence.

FIG. 11A shows the generation of an unshortened offset PN sequence. The sequence indicated by circle 500 starts at state 501, returns to state 501, and repeats. The offset sequence indicated by circle 502 starts at PN offset position 503, returns to initial state 503, and repeats. When the sequence is not shortened, this can be achieved using a simple masking operation as described earlier, which shifts the sequence to a state sooner or later in generation.

However, returning now to FIG. 11B, when the sequence is shortened, the conventional masking function is inappropriate. Arc 504 indicates an unshifted PN sequence. The generator starts at an initial state 505 and moves to a final state 507, after which the generator returns to the state 505, as described above. Arc 506 represents a shortened PN sequence 504 that is shifted using conventional masking techniques. Using conventional masking methods, the shifted sequence is
Begin at state 508 and continue to 509 before returning to state 508. This is not an exact phase shifted version of the first PN sequence. Shortened P
The correctly phase shifted version of the N sequence starts at the initial state 508, proceeds to the final state 507, returns to the initial state 505, passes through the state 508 to the final state 507, and repeats.

FIG. 12 illustrates a method for providing an accurately phase shifted version of an offset PN sequence. The shortened sequence generator 600 includes a PN generator 306
Generate a shortened PN sequence as described for. The shortened PN sequence is provided to mask operator 606 on line 608. The mask operator 608 selects the multiplexer 6 as described with respect to the masking element 428.
The masking operation is performed using the mask provided from the mask 04. The mask operator 608 shifts the shortened PN sequence as described above.

First, the mask operator 606 uses the first mask, denoted M1, to provide a shift in the sequence corresponding to the shift denoted by the PN offset in FIG. 11 (b). Final state of the shortening sequence (505 in FIG. 11B)
, The second mask is used by the mask operator 606 to phase shift the shortened sequence S to provide an initial portion of the shortened sequence.

The phase detector 602 is used to detect time to switch between masks. Returning to FIG. 11B, the phase detector 602 detects points 507 and 508 of the mask transition. Mask 1 is used to generate a sequence that proceeds from state 508 to state 507. Upon detecting state 507, mask 2 is used to generate a delayed sequence between states 505 and 508. Upon detecting state 508, mask 1 is used again.

In an exemplary embodiment, the state of the shortened sequence generator 600 is line 6
At 09, it is provided to the phase detector 602. Phase detector 602 compares the state to the appropriate state at which to change the next mask, as described above. In the exemplary embodiment, the state where the mask change is to be performed is stored in memory element 608 and provided by bus 610 to microprocessor 610 to multiplexer 612 which provides phase information to phase detector 602. You.

Upon detecting a request to change the mask, the phase detector 602 sends a signal to the multiplexer 604. In response to the signal from the phase detector 602, the multiplexer 604 switches the mask provided to the mask operator 606. In an exemplary embodiment, the masking value is stored in memory 608 and provided to register 612 from the microprocessor on a common microprocessor bus line. Register 612 provides mask 1 to a first input of multiplexer 604, and provides mask 2 to a second input of multiplexer 604.

Although described in the context of the LFSR PN sequence, the present invention
It can be extended for use with other classes of PN sequences, such as ld codes, and other configurations, such as Fibonacci.

The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein are:
It can be applied to other embodiments without using the power of the present invention. Thus, the present invention is not intended to be limited to the embodiments shown, but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

[Brief description of the drawings]

FIG. 1 is a diagram illustrating a multipath signal in a CDMA environment.

FIG. 2 is a diagram illustrating a rake receiver that receives a CDMA signal.

FIG. 3 is a block diagram of CDMA signal initial processing in a third generation CDMA communication system.

FIG. 4 is a block diagram of final processing of a CDMA signal in a third generation CDMA communication system.

FIG. 5 is a block diagram illustrating generation of a mask PN sequence used in an IS-95 system.

FIG. 6 is a diagram showing a general PN generator.

FIG. 7 is a diagram showing a state of a shortened PN sequence and a state of an offset shortened PN sequence.

FIG. 8 is a block diagram of an exemplary embodiment of a circuit used to generate a shortened PN sequence.

FIG. 9 illustrates an exemplary embodiment of a register used in the LFSR of FIG.

FIG. 10 is a block diagram of an exemplary embodiment of a circuit used to generate a shortened PN sequence and a second shortened PN sequence using a single PN generator.

FIG. 11 illustrates a problem that occurs in an attempt to phase shift a shortened PN sequence.

FIG. 12 is a block diagram of a circuit that provides a phase-shifted version of a shortened PN sequence.

FIG. 13 illustrates a preferred embodiment of the present invention in which the initial phase of the PN sequence is the same regardless of the resulting chip rate.

──────────────────────────────────────────────────続 き Continuation of front page (81) Designated country EP (AT, BE, CH, CY, DE, DK, ES, FI, FR, GB, GR, IE, IT, LU, MC, NL, PT, SE ), OA (BF, BJ, CF, CG, CI, CM, GA, GN, GW, ML, MR, NE, SN, TD, TG), AP (GH, GM, KE, LS, MW, SD, SL, SZ, TZ, UG, ZW), EA (AM, AZ, BY, KG, KZ, MD, RU, TJ, TM), AE, AL, AM, AT, AU, AZ, BA, BB, BG, BR, BY, CA, CH, CN, CR, CU, CZ, DE, DK, DM, EE, ES, FI, GB, GD, GE, GH, GM, HR, HU, ID , IL, IN, IS, JP, KE, KG, KP, KR, KZ, LC, LK, LR, LS, LT, LU, LV, MA, MD, MG, MK, MN, MW, MX, NO, (72) Invention NZ, PL, PT, RO, RU, SD, SE, SG, SI, SK, SL, TJ, TM, TR, TT, TZ, UA, UG, UZ, VN, YU, ZA, ZW Agrawal, Abnish United States 94086 Sunnyvale, South Fair Oaks Number 109,655 (72) Inventor Jaw, U-Cheun United States 92129 San Diego, California Riverhead Drive 9979 (72) Inventor Hamms Brian, United States 80301 Boulder, Colorado, Barbados Place 3309 (72) Inventor Butler, Brian K. America Country, California 92037 La Jolla, Glen wick lanes 8736 F-term (reference) 5K022 EE02 EE25

Claims (1)

[Claims] 1. An apparatus for providing a shortened PN sequence, wherein the PN generates a PN sequence including more states than the shortened PN sequence.
A sequence generator; final state detector means for detecting a final state in the state of the PN sequence generator and providing a signal indicating that the final state has been detected; and indicating that the final state has been detected. Initial state loading means for loading an initial state into the PN sequence generator in response to the signal.