JPH07307930A - Multiplex transmitted signal reproducing device - Google Patents

Abstract

PURPOSE:To stably receive an amplitude modulated and multiplex transmitted signal by detecting synchronism by reproducing a carrier wave from the multiplex transmitted signal, and demodulating multiplex transmitted signal data from that output signal. CONSTITUTION:A high frequency amplifier circuit 402 amplifies an inputted television signal and this signal is converted into an intermediate frequency for demodulation by a frequency converting circuit 403. Afterwards, the multiplex transmitted and digital coded signal of a sound signal band is extracted from the converted signal by a band pass filter 414 at the frequency converting circuit 403. Then, a synchronism detecting circuit 415 detects synchronism corresponding to the carrier wave reproduced by a carrier wave reproducing circuit 416, the quadrature multiplex transmitted signal is detected and demodulated, and a spectrum suppression processing signal reproducing circuit 417 processes the demodulated signal and returns it to the signal before spectrum suppression processing. Afterwards, concerning that signal, an error generated in the middle of transmission is detected and corrected by a digital signal processing circuit 418, and that signal is returned to an analog signal by a DAC 419 and outputted from an output terminal 420.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a multiplex transmission system, and more particularly to a signal reproducing device of a transmission system effective for multiplexing and transmitting a digitally encoded audio signal and the like in a system for transmitting a video signal.

[0002]

2. Description of the Related Art Conventionally, a method of transmitting a digitally encoded audio signal (hereinafter referred to as PCM audio) by multiplexing it with a video signal is described in Satellite Broadcasting by the Institute of Radio Engineers of Japan, published in June 1983. It is reported in the Technical Survey Report Part 1 "Satellite Broadcast Receiver". The contents are shown below.
5.72 for NTSC video signals up to 4.2 MHz
72 MHz subcarriers are multiplexed at different frequencies. The subcarrier is QPSK modulated with PCM voice. However, this system cannot be implemented while achieving compatibility with the current terrestrial television broadcasting because the subcarrier frequency is outside the band of the current terrestrial television broadcasting.

Regarding the method of multiplexing other signals for the current terrestrial television broadcasting, refer to the Broadcasting Technology Sosho 2 "Broadcasting Method" published by the Japan Broadcasting Corporation in January 1983 and edited by the Japan Broadcasting Corporation. It is described on pages 205 to 208. However, a method for obtaining the transmission rate of about 1 megabit / second or more required for transmitting high quality PCM voice has not been described.

[0004]

SUMMARY OF THE INVENTION The above prior art is about 1
No consideration has been given to the fact that a signal having a transmission rate of megabits / second is multiplexed with the current terrestrial television broadcasting and transmitted, and there is a problem that high quality PCM voice cannot be multiplexed.

It is an object of the present invention to provide a signal reproducing apparatus of a transmission system which multiplexes another signal on a signal which is amplitude-modulated and transmitted and transmits the signal. In particular, it is to provide a reproducing device for stably receiving a signal multiplexed and transmitted in a transmission system in which signal data such as PCM audio is multiplexed and transmitted to current terrestrial television broadcasting. In the device, interference from a video signal demodulated by a current television receiver is reduced and stable reception is obtained.

[0006]

The above object is to provide a carrier wave which is modulated by a signal in which an amplitude-modulated carrier wave and a quadrature-phase carrier wave are multiplexed and is amplitude-modulated at a lower level than the amplitude-modulated carrier wave. A signal to be multiplexed and transmitted after being combined is used as digitally encoded signal data different from the video signal, and is repeated in units of horizontal scanning periods of the video signal, and at least at the same timing of two adjacent horizontal scanning periods. In the signal regenerator for demodulating the multiplexed signal data transmitted by performing the spectrum processing to obtain the same signal data returned in the opposite phase relation, the signal multiplexed and transmitted in the quadrature phase relationship by the carrier regeneration means and the synchronous detection means. Delaying means for delaying the signal after detecting the signal for a time corresponding to the repetition of signal data, and the input and output signals It is achieved by providing a spectrum processed signal reproducing means for arithmetic processing such as subtraction and.

[0007]

Since the amplitude-modulated carrier wave and the quadrature-phase carrier wave are modulated by the signal to be multiplexed and transmitted, and then combined with the amplitude-modulated carrier wave and transmitted, the transmission signal is transmitted in an apparatus for receiving and reproducing the transmission signal. By synchronously detecting the transmission signal by each carrier regenerated from the signal,
The amplitude-modulated signal and the multiplexed signal are reproduced. The interference between the reproduced signals in this case is very small.
In addition, even when the amplitude-modulated video signal is obtained by demodulating the transmission signal by the envelope detection method or the pseudo-synchronization detection method as the detection method of the television receiver, the carrier wave modulated by the multiplex-transmitted signal is low. Since they are combined at the level, there is little interference with the reproduced video signal. In addition, since the same signal data is transmitted in an opposite phase relationship in two adjacent horizontal scanning periods of a video signal by spectrally processing the signal data, the detection method of the television receiver is envelope detection. In the case of the system or the pseudo-synchronous detection system, the phase variation of the color subcarrier caused by the signals multiplexed in the orthogonal relationship becomes the opposite phase, and the hue variation on the screen of the television receiver can be reduced. On the other hand, in a signal reproducing apparatus that demodulates signal data such as PCM voice from a signal that is multiplexed and transmitted, a spectrum processing signal reproducing means delays the signal data and a delay means by a time corresponding to the repetition of the signal data. Since the subtraction processing is performed with the same signal data, the amplitude of the signal data is doubled when it is transmitted by repeating twice, but the noise increases only √2 times due to its randomness, so that the reception reproduction is performed. The signal-to-noise ratio can be improved by 3 dB (the greater the number of repetitions, the more the signal-to-noise ratio is improved). Also, since the signal that amplitude-modulates the carrier wave is a video signal,
When a signal that transmits the same signal data in an antiphase relationship in a period corresponding to two horizontal scanning periods adjacent to the video signal in a period corresponding to the horizontal scanning period of the video signal is received, Since the signal data is subjected to subtraction processing with the same and opposite phase signal data delayed by one horizontal scanning period, the ghost may be caused by the correlation (horizontal correlation on the television screen) of each horizontal scanning period of the video signal. Cancels and reduces interference from leaked video signals.

[0008]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is an example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention.

Reference numeral 101 is an audio signal input terminal, and 102 is an FM.
Modulator, 103 audio signal carrier generation circuit, 104 video signal input terminal, 105 matrix circuit, 106 luminance signal processing circuit, 107 color difference signal processing circuit, 108
Is an addition circuit, 109 is a video modulation circuit, 110 is a video signal carrier wave generation circuit, 111 is an input terminal of an audio signal to be digitally encoded and transmitted, 112 is an analog / digital conversion circuit (hereinafter abbreviated as ADC), 113 is a digital signal Processing circuit 114 is a spectrum suppression processing circuit 115
Is a low pass filter, 116 is a phase shift circuit, 117 is a modulation circuit, 118 is an equalizer, 119 is an adder circuit, and 120
Is a transmission VSB filter for vestigial sideband amplitude modulation, 121
Is an adder, and 122 is an antenna.

The carrier for the audio signal generated by the audio signal carrier generation circuit 103 is frequency-modulated by the audio signal applied to the audio signal input terminal 101 in the FM modulator 102. The RGB primary color signals of the video signal applied to the video signal input terminal 104 are transferred to the matrix circuit 10.
At 5, it is divided into a luminance signal and a color difference signal. These signals are processed by the luminance signal processing circuit 106 and the color difference signal processing circuit 107, respectively, and then added by the adding circuit 108 to obtain NT.
It is processed into a composite video signal of the SC system or the like. The video signal carrier generated by the video signal carrier generation circuit 110 is amplitude-modulated in the video modulation circuit 109 by the processed video signal. The modulated signal is limited to the television transmission bandwidth by the transmission VSB filter 120 for vestigial sideband amplitude modulation, and then added to the voice signal carrier modulated with the voice signal by the adder circuit 121.
It is transmitted from the antenna 122. The above description is for the part of the transmission signal generator of the current terrestrial television broadcasting.

The part related to the present invention will be described below.
The multiplex-transmitted audio signal applied to the input terminal 111 is converted into a digital code by the ADC 112. The converted signal is subjected to so-called digital signal processing in which an error detection / correction code is added or interleave processing is performed in the digital signal processing circuit 113 so that an error occurring during transmission can be detected and corrected by the receiver on the reproducing side. It The spectrum suppression processing circuit 114 suppresses low-frequency components and the like of the processed signal. Unwanted high frequency components of the suppressed signal are deleted by the low pass filter 115. The phase of the video signal carrier generated by the video signal carrier generation circuit 110 is changed by 90 degrees by the phase shift circuit 116, and then the modulation circuit 11 is generated.
In 7, the output of the low pass filter 115 is modulated by the signal from which the unnecessary high frequency component is deleted. The amplitude characteristics of the modulated signal are corrected by the equalizer 118.
The corrected signal is added and synthesized with the carrier wave for the video signal modulated in the adding circuit 119, and is transmitted from the antenna 122. The amplitude characteristic for correction of the equalizer 118 is symmetrical with the amplitude characteristic of the IF Nyquist filter provided in the intermediate frequency stage of the video signal of the television receiver and the video carrier frequency. Equalizer 1
18 is for correcting the change from the quadrature phase of the multiplex transmission wave by the IF Nyquist filter of the television receiver on the transmission side. The phase relationship of the signal before detection of the video signal after passing through the IF Nyquist filter of the television receiver is that the modulated wave of the signal multiplexed and transmitted with respect to the video signal carrier has a quadrature phase. Also, if the addition circuit 119 adds a small amount of the multiplexed signal to the video signal, it is possible to reduce interference from the multiplexed signal with the video signal detected by the television receiver.

FIG. 2 is a spectrum diagram of the transmission signal and the like generated in FIG. Reference numeral 201 is a spectrum of a transmission signal of a video signal, 202 is a spectrum of an audio signal which is FM-modulated and transmitted, 203 is a spectrum of a multiplexed signal after spectrum suppression processing and unnecessary high frequency components are removed, 204
Is a spectrum of a multiplexed signal after being modulated, 205 is a spectrum of a multiplexed signal whose amplitude characteristic is corrected by the equalizer 118, 206 is a Nyquist filter characteristic of a receiver, and 207 is a general pseudo-synchronous detection type television receiver. It is an amplitude characteristic of a bandpass filter for carrier wave reproduction.

The spectrum 201 of the transmission signal of the video signal
Is a low frequency f ₀ − with respect to the video signal carrier frequency f ₀ .
Below f ₂ and above high frequency f ₀ + f _{3, the} characteristics are attenuated by the amplitude characteristics of the VSB filter 120 for forming the vestigial sideband amplitude modulated wave. The audio signal spectrum 202 which is FM-modulated and transmitted exists around the audio signal carrier f ₁ . On the other hand, in the spectrum of the signal to be multiplexed and transmitted, as shown in spectrum 203, the spectrum suppression processing circuit 114 suppresses low-frequency components of frequency f ₄ or lower and the low-pass filter 115 generates unnecessary high-frequency components of frequency f ₅ or higher. It has been deleted.
This is the so-called processed spectrum of the baseband signal. The spectrum after being modulated by the modulation circuit 117 has a symmetrical characteristic about f ₀ as shown in the spectrum 204. The amplitude characteristic of the equalizer 118 is frequency f
_Since the IF Nyquist filter characteristic 206 of the television receiver having a slope from ₀ −f ₆ to f ₀ + f ₆ and the characteristic symmetrical about the video carrier frequency f ₀ , the spectrum of the output signal of the equalizer 118 is the spectrum 205. As shown, the higher the frequency, the lower the attenuation. Spectrum 2
The video signal modulated wave indicated by 01 and the signal modulated wave multiplexed and transmitted by the spectrum 205 are added by the adding circuit 119. Further, the audio signal modulated wave indicated by the spectrum 202 is added by the adding circuit 121 and transmitted.

FIG. 3 is a vector diagram showing a phase relationship between a video signal modulated wave and a signal modulated wave which is multiplexed and transmitted. Reference numeral 301 is a video signal carrier vector, 302 and 303 are signal modulated wave vectors to be multiplexed and transmitted, and 304 and 305 are composite vectors thereof. At frequencies within ± f _{2 with} respect to the video carrier frequency f ₀ , the video signal modulation wave is a general amplitude modulation wave and is shown by a vector 301. In a signal to be multiplexed and transmitted in a frequency band of f ₂ or less, orthogonal carrier waves are modulated into amplitudes A and −A corresponding to digital codes 1 and 0, respectively, and shown by vectors 302 and 303, respectively. When the amplitude of the carrier of the video signal is 1, the combined signal E is E = cosωct ± Asinωct (1). Here, ωc is the angular frequency of the image carrier, and t is time. In this way, the composite vectors 304, 30 of FIG.
When viewed corresponding to 5, it is also shown as E = √ (1 + A ² ) cos (ωct− (± θ)) (2). Here, θ = tan to ¹ (A) (3).

Here, let us consider the interference caused by signals multiplexed and transmitted by the signals received and reproduced by the television receiver. When the video signal detection circuit is a synchronous detection circuit, the video signal detection output signal is cosω regardless of the value of A.
Since the signal is only the coefficient of ct (only the video signal), it is not disturbed in principle. When the video signal detection circuit is an envelope detection circuit, the interference with the video signal detection output signal changes depending on the value of A, and the lower A is, the less interference occurs. As an example, when A is 0.1, E = √ (1 + A ² ) ≈1.005 (4) This is 0.005 for 1 (about -46d
It shows that the signal of B) is affected. We think that this interference does not pose a practical problem because the SN ratio of the video signal can be 40 dB or more. On the other hand, the interference from the video signal to the digitally encoded voice is eliminated by using the synchronous detection circuit as the detection circuit for reproducing the digitally encoded voice. We also consider here the transmission signal-to-noise ratio of the multiplexed signal, whether the multiplexed signal is regenerated at the receiver due to the low level of the multiplexed signal. Video signal SN ratio is 40
In the case of dB, since the transmission bandwidth of the digitally encoded audio signal is about ¼ of the bandwidth of the video signal, the transmission SN ratio of the digitally encoded audio signal is 46 dB, and A is 0.1. Therefore, the transmission SN is 26 dB. The relationship between the SN ratio of a digital signal and the bit error rate is 10 ~ ⁴ for a general binary signal with an SN ratio of 17.4 dB.
Therefore, the signal having the SN ratio of 26 dB is a signal of good quality which can be sufficiently reproduced.

As shown by the amplitude characteristic 207, the amplitude characteristic of the band-pass filter of the carrier regenerating circuit for the pseudo-synchronous detection of the television receiver having the video signal detecting circuit of the pseudo-synchronous detection system is the frequency f ₀ ±. This is a characteristic of passing f ₇ . The spectrum 205 after modulation of the signal that has been subjected to processing such as spectrum suppression processing and has been multiplexed and transmitted has frequency f ₀ ± f ₄
Since the signal component is suppressed in the frequency band within the range, the detected video signal is less disturbed by the multiplex signal.

As described above, according to the present embodiment, the signals to be multiplexed are spectrally processed and the multiplexing level is set to be smaller than that of the video signal. There is.

FIG. 4 is a block diagram of a television transmission signal reproducing apparatus embodying the present invention. 401 is an antenna, 402 is a high frequency amplification circuit, 403 is a frequency conversion circuit, 404 is an IF Nyquist filter, 405 is an intermediate frequency amplification circuit, 406 is a video signal detection circuit, 407 is a video signal amplification circuit, 408 is a color difference signal demodulation circuit, Reference numeral 409 is a primary color signal demodulation circuit, 410 is a cathode ray tube, 411 is an audio intermediate frequency amplification circuit, 412 is an audio FM detection circuit, 413 is an audio signal output terminal, 414 is a band pass filter, 415.
Is a synchronous detection circuit, 416 is a carrier recovery circuit, 417 is a spectrum suppression processing signal reproduction circuit, 418 is a digital signal processing circuit, 419 is a digital-analog conversion circuit (hereinafter abbreviated as DAC), and 420 is digitally encoded and transmitted. It is an output terminal for a reproduced audio signal.

A television signal input from the antenna 401 is amplified by a high frequency amplifier circuit 402 and then converted into an intermediate frequency for demodulation by a frequency conversion circuit 403. The converted signal passes through the IF Nyquist filter 404 and is amplified by the intermediate frequency amplifier circuit 405. The channel selection is performed by changing the oscillation frequency of the local oscillator inside the frequency conversion circuit 403. The video signal from the signal amplified by the intermediate frequency amplifier circuit 405 is converted into a video signal detection circuit 406.
Is detected by. The obtained video signal is the video signal amplification circuit 4
The signal is amplified by 07 and a color difference signal is obtained from the signal by the color difference signal demodulation circuit 408. The signal and video signal amplification circuit 407
Output signal is converted into RGB primary color signals by the primary color signal demodulation circuit 409. The three primary color signals are applied to the cathode ray tube 410, and an image is obtained on the cathode ray tube. The voice band signal of the output signal of the intermediate frequency amplifier circuit 405 is amplified by the voice intermediate frequency amplifier circuit 411. The output signal is voice FM
The detection circuit 412 performs FM detection to produce a voice signal. The audio signal is output from the audio signal output terminal 413. The above description is for the part of the existing terrestrial television broadcasting reproducing apparatus.

The part related to the present invention will be described below.
From the signal converted to the intermediate frequency by the frequency conversion circuit 403, the signal of the digitally encoded voice signal band which is multiplex-transmitted by the band pass filter 414 is extracted. In the synchronous detection circuit 415, the extracted signal is synchronously detected by the carrier wave reproduced by the carrier wave reproduction circuit 416, and the orthogonally multiplexed signal is detected and demodulated. The demodulated signal is processed by the spectrum suppression processing signal reproduction circuit and is returned to the signal before the spectrum suppression processing. After that, the digital signal processing circuit 418 detects and corrects an error generated during the transmission of the signal. The corrected digital signal is converted back into an analog signal by the DAC 419. The analog signal is output from the output terminal 420.

As described above, according to this embodiment, since the reproducing apparatus corresponding to the transmission signal generating apparatus is provided, it is possible to demodulate the multiplexed signal.

FIG. 5 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention.
The same reference numerals as those in FIG. 1 indicate the same functions. Reference numeral 114 is a spectrum suppression processing circuit, 501 is a delay circuit, and 502 is a subtraction circuit. A specific example of the spectrum suppression processing circuit 114 is shown by a delay circuit 501 and a subtraction circuit 502.

FIG. 6 is a waveform diagram for explaining the operation of FIG.
601 is general binary digital data input, 6
02 is the output waveform of the delay circuit 501, and 603 is the subtraction circuit 5.
02 is an output waveform. Here, the delay time τ is the same as or shorter than the minimum inversion interval time T of digital data.

Τ ≦ T (5) When τ is equal to T, the required transmission band does not increase. FIG. 6 shows an example. The input binary digital data 601 is output to the output waveform 603 by subtracting the output waveform 602 which is the output of the delay circuit 501 in the subtraction circuit 502. Waveform 603 is digital data 601
Time T _1, T _5, and the subsequent rising edge, such as T ₆ only becomes time tau High, a time T _3, T _6, etc. Low and will continue for a time tau, the other time T _2, from time T ₃ or time The period from T ₄ to time T ₅ is the midpoint. The low frequency component of the waveform 603 is smaller than that of the data 601. The carrier wave which is the output of the phase shift circuit 116 is modulated by the modulation circuit 117 so that it becomes non-modulated at + A at the high of the waveform 603 and at the midpoint of the -A at the low of the waveform 603. The component near the carrier frequency of is suppressed.

According to the present embodiment, the components near the carrier frequency of the multiplex-transmitted signal are suppressed, so that there is an effect that the interference with the current terrestrial television broadcasting given by the multiplex-transmitted signal can be reduced.

FIG. 7 is a block diagram showing another example of a reproducing apparatus for a television transmission signal according to the present invention. Figure 4
The same reference numerals as those in FIG. Reference numeral 417 is a spectrum suppression processing signal reproducing circuit, 701 is a ternary discriminating circuit, 70
2 is a three-value binary conversion circuit, 703 is a code identification circuit, 704
Is a clock recovery circuit. Since the transmitted signal is ternary as shown by the waveform 603 in FIG. 6,
The reproducing device is provided with a three-value binary conversion circuit and the like. The waveform detected by the synchronous detection circuit 415 is the ternary discrimination circuit 70.
The value of 1 is converted into ternary digital data of +1, 0, -1, and the ternary binary conversion circuit 702 converts it into binary digital data. The digital signal is a code identification circuit 70.
3, the clock reproduced by the clock reproduction circuit 704 is strobed.

FIG. 8 shows a specific example of the ternary discrimination circuit 701 shown in FIG. FIG. 9 is a waveform diagram for explaining the operation of FIGS. 7 and 8. 701 is a three-value discrimination circuit, 702 is a three-value binary conversion circuit, 801 is an input terminal, 802 is a capacitor, and 8
Reference numeral 03 is a resistor, 804 is an amplifier, 805 and 806 are voltage comparison circuits, 807 and 808 are reference voltage sources, 809 is a set / reset circuit, and 810 is a binary digital data output terminal. 901 is a waveform, 902 and 903 are reference power supply voltage values, 904 is an output waveform of the voltage comparison circuit 805, and 905.
Is an output waveform of the voltage comparison circuit 806, 906 is an output waveform of the set / reset circuit 809, 907 is a clock timing pulse obtained by the clock recovery circuit 704, and 908.
Is the output waveform of the code identification circuit 703.

The output signal of the synchronous detection circuit 415 is applied to the input terminal 801. The DC voltage of the signal is cut off by the capacitor 802 and the resistor 803, and is amplified by the amplifier 804 to become a signal as shown by a waveform 901. The signal of this waveform 901 is the voltage comparison circuit 805 and the voltage comparison circuit 8.
06, the voltage is compared with the reference voltage V ₁ or the reference voltage V _2, and the signal indicated by the waveform 904 is converted into the voltage comparison circuit 80.
5 and the signal shown in waveform 905 is obtained at the output of the voltage comparison circuit 806. These signals are applied to the set / reset circuit 809, and the signal represented by the waveform 906 is obtained at the output of the set / reset circuit 809. This signal is obtained at the output terminal 810 as the output signal of the ternary binary conversion circuit 702. At time T ₁ , the waveform 901 exceeds the reference voltage V ₁ , the waveform 904 becomes High, the set / reset circuit 809 is set, and the waveform 906 becomes High.
becomes h. At time T ₃ , the waveform 901 becomes lower than the reference power source V ₁ and the waveform 904 becomes Low, but the waveforms 905 and 90
6 remains the same. At time T ₄ , the waveform 901 becomes lower than the reference voltage V ₂ and the waveform 905 becomes High, so that the set / reset circuit 809 is reset and the waveform 906 becomes Low. At time T ₆ , the waveform 901 exceeds the reference voltage V ₂ and the waveform 905 becomes Low, but the waveform 904
Remains low, waveform 906 continues low.
At time T ₇ , the waveform 901 exceeds the reference voltage V ₁ and the waveform 904
Becomes high, and the waveform 906 becomes high. By the above operation, the binary digital data as shown by the waveform 906 can be demodulated. The binary digital data indicated by the waveform 901 is generated by the code discriminating circuit 703 and reproduced by the clock reproducing circuit 704.
7 is strobed to form binary digital data of waveform 908. The waveform 906 is strobed at the time indicated by the arrow of the clock timing pulse 907 indicated by the times T ₂ , T ₄ , T _{8 and the} like to become the waveform 908. Strobing the data at the time indicated by this arrow means that the waveform 901 is the reference voltage V ₁
Is the time most distant from the reference voltage V ₂ and is the time at which the probability of error occurrence due to noise or the like is least.

As described above, according to this embodiment shown in FIGS. 7 to 9, the transmission signal shown in FIGS. 5 and 6 can be reproduced.

FIG. 10 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 1 indicate the same functions. Reference numeral 114 is a spectrum suppression processing circuit, and 1001 is a digital modulation circuit. The output signal of the digital signal processing circuit 113 is subjected to digital modulation such as a digital FM modulation method by the digital modulation circuit 1001 and the low frequency component of the signal is suppressed. As a result, according to this embodiment, as in the case of FIG. 1, there is an effect that interference with the current terrestrial television broadcasting can be reduced. A detailed description of a digital modulation method such as a digital FM modulation method in which low frequency components are suppressed is described in "Modulation method of digital magnetic recording" of Nikkei Electronics P126 to P164, No. 1978, 12, 11. .

FIG. 11 is another example of a block diagram of a transmission signal reproducing apparatus for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 4 indicate the same functions. Reference numeral 417 is a spectrum suppression processing signal reproduction circuit, 1101 is a code identification circuit, 1102 is a clock reproduction circuit, and 1103 is a digital demodulation circuit. The output signal of the synchronous detection circuit 415 is supplied from the code identification circuit 1101 to the clock recovery circuit 11
The clock signal regenerated in 02 is strobed into a digital signal at a timing at which an error occurrence probability caused by noise or the like is low. This digital signal is returned to digital data before digital modulation processed by the transmitting side in the digital demodulation circuit 1103. By the operation of the spectrum suppression processing signal reproducing circuit 417, the signal which has been subjected to the spectrum suppression processing and transmitted can be reproduced.

According to this embodiment, there is an effect that the transmission signal shown in FIG. 10 can be reproduced.

FIG. 12 shows the spectrum suppression processing circuit 1 of FIG.
14 is a block diagram of another specific example of 14. FIG. Reference numeral 114 is a spectrum suppression processing circuit, 1201 is an input terminal, 1202 is a time axis compression circuit, 1203 is a timing generation circuit, and 120.
Reference numeral 4 is an inverter, 1205 is a delay circuit, 1206 is a changeover switch, and 1207 is an output terminal.

As a concrete example of the spectrum suppression processing circuit,
The continuous data applied to the input terminal 1201 is converted into intermittent data by the time axis compression circuit 1202, and the inverter 120
4, the delay circuit 1205 and the change-over switch 1206 can obtain the same data inverted during the intermittent data. As a result, the same data is inverted and transmitted at fixed intervals, so low-frequency components near the DC of the averaged signal and components near the frequency indicated by the reciprocal of the fixed-interval period are suppressed. To be done.

FIG. 13 shows an example of a transmission data string for explaining the operation of FIG. 1301 is a data string of the input terminal 1201, 1302 is an output data string of the time axis compression circuit 1202, 1303 is an inverter 1204 and a delay circuit 1205.
The output data sequence of the delay circuit 1205 that has passed through 1304 is the output data sequence of the changeover switch 1206, and 1305 is the timing waveform.

Data string 1 applied to input terminal 1201
301 is time-axis compressed into a data string 1302 according to the timing of the time-axis compression circuit 1202 and the timing generation circuit 1203. Data string 1 from time T ₁ to time T ₅
301 is 1 / of the data sequence 1302 from time T ₃ to time T _5.
Compressed in 2 periods.

This intermittent data 1302 is the inverter 12
04 and delay circuit 1205 inverts the data and delays it by time τ to form a data string 1303.
The data sequence 1302 and the data sequence 1303 are switched by the changeover switch 1206 according to the timing waveform 1305 as at times T ₁ , T ₃ , T ₅ , and T ₆ to form the data sequence 1304. The data sequence 1302 and the data sequence 1303 are intermittent data, and since the data sequences do not coincide with each other in their signal periods, the data sequence 1304 can be obtained even if they are digitally added. Therefore, in this case, the changeover switch 1206 can also be configured by a digital addition circuit (OR circuit). FIG. 14 is a simulation diagram showing a transmission pattern.
This figure shows the appearance of the multiplex-transmitted signals on the television screen when the timing waveform 1305 of FIG. 13 is synchronized with the horizontal synchronizing signal of the television signal. The data from a ₁ to a _{5 in} the first horizontal scanning period
The data from a ₁ bar to a ₅ bar in the second horizontal scanning period, the data from b ₁ to b ₅ in the third horizontal scanning period,
In the fourth horizontal scanning period, data from b ₁ bar to b ₅ bar is transmitted. As a result, the data in the two horizontal scanning periods, such as the first and second horizontal scanning periods and the third and fourth horizontal periods, have the same opposite phase.

Here, we consider the video color subcarrier of current television broadcasting. FIG. 15 is a vector diagram showing a change of a video carrier due to a color subcarrier. Figure 15
FIG. 15A is a video carrier vector diagram when there is no orthogonal component multiplexing, and FIG. 15B is a video carrier vector diagram when there is orthogonal component multiplexing. ωs is the angular frequency of the color subcarrier, and the phase changes from 1 to m, n, o ..., s.
Further, ωs ′ is the angular frequency of the color subcarrier in the adjacent horizontal scanning period, and the phase is π deviated from ωs by l ′, m ′, n.
The phase changes as ', ... s'. A and -A indicate multiplexing on orthogonal components. In current television broadcasting, the phases of the color subcarriers in adjacent horizontal scanning periods are deviated by π from the relationship between the frequency of the color subcarrier and the horizontal scanning frequency. In the orthogonal multiplexing, a vector diagram in the case where the signal A is multiplexed in a certain horizontal scanning period having a subcarrier having a phase of ωs and the signal −A is multiplexed in the horizontal scanning period of the adjacent ωs ′ is shown in FIG. Corresponds to b). As shown in FIG. 4, the orthogonal multiple components cause a phase variation of the video carrier, and when the detection method of the television video signal is the envelope detection method or the pseudo-coherent detection method, the video detection output signal includes the video carrier. A signal corresponding to the envelope of is obtained. In that case, the phase of the color sub-carrier included in the video detection output signal does not have orthogonal multiplexing.
In the case shown in FIG. 5 (a), it is l or p ', but in the case shown in FIG. 15 (b) where there is orthogonal multiplexing, φ ₁ or φ ₂
Is. The phase difference between them is φ. The phase change of the color sub-carrier changes the hue on the reproduced video screen.

When the video detection method is synchronous detection, only the directional component of cosωct shown in the figure is detected.
Even if A is multiplexed, the maximum amplitude phase is 1 or p ', and the phase is not changed. In the case of envelope detection, signals to be multiplexed are A and -A according to the code to be multiplexed.
When the phase is inverted as shown in, the phase directions of the color subcarriers become opposite. The signal multiplexed in the adjacent horizontal scanning period is A
When the phases of the color subcarriers are opposite to each other as shown in FIG. 15A, the color subcarriers have the relationship of ωs and ωs ′ shown in FIG. 15B, the phase change amounts of the color subcarriers are equal, and the phases are opposite phases. The up and down hue changes on the screen of the television in the adjacent horizontal scanning periods become opposite phases. A person who sees such a screen is unlikely to feel a hue change due to the low chromaticity-sensitivity frequency characteristic of the visual sense or the integration effect of the eyes.

That is, a ₁ to a ₅ and a shown in FIG.
_{Between the 1} bar to a ₅ bar or the b ₁ bar to b ₅ bar and the horizontal scanning period such as b ₁ to b ₅ , it is difficult for a person to perceive a hue change. However, there is no effect of making it difficult to perceive the hue change in the areas where the data of the same reverse phase such as a ₁ bar to a ₅ bar and b ₁ to b ₅ are not provided.

Further, when data of the same opposite phase is transmitted in adjacent horizontal scanning periods, a "comb filter" having a correlation (so-called line correlation) in the horizontal scanning periods is used for separating the luminance signal and the color difference signal. In some television receivers, the phase change of the color sub-carrier wave is canceled by the circuit, and the hue change does not occur.

FIG. 16 is a block diagram of a comb filter for extracting a color difference signal for general luminance signal color difference signal separation. 1
Reference numeral 601 is an input terminal, 1602 is a delay circuit, 1603 is a subtraction circuit, and 1604 is an output terminal. Input terminal 1601
Is added to the color difference signal delayed by the time of one horizontal scanning period in the delay circuit 1602 and the subtraction circuit 1603, and the result is obtained at the output terminal 1604.

FIG. 17 is a waveform diagram for explaining the operation. 17
01 is a waveform when not multiplexed, 1702 is a waveform when the color sub-carrier of ωs is subjected to A multiplexing, and is on the right side of FIG. 15, 17
03 indicates that the color subcarrier of ωs ′ has undergone the multiplexing of −A.
A waveform on the left side, 1704 is an inverted waveform of the waveform 1703. The color subcarrier waveform 1701 when there is no multiplexing has the maximum amplitude at time l. The waveform 1702 of the color subcarrier when the multiplexed signal A is orthogonally multiplexed receives the phase change of φ ₁ and has the maximum amplitude between the time s and the time l. In addition, the waveform 1703 of the color subcarrier in the case of orthogonal multiplexing of −A in the adjacent horizontal scanning period receives the phase change of φ ₂ , and the time p ′
And the time q ′, the amplitude becomes maximum. Delay circuit 1602
The waveform 1702 delayed by one horizontal scanning period via the subtraction circuit 1603 is subtracted by the waveform 1703.
The inverted waveform 1704 of 703 is to be added to the waveform 1702, and when the waveform after addition is halved in amplitude, it becomes the same as the waveform 1701. That is, the color subcarrier obtained by the comb filter does not undergo phase change due to the orthogonally multiplexed signals even if the video signal detection circuit is an envelope detection circuit. Note that in this case as well, FIG.
As shown in FIG. 14, since there is no phase change only when the upper and lower data are in the same opposite phase in the adjacent horizontal scanning periods like a ₁ to a ₅ and a ₁ bar to a ₅ bar shown in FIG. Does not have a phase change for each horizontal scanning period.

As described above, according to the present embodiment in which the spectrum suppression processing circuit 114 of FIG. 1 is composed of the circuit shown in FIG. 12, according to the hue change of the video signal due to the multiplex-transmitted signal. It also has the effect of reducing interference.

Note that the time axis compression circuit 1202 is used because the transmission data is assumed to be continuous data in FIG. 12, but is unnecessary when the transmission data is intermittent discontinuous data.

FIG. 18 is a block diagram showing another example of a reproducing apparatus for a television transmission signal embodying the present invention. The same reference numerals as those in FIG. 4 indicate the same functions. Reference numeral 417 is a spectrum suppression processing signal reproduction circuit, 1801 is a code identification circuit,
1802 is a clock regeneration circuit, 1803 is a switching circuit, 1
Reference numeral 804 is a time axis expansion circuit, and 1805 is a timing reproduction circuit. The waveform detected by the synchronous detection circuit 415 is converted into a digital code in the code identification circuit 1801 by the clock timing pulse reproduced by the clock reproduction circuit 1802.

Data of a necessary period of the signal returned to the digital code is selected and taken out by the switching circuit 1803 and the timing back circuit 1804. After that, the time axis expansion circuit 1804 restores the original transmission data.

FIG. 19 shows an example of a data string for explaining the operation of FIG. 1901 is a timing waveform for synchronizing the horizontal scanning period, 1902 is a data string of the transmitted and received signals shown in FIGS. 12, 13 and 14, 1903 is a timing waveform obtained from the timing waveform 1901, 190
Reference numeral 4 is an output data string of the switching circuit 1803, and 1905 is an output data string of the time axis expansion circuit 1804. The transmitted and received data string 1 which is the output of the code identification circuit 1801.
902 is a switching circuit 180 according to a timing waveform 1903 obtained from the timing waveform 1901 of the horizontal synchronizing signal.
3 is repeated from time T ₁ to time T ₂ from conduction T ₂ to time T ₃ like a cutoff to form a data string 1904.
By the time base expansion circuit 1804, the data string 1904 intermittent data from time T ₁ until T ₂ is extended, the time T ₁
The data is from T ₃ to T ₃ , and this operation is repeated to form a data string 1905. As a result, the data string before the spectrum suppression processing shown in the data string 1301 of FIG. 13 was reproduced.

As described above, according to this embodiment, there is an effect that the transmission signals shown in FIGS. 12, 13 and 14 can be reproduced.

FIG. 20 is a block diagram of still another example of a reproducing apparatus for a television transmission signal according to the present invention.
The same reference numerals as those in FIGS. 4 and 18 indicate the same functions. 41
Reference numeral 7 is a spectrum suppression processing signal reproducing circuit, 2001 is a subtracting circuit, and 2002 is a delay circuit. As shown in FIG. 12, FIG. 13, and FIG. 14, since the same data is multiplexed and transmitted in the opposite phase in the two horizontal scanning periods, the output signal of the synchronous detection circuit 415 is delayed by the delay circuit 2002 for one horizontal scanning period. When this signal is subtracted from the subtraction circuit 2001, doubled data amplitude is obtained. White noise added in the transmission system is √2
It only doubles. The interfering signal from the video signal due to the ghost of the video signal is canceled and removed because the video signal has a correlation for each horizontal period. The interference signal is removed from the video signal in the following process. Data X is sent at a certain timing in a certain horizontal scanning period, and X is transmitted at a timing after one horizontal scanning period time has elapsed.
Assuming that X-bar data, which is the inverted data of the above, is sent, the received signal X-bar and the data X delayed by one horizontal scanning period in the delay circuit 2002 are added to the subtraction circuit 2001 and subtracted at the same timing. Therefore, the output of the subtraction circuit 2001 is X− (X bar) = 2X (6), and a double signal is obtained. Next, assume that the disturbance G from the video signal is added during transmission. Since the video signal has a high correlation for each horizontal scanning period (images such as vertical stripes are particularly strong), the interference from the video signal is the interference G at the data X timing and the data X timing. The output of the subtraction circuit 2001 becomes (X + G)-(X bar + G) = 2X (7), and the disturbance G is canceled. However, the offset effect is small at a portion where the correlation of the video signal for each horizontal scanning period is small.

FIG. 21 is an example of a data string for explaining the operation of FIG. 2101 is a timing waveform for synchronization in the horizontal scanning period, 2102 is a data sequence after the transmission signals shown in FIGS. 12, 13 and 14 are received, 2103 is an output data sequence of the delay circuit 2002, and 2104 is subtraction. Circuit 200
1 output data string, 2105 is a timing waveform 2101
2106 is a timing waveform obtained from
An output data sequence 03, 2107 is a time axis expansion circuit 180
4 is an output data string.

The transmitted and received data string 2102 output from the synchronous detection circuit 415 is delayed by the delay circuit 2002 for one horizontal scanning period to become the data string 2103.
The data at time T ₀ is delayed at time T ₁ . The delayed data string 2103 is subtracted from the data string 2102 by the subtraction circuit 2001 to become a data string 2104. In accordance with the timing waveform 2105 obtained from the timing waveform 2101 of the horizontal synchronizing signal, the changeover switch 1803 conducts from time T ₁ to T ₂ and is cut off from T ₂ to T ₃ , and the data sequence 2104 becomes a data sequence. 21
It will be 06. Here, the data sequence 2104 and the data sequence 210
In the data _No. 6, double description such as data 2a ₁ and a ₁ is omitted.

The data from the time T ₁ to T ₂ is expanded in the data string 2106 by the time axis expansion circuit 1804,
The data from time T ₁ to T ₃ is obtained, and this operation is repeated to obtain data 2107. The results are shown in Figure 1.
The data string before the spectrum suppression process shown in the data string 1301 of No. 3 was reproduced.

According to this embodiment, FIG. 12, FIG. 13, and FIG.
The transmission signal shown in FIG. 4 can be reproduced, and further, the interference from the video signal can be reduced. FIG. 22 is a graph showing the effect. The horizontal axis is the disturbance level due to ghosts (the right is less). The vertical axis is the bit error rate representing the error rate associated with transmission. White noise was added during transmission during the experiment to generate an error. In that case, the carrier-to-noise ratio of the signals multiplexed and transmitted orthogonally is shown by CIN. 2201 is a curve connecting measured values when the subtraction circuit 2001 is not used,
2202 is a curve connecting measured values when the subtraction circuit 2001 is used. The ghost level that gives the same bit error rate of 10 to ⁴ is about 15 dB to the left on the curve 2202. The fact that the ghost has the same error rate when the ghost is about 15 dB more shows that the interference from the video signal due to the ghost and the like can be reduced. Further, when the ghost level is as small as −50 dB, the bit error rate of C / N = 15 dB of the curve 2202 and the curve 2201.
C / N = 18 dB has a similar bit error rate, it means that the signal amplitude by the subtraction circuit is double and the noise is √2.
By doubling the value, it shows that the C / N ratio is improved by 3 dB.

FIG. 23 shows the spectrum suppression processing circuit 1 of FIG.
14 is a block diagram showing still another example of 14. FIG. Reference numeral 114 is a spectrum suppression processing circuit, 2301 is an input terminal, and 230
2 is a timing generation circuit, 2303 is an inverter, 23
Reference numeral 04 is a delay circuit, 2305 is a changeover switch, and 2306 is an output terminal.

As a concrete example of the spectrum suppression processing circuit,
The data added to the input terminal 2301 is the inverter 23.
03, and the delay circuit 2304 performs the inversion delay. Depending on the signal generated by the timing generation circuit 2302, the input data and the inverted and delayed data are changed over to the changeover switch 230.
5 to obtain the output terminal 2306.

FIG. 24 shows an example of a transmission data string for explaining the operation of FIG. 2401, 2404, and 2406 are timing waveforms, 2402 is an input data string applied to the input terminal 2301, 2403 is an output data string of the delay circuit 2304, and 2405 is an output data string.

Data string 2 applied to input terminal 2301
The data 402 is inverted by the inverter 2303 and the delay circuit 2304 and delayed by the time τ to become the data string 2403.
The timing waveform 2401 is inverted every time τ. (Times T ₁ , T ₂ , T ₃ ) Timing waveform 2404
Is inverted in the middle of the data string. The changeover switch 2305 is connected to the (a) side in the state of (a) indicated by High in the drawing (in the period from time T ₂ to time T ₄ ). L
In the state of (b) indicated by ow (from time T ₄ to time T ₅
The changeover switch 2305 is connected to the (b) side during the period (2). The output data string 2405 of the changeover switch 2305 is
It is available at the output terminal 2306.

FIG. 25 is a simulation diagram showing a transmission pattern. This figure shows the appearance of the multiplex-transmitted signal on the television screen when the timing waveform 2406 of FIG. 24 is synchronized with the horizontal synchronizing signal of the television signal. The horizontal scanning direction is shown horizontally and the vertical scanning direction is shown vertically. As indicated by the circled frame in FIG. 25, the inverted data is the upper and lower data for each data in the adjacent horizontal scanning period. Since the data is inverted in the adjacent horizontal scanning period, the multiplexed signal to the quadrature component of the video carrier has an opposite phase, and the effect of reducing the interference to the hue change of the video signal due to the multiplexed signal is reduced. is there.

As shown in the above description, according to the present embodiment in which the spectrum suppression processing circuit 114 of FIG. 1 is composed of the circuit shown in FIG. 23, in the horizontal scanning period in which the signals to be multiplexed are adjacent to each other. Since the phases are opposite to each other, there is an effect that interference caused by a hue change of an image is reduced. Further, in all the horizontal scanning periods, as shown by the circled frame in FIG. 25, the phase of each data is opposite to that of the upper and lower adjacent scanning periods in every mesh, so that it affects the hue change of the image. The interference is fine, and the effect is that the interference is reduced by the low sensitivity frequency of the visual chromaticity.

The above transmission signal can be reproduced by the configuration shown in FIG. Since some operation timings are different, a data string for explaining the operation is shown in FIG. 2601 is a timing waveform for synchronization in the horizontal scanning period, 2602 is a data string after the transmission signals shown in FIGS. 23, 24, and 25 are received, 2603 is an output data string of the delay circuit 2002, and 2604 is Output data string of subtraction circuit 2001, 26
05 is a timing waveform, 2606 is a changeover switch 180
3 output data sequence 2607 is a time axis expansion circuit 1804
Is an output data string of. The transmitted and received data string 2602 which is the output of the synchronous detection circuit 415 is the delay circuit 20.
The data string 260 is delayed by one horizontal scanning period by 02.
It will be 3. Data at time T ₁ is at time T ₂ or at time T
The data of ₂ is delayed at time T ₃ . The delayed data string 2603 is converted into the data string 26 by the subtraction circuit 2001.
02 is subtracted to form a data string 2104. In the data string 2604, the changeover switch 1803 is cut off from the time T ₂ to T ₄ according to the timing waveform 2605, and the data string 2604 is turned on from the time T ₄ to the T ₅ to repeat the data string 26.
04 becomes a data string 2606. Here, the data string 260
4 and the data string 2606, double description is omitted, such as data 2f ₁ and f ₁ . Time axis expansion circuit 1804
Therefore, the data from time T ₄ to time T ₅ is changed to time T ₄
Data string 2606 as extended data up to time T ₆ from is extended two hours axis data column 2607. As a result, the data string before spectrum suppression processing shown in the data string 2402 of FIG. 24 was reproduced.

In FIG. 14 and FIG. 25 described above, the signals multiplexed and transmitted on the screen of the television are drawn as simulated. In these cases, it is explained that the signals to be multiplexed and transmitted have a fixed number of data in the horizontal scanning period and are synchronized signals, but even if they are not perfectly synchronized, if they are substantially the same, the same effect can be obtained. Is obtained. In addition, the transmission speed of the signal to be multiplexed does not match the horizontal scanning period of the television because the last data period of the horizontal scanning period is arbitrarily set or the number of data in a certain pair of horizontal scanning periods is increased or decreased. Even in the case, signal transmission is possible.

FIG. 27 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 1 indicate the same functions. 2701
Is a control circuit, 2702 is a control signal generation circuit, and 2703 is a changeover switch. In the present embodiment, the above-mentioned increase / decrease in the number of data in the pair of horizontal scanning periods, a signal indicating the increasing / decreasing horizontal scanning period, a signal indicating whether the top data of the horizontal scanning period is the same as the upper or lower horizontal scanning period data, or A signal indicating the positional relationship with the vertical scanning period is added to the signal to be multiplexed. The output signal of the digital signal processing circuit 113 and the output signal of the control signal generation circuit 2702 are time-division multiplexed by the changeover switch 2703 controlled by the control circuit 2702. Since the output of the digital signal processing circuit 113 is not transmitted only during the period in which signals such as control signals are multiplexed, the data is time-axis compressed in the digital signal processing circuit 113.

FIG. 28 shows an example of a signal multiplexed by time division in FIG. 2801 is a vertical synchronizing signal, 2802
Is a horizontal synchronization signal, 2803 is an example of a multiplexed signal, 280
4 is an example of a control signal. Horizontal sync signal 2 for television
During the two horizontal scanning periods subsequent to the vertical synchronizing equalization pulse period of 802, the control signals indicated by C and C bar change from time T ₁ to time T _2.
Is transmitted in the positive phase during the horizontal scanning period of, and is transmitted in the reverse phase during the next horizontal scanning period from time T ₂ to T ₃ . The control signal is a control signal example 280.
As shown by 4, it is composed of a 16-bit synchronizing signal, a 32-bit control signal, 48-bit data number information, or other information. 2 in 1 vertical scanning period
The horizontal scanning period is used for transmitting this control signal.
Since 4 horizontal scanning periods are used for transmission of this control signal within 2 vertical scanning periods which are 525 horizontal scanning periods of the current NTSC television, the digital signal processing circuit 1
The time base compression rate performed in 13 is 525/521 or more.

According to the present embodiment shown in FIGS. 27 and 28, the polarities of signals multiplexed and transmitted orthogonally to the adjacent horizontal scanning periods in the horizontal scanning period and the transmission of multiplexed signals in the horizontal scanning period. Since the control signals such as the increase / decrease in the number of data and the increased / decreased horizontal scanning period number are transmitted in a time division manner, there is an effect that the receiver for receiving and reproducing the signals multiplexed and transmitted orthogonally stably operates.

FIG. 29 is a transmission pattern diagram schematically showing an example of interleave processing of the digital signal processing circuit 113 corresponding to the television screen of FIG. The 0th, 1st, and 2nd sampled data of the left channel of the audio signal are denoted by L ₀ , L ₁ , and L ₂ . The 0th, 1st and 2nd sampled data of the right channel are R ₀ ,
It is represented by R ₁ and R ₂ . L ₀ bar, L ₁ bar, L ₂ bar,
R ₀ bar, R ₁ bar, R ₂ bar are L ₀ , L ₁ , L ₂ , R ₀ , R
_This is the inverted data of ₁ and R ₂ . As shown in the figure, the first sampled data L _{1 is characterized in} that the previous L ₀ and the subsequent L ₂ are interleaved so that they are separated by the same horizontal scanning period or more.

According to this embodiment, adjacent sampled data before and after is transmitted without intervening in the same horizontal scanning period by the interleave process, so that adjacent horizontal signals are reproduced in the audio signal reproduction by the receiver shown in FIG. For an image (horizon, etc.) in which the upper and lower phases are small in the scanning period, the subtraction circuit 2001 has a small effect of reducing interference from the image signal to the signal multiplexed and orthogonally transmitted, and an error is likely to occur in certain sampled data. By interpolating the sampled data in which this error has occurred from the sampled data before and after, it is possible to prevent the generation of abnormal sounds. In the above description, one sampled data is indicated by 1 bit, but similarly for N-bit data, the preceding and following sampled data are separated from the same horizontal scanning period to obtain the same effect.

FIG. 30 is a block diagram of still another example of a reproduction apparatus for a television transmission signal according to the present invention.
30 is an example of a receiver that receives and reproduces the transmission signals shown in FIGS. 27, 28, and 29. The same reference numerals as those in FIG. 20 indicate the same functions. 3001 is a control signal reproducing circuit, 300
Reference numeral 2 is an interpolation control circuit.

The control signal reproducing circuit 3001 extracts the control signal from the output signal of the code identifying circuit 1802. According to the control signal, the timing reproduction circuit 1805
The changeover switch 1803 is changed over by the signal from, and a necessary signal among the output signals of the code identification circuit 1802 is obtained. The output signal of the changeover switch 1803 is expanded by the time axis expansion circuit 1804. The decompressed signal is deinterleaved in the digital signal processing circuit 418 to detect and correct an error occurring during transmission, and is returned to the original transmission data. In the error detection and correction, it is not possible to correct errors that are intensively generated in a portion where the correlation of the video signal is small due to an increase in interference from the video signal. The sampling data having the error that cannot be corrected is interpolated from the adjacent sampling data in the digital signal processing circuit 418 controlled by the interpolation control circuit 3002.

According to the present embodiment, since the reproduction by the control signal and the interpolation from the adjacent sampling data are performed, stable reproduction can be performed.

FIG. 31 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 1 indicate the same functions. Reference numeral 114 is a spectrum suppression processing circuit, 3101 is a subtraction circuit, and 310.
2 is a delay circuit.

The signal applied to the input terminal 111 is subtracted by the subtraction circuit 3101 from the signal delayed by the delay time τ in the delay circuit 3102. For example, if the input signal is 20K
If the signal is Hz or less, the signal can be transmitted by setting the delay time τ to about 10 μsec. Since the subtraction circuit 3101 obtains the difference between the input signal and the delayed input signal, the lower the frequency component of the input signal, the smaller the difference. As a result, a signal with few low frequency components is obtained at the output of the spectrum suppression circuit 114. Since the carrier wave is modulated by the modulation circuit 117 by the signal, the component in the vicinity of the carrier wave frequency of the modulated signal is suppressed and transmitted.

According to the present embodiment, the component near the carrier frequency of the multiplex-transmitted signal is suppressed, so that there is an effect that interference with the existing terrestrial television broadcasting given by the multiplex-transmitted signal can be reduced.

FIG. 32 is a block diagram showing another example of a reproducing apparatus for a television transmission signal according to the present invention. The same reference numerals as those in FIG. 4 indicate the same functions. Reference numeral 417 is a spectrum suppression processing signal reproducing circuit, 3201 is an adding circuit, 32
Reference numeral 02 is a delay circuit.

The output signal of the synchronous detection circuit 415 is processed by the spectrum suppression processing signal regenerating circuit 417 and the spectrum suppression processing circuit 11 of the transmission signal generator shown in FIG.
4 input signal, which is obtained at the output terminal 420. In the spectrum suppression processing signal reproduction circuit 417, the addition circuit 320
The reproduction signal which is the output signal of 1 is delayed by the delay time τ in the delay circuit 3202 and then added with the output signal of the synchronous detection circuit 415 in the adding circuit 3201. Synchronous detection circuit 415
In, the signal of the difference from the signal transmitted before the time τ transmitted by the transmission signal generator is output. The signal is added back to the signal before time τ to restore the original signal.

According to this embodiment, the transmission signal shown in FIG. 32 can be reproduced.

FIG. 33 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 1 indicate the same functions. Reference numeral 114 is a spectrum suppression circuit and 3301 is a pre-emphasis circuit.

The signal applied to the input terminal 111 is applied to the pre-emphasis circuit 3301 and the low frequency component of the signal is reduced. Since the carrier wave which is the output of the phase shift circuit 116 is modulated by the signal whose low frequency component is reduced by the modulator circuit 117, the component in the vicinity of the carrier frequency of the modulated signal is suppressed and transmitted.

According to this embodiment, the components near the carrier frequency of the multiplex-transmitted signal are suppressed, so that there is an effect that the interference with the existing terrestrial television broadcasting given by the multiplex-transmitted signal can be reduced.

FIG. 34 is a block diagram showing another example of a reproducing apparatus for a television transmission signal according to the present invention. The same reference numerals as those in FIG. 4 indicate the same functions. Reference numeral 417 is a spectrum suppression processing signal reproducing circuit, and 3401 is a de-emphasis circuit.

The output signal of the synchronous detection circuit 415 is the de-emphasis circuit 340 in the spectrum suppression processing circuit 417.
1, the reverse processing of the pre-emphasis circuit 3301 of the transmission signal generator shown in FIG. 33 is performed to restore the original signal.

According to this embodiment, the transmission signal shown in FIG. 33 can be reproduced.

FIG. 35 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. The same reference numerals as those in FIG. 4 indicate the same functions. Reference numeral 3501 is a bandpass filter, and 3502 is a frequency conversion circuit.

The difference from FIG. 4 is that the bandpass filter 3501.
And the frequency of the intermediate frequency signal of the signal multiplexed and transmitted orthogonally by the frequency conversion circuit 3502 is set lower than the frequency of the intermediate frequency signal of the video signal. The intermediate frequency signal output from the frequency conversion circuit 403 is detected, and the video signal is demodulated. The frequency of this intermediate frequency signal is often 58.75 MHz in the case of a receiver for terrestrial television broadcasting in Japan. The output signal of the frequency conversion circuit 430 is further converted by the frequency conversion circuit 3502 and then detected by the synchronous detection circuit 415. The frequency of the intermediate frequency signal detected by the frequency conversion circuit 3502 to, for example, about 5 MHz is lowered.

According to this embodiment, since the frequency of the signal used in the synchronous detection circuit 415 is low, the carrier recovery circuit 4
A phase error caused by a delay or the like caused by passing the carrier wave regenerated in 16 through the circuit can be reduced, and there is an effect that a signal which is stably multiplexed in quadrature is reproduced.

FIG. 36 is a block diagram showing another example of a reproducing apparatus for a television transmission signal according to the present invention. The same reference numerals as those in FIG. 4 or FIG. 35 indicate the same functions.
Reference numeral 3502 is a frequency conversion circuit, 3601 is a reference signal generation circuit, 3602 is a mixing circuit, 3603 is a voltage-controlled local oscillation circuit, and 3604 is a low-pass filter.
The frequency conversion circuit 3502 is composed of a mixing circuit 3602 and a voltage controlled local oscillation circuit 3603.

Detection is performed by utilizing the fact that the modulation by the multiplex-transmitted signal is orthogonal to the modulation by the video signal. The phase difference between the output signal of the band pass filter 414 and the output signal of the reference signal generation circuit 3601 is the synchronous detection circuit 41.
5 and the low pass filter 3604, and is fed back to the voltage controlled local oscillation circuit 3603. As a result, the carrier of the intermediate frequency signal that is the output of the band pass filter 414 and the output signal of the reference signal generation circuit 3601 are synchronized, and a signal that is orthogonally transmitted to the output of the synchronous detection circuit 415 and is multiplexed is obtained.

According to this embodiment, the reference signal generating circuit 36
Since the negative feedback is performed so that the frequency of the intermediate frequency signal detected to the signal frequency of 01 matches, the deviation of the tuning frequency of the band pass filter 414 due to the frequency drift of the local oscillation signal of the frequency conversion circuit 403 is small, This has the effect of stably demodulating signals that are orthogonally multiplexed and transmitted.

FIG. 37 shows the spectrum suppression processing circuit 1 of FIG.
14 shows still another example.

23 to 25, the case of odd data is shown as an example of 7 data in one horizontal scanning period as a transmission data string.
It shows in FIG. FIG. 38 is a diagram for explaining an operation such as a transmission data string example, and FIG. 39 is a simulation pattern example of the transmission data of the present invention. Reference numeral 114 denotes a spectrum suppression processing circuit, 230
2 is a timing generation circuit, 3701 is a timing input terminal, 3702 is a timing generator, 3703 is exclusive OR (hereinafter abbreviated as EOR) 3801 and 380.
4, 3807 is a timing waveform in the timing generation circuit 2302, 3802 is an input data string of the input terminal 2301, 3803 is an output data string of the delay circuit 2304, and 38.
Reference numeral 05 designates a timing waveform of the output of the timing generation circuit 2302, reference numeral 3806 designates an embodiment of the transmission data sequence of the present invention, 37
03 is EOR. The same reference numerals as those in FIG. 23 denote the same functions.

The difference from FIG. 23 is the timing generation circuit 230.
2 is provided with an exclusive OR 3703, and the timing generation circuit 2 is configured by the timing waveforms 3801 and 3804.
The purpose is to obtain the timing waveform 3805 at the output of 302 and control the changeover switch 2305. EOR3703
Inverts the control timing of the changeover switch 2305 every horizontal scanning period, and a transmission data string 3806 is obtained, which is a pattern of transmission data shown on a television screen in a simulated manner in FIG.

In the above-described embodiment, as in FIGS. 23 to 25, it is possible to reduce the interference caused by the multiplexed signal on the hue change of the image.

Interference can be reduced by adopting a form in which the same multiplex signal is transmitted twice in opposite phase, but on the other hand, if the transmission band of the multiplex signal is fixed, the transmission capacity is reduced to 1/2, so that the multilevel method or duo It is also possible to revive by a partial response method or the like, in which inter-code interference such as binary code is positively used to compress the transmission band. In addition,
Regarding the partial response method, pages 137 to 1 of Ohmsha's version of modern digital communication method issued in September 1981
The details are omitted because it is shown on page 42.

FIG. 40 shows a demodulation operation when the transmission data string 3806 is received. Reference numeral 4001 is a timing waveform for synchronization in the horizontal scanning period, 4002 is a data string transmitted and received, 4003 is a data string output from the delay circuit 2002, 4004 is a data string output from the subtractor 2001, 40
Reference numeral 05 is a timing waveform, 4006 is a timing waveform inverted every horizontal scanning period, and 4007 is a timing waveform 4.
Timing waveform obtained from 005 and timing waveform 4006, 4008 is a data string holding the value of the switch 1803, 4009 is a timing waveform, and 4010 is an output data string of the time axis expansion circuit 1804.

The received data string 4002 is the delay circuit 20.
A data string 4003 is created by 02. Data string 4003
When the data string 4002 is subtracted by the subtractor 2001, a data string 4004 is obtained. Timing waveform 40
05 and the timing waveform 4006 are exclusive ORed (the same operation as EOR3703 in FIG. 37). The switch 1803 is made conductive above the obtained timing waveform 4007, and the value of the conduction period is held during the cutoff period of the switch 1803. A data string 4008 is obtained. It can be composed of digital circuits that are latched above the timing waveform 4007. By latching this data string 4008 at the falling edge of the timing waveform 4009, the data string 4010 is obtained at the output of the time axis expansion circuit 1804. This data string 4010 matches the original data string 3802 on the transmitting side shown in FIG. The data string 4008
In the data string 4010, double display such as 2f ₁ is omitted.

As can be seen from the above description, interference can be reduced by adopting a form in which the same multiplex signal is transmitted twice in antiphase, but on the other hand, if the transmission band of the multiplex signal is constant, the transmission capacity is 1
In order to reduce the number to / 2, it is possible to revive by a multi-valued method or a partial response method in which the inter-code interference such as duobinary code is positively used to compress the transmission band. The partial response method is shown in pages 137 to 142 of Ohmsha's version of modern digital communication method issued in September, 1981, and so its details are omitted.

Further, in FIGS. 14, 25 and 39,
The modulation direction of the multiplex signal was simulated in correspondence with the screen of the television video signal. In these cases, the multiple signals are
Although the description has been given of the synchronized signal in which a fixed number is included in the horizontal scanning period, when the transmission rate of the multiplexed signal and the horizontal scanning period are not synchronized, the horizontal scanning period of the multiplexed signal and the horizontal scanning period of the video signal are almost equal to each other. If they match, the same effect of the low frequency interference is obtained on the video signal. It is also possible to absorb the data by making the last data time of the horizontal scanning period arbitrary or increasing or decreasing the number of data in a certain pair of horizontal scanning periods.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIGS. 20 and 35 denote the same functions. A difference from FIG. 20 is that a bandpass filter 3501 and a frequency conversion circuit 3502 are provided in order to lower the frequency for demodulating a digitally encoded and multiplex-transmitted audio signal below the frequency for video signal demodulation.

According to the present embodiment, the frequency conversion circuit 403
The video signal is demodulated at the intermediate frequency of the output (58.75 MHz is generally used in Japanese terrestrial television), and the intermediate frequency of the output of the frequency conversion circuit 4102 is lower (for example, about 5 MHz). Since the audio signal transmitted after being digitally encoded by the digital signal is demodulated by the carrier wave recovery circuit 41 used in the synchronous detection circuit 415.
The phase error due to the circuit delay time of the carrier wave reproduced in 6 is reduced by lowering the frequency, and there is an effect that the audio signal stably digitally encoded and transmitted can be demodulated.

FIG. 42 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The same reference numerals as those in FIGS. 20, 36 and 41 indicate the same functions. The frequency conversion circuit 3502 in FIG. 41 is composed of a mixing circuit 3602 and a voltage control type local oscillation circuit 3603.

41 is different from FIG. 41 in that it is detected by the synchronous detection circuit 415 in synchronism with the audio signal which is reproduced by the carrier wave reproducing circuit 416 and is orthogonally modulated with the carrier wave video signal and is digitally encoded and transmitted. On the other hand, in FIG. 42, by utilizing the fact that the modulation by the digitally encoded audio signal and the modulation by the video signal are in the orthogonal relationship,
The phase difference between the reference signal generation circuit 3601 and the intermediate frequency signal including the carrier wave is detected by the synchronous detection circuit 415 and the low pass filter 3.
This is because the carrier wave of the intermediate frequency is synchronized with the output of the reference signal generation circuit and the output of the synchronous detection circuit 415 is used as the detection output by being detected by 604 and fed back to the voltage controlled local oscillation circuit 3603.

According to this embodiment, the reference signal generation circuit 36
Since it is a negative feedback loop in which the intermediate frequency for demodulation matches the frequency of 01, the frequency shift of the band pass filter 414 and the demodulation frequency drift due to the frequency drift of the frequency conversion circuit 403 and the like are small, and the embodiment shown in FIG. There is an effect that demodulation can be performed more stably.

FIG. 43 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The same reference numerals as those in FIG. 42 indicate the same functions. 4301 is a sample and hold circuit. The difference from FIG. 42 lies in the sample and hold circuit 430.
1 is in the loop of the carrier recovery circuit, and the timing recovery circuit 1805 controls the sample and hold circuit 4301.

According to this embodiment, when the amplitude of the video signal in the output of the synchronous detection circuit 415 becomes large, such as the horizontal sync pulse feedback of the video signal, the period is held and the system is stabilized.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIG. 42 indicate the same functions. Reference numeral 4401 is a synchronization signal detection circuit and 4402 is a delay circuit. The difference from FIG. 42 is that the synchronization signal detection circuit 440
By detecting sync signals such as vertical sync and horizontal sync of the video signal by 1 and sending them to the timing reproduction circuit 1805, generation of control timing waveforms of the switch 1803 and the time axis expansion circuit 1804 is facilitated. Delay circuit 440
2 is for absorbing the delay time difference between the video detection circuit 406 and the code identification circuit 1801.

According to this embodiment, the timing reproduction circuit 1
This has the effect of facilitating the configuration of 805.

FIG. 45 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The same reference numerals as those in FIG. 44 indicate the same functions. Reference numeral 4501 is a synchronous detection circuit, and 4502 is a phase shift circuit. The difference from FIG. 44 is the bandpass filter 44.
Is to use the synchronous detection circuit 4501 for the output of the above, and to use the video signal for which the output signal of the reference signal generation circuit 3601 is subjected to the synchronous detection with the signal which is phase-shifted by π / 2 in the phase shift circuit 4502. As a result, the synchronizing signal detection circuit 440 for the input signal
The time up to 1 and the time up to the synchronous detection circuit 415 substantially coincide with each other, and the time of the delay circuit 4402 can be reduced.

According to the present embodiment, the time of the delay circuit 4402 can be shortened, so that the effect of more stable reception is increased.

FIG. 46 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The same reference numerals as those in FIG. 42 indicate the same functions. Reference numeral 4601 is a control signal reproducing circuit. Figure 4
2 is different from the received control signal in the code identification circuit 18
The reproduction is performed from the output of 01 and the switch 1803 and the time axis expansion circuit are controlled via the timing reproduction circuit 1805 according to the control signal.

According to this embodiment, since the timing is generated by the control signal, there is an effect that more stable reception and reproduction can be performed.

FIG. 47 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. The same reference numerals as those in FIG. 1 indicate the same functions. 11
Reference numeral 4 is a spectrum suppression processing circuit, and 4701 is a ternary conversion circuit.

The digital code output from the digital signal processing circuit 113 is converted by a ternary conversion circuit 4701 from a binary digital signal of +1,0 into a ternary digital signal of + 1,0, -1 and converted into a ternary digital signal. Unnecessary high-pass components are deleted via the low-pass filter 115 suitable for the spectral band. A video signal carrier phase-shifted by 90 degrees is modulated by the digitally encoded voice. In order to prevent the effect of the characteristics of the IF Nyquist filter on the orthogonality of the receiver, the signal is passed through the equalizer 118 having the inverse characteristics of the IF Nyquist filter and added by the adder 119 to the carrier wave modulated with the video signal. As a result, the carrier wave for video is
The video signal and the digitally encoded audio signal are modulated in an orthogonal relationship.

FIG. 48 shows an example of the ternary conversion circuit 4701. 4801 is a binary digital data input, 4802 is a delay circuit, 4803 and 4804 are inverters, 480
5, 4806 is an AND circuit, 4807 is an inverter, 4
Reference numeral 808 is an adder circuit, and 4809 is ternary digital data output. The operation of FIG. 48 will be described with reference to the timing chart of FIG. FIG. 49A shows a binary digital data waveform, FIG. 49B shows a delay circuit 4802 output, and FIG. 49C shows AN.
D circuit 4805 output, (d) inverter 4807 output, (e) ternary digital data waveform (adder 480
8 outputs). The binary digital data shown in (a) is delayed by the time τ by the delay circuit 4802 (b).
The timing waveform shown in the figure is obtained. Here, the time τ is shorter than one data length T. 2 with AND circuit 4805
The inversion of the value digital data (a) and the delay circuit output (b) is ANDed to detect the rising edge of the binary digital data (a) as shown in FIG. Similarly A
The ND circuit 4806 inverts the binary digital data and the AND of the delay circuit output (b) to detect the falling edge of the binary digital data (a), which is inverted by the invert 4807 to invert the waveform in the diagram (d). To get The addition circuit 4808 adds the waveforms shown in FIG. 7C and the waveform shown in FIG. 4D to obtain ternary digital data shown in FIG.
Comparing Figures (a) and (e), the three-valued digital data is highlighted at the rising edge of the two-valued digital data.
It can be seen that a pulse of Low (-1) is generated with a pulse width τ at (+1), the falling edge, and is at an intermediate potential (0) between High and Low in other cases. By converting the binary digital data into the ternary digital data in this way, it is possible to suppress the low-frequency component of the baseband digital signal. From this, unnecessary high-frequency components are removed by the low-pass filter 115, and digital coding is performed. A signal in which the spectrum in the vicinity of the carrier frequency is suppressed can be obtained by modulating with the modulation circuit 117 for the audio signal. The ternary conversion circuit of FIG. 48 can suppress the low frequency component without reducing the transmission capacity when the transmission band is considered to be constant, and the circuit configuration of FIGS. There is an effect that the component can be reduced.

FIG. 50 shows another example of the ternary conversion circuit 4701. 48, the same reference numerals indicate the same functions, 5001 is a clock input terminal, 5002 is an inverter, 5003.
5004 is a D-flip-flop. FIG. 50 is FIG.
The delay time .tau. Of the delay circuit 4802 of FIG. Therefore, only the operation different from that of FIG. 48 will be described with reference to the timing chart of FIG. In FIG. 51, (a)
Is a binary digital data waveform, (b) is a clock signal,
(C) D-flip-flop 5003 output, (d) D-flip-flop 5004 output, (e) AND circuit 4805 output, (f) inverter 4807 output,
(G) is a ternary digital data waveform (adding circuit 4808
Output). The binary digital data shown in FIG. 9A is first transferred by the D-flip-flop 5003 to the one data length T.
Is delayed by T / 2, which is half of (see FIG. 51 (C)),
Next, it is further delayed by T / 2 which is a half of one data length by the D-flip-flop 5004, so that the output of the D-flip-flop 5004 is one data length longer than the binary digital data of the binary digital data input 4801. The signal is delayed by a certain T (see FIG. 51 (d)).
The subsequent operation is similar to that of FIG. 48, and the ternary digital data output 4809 outputs ternary digital data having a pulse width of 1 data length T (see FIG. 51 (g)).

According to the ternary conversion circuit of FIG. 50, the fundamental wave component of the ternary digital data to be output is lower than that in the case where the delay time τ of the delay circuit 402 of FIG. 48 is shorter than one data length T. Since the frequency is used, the baseband band is narrowed, and as a result, the transmission bandwidth after modulation is also narrowed.

FIG. 52 shows still another embodiment of the transmission signal generator of the present invention. The same reference numerals as those in FIGS. 1 and 10 denote the same functions, and 5201 is a multi-level modulation circuit. The digital modulation circuit 1001 operates in the same manner as in FIG.
Although the low frequency component can be suppressed by performing digital modulation in the embodiment of No. 0, the transmission band is widened when the transmission capacity is fixed as compared with the embodiment of FIG. Therefore, the multilevel modulation circuit 5201 is used to
The transmission capacity can be restored by compressing the transmission band using the partial response method that positively utilizes intersymbol interference such as duobinary code. The multilevel modulation circuit 5201 is input to the low pass filter 115,
The following is the same operation as in FIG.

An example of a receiver capable of receiving the signal transmitted from the signal transmission device of FIG. 52 is shown in FIG. 53, FIG.
The same reference numerals denote the same functions, and 5301 is a multilevel demodulation circuit.

The receiving circuit of FIG. 53 receives the transmission signal from FIG. 52, and the signal detected and demodulated by the synchronous detection circuit 415 is input to the multilevel demodulation circuit 5301 and demodulates the digitally modulated digital data. The subsequent operation is similar to that of FIG. Since the partial response method is shown in P137 to P142 of "Modern Digital Communication Method" of Ohmsha, Ltd., issued in September, 1981, its details are omitted. According to the embodiment of FIG. 52, there is an effect that the low frequency component can be suppressed without reducing the transmission capacity.

FIG. 54 shows an example of a circuit having the functions of the ternary discrimination circuit 701, the ternary binary conversion circuit 702 and the code discrimination circuit 703. 7 and 8 indicate the same functions, and 5400 is a ternary code identifying circuit having the functions of the ternary identifying circuit 701 and the code identifying circuit 703. 5
Reference numeral 401 is a sample and hold circuit (hereinafter abbreviated as S / H circuit), and 5402 is a clock signal. 54 will be described with reference to FIG. 55.

FIG. 55A shows ternary digital data,
(B) is a clock signal, (c) is S / H circuit 5401 output, (d) is comparator 805 output, (e) is comparator 806 output, (f) is binary digital data (R
S-flip-flop 809 output). The S / H circuit 5401 uses the clock signal 54 as shown in FIG.
Sample at the rising edge of 02 and hold that value until the next sample. The clock signal 5402 reproduced by the clock reproduction circuit 704 is a signal in which one data length T is one cycle, and the rising edge of the clock is located at a point where the code error rate is small (so-called maximum opening of eye pattern). The output of the S / H circuit 5401 is shown in FIG. 55 (c), and the ternary digital data input at the input terminal 801 is identified as a ternary digital code synchronized with the clock signal 5402. Thereafter, the ternary digital code is input to the comparators 805 and 806, and the ternary digital code is identified as +1, 0, -1 and converted into the binary digital code (FIG. 55 (f)) in the same manner as the operation described in FIG. . By using the circuit of FIG. 54, even when noise such as that shown in 5501 of FIG. 55 occurs, the position of the noise is S /
If it deviates from the sampling point of the H circuit 5401, it does not affect the demodulated binary digital code at all, and has an effect of not deteriorating the code error rate characteristic.

FIG. 56 shows another circuit example having the functions of the ternary discrimination circuit 701, the ternary binary conversion circuit 702 and the code discrimination circuit 703. 54, the same reference numerals indicate the same functions, 5601 is an S / H circuit, 5602 is a window comparator, 5603 and 5604 are addition circuits, and 5605.
Is an intermediate level detection signal.

Since the basic operation of FIG. 56 is the same as that of FIG. 54, a portion which operates differently from FIG. 54 will be described with reference to FIG.

In FIG. 57, (a) is ternary digital data, (b) is a clock signal, and (c) is an S / H circuit 54.
01 output, (d) is a window comparator output (intermediate level detection signal 5602). Now, the signal received by the antenna 401 is distorted due to the spatial transmission path and other causes, so that the ternary digital data input to the input terminal 801 is at an intermediate level as shown in FIG. 57 (a). Consider a case where the high level pulse is higher than the low level pulse. At this time, the ternary digital data becomes a signal containing a DC component and the capacitor 80
When the DC is cut by 2 and the operating point is determined by the resistor 803, the signal is such that the intermediate level does not become 0 volt as shown in FIG. This signal is sampled using the S / H circuit 5401 at the rising edge of the clock signal 5402 in FIG. 57 (b), and its value is held until the next sampling point, resulting in the waveform shown in FIG. 57 (c), which is Δ at the intermediate level. The signal has an offset of V. The signal in FIG. 57 (c) is input to the comparators 805 and 806, and also to the window comparator 5602 and S / H circuit 5601. The window comparator 5602 is shown in FIG.
The intermediate level portion is detected from the signal in (c), and the high level is output only during that period as shown in FIG. 57 (d). In addition,
The intermediate level detection signal 5605 is output to the comparators 805 and 8
It is possible to bring from the 06 output. S / H circuit 5
601 receives the output of the window comparator 5602, samples the output of the window comparator 5602 during the period of High, and holds it during the period of Low.

By operating in this way, the S / H circuit 56
01 can extract the intermediate level offset ΔV of the ternary digital data. Here, the reference voltage source 807
Output V ₁ and reference voltage source 808 output V ₂ are 0 V (G
Since it is set with ND) as a reference, if there is an offset of ΔV in the intermediate level of the three-valued digital data, an error is caused accordingly. Therefore, if the error component ΔV is added to the reference voltage source 807 output V ₁ and the reference voltage source 808 output V ₂ using the adder circuits 5603 and 5604, the optimum reference voltage can be given to the comparators 805 and 806. it can. As described above, according to the circuit configuration of FIG. 56, Hig for the intermediate level of ternary digital data is
It is possible to cancel the influence of the imbalance of the pulse heights of h and Low, and it is possible to perform the ternary discrimination of the ternary digital signal by using the optimum reference voltage. The error voltage canceling circuit of FIG. 56 can also be used for the ternary discriminating circuit of FIG.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIG. 7 represent the same functions, and the difference from FIG. 7 is that the input of the clock recovery circuit 704 is obtained from the ternary binary conversion circuit 702. According to this configuration, there is an effect that the clock recovery circuit 704 can be configured by a digital circuit.

FIG. 59 shows another embodiment of the transmission signal reproducing apparatus of the present invention. The same reference numerals as those in FIG. 7 indicate the same functions, 5901 is a video signal AGC circuit, and 5902 is a digital audio system AGC circuit. When the radio wave input through the antenna 401 has strength or weakness, the three-value discrimination circuit 7 is accordingly
The input of 01 also fluctuates, and as a result, the reference voltage source 80 of the comparators 805 and 806 forming the three-value discrimination circuit 701.
In the embodiment of FIG. 7, the problem that the values of the generated voltages V ₁ and V ₂ of 7,808 are not the optimum reference voltages is considered. FIG. 59.
The above measures have been taken by installing an AGC circuit in the digital audio system of the receiver shown in FIG. Also video signal AGC
The circuit 5901 is used in a conventional television receiver, and is added to the drawing for convenience of explanation. The video signal AGC circuit 5901 determines the strength of the input radio wave by using the detected video signal, and controls the gains of the high frequency amplification circuit 402 and the intermediate frequency amplification circuit 405 accordingly. Since the strength of the radio wave of the video signal is proportional to the strength of the digitally encoded audio signal that is modulated orthogonally to the video signal, digital audio system AGC is also applied using the AGC control voltage of the video signal AGC circuit 5901. be able to. The AGC circuit 5902 is a video signal AGC circuit 59.
The AGC control signal of 01 is received, and the gain is controlled so that the input level of the ternary discrimination circuit 701 is kept constant. According to this embodiment, AGC can be applied to the detected digital data with a simple circuit configuration, and the operation band of the AGC circuit 5902 can be set to the base band band.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIG. 59 indicate the same functions, and 6001 is a digital voice system AGC circuit.
FIG. 60 also shows an AGC circuit provided in the digital audio system of the receiver shown in FIG.
The point of using the GC control voltage is similar to that of FIG. 59, but the AGC is used.
It differs from the example of FIG. 59 in that the insertion position of the circuit is between the BPF 414 and the synchronous detection circuit 415. According to the embodiment of FIG. 60, digital audio system AGC can be applied with a simple circuit configuration, and the AGC circuit 6001 is controlled so that the input level of the synchronous detection circuit 415 becomes constant. The optimum operation level of the synchronous detection circuit 415 has an effect that the synchronous detection circuit 415 can always operate in the best state. In addition, BPF414
It is also conceivable to provide an AGC circuit in front of the above, or to apply AGC to both the conventional television receiving circuit and the digital audio circuit only by changing the gain of the high frequency amplifier circuit 402.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. This embodiment also uses a digital voice system AGC.
Regarding the circuit. The same symbols as those in FIG. 59 indicate the same functions, and 6101 is an envelope detection circuit. In FIG. 61, the AGC control signal of the AGC circuit 5902 is generated using the envelope detection circuit 6101. The operation of the envelope detection circuit 6101 will be described with reference to FIGS. 62 and 63. The envelope detection circuit of FIG. 62 is one of the three-value discrimination circuit.
It is made up of parts. The same reference numerals as those in FIG. 54 represent the same functions, 6201 is an S / H circuit, 6202 is an AGC control signal, and 6203 is an AGC control signal output terminal. The operation as the ternary discrimination circuit of FIG. 62 is the same as that of FIG. 54, and the envelope detection operation will be described with reference to FIG. 63. In FIG. 63, (a) is ternary digital data input from the input terminal 801, (b) is clock signal 5402 (c) is S / H circuit 5401 output, (d) is comparator 805 output, and (e) is S. / H circuit 6201
The output is the AGC control signal 6202. Now, assuming that the positive and negative pulse heights of the ternary digital data change according to the strength of the radio wave input to the antenna 401 as shown in FIG. 63A, the ternary digital data is sampled and held by the clock signal 5402. The ternary digital data is also shown in Figure 6.
The pulse height changes as shown in 3 (c). Comparator 8
The section of the pulse High extracted in 05 is shown in FIG.
As shown in (d), the S / H during the High period of this signal
The circuit 6201 performs a hold operation during the sample operation Low period. As a result, the S / H circuit 6201 displays the high level envelope of the ternary digital data as shown in FIG. 63 (e).
This signal can be used as the AGC control signal. Similarly, if the output of the comparator 806 is used as the control signal of the S / H circuit 6201, the Low level envelope of the ternary digital data can be detected.
The value obtained by R is used as the input of the S / H circuit 6201, and S / H
If the output of the H circuit 6201 is full-wave rectified, both the High level and the Low level of the ternary digital data can be used as the AGC control signal.

According to the embodiment shown in FIG. 61, since AGC is applied while observing the output of the digital voice system, it is possible to operate in an optimum state for the digital voice system, and
The operation band of the GC circuit 5902 has an effect that it can be set to the band of baseband digital data.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIG. 60 or FIG. 61 indicate the same functions. In this embodiment, the digital voice system A is also used.
Regarding the GC circuit, an envelope detection circuit 6101 is used in the same way as in FIG. 61, but the insertion position of the AGC circuit is between the BPF 414 and the synchronous detection circuit 415 in FIG.
Different from the example of 1. According to the embodiment of FIG. 64, since AGC is applied while observing the output of the digital voice system, the operation can be performed in the optimum state for the digital voice system, and the AGC is performed so that the input level of the synchronous detection circuit 415 becomes constant. Since the circuit 6001 is controlled, there is an effect that the synchronous detection circuit 415 can always operate in the best state as in FIG.

The example of the AGC circuit shown in FIGS. 59, 60, 61, and 64 has been described above with respect to the embodiment shown in FIG. 7. However, the embodiment shown in FIG. 4, FIG. 11, and FIG. It can also be used against. Further, the comparator 805, which constitutes the ternary discrimination circuit 701 using the AGC control signal,
It is also possible to control the reference voltage of 806 to optimize the slice level according to the strength of the input level. FIG. 65 shows the embodiment, and the same reference numerals as those in FIG. 8 indicate the same functions.
Reference numeral 6501 denotes a data slice level signal input terminal,
It is similar to the AGC control signal. Reference numerals 6502 and 6503 denote reference voltage control circuits, which are used for data slice level signal 65.
01 is adjusted so that the reference voltages of the comparators 805 and 806 are optimized. According to the example of FIG. 65, there is an effect that ternary discrimination can be performed at the optimum slice level.

Another embodiment of the transmission signal reproducing apparatus of the present invention is shown in FIG. The same reference numerals as those in FIG. 7 or FIG. 35 indicate the same functions. The difference from FIG. 7 is that a filter 3501 and a frequency conversion circuit 3502 are provided in order to lower the demodulation frequency of a digitally encoded and multiplex-transmitted audio signal below the frequency for video signal demodulation.

According to this embodiment, the frequency conversion circuit 403
The video signal is demodulated at the intermediate frequency of the output (58.75 MHz is commonly used in Japanese terrestrial television), and the intermediate frequency of the output of the frequency conversion circuit 3502 is lower (for example, about 5 MHz). Since the audio signal transmitted after being digitally encoded by the digital signal is demodulated by the carrier wave recovery circuit 41 used in the synchronous detection circuit 415.
The phase error due to the circuit delay time of the carrier wave reproduced in 6 is reduced by lowering the frequency, and there is an effect that the audio signal stably digitally encoded and transmitted can be demodulated.

FIG. 67 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The received signal is the same as that in the case of FIG. 7, and the same reference numerals as those in FIG. 7 or FIG. 36 indicate the same functions. The frequency conversion circuit 3502 shown in FIG.
602 and a voltage control type local oscillation circuit 3603.

66 is different from FIG. 66 in that the signal transmitted orthogonally to the video signal is detected by the synchronous detection circuit 415 using the carrier reproduced by the carrier reproduction circuit 416 in FIG. In 67, the reference signal generator 3601 and the carrier wave are included by utilizing the fact that the modulation by the digitally encoded audio signal and the modulation by the video signal have an orthogonal relationship and the DC component of the modulation by the digitally encoded audio signal is small. Synchronous detection circuit 4 for detecting the phase difference from the intermediate frequency signal
15 and the low-pass filter 3604, and by feeding back to the voltage-controlled local oscillator 3602, the carrier of the intermediate frequency is synchronized with the output of the reference signal generator and the output of the synchronous detection circuit 415 is used as the detection output. Especially.

According to this embodiment, the reference signal generator 360
Since this is a negative feedback loop in which the intermediate frequency for demodulation matches the frequency of 1, the frequency shift of the band-pass filter 414 and the demodulation frequency drift due to the frequency drift of the frequency conversion circuit 403 and the like are small, and further more than the embodiment shown in FIG. 66. It has the effect of stable demodulation.

Although the examples of FIGS. 66 and 67 have been described with respect to the embodiment of FIG. 7, they can be used for other embodiments of the receiving apparatus.

FIG. 68 shows another circuit example having the functions of the three-value discrimination circuit 701, the three-value binary conversion circuit 702 and the code discrimination circuit 703. 7 and 8 indicate the same functions, and 6801 and 6802 are latches.
68 will be described with reference to FIG. 69. In FIG. 69,
(A) is ternary digital data, (b) is comparator 805 output, (c) is comparator 806 output, (d)
Is a clock signal, (e) is a latch 6801 output, (f)
Is a latch 6802 output, and (g) is binary digital data (RS-flip-flop 809 output). The operation until the outputs of the comparators 805 and 806 are obtained is the same as in FIG. The outputs of the comparators 805 and 806 are output to the clock recovery circuit 704 by the latches 6801 and 6802.
It is latched at the timing of (d) in FIG. 69 by using the clock signal reproduced in step S6, and becomes a digital signal synchronized with the clock signal. The outputs of the latches 6801 and 6802 are input to the RS-flip-flop 809 as shown in FIG. 68, and are identified as digital signals by the same operation as in FIG.
Demodulate the value digital code. According to FIG. 68, the circuit configuration is simple and the ternary digital data is converted into the one shown in FIG.
Even if unnecessary noise such as 6901 and 6902 of (a) is mixed, it does not affect the demodulated binary digital code at all if it is not at the rising edge of the clock signal, and does not deteriorate the code error rate characteristic. effective.

FIG. 70 shows another circuit example of the three-value discrimination circuit 701 and the three-value binary conversion circuit 702. 7 or 8
The same reference numerals as those in FIG.
2 is a gate, and 7003 is a gate control circuit. FIG. 71
70 is a timing chart for explaining FIG. 70, (a) is ternary digital data, (b) is comparator 805 output, (c) is comparator 806 output, (d) is gate control signal, (e) is Gate 7001 output, (f) gate 7002 output, (g) binary digital data (RS
-Flip-flop 809 output). The operation until the outputs of the comparators 805 and 806 are obtained is the same as in FIG. Comparator 805,806 output is gate 7
001 and 7002 are input and gated. The gate signal is generated by the gate control circuit 7003 using the clock obtained from the clock reproduction circuit 704, and the comparators 805, 8 are provided as shown in FIG. 71 (d).
It is assumed that the normal rising edge of 06 output is captured. As a result, the outputs of the comparators 805 and 806 are shown in FIG.
It is gated as in (f) and sent to the RS-flip-flop 809. After that, the same operation as in FIG.
Value Demodulate digital data. According to the example of FIG. 70,
71A, 71 of FIG.
Even when unwanted noise such as 02 is mixed in, unless it is during the gate ON of the gate signal, it has no effect on the demodulated binary digital code and has the effect of not deteriorating the code error rate characteristic. Note that the gate control circuit 7003
In this case, it is possible to provide a plurality of gate signals having different gate pulse intervals, and monitor the code error rate to determine which pulse interval to select to bring the code error rate to the optimum state. it can. It is also possible to change the gate pulse timing by monitoring the code error rate and the like to bring the code error rate to the optimum state.

FIG. 72 shows another circuit example of the ternary discrimination circuit 701 and the ternary binary conversion circuit 702. The same reference numerals as those in FIG. 7 or FIG. 8 indicate the same functions.
Is a memory circuit, 7203 is a memory control circuit, and 7204 is a digital signal processing circuit. Similarly to FIG. 70, the example of FIG. 72 is also an example having a noise removing function when unnecessary noise is mixed in the ternary digital data. The clock reproduction circuit 704 reproduces a clock signal that is n times the data transmission cycle, and using this, the memory control circuit 7203 divides the outputs of the comparators 805 and 806 into digital data by dividing them by 1 clock.
It is stored in 02. Then, the digital signal processing circuit 72
At 04, a set pulse indicating a High portion of the ternary digital data or a reset pulse indicating a Low portion of the ternary digital data close to the normal data sample point is selected to remove unnecessary noise mixed in the ternary digital data. After that, the RS-flip-flop 809 converts it into binary digital data. According to the embodiment of FIG. 72, there is an effect that various digital processes are performed and unnecessary noise mixed in the ternary digital data is removed. It is also possible to adjust the normal data sample points with reference to the code error rate and the like to select the optimum sample point that gives the best code error rate. In addition, the comparator 80
5,806 outputs are input to the memory control circuit 7203 and the digital signal processing circuit 7204, and when a plurality of set pulses appear before the reset pulse comes, and a plurality of reset pulses appear before the set pulse come. Sometimes you can choose a regular data sampling point.

[0141]

According to the present invention, when a carrier wave of a signal transmitted by amplitude modulation and a carrier wave of quadrature phase are modulated by a multiplexed signal, signal data is spectrally processed and multiplexed signal is transmitted. In demodulating the signal, the spectrum-processed signal reproducing means subtracts the signal data and the same signal data of the opposite phase delayed by the time corresponding to the repetition of the signal data by the delay means. It has the effect of improving
Due to the correlation of the video signal for each horizontal scanning period, interference from the video signal can be offset and reduced, and stable signal data can be received.

[Brief description of drawings]

FIG. 1 is a block diagram showing an example of a transmission signal generator of the present invention,

FIG. 2 is a spectrum diagram of a transmission signal of the present invention,

FIG. 3 is a vector diagram of a transmission signal of the present invention,

FIG. 4 is a block diagram showing an example of a transmission signal reproducing apparatus according to the present invention;

FIG. 5 is a block diagram showing another example of the transmission signal generator of the present invention,

6 is a waveform diagram explaining the operation of the apparatus of FIG.

FIG. 7 is a block diagram showing another example of the transmission signal reproducing apparatus of the present invention;

8 is a block diagram showing an example of the three-value discrimination circuit shown in FIG.

9 is a waveform chart explaining the operation of FIG. 7 and FIG.

FIG. 10 is a block diagram of still another example of the transmission signal generator of the present invention,

FIG. 11 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 12 is a block diagram of another example of the spectrum suppression processing circuit of the transmission signal generator of the present invention,

13 is a schematic diagram showing a data sequence for explaining the operation of FIG. 12,

FIG. 14 is a schematic diagram of an example of a transmission data pattern of the present invention shown in FIGS. 12 and 13.

FIG. 15 is a vector diagram of a video color subcarrier for explaining the present invention;

FIG. 16 is a block diagram showing a general configuration of a comb filter,

FIG. 17 is a waveform diagram explaining the operation,

FIG. 18 is a block diagram showing still another example of the transmission signal reproducing apparatus of the present invention;

19 is a schematic diagram showing a data sequence for explaining the operation of FIG.

FIG. 20 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 21 is a schematic diagram showing a data sequence for explaining the operation of FIG. 20;

22 is a characteristic diagram showing characteristics of the device of FIG. 20,

FIG. 23 is yet another block diagram of the spectrum suppression processing circuit of the transmission signal generator of the present invention;

24 is a schematic diagram showing a data sequence for explaining the operation of FIG. 23,

FIG. 25 is a schematic diagram of an example of a transmission data pattern of the present invention shown in FIGS. 23 and 24;

FIG. 26 is a schematic diagram of a data sequence for explaining the operation,

FIG. 27 is a block diagram of still another example of the transmission signal generator of the present invention,

28 is a waveform chart showing an example of signals for explanation of FIG.

FIG. 29 is a schematic diagram of an example of interleaving processing of the transmission signal generator of the present invention,

FIG. 30 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 31 is a block diagram of still another example of the transmission signal generator of the present invention,

FIG. 32 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 33 is a block diagram of still another example of the transmission signal generator of the present invention,

FIG. 34 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 35 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 36 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 37 is a block diagram of still another example of the spectrum suppression processing circuit of the transmission signal generator of the present invention;

38 is a schematic diagram of a data sequence for explaining the operation of FIG. 37,

FIG. 39 is a schematic diagram of an example of a transmission data pattern of the present invention shown in FIGS. 37 and 38;

FIG. 40 is a schematic diagram of a data sequence for explaining the operation;

FIG. 41 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 42 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 43 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 44 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 45 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 46 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 47 is a block diagram of still another example of the transmission signal generator of the present invention,

FIG. 48 is a block diagram of an example of a ternary conversion circuit of the transmission signal generator of the present invention;

49 is a waveform chart explaining the operation of the circuit of FIG. 48;

FIG. 50 is a block diagram of another example of the three-value conversion circuit of the transmission signal generator of the present invention,

51 is a waveform chart explaining the circuit operation of FIG. 50;

52 is a block diagram of still another example of the transmission signal generator of the present invention, FIG.

FIG. 53 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 54 is a block diagram of another example of a ternary identification circuit of the transmission signal reproducing apparatus of the present invention,

55 is a waveform chart explaining the circuit operation of FIG. 54;

FIG. 56 is a block diagram of still another example of a three-value identifying circuit of the transmission signal reproducing apparatus of the present invention;

57 is a waveform chart explaining the operation of FIG. 56;

FIG. 58 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 59 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 60 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 61 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 62 is a block diagram of an example of an envelope detection circuit of the transmission signal reproducing apparatus of the present invention;

63 is a waveform chart explaining the operation, FIG.

FIG. 64 is a block diagram of still another example of the transmission signal reproducing apparatus of the present invention;

FIG. 65 is a block diagram of still another example of a ternary identification circuit of the transmission signal reproducing apparatus of the present invention;

FIG. 66 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 67 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 68 is a block diagram of still another example of a three-value identifying circuit of the transmission signal reproducing apparatus of the present invention;

69 is a waveform chart explaining the operation of the circuit of FIG. 68;

FIG. 70 is a block diagram of still another example of a ternary identification circuit of the transmission signal reproducing apparatus of the present invention;

71 is a waveform diagram explaining the operation of FIG. 70,

FIG. 72 is a block diagram showing still another example of a three-value identification circuit of the transmission signal reproducing apparatus of the present invention.

[Explanation of symbols]

 110 ... Video signal carrier generation circuit, 114, 114 ... Spectral suppression processing circuit, 116 ... Phase shift circuit, 117 ... Modulation circuit, 118 ... Equalizer, 119 ... Addition circuit, 120 ... Transmission VSB filter, 201 ... Video signal transmission signal , 205 ... Multiplex signal transmission signal spectrum, 301 ... Video signal carrier vector, 302 ... Modulated wave vector of multiplex transmitted signal, 414 ... Bandpass filter, 415 ... Synchronous detection circuit, 416 ... Carrier recovery Circuits, 417, 417 ... Spectrum suppression processing signal reproduction circuit, 501 ... Delay circuit, 502 ... Subtraction circuit, 701, 701 ... Three-value discrimination circuit, 702, 702 ... Three-value binary conversion circuit, 703 ... Code discrimination circuit, 704 ... Clock recovery circuit, 1001 ... Digital modulation circuit, 1101 ... Code identification circuit Reference numeral 1102 ... Clock recovery circuit, 1103 ... Digital demodulation circuit, 1202 ... Time axis compression circuit, 1204 ... Inverter, 1205 ... Delay circuit, 1206 ... Changeover switch, 1801 ... Code identification circuit, 1802 ... Clock recovery circuit, 1803 ... Changeover Circuit 1804 ... Time axis expansion circuit 1805 ... Timing recovery circuit 2001 ... Subtraction circuit 2002 ... Delay circuit 2303 ... Inverter 2304 ... Delay circuit 2305 ... Changeover switch 2702 ... Control signal generation circuit 3001 ... Control signal Reproduction circuit, 3101 ... Subtraction circuit, 3102 ... Delay circuit, 3201 ... Addition circuit, 3202 ... Delay circuit, 3301 ... Pre-emphasis circuit, 3401 ... De-emphasis circuit, 3502, 3502 ... Frequency conversion circuit, 3601 ... Reference signal generation times , 3602 ... mixing circuit, 3603 ... voltage control type local oscillator, 3604 ... low-pass filter.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Akihide Okuda 292 Yoshida-cho, Totsuka-ku, Yokohama-shi, Kanagawa Inside Home Appliances Research Laboratory, Hitachi, Ltd. (72) Inventor Yoshitaka Hotta 292 Yoshida-cho, Totsuka-ku, Yokohama-shi, Kanagawa Hiritsu Video Engineering Co., Ltd.

Claims (7)

[Claims] 1. A multiplex in which digitized signal data other than a video signal obtained by subjecting a quadrature carrier having a quadrature phase relationship to a carrier subjected to vestigial sideband amplitude modulation with a video signal to spectral processing of a signal other than the vestigial sideband amplitude modulated video signal. A device for receiving and reproducing a multiplex transmission signal, which is subjected to carrier suppression amplitude modulation with a signal, is synthesized with the carrier wave subjected to the vestigial sideband amplitude modulation and is transmitted, and carrier wave reproduction means for reproducing a carrier wave from the multiplex transmission signal. A synchronous detection means for synchronously detecting the multiplex transmission signal by the output signal of the carrier wave reproduction means, and a spectrum processing signal reproduction means for demodulating multiplex signal data from the output signal of the synchronous detection means. Multiplex transmission signal reproduction device. 2. A multiplex in which digitized signal data other than a video signal obtained by performing the vestigial sideband amplitude modulation on a quadrature carrier having a quadrature phase relationship with a carrier subjected to vestigial sideband amplitude modulation by a video signal has been spectrally processed. A device for receiving and reproducing a multiplex transmission signal, which is subjected to carrier suppression amplitude modulation with a signal, is synthesized with the carrier wave subjected to the vestigial sideband amplitude modulation and is transmitted, and carrier wave reproduction means for reproducing a carrier wave from the multiplex transmission signal. Variable gain amplifying means for amplifying the multiplex transmission signal, synchronous detecting means for synchronously detecting the output signal of the variable gain amplifying means with the output signal of the carrier wave reproducing means, and multiplexed signal data from the output signal of the synchronous detecting means And a variable gain amplification means for detecting an amplitude level from an input signal or an output signal of the synchronous detection means. Multiplex transmission signal reproducing apparatus characterized by digit. 3. Multiplexing in which digitized signal data other than a video signal obtained by subjecting a quadrature carrier having a quadrature phase relationship to a carrier subjected to vestigial sideband amplitude modulation with a video signal to spectrum processing to a signal other than the video signal subjected to vestigial sideband amplitude modulation. A device for receiving and reproducing a multiplex transmission signal, which is subjected to carrier wave suppression amplitude modulation with a signal, and is synthesized with the carrier wave subjected to the residual sideband amplitude modulation, and is transmitted, comprising: first frequency conversion means for channel selection frequency conversion, Detection means for detecting the output signal of the first frequency conversion means to demodulate the video signal subjected to the residual sideband amplitude modulation, and a second frequency for further frequency-converting the output signal of the first frequency conversion means. Conversion means, carrier wave reproduction means for reproducing the carrier wave after frequency conversion from the output signal of the second frequency conversion means, and the second frequency conversion with the output signal of the carrier wave reproduction means Synchronous detection means for synchronously detecting the output signal of the stage, multiplex transmission signal reproducing apparatus characterized in that a spectrum processed signal reproducing means for demodulating a multiplex signal data from the output signal of the synchronous detection means. 4. The spectrum processed signal reproducing means,
A delay unit that delays the output signal of the synchronous detection unit for a certain period related to a period corresponding to the horizontal scanning period or the vertical scanning period of the video signal; an output signal of the delay unit and an output signal of the synchronous detection unit. An arithmetic processing unit for performing a subtraction process is provided.
The multiplex transmission signal reproducing device according to any one of items 1 to 3. 5. The spectrum processed signal reproducing means,
A delay unit that delays the output signal of the synchronous detection unit for a certain period related to a period corresponding to the horizontal scanning period or the vertical scanning period of the video signal; an output signal of the delay unit and an output signal of the synchronous detection unit. 4. An arithmetic processing means for performing a subtraction process, and a cutoff means for conducting only signal data of an output signal of the arithmetic processing means for a necessary period, provided in claim 1. Multiplex transmission signal reproduction device. 6. The spectrum processed signal reproducing means,
A delay unit that delays the output signal of the synchronous detection unit for a certain period related to a period corresponding to the horizontal scanning period or the vertical scanning period of the video signal; an output signal of the delay unit and an output signal of the synchronous detection unit. Arithmetic processing means for performing a subtraction process, interruption means for conducting only signal data of the output signal to the arithmetic processing means for a required period, and time axis expansion means for expanding the output signal of the interruption means in the time axis are provided. The multiplex transmission signal reproducing device according to any one of claims 1 to 3, wherein: 7. The spectrum processed signal reproducing means is,
A delay unit that delays the output signal of the synchronous detection unit for a certain period related to a period corresponding to the horizontal scanning period or the vertical scanning period of the video signal; an output signal of the delay unit and an output signal of the synchronous detection unit. Arithmetic processing means for performing subtraction processing, and cutoff means for conducting only signal data of an output signal from the arithmetic processing means for a necessary period are provided, and spectrum processing is performed in accordance with a control code transmitted every vertical scanning period of a video signal. 4. The multiplex transmission signal reproducing device according to claim 1, wherein the signal is reproduced.